6.012 - Microelectronic Devices and Circuits - Fall 2005
Lecture 14-1
Lecture 14 - Digital Circuits (III)
CMOS
October 27, 2005
Contents:
1. Complementary MOS (CMOS) inverter: introduction
2. CMOS inverter: noise margins
3. CMOS inverter: propagation delay
4. CMOS inverter: dynamic power
Reading assignment:
Howe and Sodini, Ch. 5, §5.4
Announcements:
• Cadence tutorial by Kerwin Johnson in place of regular recitations on Friday 10/28
6.012 - Microelectronic Devices and Circuits - Fall 2005
Lecture 14-2
Key questions
• How does CMOS work?
• What is special about CMOS as a logic technology?
• What are the key design parameters of a CMOS inverter?
• How can one estimate the propagation delay of a
CMOS inverter?
• Does CMOS burn any power?
Lecture 14-3
6.012 - Microelectronic Devices and Circuits - Fall 2005
1. Complementary MOS (CMOS) Inverter
Circuit schematic:
VDD
VIN
VOUT
CL
Basic operation:
•VIN = 0 ⇒ VOU T = VDD
VGSn = 0 < VT n ⇒ NMOS OFF
VSGp = VDD > −VT p ⇒ PMOS ON
•VIN = VDD ⇒ VOU T = 0
VGSn = VDD > VT n ⇒ NMOS ON
VSGp = 0 < −VT p ⇒ PMOS OFF
Lecture 14-4
6.012 - Microelectronic Devices and Circuits - Fall 2005
Output characteristics of both transistors:
IDn
-IDp
VGSn
VSGp
VGSn=VTn
VSGp=-VTp
0
0
0
VDSn
0
VSDp
Note:
VIN = VGSn = VDD − VSGp ⇒ VSGp = VDD − VIN
VOU T = VDSn = VDD − VSDp ⇒ VSDp = VDD − VOU T
IDn = −IDp
Combine into single diagram of ID vs. VOU T with VIN as
parameter.
Lecture 14-5
6.012 - Microelectronic Devices and Circuits - Fall 2005
VDD
ID
VIN
VOUT
VDD-VIN
VIN
CL
0
0
VOUT
no current while id
idling
ling in any lo
logic
gic state
state..
Transfer function:
NMOS cutoff
PMOS triode
VOUT
NMOS saturation
PMOS triode
VDD
NMOS saturation
PMOS saturation
NMOS triode
PMOS saturation
NMOS triode
PMOS cutoff
0
0
VTn
VDD+VTp VDD VIN
ail-to-rail” lo
”r
”rail-to-rail”
logic:
gic: lo
logic
gic levels are 0 and VDD
high |Av | around logic threshold ⇒ good noise margins
6.012 - Microelectronic Devices and Circuits - Fall 2005
Lecture 14-6
Transfer characteristics of CMOS inverter in WebLab:
Lecture 14-7
6.012 - Microelectronic Devices and Circuits - Fall 2005
2. CMOS inverter: noise margins
VOUT
NML
VDD
Av(VM)
VM
0
0
VILVM VIH
VDD VIN
NMH
• Calculate VM
• Calculate Av (VM )
• Calculate N ML and N MH
2 Calculate VM (VM = VIN = VOU T )
At VM both transistors saturated:
1 Wn
IDn =
µnCox (VM − VT n)2
2 Ln
−IDp =
1 Wp
µpCox (VDD − VM + VT p )2
2 Lp
Lecture 14-8
6.012 - Microelectronic Devices and Circuits - Fall 2005
Define:
kn =
Wn
Wp
µnCox, kp =
µpCox
Ln
Lp
Since:
IDn = −IDp
Then:
1
1
kn(VM − VT n )2 = kp(VDD − VM + VT p )2
2
2
Solve for VM :
VM =
VT n +
�
�
� kp
�
kn
(VDD + VT p )
1+
�
�
� kp
�
kn
Usually, VT n and VT p fixed and VT n = −VT p ⇒ VM engineered through kp/kn ratio
Lecture 14-9
6.012 - Microelectronic Devices and Circuits - Fall 2005
• Symmetric case: kn = kp
VM =
VDD
2
This implies:
kp
=1=
kn
Wp
µC
Lp p ox
Wn
Ln µn Cox
Wp
µ
Lp p
Wn
Ln 2µp
Wp
Wn
⇒
2
Lp
Ln
Since usually Lp Ln ⇒ Wp 2Wn.
VOUT
VDD
VIN=VOUT
kn=kp
0
0
VTn
VM VDD+VTp VDD
VIN
Lecture 14-10
6.012 - Microelectronic Devices and Circuits - Fall 2005
• Asymmetric case: kn kp , or
Wn
Ln
Wp
Lp
VM VT n
NMOS turns on as soon as VIN goes above VT n .
• Asymmetric case: kn kp , or
Wn
Ln
Wp
Lp
VM VDD + VT p
PMOS turns on as soon as VIN goes below VDD + VT p .
Can engineer VM anywhere betw
between
een VT n and VDD + VT p .
VOUT
VDD
VIN=VOUT
kn>>kp
kn=kp
kn<<kp
0
0
VTn
VM VDD+VTp VDD
VIN
Lecture 14-11
6.012 - Microelectronic Devices and Circuits - Fall 2005
2 Calculate Av (VM )
VDD
Small-signal model:
VIN
S2
+
vsg2=-vin
-
+
vin
-
+
gmpvsg2
rop
G2
D2
G1
D1
+
vgs1
gmnvgs1
vout
ron
-
-
S1
G1=G2
D1=D2
+
vin
+
gmnvin
-
gmpvin
ron//rop
vout
-
S1=S2
Av = −(gmn + gmp )(ron //rop )
This can be rather large.
VOUT
Lecture 14-12
6.012 - Microelectronic Devices and Circuits - Fall 2005
2 Noise margins
VOUT
NML
VDD
Av(VM)
VM
0
0
VILVM VIH
VDD VIN
NMH
• Noise-margin-low:
VIL
VDD − VM
= VM −
|Av |
Therefore:
N ML = VIL − VOL = VIL = VM −
In the limit of |Av | → ∞:
N ML → VM
VDD − VM
|Av |
Lecture 14-13
6.012 - Microelectronic Devices and Circuits - Fall 2005
• Noise-margin-high:
VOUT
NML
VDD
Av(VM)
VM
0
0
VILVM VIH
VDD VIN
NMH
VIH = VM (1 +
1
)
|Av |
and
N MH = VOH − VIH
1
= VDD − VM (1 +
)
|Av |
In the limit of |Av | → ∞:
N MH → VDD − VM
When VM =
VDD
2
⇒ N ML = N MH =
VDD
2
Lecture 14-14
6.012 - Microelectronic Devices and Circuits - Fall 2005
3. CMOS inverter: propagation delay
Inverter propagation delay: time delay between input and
output signals; key figure of merit of logic speed.
Typical propagation delays: < 1 ns.
Complex logic system has 20-50 propagation delays per
clock cycle.
Estimation of tp : use square-wave at input
VIN
VDD
tCYCLE
0
VOUT
t
tPHL
tPLH
VDD
50%
tCYCLE
0
t
Average propagation delay:
1
tp = (tP HL + tP LH )
2
Lecture 14-15
6.012 - Microelectronic Devices and Circuits - Fall 2005
2 Propagation delay high-to-low:
VDD
VOUT:
HI LO
VIN:
LO HI
CL
VDD
VIN=0
VDD
VDD
VOUT=VDD
VOUT=0
VIN=VDD
VIN=VDD
VOUT=VDD
CL
CL
t=0-
t=0+
CL
t
During early phases of discharge, NMOS is saturated and
PMOS is cut-off.
Time to discharge half of CL:
tP HL 1
2 charge
of CL@t = 0−
discharge current
Lecture 14-16
6.012 - Microelectronic Devices and Circuits - Fall 2005
Charge in CL at t = 0−:
QL(t = 0−) = CLVDD
Discharge current (NMOS in saturation):
IDn =
Wn
µnCox (VDD − VT n )2
2Ln
Then:
tP HL CLVDD
Wn
2
Ln µnCox (VDD − VT n )
ID
VDD-VIN
VIN
0
0
VOUT
Lecture 14-17
6.012 - Microelectronic Devices and Circuits - Fall 2005
2 Propagation delay low-to-high:
VDD
VOUT:
LO HI
VIN:
HI LO
CL
VDD
VDD
VDD
VOUT=0
VIN=VDD
VOUT=VDD
VIN=0
VIN=0
VOUT=0
CL
CL
t=0-
t=0+
CL
t
During early phases of charge, PMOS is saturated and
NMOS is cut-off.
Time to charge half of CL:
tP LH 1
2 charge
of CL@t = ∞
charge current
Lecture 14-18
6.012 - Microelectronic Devices and Circuits - Fall 2005
Charge in CL at t = ∞:
QL(t = ∞) = CLVDD
Charge current (PMOS in saturation):
−IDp
Wp
=
µp Cox(VDD + VT p )2
2Lp
tP LH
CLVDD
Wp
2
µ
C
(V
+
V
)
p
ox
DD
T
p
Lp
Then:
ID
VDD-VIN
VIN
0
0
VOUT
6.012 - Microelectronic Devices and Circuits - Fall 2005
Key dependencies of propagation delays:
• VDD ↑⇒ tp ↓
Reason: VDD ↑⇒ Q(CL) ↑, but also ID ↑
Trade-off: VDD ↑, more power usage.
• L ↓⇒ tp ↓
Reason: L ↓⇒ ID ↑
Trade-off: manufacturing costs!
Lecture 14-19
6.012 - Microelectronic Devices and Circuits - Fall 2005
Lecture 14-20
Components of load capacitance CL:
• following logic gates: must add capacitance presented
by each gate of every transistor the output is connected to
• interconnect wire that connects output to input of
following logic gates
• own drain-to-body capacitances
CL = CG + Cwire + CDBn + CDBp
VDD
VIN
VOUT
Cwire
[See details in Howe & Sodini §5.4.3]
6.012 - Microelectronic Devices and Circuits - Fall 2005
Lecture 14-21
4. CMOS inverter: dynamic power
• In any of the two logic states: one transistor always
OFF ⇒ zero static power dissipation.
• Dynamic power?
Every complete transient, CL is charged up to VDD and
then discharged to 0
⇒ energy dissipated
⇒ clock frequency ↑ ⇒ dissipated power ↑
Lecture 14-22
6.012 - Microelectronic Devices and Circuits - Fall 2005
2 Dynamic power dissipated while charging load
VDD
VOUT
iDD(t)=iC(t)
VIN=HI
LO
VOUT=LO
CL
VDD
HI
0
t
0
1. Energy provided by battery during transient:
ES =
=
�
dvOU T
∞
∞
dt
V
i
(t)dt
=
V
C
DD C
DD 0
L
0
dt
�
VDD
2
CLVDD 0
dvOU T = CLVDD
�
=
2. Energy stored in capacitor during transient:
1
2
∆EC = EC (t = ∞) − EC (t = 0) = CL VDD
2
3. Energy dissipated in PMOS during transient:
1
2
EP = ES − ∆EC = CLVDD
2
Lecture 14-23
6.012 - Microelectronic Devices and Circuits - Fall 2005
2 Dynamic power dissipated while discharging load
VDD
iDD(t)=0
VOUT
VDD
VIN=LO
HI
VOUT=HI
iC(t)
LO
CL
0
0
1. Energy provided by battery during transient:
ES =
�
∞
0 VDD iDD (t)dt
=0
2. Energy removed from capacitor during transient:
1
2
∆EC = EC (t = 0) − EC (t = ∞) = CL VDD
2
3. Energy dissipated in NMOS during transient:
1
2
EN = ∆EC = CLVDD
2
t
6.012 - Microelectronic Devices and Circuits - Fall 2005
Lecture 14-24
2 Energy dissipated in complete cycle
2
ED = EP + EN = ΣES = CLVDD
2 Power dissipation
If complete switching cycle takes place f times per second:
2
PD = f ED = f CLVDD
Fundamental
F
undamental trade-off between switching
switching speed
speed ant power
power
dissipation!
Key dependencies in dynamic power:
• f ↑⇒ PD ↑, charge and discharge CL more frequently
• CL ↑⇒ PD ↑, more charge being shuttled around
• VDD ↑⇒ PD ↑, more charge being shuttled around
6.012 - Microelectronic Devices and Circuits - Fall 2005
Lecture 14-25
Key conclusions
• Key features of CMOS inverter:
– no current while idling in any logic state
– ”rail-to-rail” logic: logic levels are 0 and VDD
– high |Av | around logic threshold ⇒ good noise
margins
• CMOS inverter logic threshold and noise margins engineered through Wn/Ln and Wp/Lp .
• Key dependences of propagation delay:
– VDD ↑ ⇒ tp ↓
– L ↓ ⇒ tp ↓
• Dynamic power dissipated in CMOS:
2
PD = f ED = f CLVDD
Fundamental trade-off between switching speed and
power dissipation.