A Five-Phase Brushless DC-Machine Direct Drive System
A Five-Phase Brushless DC-Machine Direct Drive System
M. Godoy Simões, Colorado School of Mines, Golden, Colorado, USA
P. Vieira Jr., Federal University of Pará Belém, Brazil
Abstract
The paper describes the design, analysis, simulation, modeling and control implementation of a high-torque, low-speed,
multiphase, permanent magnet, brushless dc machine. The main focus is on issues regarding the high-level modeling,
comprised of a transient model, in conjunction with corresponding experimental evaluation. The general assumption of
ideal rectangular current waveforms for brushless-dc machines is not encountered in practice; the existing distortions
can be modeled by incorporating mutual inductance and armature reaction in order to avoid erroneous control strategy
development. Analyses are made to put together modeling efforts with the expected behavior so as to build a model of the
expected behavior so realistic simulation results can be verified. Coherent and consistent results were observed by comparing simulation and experimentation. A digital signal processing (DSP) system control was developed to implement the
strategies that corroborate the work.
Introduction
The majority of electrical drive systems are three-phase systems.
Recently some quasi-four-phase systems employing neutral leg
also have been used for harmonic optimization and fault-tolerant
drives. Three-phase drive systems have been widely used for
years because of the availability of such machines, their inverters,
modeling and control. However, polyphase schemes have been
used in the past in drive systems where an induction machine with
asymmetric windings has three-phase sets advanced by 30 degrees
for twelve-step industrial applications. Such multiphase drives are
likely to be limited to specialized applications where high performance and reliability are required (such as EV, HEV, aerospace,
ship propulsion and high power applications) and when cost
requirements are not so oppressive when compared to the overall
environment.
The recent literature indicates several advantages for using a multiphase multi-pole electrical machine in hub-wheel systems –
high-torque low-speed motors can directly drive systems, avoiding mechanical losses incurred by the clutch, reduction and
differential gear during power transmission from the motor to the
wheels. This work presents the design, analysis, simulation,
modeling and control implementation of a high-torque, lowspeed, multiphase, permanent magnet, brushless dc-machine. The
paper focuses on issues regarding the high-level modeling,
comprised of a transient model, in conjunction with corresponding experimental evaluation. Analyses were made to put together
the modeling efforts with the expected behavior in order to have
realistic simulation results verified by the experimental setup;
comprehensive experimental results corroborate the work.
DSP microcontroller offers the advantages of a single chip system
combined with the power of a high performance DSP core and is
actually the ideal device to implement the complex control laws
required for high performance drives where a full set of functions
required for ac drive control systems include transfer functions,
filter algorithms, and some special ac motor control functions.
Loop compensator and matrix vector multiplication that is
required for state space control and generation can be implemented
within some hundreds of microseconds.
The second important issue about the high cost of multiphase
drives can be properly addressed with the support of modeling
multi-machine multi-converter systems (MMS) [1]. The MMS
formalism is used to build a representation of a system where the
power structure can have several couplings (electrical, magnetic
and mechanical); there are four conversion structures, namely
mono-structures, multi-structures, upstream and downstream.
Within such a MMS analysis and modeling approach several
applications have been described. A five-leg inverter has been
shown to be able to supply two three-phase induction motors; such
power structure allows reduced global cost and weight. Other
structures allow a back-to-back connection of a generator (or a
three-phase grid) with a three-phase machine with just one fivephase inverter [2]. The torque control of electric vehicles with
separated wheel drives has been recently addressed under the
MMS approach [3, 4] and a polyphase cartesian vector approach
to control polyphase machines demonstrated that a single inverter
with (2N + 1) phases can independently control (N) ac-machines
connected in series with appropriate phase swapping [5]. Such
recent MMS initiatives have been driving forces for further
research and development of multiphase machines as the one
presented in this paper.
Five-phase brusheless dc-machine
A high number of phases yield a smaller magnetic yoke and
decreased volume and weight. However, the number of poles is
restricted physically to the size of the permanent magnets and the
rotor diameter. Multiple phase arrangement for electrical
machines minimizes torque ripple, increases power density and
improves fault-tolerance in respect to open-circuit legs. The criticism against a higher number of phases is because of its more
complex inverter control scheme and higher cost.
Control complexity is easily managed with the new generation of
DSP controllers targeted to high performance motion systems. A
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
Analysis and modeling
A radial flux-based motor was designed to be applied as an inwheel, high-torque, and low-speed motor direct drive system [6].
The machine is composed of a rotor with 275 mm of external
diameter. Magnets are bonded on the internal surface making up
the twelve pole structure [6, 7], where five phases were accommodated within 60 slots (12 coils for each phase) in a double-layer
lap winding construction. With a built-in shaft optical sensor, the
motor phase windings are excited sequentially, the optical position
system addresses a lookup table, a phase is turned on at the same
15
M. Godoy Simões, P. Vieira Jr.
Fig. 1: Five-phase brushless machine control block diagram.
time that one coil leaves the polar section. The permanent magnet
flux produces a trapezoidal back-EMF and the currents must be
commanded to be ideally in phase with the back-EMF voltages.
Table 1 presents the nominal parameters for the machine under
consideration. A Motorola 56824 embedded board is integrated
into the feedback control system as indicated in Fig. 1.
a)
b)
Fig. 2: Linear commutated equivalent circuits, (a) Forced
excitation, (b) Free-wheeling path
Table I: Motor Parameters
External diameter:
Axial length:
Power rating:
Poles:
Phases:
Nominal voltage:
Nominal current:
Rated speed:
Rated torque:
16
275 mm
130 mm
3.2 HP
12
5
140 V
7.5 A
750 rpm
30 Nm
The DSP board was specially designed with a dual-port memory
with shared addresses with the PC host, easing the development
and high-speed communication needs. The torque control loop is
described later. Detailed simulation studies have been performed
initially in order to fully develop the control strategy. The concept
of commutated linear equivalent circuits was applied to the
machine, i.e., every 72 electrical degrees two equivalent circuits
valid for forced and freewheeling conditions were devised. Figs. 2
(a) and (b) show the first driving stage where transistors Q1, Q3,
Q2 and Q4 are on, impressing excitation for phases A, B, D and
E, while there is still current flowing on the machine coils due to
the last driving stage as indicated by the free-wheeling path. The
equations (1) to (7) represent this condition. Such an approach is
applied subsequently to all the phases. Therefore, the modeling
was extended as indicated by the equivalent circuits per stage in
Fig. 3. A mechanism of switching all the equations, saving initial
conditions for next circuit and retrieving the currents from all
those difference equations was implemented in Simulink/Matlab.
Even though such a modeling approach has been used in the
literature [8-11], other important issues were found for accurate
machine mathematical modeling: the mutual inductance between
phases and the armature reaction because of the distortion that
occurs in existing brushless dc-machines [6, 7].
The mutual inductance can be considered by observing how the
air-gap flux is composed by all five contributions and building up
the inductance matrix (through experimental parameter identification of self and mutual inductance parameters). Equation (8)
shows that currents have been identified as state variables.
Equation (8) also shows that the inductance matrix needs to be
inverted and there is one phase with null current every time, i.e.
the five-phase system is reduced to a fourth-order system because
every 36 electrical degrees there are only four phases conducting,
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
A Five-Phase Brushless DC-Machine Direct Drive System
Fig. 3: Equivalent circuits for each commutation state
while a fifth phase is kept off. For dynamic simulation the inductance matrix must be numerically inverted each simulation stepsize and a Cholesky decomposition helped the matrix to be
expressed analytically permitting the dynamic simulation. The
same fourth-order system is equivalent every 36 electrical
degrees, as long as the equations have their variables redefined in
accordance with the flowing currents and the four integrator initial conditions introduced from the previous stage, i.e., each stage
has its own current matrix, back-EMF matrix, mutual matrix, transistor and diode-drop matrix.
As described in [6] and [7], the incorporation of armature reaction
in the simulation studies is absolutely necessary in order to have a
more realistic system response. From the energy-modeling point of
view, the induced voltage due to the changing electromagnetic
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
stored energy (back-EMF) delivers mechanical energy. Thus, the
torque for the five-phase BPM machine has to account for the airgap flux distortion by the armature reaction as shown in Equation
(9). The actual experimental evaluation of air-gap back-EMF was
used for torque calculation. Equation (9) needs two lookup tables, a
trapezoidal normalized function (Γω) due to the permanent magnet
induced motional voltage, and a triangular normalized function (Λc)
for the armature reaction voltage. Therefore, two experimental tests
for back-EMF were required – open-circuit back-EMF in the whole
speed operating range and full-loaded shaft machine in order to
obtain Γω and Λc. The experimental lookup tables for obtain Γω and
Λc can easily incorporate the effect of rotor position shift in respect
to the actual value, a problem that arises in real world. The drive
model is given with the following equations:
17
M. Godoy Simões, P. Vieira Jr.
(
)
(
)
dia
1
=
−4 Ria − 3ea′ + eb′ + ed′ + ee′ + 2Vs − 4Vq
dt
4L
dib
1
=
−4 Rib + ea′ − 3eb′ + ed′ + ee′ + 2Vs − 4Vq
4L
dt
(1)
( )
(
)
Lae = M cos 72 o = Lea = M cos 288o = M1
– mutual inductance between phase a and e
(2)
(
)
( )
Lbc = M cos 288o = Lcb = M cos 72 o = M1
– mutual inductance between phase b and c
(
did
1
=
−4 Rid + ea′ + eb′ − 3ed′ + ee′ − 2Vs + 4Vq
4L
dt
(
die
1
=
−4 Rie + ea′ + eb′ + ed′ − 3ee′ − 2Vs + 4Vq
dt 4 L
)
)
(3)
(
)
(
)
Lbd = M cos 216 o = Ldb = M cos 144 o = M2
– mutual inductance between phase b and d
(4)
(
)
(
)
Lbe = M cos 144 o = Lbe = M cos 216 o = M2
dia 1  2
=
− ea − Ria 

dt
L 3
(5)
did 1  1
=
ea − Rid 

dt
L3
(6)
die 1  1
=
ea − Rie 

dt
L3
(7)
– mutual inductance between phase b and e
(
)
( )
Lcd = M cos 288o = Ldc = M cos 72 o = M1
– mutual inductance between phase a and d
(
)
(
)
Lce = M cos 216 o = Lec = M cos 144 o = M2
– mutual inductance between phase e and c
where:
ea′ = ea + Lab
eb′ = eb + Lba
dib
di
di
di
+ Lac c + Lad d + Lae e
dt
dt
dt
dt
dia
di
di
di
+ Lbc c + Lbd d + Lbe e
dt
dt
dt
dt
(
)
( )
Lde = M cos 288o = Led = M cos 72 o = M1
– mutual inductance between phase d and e
ea, eb, ec, ed and ie
– back-EMF developed by phases a, b, c, d and e respectively.
di
di
di
di
ec′ = ec + Lca a + Lcb b + Lcd d + Lce e
dt
dt
dt
dt
ed′ = ed + Lda
ee′ = ee + Lea
dia
di
di
di
+ Lbd b + Ldc d + Lde e
dt
dt
dt
dt
dia
di
di
di
+ Leb b + Lec c + Led d
dt
dt
dt
dt
( )
Lab = M cos 288o = Lba = M cos 72 o = M1
– mutual inductance between phase a and b
(
)
(
)
Lac = M cos 218o = Lca = M cos 144 o = M2
– mutual inductance between phase a and c
(
)
(
)
Lad = M cos 144 o = Lda = M cos 216 o = M2
– mutual inductance between phase a and d
18
Lls
M
R
Vs
Vq
– leakage inductance
– air-gap inductance
– stator resistance
– inverter voltage supply
– diode and transistor voltage drop
−1
– self inductance of phase a, b, c, d and e respectively.
)
– current in phases a, b, c, d and e respectively.
M
d
Iabde ] = [ I ] −
⋅ [ Mabde ] ⋅
[
4L
dt


Laa = Lbb = Lcc = Ldd = Lee = Lls + M
(
ia, ib, ic, id and ie

−3 1 1 1 

 1 −3 1 1 
1 

 ⋅ [ E ] + Vs Vq
−
R
I
⋅
+
4
[
]
[
]

abde
 1 1 −3 1  abde
4L 



 1 1 1 −3
[






(8)
]
where :
ia 
i 
[ Iabde ] = ib 
d
 
ie 
– is the current vector for each phase for stage I
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
A Five-Phase Brushless DC-Machine Direct Drive System
Fig. 4: System modeling approach
ea 
e 
[ Eabde ] = eb 
d
 
e
 e
– is the back-EMF vector for each phase for
stage I
a)
[ Mabde ] =
 (2 M1 + M2 )

( −2 M1 + M2 )
 (2 M1 − 3M2 )

( −2 M1 + M2 )
(−3M1 + 2 M2 )
( M1 + 2 M2 )
( M1 − 2 M2 )
( M1 − 2 M2 )
( M1 − 2 M2 ) (−2 M1 + M2 )
( M1 − 2 M2 ) (2 M1 − 3M2 ) 
( M1 + 2 M2 ) (−2 M1 + M2 )
(−3M1 + 2 M2 ) (2 M1 + M2 ) 
– is the inductance matrix for state I.
0
0
0 
−4 R
 0
−4 R
0
0 

[ R] = 
 0
0
−4 R
0 


0
0
−4 R
 0
b)
– is the resistance matrix
1
0
[I] = 
0

0
0
0

0

1
– is the identity matrix
0
1
0
0
0
0
1
0
c)
[Vs Vq]
– is the voltage matrix for transistors and diodes
The torque equation is



 n 
 K v ω (t ) Γ ω  t − 2 π +


5
1


Te =
 in (t ) (9)

ω (t ) n = 0 
n

Ka in (t ) Λc  t − 2 π 
 5 


4
∑
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
Fig. 5: Simulation study of displacement effect of position
encoder in electrical degrees for fixed PWM duty-cycle;
(a) 3 electrical degrees displacement, (b) 12 electrical degrees
displacement, (c) 18 electrical degrees displacement.
19
M. Godoy Simões, P. Vieira Jr.
where :
Te
ω
n
Kv
Γω
a)
Ka
In
Λc
– electromagnetic torque (Nm)
– electrical speed (rad/s)
– phase contributing with torque
– velocity constant
– contribution of flux from armature current for the
armature reaction
– torque constant
– current at phase n
– contribution of flux from permanent magnet for the
armature reaction
The equations (1)–(9) were implemented in Matlab/Simulink; the
differential equations are algebraically commanded by the rotor
angle so as to reinitialize the initial conditions and redefine variables and system matrices in accordance to the absolute angle
position of the rotor. The model needs the machine electrical
parameters and the back-EMF experimental characterization as
presented before.
b)
c)
Fig. 4 shows the complete system model where initially an open
loop operation was evaluated in order to observe steady state
response and allow the understanding of the full operating range
of such drive. The effect of rotor position displacement was studied
in order to validate the inclusion of armature reaction. Figs. 5 (a)
and (b) depict the simulated waveforms for two different conditions of displacement showing that for 3 degrees of displacement the current can barely be established with a very sluggish
response, while for 27 degrees the current will establish much
faster at the expense of strong distortion. An open loop validation
of the drive system was performed in order to substantiate the
study of the variables’ range for a closed loop control. Figs. 6 (a),
(b), (c) and (d) show phase current, phase voltage, torque and
speed for open loop operation, imposing a variation in the dutycycle for all the five phases in the PWM modulator. There is a
torque and speed oscillation due to duty-cycle change. The torque
response is not optimized since it is an open loop operation, but
overall the system is well behaved with this open loop command.
A closed loop system was designed in accordance with the block
diagram of Fig. 4, and the proportional-integral controller was
fine-tuned by imposing several transient conditions. Fig. 7 shows the
system response for a torque reference step command from 5 Nm
to 20 Nm. The system response is so fast that only two electrical
cycles are required for establishing the operation, as indicated by
the current and voltage waveforms. The simulation does not
consider any current limitations but in the real application such
high current peak will be avoided through protection circuitry. The
output torque has an overshoot as indicated in Fig. 7 (c) but shaft
and load friction impose a damped speed response, as indicated by
Fig. 7 (d).
Control system implementation
d)
Fig. 6: Open loop behavior. The transient is because of the
prescribed PWM duty-cycle variation, (a) Terminal current,
(b) Terminal voltage, (c) Machine torque, (d) Machine speed.
20
A custom made Motorola 56824-based DSP board was designed
and implemented for this project. The board is represented in Fig. 8.
It is connected to a personal computer with the well-documented
ISA bus. The description of such a board is outside the scope of
this paper and is available in [12]. CodeWarrior user-friendly tools
were used for software development and debugging [13] [14]. A
dual-access 64K memory on the DSP board, addressed as locations 0 × 030000 through 0 × 3FFFF, provide the background
communication. Although the PC can read and write other memory areas on the board, it incurs more overhead because the DSP
chip must be “held” during PC accesses. The variables or arrays
that are being passed must be defined as global variables. Those
locations can hold pointers to variables or arrays of pointers;
therefore, they are used to communicate between the Processor
Boards and the PC.
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
A Five-Phase Brushless DC-Machine Direct Drive System
a)
c)
b)
d)
Fig. 7: Closed loop behavior. An outer torque loop commands the machine operation (a) Terminal current, (b) Terminal voltage,
(c) Machine torque, (d) Machine speed.
Fig. 8: DSP-PC system
The control software that runs in the DSP-PC Board is a multitasking, real-time program as indicated by Fig. 9. Therefore, it is
necessary to trigger the interruptions by software in such a way
that lower level interruptions are able to be interrupted by higher
level ones with automatic context save/restoring of variables. The
most critical time assignment task is the operation of the PWM
timers and the ADC converters. The eight channel ADC conversion is triggered at the end of Task 1 and the complete conversion
of the four channels takes 32 ms, which is inside the 100 ms setup
for the PWM. Two timers activate the PWM, one decrementing
from an initial value and another one setting up it again to recount. For the Motorola DSP56824 with the duty-cycle resolution
ranges from 100ns to 13.1msec.
A semaphore handler coordinates Task 1 for acquisition and PWM
setup plus Task 2 running at 1 msec for torque control loop which
contains a first order IIR filter for torque estimation plus a discrete
PI controller. The timing is roomy enough for background com-
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
Fig. 9: Tasks coordination chart for the DSP control code
munications within the control loop structure indicated in Fig. 10.
The control loop structure has four main routines: (1) input/output
of analog data, (2) input/output of digital data (PWM five phase
output included), (3) signal filtering, and (4) torque control.
The algorithm procedure acquires and filters the five-phase currents, estimates the average value proportional to the instantaneous torque, estimates torque by using the experimental torque
parameter, computes error with the reference, processes a proportional-integral difference equation, and programs the PWM initialization with the PI result bounded to internal scaling.
Before closing the loop, an experimental evaluation, depicted in
Fig. 11, of the relationship of duty-cycle with respect to the average current was performed so as to have a clear indication of the
wide machine range. Fig. 11 is a family of curves taken from
experimental evaluation of the open loop response of the armature
current in respect to the PWM duty-cycle imposed to the five
21
M. Godoy Simões, P. Vieira Jr.
a)
b)
c)
Fig. 10: Torque control implementation structure
d)
Fig. 11: Open loop experimental evaluation of duty-cycle relation with
current
phase converter for several operating speeds. Such family of
curves helped to understand the expected range of operation of the
drive system in closed loop control.
A mechanical optimization of the absolute encoder position was
carefully conducted in the laboratory in order to optimize alignment. Figs. 12 (a), (b), (c) and (d) show the response for a
mechanical shift of the absolute encoder set at 5 degrees, 10
degrees, 15 degrees and 20 degrees. A heuristic procedure was
used in this mechanical adjustment by looking at the best current
rise and fall time response, bounded current spikes, and personal
evaluation of vibrating noise. The experimental displacement was
also used to validate the simulation studies where the misalignment was simulated with lookup tables, as previously described.
A perfect comparison of the simulated strategy (Fig. 5) with the
22
Fig. 12: Experimental evaluation of current
waveform due to the optical disc mechanical
displacement;
(a) 5 electrical degrees displacement,
(b) 10 electrical degrees displacement,
(c) 15 electrical degrees displacement,
(d) 20 electrical degrees displacement
real implementation (Fig. 12) was not possible because that would
require much more elaborated instrumentation that is only available in Fine Mechanics laboratories. However, the overall simulated behavior was very close to the observed electrical response,
validating our simulation strategy of the optical encoder mechanical adjustment.
A torque controller was implemented in the DSP system in accordance with Fig. 10; the PI parameters were fine tuned to Kp = 1
and Ki = 0.01. Fig. 13 (a) shows a step torque response from
5 Nm to 15 Nm, where the real mechanical torque is presented
with the reference value, showing a very fast response. Figs. 13
(b) and (c) show the steady-state current and voltage waveforms
before and after the step torque command. The five-phase
machine was connected to a dc-machine with shunt resistances to
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
A Five-Phase Brushless DC-Machine Direct Drive System
a)
b)
c)
Fig. 13: Torque control (a) step response, (b) terminal voltage
and current for low torque level, (c) terminal voltage and
current for high torque level
absorb incoming power and the speed was kept within the range
650 rpm to 750 rpm. The steady-state voltage and current waveform indicated in Figs. 13 (b) and (c) confirmed the expected
duty-cycle and average current evaluation in open-loop as indicated
previously by Fig. 11.
[2] P. Delarue, A. Bouscayrol and B. Francois: Control implementation of a five-leg voltage-source-inverter supplying two threephase induction machines, IEEE International Electric Machines
and Drives Conference, Villeneuve d'Ascq, June 1-4, 2003, France,
vol. 3, pp. 1909–1915.
[3] B. Arnet and M. Jufer: Torque control on electric vehicles with separate wheel drives, EPE '97, Trondheim, Norway, September 1997,
pp. 659–664.
[4] B. Hredzak and P.S.M. Chin: Design of a novel multi-drive system
with reduced torque pulsations for an electric vehicle, Power
Engineering Society Winter Meeting, Jan. 23–27, 2000, Singapore,
vol. 1, pp. 208–212.
[5] S. Gataric: A polyphase cartesian vector approach to control of
polyphase ac machines, Conf. Rec. of IEEE IAS Annual Meeting,
Oct. 8–12, 2000, Rome, Italy, vol. 3, pp. 1648–1654.
[6] M. Godoy Simões and Petronio Vieira Jr.: A high torque low-speed
multiphase brushless machine – A Perspective application for electric vehicles, IEEE Trans. on Industrial Electronics, October 2002,
vol. 49, no. 5 pp. 1154–1164.
[7] M. Godoy Simões and P. Vieira Jr.: Model development and design
of a wheel-motor drive system, Proceedings of EPE – PEMC,
September 5–7, 2000, Kosice, Slovak Republic, pp. 74–79.
[8] P. Pillay and R. Krishnan: Modeling of permanent magnet motor
drives, IEEE Trans. on Industrial Electronics, November/
December 1988, vol. 35, no. 4, pp. 537–541.
[9] P. D. Evans and D. Brown: Simulation of brushless dc drives, IEE
Proceedings – B, vol. 137, issue 6, November 1990, pp. 299–308.
[10] T. Kenjo and S. Nagamori. Permanent-Magnet and Brushless DC
Motors, Oxford Science Publications, 1985.
[11] T. S. Low, K. J. Tseng, T. H. Lee, K. W. Lim, K. S. Lock: Strategy
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[13] Motorola, DSP 56L811 User’s Manual, Motorola Inc., 1996
[14] Motorola, Motorola DSP 56800 Family Manual, Motorola Inc.,
1996
The Authors
Conclusion
The paper showed that the design and evaluation of a polyphase
brushless dc-machine direct drive system suitable for high performance and reliability is critical for applications such as EV, HEV,
aerospace and the requirements are not cost oppressive. This work
presented the design, analysis and issues regarding the high-level
modeling comprised of a transient model in conjunction with their
corresponding experimental evaluation. Analysis was made to put
together the modeling efforts with the expected behavior so as to
have realistic simulation results verified by the experimental
setup. The dynamic modeling permitted the control strategy
design and validation with a DSP-based torque loop control and
comprehensive experimental results validated the work.
References
[1] A. Bouscayrol, B. Davat, B. B. De Fornel, B. Francois, J.P.
Hautier, J.P. F. Meibody-Tabar, M. Pietrzak-David: Multi-machine
multi-converter system for drives: analysis of coupling by a global
modeling, Conf. Rec. of IEEE IAS Annual Meeting, Oct. 8–12,
2000, Rome, Italy, vol. 3, pp. 1474–1481.
EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004
M. Godoy Simões, Ph.D., earned the B.Sc. and
his M.Sc. degrees from the University of São
Paulo, Brazil, and his Ph.D. degree from the
University of Tennessee, in 1985, 1990 and 1995,
respectively. In 1998, he also received a D.Sc.
degree (Livre-Docência) from the University of
São Paulo. Dr. Simões joined the faculty of the
University of São Paulo from 1989 to 2000 and
Colorado School of Mines in April 2000. He has
been working to establish research and education
activities in the development of intelligent control for high-power electronics applications in
renewable and distributed energy systems. Dr. Simões is currently serving as IEEE Power Electronics Society Intersociety chairman. He is
associate editor of Energy Conversion as well as editor of Intelligent
Systems of IEEE Transactions on Aerospace and Electronic Systems.
He is also associate editor of Power Electronics in Drives of IEEE
Transactions on Power Electronics . He has been actively involved in the
Steering and Organization Committee of the IEEE 2005 International
Future Energy Challenge. Dr. Simões is IEEE Senior-Member, EPE,
IEE and Cigré Member. He was the recipient of a National Science
Foundation (NSF) Faculty Early Career Development (CAREER) in
2002. It is the NSF’s most prestigious award for new faculty members,
recognizing activities of teacher-scholars who are considered most
likely to become the academic leaders of the 21st century.
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M. Godoy Simões, P. Vieira Jr.
Petronio Vieira Jr. received his B.Sc degree
(1985) in Electrical Engineering from Federal
University of Pará (UFPA), his M.Sc. degree
(1996) in Power Electronics from Federal
University of Santa Catarina (UFSC) and his
Ph.D. degree (2000) in Mechatronics from
University of São Paulo. He was a visiting
research fellow at Colorado School of Mines,
USA, in 2000. Dr. Vieira is member of Brazilian
Power Electronics Society (SOBRAEP), the
Brazilian Society of Maintenance (ABRAMAN)
and member of the Institute of Electrical and
Electronics Engineers. (IEEE). He worked for Centrais Elétricas do
Norte do Brasil S.A from 1985 to 1987 in the maintenance of synchronous 300MW generators. He has been a faculty member at the
Computer and Electrical Engineering Department (DEEC) of Federal
University of Pará since 1987. Dr. Vieira has been involved in teaching
undergraduate courses in energy conversion and power electronics, and
graduate courses in electronic drives where he coordinates and
supervises the research of efficiency issues of electrical drives. He is the
head of the Power Electronics Laboratory of DEEC. His main research
interests are control, instrumentation and electronic drives for industrial
applications and systems.
European Power Electronics and Drives Journal
Guidelines for the preparation of a paper to be published in EPE journal
Submitting a paper
The papers will be submitted by e-mail (Word or .pdf), one paper
copy will be sent as well, including full coordinates. E-mail
address: bsneyers@vub.ac.be. Mail address: EPE Secretariat, c/o
VUB-TW, Pleinlaan 2, B-1050 Brussels, Belgium
The text will be typewritten without page set-up (no columns, no
styles).
Paragraphs are separated by a line space.
Titels are in bold with level indication: size or number. They finish without a period(.).
Figure captions will be included in the text and will be of the kind:
Fig. 1: Typical figure caption
Writing the paper.
The paper will be written in English.
The originals of the figures will be given: ink drawings, photographs, lazer print out...
Figures can be included in the general file.
The title will be chosen short and expressive. It will be followed
by the authors' names, title, function and address and by keywords.
Figures and photographs will be sent back after use, upon request.
Deadlines
The paper will start with a summary that will allow non-specialists to understand the problem, have an idea of the state of the art,
and understand the originality of the described solutions. The
summary will be half a typewritten page long and refer to a figure.
The figure will be particularly clear and eloquent.
EPE Journal is published quarterly. Submitted papers undergo a
reviewing process of about 6 months. Publication dates are as follows:
Vol. 13 no 1: out 15/02/2003, special issue on Matrix Converters
The end of the paper will include a list of references and the
authors' curriculi and photographs.
Vol. 13 no 2: out 15/05/2003, normal issue with peer reviewed
papers
When the paper is accepted
Vol. 13 no 3: out 15/08/2003, special issue on Sensorless Drives
Texts are sent on floppy disks with indication of the format and a
paper copy.
Vol. 13 no 4: out 30/11/2002, normal issue with peer reviewed
papers
EPE 2005, 11 – 14 September in Dresden, Germany
http://www.epe2005.com
http://www.epe-association.org
Call for papers on p. 5 of this issue.
Don’t miss the event!
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EPE Journal ⋅ Vol. 14 ⋅ no 3 ⋅ August 2004