Experimental Validation of High-Voltage-Ratio Low-Input-Current-Ripple Converters for Hybrid Fuel Cell Supercapacitor Systems

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IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 8, OCTOBER 2012
Experimental Validation of High-Voltage-Ratio
Low-Input-Current-Ripple Converters for Hybrid
Fuel Cell Supercapacitor Systems
Mohammad Kabalo, Student Member, IEEE, Damien Paire, Benjamin Blunier, David Bouquain,
Marcelo Godoy Simões, and Abdellatif Miraoui, Senior Member, IEEE
Abstract—Electric vehicle technology has been adopting fuel
cells (FCs) for hybrid applications over the past few years. Therefore, the development of advanced power electronic systems for
the integration of fuel cells with on-board energy management
is fundamental for achieving high-performance systems. An FC
for vehicular applications is usually a low-voltage current-source
like device that produces electricity and heat directly from input
hydrogen and oxygen. Most often, it is required that the FCs be
stacked for high-voltage dc-link in order to supply the input power
for the drivetrain and electric motor drive system. The FC has a
nonlinear nature, and it must be controlled to operate in the highefficiency operating range. Hybrid electric vehicles have physical
constraints such as volume and weight under limited cost and expected lifetime. There is a need for high-voltage input/output ratio
of dc-dc boost converters to be connected between the FC to the
motor drive dc-link. In addition, it is necessary to have low input
ripple at the dc-dc boost converter in order to maximize the FC
lifetime, and the traditional dc-dc boost converter topologies have
poor performance on these specifications. This paper proposes a
new dc-dc converter family of topologies aimed at improving the
application to electric vehicle power control. This family is defined
as floating-interleaving boost converters (FIBCs). The paper will
thoroughly show analysis and experimental verification of FIBC’s,
and they will be compared with conventional boost converter characteristics. The paper supports how performance figures related to
the passive components, i.e., the inductor and capacitor, will have
better volume and weight, extremely low input current ripple,
and improved efficiency and transfer ratio. The analysis presented
in this paper shows how to choose the most suitable topology in
order to achieve the desired specifications. The selected topology
is fully validated experimentally using advanced nonlinear sliding
mode control, which has the additional feature of operating even
in faulty conditions.
Index Terms—Boost converter, dc-dc converter, fuel cell (FC)
hybrid vehicle, sliding mode control.
Manuscript received December 8, 2011; revised March 28, 2012; accepted
June 20, 2012. Date of publication July 11, 2012; date of current version
October 12, 2012. The review of this paper was coordinated by Dr. Z. Nie.
M. Kabalo, D. Paire, and D. Bouquain are with the Université de Technologie de Belfort-Montbéliard, 90010 Belfort, France (e-mail: mohammad.
[email protected]; [email protected]; [email protected]).
B. Blunier, deceased, was with the Université de Technologie de BelfortMontbéliard, 90010 Belfort, France.
M. G. Simões is with Colorado School of Mines, Golden, CO 804010-1887
USA (e-mail: [email protected]).
A. Miraoui is with Cadi Ayyad University, Marrakech 511-40000, Morocco
(e-mail: [email protected]).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TVT.2012.2208132
I. I NTRODUCTION
A
FUEL cell (FC) may be one of the promising solutions to
decrease carbon dioxide emissions under the assumption
that the hydrogen can be produced from renewable-energy
sources such as photovoltaic and wind energy or as a subproduct of currently wasted energy of large power plants under a
low-load situation (e.g., when base power plants are producing
higher power than the demand). In automotive applications,
proton exchange membrane FCs appear to be the most suitable,
because their working conditions at low temperature allow the
system to start up faster than those technologies using hightemperature FCs; moreover, the solid state of their electrolyte
(no leakage and low corrosion) and their high power density
make them fit for transport applications. Finally, they also
provide very good tank-to-wheel efficiency, compared with internal combustion engines [1], [2]. FCs are low-voltage current
intensive sources. A single cell produces a voltage of approximately 1 V; therefore, several cells must be stacked to achieve
high voltage output. FC stacking reduces its reliability and
lifetime as a chain of series-connected cells is as strong as the
weakest cell. Due to reliability and lifetime reasons, practical
FC stack output voltage is reduced to approximately 100 V. On
the other hand, the vehicle powertrain dc bus has a high voltage
of a few hundred volts. Therefore, a dc-dc converter is required
to interface the FC stack with the powertrain dc-bus voltage and
to achieve good power management of the input power source
[3]–[8]. The FC dc-dc converter is also required for voltage
conditioning as the FC output voltage strongly varies with the
load. Ideally, the power conditioner must have minimal losses,
leading to higher efficiency [9]. Power-conditioning efficiency
values can typically be higher than 90% [10].
The most important requirements expected from dc-dc converter for FC applications are high voltage ratio and low current
ripple [11]. The lower the current ripple, the longer the FC
lifetime [12], [13]. However, in an FC electric vehicle (FCEV),
the high voltage ratio and low current ripple, which are associated with volume, weight, reliability, and efficiency constraints,
are very important requirements. A cascade dc-dc converter
composing of two phase-interleaved boost converters and three
level series boost converters is proposed in [14]. This solution
suffers from low efficiency and reliability problem. In [15] and
[16], a parallel resonnant converter resonant topology with a
capacitor as output filter is proposed. However, in this topology,
determination of the leakage inductance and capacitance, as
0018-9545/$31.00 © 2012 IEEE
KABALO et al.: VALIDATION OF RIPPLE CONVERTER FOR HYBRID FC SUPERCAPACITOR SYSTEM
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well as the topology modeling, is very complex. A multiphase
interleaved boost converter for FC applications is proposed
in [17]. The proposed topology has a voltage ratio identical
to that of the classic boost converter, which leads to low
efficiency for applications where high voltage ratio is required.
A review of isolated and nonisolated boost dc-dc converters
suitable for FC and photovoltaic grid-connected applications is
proposed in [18] and [19]. The limitations of the conventional
boost converters in these applications are analyzed. Furthermore, the advantages and disadvantages of these converters are
discussed.
In this paper, new dc-dc converter topologies defined as
the floating-interleaving boost converter (FIBC) family will be
presented. These topologies are compared to conventional boost
converters, and the analysis will support the choice for the best
topology corresponding to specified constraints. The selected
topology is then experimentally validated using an advanced
nonlinear sliding-mode controller, which, for technical reasons,
will be explained later.
II. S YSTEM S PECIFICATIONS AND
P ROPOSED T OPOLOGIES
In high-power FCEV applications, the major drawbacks of
using conventional boost converters are the difficulty in the
design of magnetic components and high input current ripple,
which may lead to reduce the FC stack lifetime. Reducing
the rating current and voltage applied to passive and power
electronic components (keeping the same system rated power)
is a proposed solution. This makes the magnetic component
construction easier, giving further flexibility for the selection
of power electronic components used in the converters. Fig. 1
shows the proposed topologies in addition to the conventional
boost converter.
These topologies have a floating output and interleaving
input, which permits reduction in not only current stress but
also voltage stress, unlike conventional interleaved topologies.
The benefits of the N -phase FIBC are the following:
1) increasing the overall converter efficiency;
2) increasing the input and output ripple frequency without
increasing the switching frequency;
3) decreasing the input ripple current;
4) enhancing the system reliability by paralleling phases and
not by paralleling multiple devices;
5) decreasing current and voltage ratings of power electronic
devices;
6) reducing the size and weight of the passive components.
The system specifications are presented in Table I.
Table II shows that the current and voltage ratings of the
power electronic devices of FIBCs are smaller than those of
the boost and interleaving boost converters.
The duty cycle of the proposed topologies is expressed as
follows:
D=
VBus − VFC
.
VBus + VFC
(1)
Fig. 1. Proposed topologies. (a) Conventional boost. (b) Two-phase FIBC.
(c) Four-phase FIBC. (d) Six-phase FIBC.
On the other hand, the conventional boost converter duty
cycle is given by
D=
VBus − VFC
.
VBus
(2)
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IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 8, OCTOBER 2012
TABLE I
S YSTEM S PECIFICATIONS
For conventional boost converter
ΔiFC = ΔiL =
DVFC
.
Lfs
(4)
The input current slope of the N -phase FIBC is expressed as
follows:
TABLE II
C URRENT AND VOLTAGE R ATINGS OF P OWER E LECTRONIC D EVICES
n=N
diL
diFC
diLoad
n
=
−
.
dt
dt
dt
n=1
(5)
The ratio of the input current ripple to the inductor current
ripple of a two-phase FIBC as a function of duty cycle M2 (D)
is given by
1−2D
1−D , 0 < D < 0.5
(6)
M2 (D) = 2D−1
D , 0.5 < D < 1.
The ratio of the input current ripple to the inductor current ripple of a four-phase FIBC as a function of duty cycle
M4 (D) is
⎧ 1−4D
0 < D < 0.25
⎪
1−D ,
⎪
⎪
⎨ 3D−4D2 −0.5 , 0.25 < D < 0.5
D(1−D)
M4 (D) = 5D−4D
(7)
2
−1.5
⎪
⎪
⎪ D(1−D) , 0.5 < D < 0.75
⎩
4D−1
0.75 < D < 1.
D ,
Fig. 2. Duty cycle comparison.
Fig. 2 shows that, for the same FC rated power, the basic
boost duty cycle is higher than the proposed topology duty
cycle. The higher the duty cycle, the lower the converter
efficiency [20].
III. I NPUT C URRENT R IPPLE E VALUATION
The mathematical expressions for input current ripple are
derived under six assumptions.
1)
2)
3)
4)
5)
6)
The resistances of inductor and capacitor are negligible.
Stray inductor and capacitor are negligible.
Switches are ideal.
Passive components are identical.
Switches in parallel operate (360/N )◦ out of phase.
The converters operate in continuous conduction mode
(CCM).
The ratio of the input current ripple to the inductor current
ripple is given by
M (D) =
ΔiFC
.
ΔiL
(3)
The ratio of the input current ripple to the inductor current
ripple of a six-phase FIBC as a function of duty cycle M6 (D)
is given by
⎧ 1−6D
0 < D < 1/6
⎪
1−D ,
⎪
⎪
⎪ 3D−6D2 −1/3
⎪
, 1/6 < D < 1/3
⎪
⎪
D(1−D)
⎪
⎪
⎨ 9D−6D2 1 ,
1/3 < D < 1/2
D(1−D)
M6 (D) = 7D−6D
(8)
2
2
⎪ D(1−D) ,
1/2 < D < 2/3
⎪
⎪
⎪
⎪
9D−6D 2 −10/3
⎪
⎪
, 2/3 < D < 5/6
⎪
D(1−D)
⎪
⎩ 6D−5
5/6 < D < 1.
D ,
The previous analysis permits having the generalized expression of the ratio of the input current ripple to the inductor
current ripple of the N -phase FIBC as a function of duty cycle
MN (D)
(X − N D) D − X−1
ΔiFC
N
MN (D) =
(9)
=
ΔiL
D(1 − D)
where X is the interval between two duty cycle values, resulting
in zero current ripple. The variation of the ratio of input current
ripple to inductor current ripple as a function of duty cycle is
shown in Fig. 3. On the one hand, by studying Fig. 3, it can be
observed that input current ripple cancelation occurs at specific
duty cycles, which are multiple duties of 1/N , such as 0.5 in
a two-phase FIBC; 0.25, 0.5, and 0.75 in a four-phase FIBC;
and 0.16, 0.33, 0.5, 0.66, and 0.83 in a six-phase FIBC. On
the other hand, it is clear that the input current ripple is always
less than the inductor current ripple. The fact that the input
current ripple is always less than the inductor current ripple
permits to increase this latter and, consequently, decrease the
inductor value according to (4). However, due to core losses
KABALO et al.: VALIDATION OF RIPPLE CONVERTER FOR HYBRID FC SUPERCAPACITOR SYSTEM
Fig. 3. Ratio between the input current ripple and the inductor current ripple
versus duty cycle.
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Fig. 4. Input current ripple as a percentage of the FC rated current according
to the number of phases.
and the CCM condition, the critical inductor current ripple for
the proposed N -phase FIBC is defined as follows:
ΔiLcritical ≤
4IFC
.
N (1 + D)
(10)
Decreasing both the inductor value and the current flowing
through it permits to reduce its volume, weight, and cost, as
will be shown in the next section.
The best way to show how the input current ripple (as a
percentage of input rated current) decreases with the number
of phases is to determine this first according to the system
specifications presented in Table I for a given inductor value.
For this evaluation, the inductor value will be chosen as 100 μH.
On the one hand, Fig. 4 shows that decreasing the input current
ripple from a four-phase to a six-phase FIBC is not important.
It is not the case from a one-phase or a two-phase to a fourphase FIBC. On the other hand, depending on the rated power
of the system, a six-phase FIBC can have input current ripple
bigger than a four-phase FIBC, which is not the case when one
compares a four-phase with a two-phase FIBC for any rated
power. Consequently, from the input current ripple reduction
point of view, we can see that the benefits of six-phase converter
are not so attractive compared with its increased complexity and
costs.
IV. E VALUATION OF THE I NDUCTOR VOLUME
To evaluate the reduction in the inductor volume for the
proposed topologies compared with that of the conventional
boost, it is necessary to go through the details of the electromagnetic used for similar magnetic cores. In this analysis, the
material of the selected core is ferrite. Fig. 5 shows the top and
frontal views of the inductor magnetic circuit. Some guidelines
of magnetic material comparison and selection for high-power
Fig. 5. Geometry of the inductor core.
high-frequency inductors in dc-dc converter can be found in
[21] and [22].
The stored energy in the inductor is proportional to the
inductor and peak current value, i.e.,
E=
1 2
LI
.
2 peak
(11)
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IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 8, OCTOBER 2012
On the other hand, the stored energy as a function of the
inductor’s dimensions for the proposed geometry is given by
1
B 2 g b2
→ −
−
→
E=
H · B dV =
(12)
2
4μo
→
−
→
−
where B and H are the magnetic field density and strength,
respectively, and μo is the magnetic permeability of air. The
volume of the geometry shown in Fig. 5 is given by
g
V = (2b + 2kb b)(b + 2kb b) b + h +
.
(13)
2
This volume can be expressed as follows [23]:
0.75
4kB L IL2
V =K
JΔB
2
kB
K=
(1 + kb ) 1 +
(1 + 2kb ).
0.75
kB
kb
(14)
ΔB = 0.2 T is the change in flux density J, which is the current
density varying between 2 and 5 A/mm2 . In our analysis, it is
chosen as 3.5 A/mm2 . kB is a coefficient greater than 1, which
takes into account the difference between the effective section
of the conductors and the required section windings. It is related
to the shape of the conductors and the presence of different
levels of isolation. In this analysis, it is chosen as (kB = 1.5).
kb is a geometric coefficient, and in our case, it is taken to be
equal to 1.
Equation (15), shown below, gives the reduction in total inductor stored energy and associated magnetic volume, as compared with that in the conventional boost converter. EInductor(1)
and VInductor(1) are the stored energy and the associated volume in the traditional boost, respectively. EInductor(N ) and
VInductor(N ) are the stored energy in the single-phase inductor
and the corresponding volume of the N -phase FIBC, respectively. E(N ) and V (N ) are the percentage of inductor stored
energy and volume reduction, respectively, given by
EInductor(1) − N × EInductor(N )
E(N )
=
100
EInductor(1)
VInductor(1) − N × VInductor(N )
V (N )
=
. (15)
100
VInductor(1)
From (15), the total stored energy and volume going from
conventional boost to a two-phase FIBC have been reduced by
62.6% and 49.3%, respectively. For a four-phase FIBC, they
are reduced by 86% and 83.7% and by 87.2% and 87.35% for
a six-phase FIBC. In conclusion, from this point of view, it can
be seen that there is no great benefit to use a six-phase FIBC,
compared with a four-phase FIBC.
V. ROOT M EAN S QUARE C APACITOR C URRENT
E VALUATION
The output filter capacitor ensures dc-bus stabilization and
also filters discontinuous diode current behavior. In the one
hand, reducing the root mean square (RMS) value of the capacitor current leads to avoiding utilization of a bulky capacitor;
on the other hand, it permits the reduction of capacitor heating
by decreasing resistive losses because of the equivalent series
internal resistance (ESR). The lower the capacitor temperature
and RMS current, the longer the capacitor lifetime [24].
The boost RMS capacitor current as a function of the duty
cycle and the FC rated current is given by
0 ≤ D ≤ 1.
(16)
IRMSC = IFC D(1 − D),
For a two-phase FIBC, the RMS capacitor currents as
a function of the duty cycle and the FC rated current are
given by
D(1 − D)
IRMSC1,C2 = IFC
,
0 ≤ D ≤ 1.
(17)
(1 + D)2
For a four-phase FIBC, the RMS capacitor currents as
a function of duty cycle and the FC rated current are
expressed as
⎧
D−2D 2
⎪
I
0 ≤ D ≤ 0.25
FC
⎪
2(1+D)2 ,
⎪
⎪
⎪
⎨
2 −0.5
IRMSC1,C2 = IFC 4D−4D
(18)
4(1+D)2 , 0.25 ≤ D ≤ 0.75
⎪
⎪
⎪
⎪
⎪
⎩I
3D−2D 2 −1
0.75 ≤ D ≤ 1.
FC
2(1+D)2 ,
Finally, a six-phase FIBC, the RMS capacitor currents
as a function of duty cycle, and the FC rated current are
expressed as
⎧
D(1−3D)
⎪
⎪
I
0 ≤ D ≤ 1/6
FC
⎪
3(1+D)2 ,
⎪
⎪
⎪
⎪
⎪
7D−9D 2 −2/3
⎪
⎪
I
1/6 ≤ D ≤ 1/3
⎪
FC
9(1+D)2 ,
⎪
⎪
⎪
⎨
2 −4/3
IRMSC1,C2 = IFC 9D−9D
1/3 ≤ D ≤ 2/3 (19)
9(1+D)2 ,
⎪
⎪
⎪
⎪
⎪
11D−9D 2 −8/3
⎪
⎪
, 2/3 ≤ D ≤ 5/6
⎪ IFC
9(1+D)2
⎪
⎪
⎪
⎪
⎪
⎪
⎩ IFC 5D−3D2 −2
5/6 ≤ D ≤ 1.
3(1+D)2 ,
The following gives the percentage of the RMS capacitor
current reduction:
IRMSC (1) − IRMSC (N )
I(N )
=
.
(20)
100
IRMSC (1)
From (20), it can be seen that the RMS capacitor current
from a traditional boost to a two-phase FIBC is reduced by
30%, by 56% from a two-phase to a four-phase FIBC, and by
60% from a four-phase to a six-phase FIBC. This reduction in
RMS current will reduce electrical stress in the output capacitor
and improve the converter’s reliability and lifetime. The RMS
capacitor current analysis shows that the four-phase FIBC is the
best choice among the proposed topologies.
VI. E FFICIENCY A NALYSIS
An N -phase FIBC is analyzed to evaluate losses in every
component and compare them with those of a conventional
KABALO et al.: VALIDATION OF RIPPLE CONVERTER FOR HYBRID FC SUPERCAPACITOR SYSTEM
boost topology. This allows determining the most suitable converter among these topologies from the efficiency point of view.
This analysis is reported according to the system specifications
presented in Table I.
given by
IS =
• Traditional boost converter losses:
VS =
1) Inductor losses: These losses include copper losses
(Pcop ), which are caused by the skin effect and
proximity effect, and core losses (Pcore ), which are
caused by the hysteresis phenomenon and eddy currents. These losses depend on the core type and the
wire type used, i.e.,
Δi2FC
2
+
Pcop = RL IFC
12
1.74
ΔB
m
(21)
Pcore = 6.5(fs )1.51
2
where RL is the equivalent series resistance of the
inductor, and m is the weight of the magnetic circuit
in kilograms.
2) Capacitor losses: They are due to ESR; in this analysis, they are neglected.
3) Switch losses: These losses include the conduction
(Psconl ) and the switching losses (Pswl )
Δi2FC
2
+
Psconl = D Rs IFC
+ Vs IFC
12
Pswl =
2
2
Vbus
tfs
IFC
2Itest Vtest
t = td(on) + tr + td(off) + tf
(22)
Pdconl = (1 − D) (Rd λ + Vd IFC )
Δi2FC
2
λ = IFC +
12
2
IFC IRM trr fs
Vbus
2Itest Vtest
(23)
where Vd and Rd are the voltage drop and the resistance in the ON-state of the diode, respectively.
IRM and trr are the reverse recovery current and
reverse recovery time, respectively. They can also be
obtained from the manufacturer’s data sheet.
• N -phase FIBC: The switch current and voltage during
the commutation of the proposed N -phase FIBC are
N
2
IFC
(1 + D)
Vbus
.
(1 + D)
(24)
(25)
By increasing the number of phases, the ratio
(IS V S/Itest Vtest ) becomes lower, and consequently, the
switching losses become lower.
1) Inductor losses: These losses are the sum of the
losses of each inductor of an N -phase FIBC, and
they are expressed as a function of phases number,
i.e.,
2
4IFC
N Δi2L
+
PNcop = RL
N (1 + D)2
12
1.74
ΔB
PNcore = N 6.5 (fs )1.51
m.
(26)
2
2) Switch losses: These losses represent the sum of the
losses of each switch of an N -phase FIBC. They are
given by
2IFC Vs
PNsconl = D Rs κ +
1+D
2
4IFC
N Δi2L
κ=
+
N (1 + D)2
12
PNswl =
where Vs are Rs are the voltage drop and the resistance in the ON-state of the switch, respectively;
and td(on) , tr , td(of f ) , and tf are the turn-on delay
time, turn-on rise time, turn-off delay time, and turnoff fall time, respectively. They can be obtained from
the manufacturer’s data sheet. Itest and Vtest are the
current and the voltage, respectively, under which the
switch was tested to determine its data sheet.
4) Diode losses: These losses include the conduction
(Pdconl ) and reverse recovery (Pdrrl ) losses
Pdrrl =
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2
2
Vbus
tfs
2IFC
.
4
N (1 + D) Itest Vtest
(27)
3) Diode losses: These losses represent the sum of the
losses of each diode of an N -phase FIBC. They are
given by
2IFC Vd
PNdconl = (1 − D) Rd κ +
1+D
PNdrrl =
2
IFC IRM trr fs
2Vbus
.
N (1 + D)4 Itest Vtest
(28)
Fig. 6 shows the proposed converter efficiency as a
function of the FC current.
By analyzing Fig. 6, one can see, on the one hand,
that a four-phase FIBC and a six-phase FIBC keep high
efficiency for a wide range of load power. However, a traditional boost and a two-phase FIBC efficiency drastically
decrease when increasing the system power. For the rated
power of the system in Table I, a four-phase and a sixphase FIBC have nearly the same efficiency. Therefore,
from the efficiency analysis point of view, we can see that
a four-phase FIBC is the best choice among the proposed
converters. It has to be noted that the previous calculations
are based on an insulated-gate bipolar transistor (IGBT)
switch (ref. 38NAB066V1) from Mitsubishi Electric company. For the diode, the antiparallel emitter-to-collector
free-wheel diode (FWDi) of the previous IGBT has been
used. The inductor series resistance numerical value has
been chosen to correspond to the one of the real inductor.
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IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 8, OCTOBER 2012
Fig. 9.
Convergence relation for control of four-phase FIBC.
A. Sliding-Mode Controller
Fig. 6. Efficiency versus input current.
Fig. 7. Implemented four-phase FIBC.
Since a sliding-mode controller is based on the large signal
model of dc-dc converters, its stability is not restricted by the
variations around the operating point, which contributes to an
overall improved controller performance. Therefore, the large
signal model of the four-phase FIBC is defined as follows:
dIL1
dt
dIL2
dt
dIL3
dt
dIL4
dt
dVC1
dt
dVC2
dt
1
((D1 + 1)VFC − 2rL1 IL1 + (D1 − 1)VBus )
2L1
1
=
((D2 + 1)VFC − 2rL2 IL2 + (D2 − 1)VBus )
2L2
1
=
((D3 + 1)VFC − 2rL3 IL3 + (D3 − 1)VBus )
2L3
1
=
((D4 + 1)VFC − 2rL4 IL4 + (D4 − 1)VBus )
2L4
1
=
((1 − D1 )IL1 + (1 − D2 )IL2 − ILoad )
C1
1
=
((1 − D3 )IL3 + (1 − D4 )IL4 − ILoad ) . (29)
C2
=
The sliding surfaces or the control laws are defined by the
following expression:
t
SILi = ILi − Ki + KILi
(ILi − Ki ) dτ
(30)
0
Fig. 8. FCEV architecture.
VII. E XPERIMENTAL VALIDATION OF F OUR -P HASE
F LOATING I NTERLEAVING B OOST C ONVERTER
The previous evaluations show that the four-phase FIBC
is top choice among all the proposed topologies. The implemented four-phase FIBC and the FCEV architecture are shown
in Figs. 7 and 8, respectively. The high-voltage battery in Fig. 8
can be replaced by a supercapacitor connected to the dc bus via
bidirectional four-phase FIBC.
To achieve instantaneous power sharing between the phases
and good power management of the input power source, each
converter phase has its own current controller. A nonlinear
sliding-mode controller is used to generate the required control
input for each switch of the four-phase FIBC [25]–[27].
where i = [1, . . . , 4], ILi is the average value of the inductor
current, Ki is the desired inductors current, and KILi is a
coefficient that defines the dynamic of convergence to zero
of the static error. The convergence dynamic of the sliding
surfaces to zero is defined as follows:
SI˙Li = −λILi SILi
(31)
where λILi are positive real numbers and are called the convergence factors. The convergence relation for control of fourphase FIBC is shown in Fig. 9.
According to (31), the larger the convergence factors, the
faster the system reaches its steady state. However, due to limits
on the system parameters such as duty cycle, it is not possible
to increase the convergence factors beyond a certain value.
KABALO et al.: VALIDATION OF RIPPLE CONVERTER FOR HYBRID FC SUPERCAPACITOR SYSTEM
3437
To design the controller, it is necessary to combine (31) with
(29) and (30). This will result in equations for control inputs in
terms of the state variables and the system parameters.
The duty cycle of each phase of the four-phase FIBC as a
function of time is shown by
Di (t) = 1 +
2 (rLi ILi − VFC + Li χ)
VFC + VBus
(32)
where χ = (K̇i − λILi SILi − KILi (ILi − Ki )).
Equation (32) shows that the control inputs are irrelevant
with the value of load resistance R. Therefore, this controller
will not be perturbed by the variations of the load. Because
each duty cycle is relevant with its own phase parameters, this
controller is able to work in degraded mode. This is a very
important feature as the reliability is a major criterion in FCEV.
At the steady state, where the state variables ILi are following the commanded references Ki , the duty cycle is given by
VBus − VFC
.
(33)
Di =
VFC + VBus
By replacing the control inputs in the large signal model of
four-phase FIBC, we get
t
y dτ = 0
(34)
ẏ + (KILi + λILi )y + KILi λILi
0
where y = ILi − Ki by deriving (34)
ÿ + (KILi + λILi )ẏ + KILi λILi y = 0.
(35)
This equation is irrelevant with the topology parameters,
which underlines the robustness of the controller. The coefficients in (35) are positive. This means that all roots of the
system have negative real parts, which ensure its stability.
The method for a second-order system can be used to determine
the coefficients KILi and the convergence factors λILi to
get the desired performance.
B. Experimental Setup
The four-phase FIBC test bench and the four-phase FIBC
converter are shown in Fig. 10. Each inductor current has zeroflux Hall effect current sensor for feedback control where the
inductor current references Ki are generated by a real-time
board dSPACE DS 1104. The control system developed in this
study has been downloaded using the Matlab-Simulink and
ControlDesk software. Experimental results have been obtained
by an emulated FC power source. The benefits of interleaving
the input current ripple make the control signals of the main
switches be shifted 90◦ from each other. In the experimental
evaluation implemented for this project, the control signals are
shifted by means of a Field-Programmable Gate Array control
card.
The specification of the implemented four-phase FIBC are
detailed in Table III.
C. Experimental Results
1) Sliding-Mode Controller Validation: The dynamic response of the sliding-mode controller for a step variation of
inductor current from 13 to 20 A shown in Fig. 11. It shows that
Fig. 10. Test bench and four-phase FIBC converter. (a) Test bench. (b) Fourphase FIBC converter.
TABLE III
F OUR -P HASE FIBC S PECIFICATION
the currents perfectly follow the reference signal with a settling
time of 0.8 ms and with no noticeable ringing or overshoot. This
indicates that the proposed controller has excellent dynamic
performance.
Fig. 12 shows the steady-state FC current, the converter
input current, and the inductor current for an inductor current
reference of 17.5 A. Similar to its dynamic performance, the
proposed sliding controller has very good steady-state response
with negligible steady-state error around 17.5 A.
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IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 8, OCTOBER 2012
Fig. 11. Sliding-mode controller dynamic response for a step variation of
inductor current from 13 to 20 A.
The value of the coefficients KILi and the convergence
factors λILi for this excellent performance of the proposed
sliding-mode controller are given by
λILi = KILi = 6000.
(36)
The steady-state waveforms underline the great benefit of the
four-phase FIBC from the current ripple point of view. One can
go from a 22-A current ripple in the inductors to 3 A at the
converter input. By using a second-order low-pass filter, the FC
current ripple is nearly zero, as shown in Fig. 12.
The steady-state switch driving signals with 70% steady-state
duty cycle and 90◦ out of phase from each other are shown
in Fig. 13.
Fig. 14 shows experimental and analytical four-phase FIBC
efficiency curves, which indicate that the implemented converter has a maximum efficiency of 95% at a power demand
of 2.5 kW and an efficiency of 94.7% at a power demand
of 5 kW (operating point). Such efficiency is very good for
intensive-current low-voltage power source and much higher
than an equivalent traditional dc-dc boost converter for the same
application.
VIII. C ONCLUSION
This paper has proposed a new family of converter topologies
for optimized hybrid integration of FCs and supercapacitors.
Three different solutions have been investigated, i.e., two-
Fig. 12. Sliding-mode controller steady-state response for inductor current
reference of 17.5 A.
phase, four-phase, and six-phase FIBCs have been compared
with the conventional dc-dc boost converter. Analysis has
shown that these proposed FIBC converters have all better
characteristics for input current ripple, inductor volume, and
capacitor stress better than classic boost converters. Studies
have been made to evaluate their operation for an increased
number of phases. As a result, it has been concluded that a
six-phase FIBC shows that is an efficiency figure comparable
with a four-phase FIBC for the same rated power and similar
circuit complexity. In addition, there are benefits in decreasing
inductor volume and capacitor stress, corroborating that all the
relevant design parameters are optimized for a four-phase FIBC
when compared with other FIBC structures. As a conclusion,
a four-phase FIBC has been selected, and nonlinear slidingmode control has been implemented for the feedback look,
KABALO et al.: VALIDATION OF RIPPLE CONVERTER FOR HYBRID FC SUPERCAPACITOR SYSTEM
3439
R EFERENCES
Fig. 13. Steady-state switch driving signals with 70% steady-state duty cycle
and 90◦ out of phase from each other.
Fig. 14. Implemented converter efficiency. The four-phase FIBC has an
efficiency of 94.7% at a power demand of 5 kW (operating point).
where experimental results showed excellent performance with
augmented characteristics such as improved input current ripple
reduction, a decrease in inductor volume, and higher efficiency
under a full-range transfer function voltage. The proposed solution demonstrates much potential and promise for utilization
in modern FCEV applications.
ACKNOWLEDGMENT
The authors dedicate this paper to the memory of their
friend and coauthor Dr. B. Blunier, Associate Professor with the
Université de Technologie de Belfort-Montbéliard, who passed
away on February 23, 2012.
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Mohammad Kabalo (S’09) received the B.S. degree
from the University of Tishreen, Lattakia, Syria, and
the M.S. degree in electrical engineering from the
Ecole Centrale de Lyon, Ecully, France, in 2005
and 2009, respectively. He is currently working toward the Ph.D. degree in electrical engineering with
the Research Institute on Transportation, Energy
and Society, Université de Technologie de BelfortMontbéliard, Belfort, France.
His current research interests include power electronics and dc-dc power converters for fuel cell
applications.
Damien Paire was born in France in 1979. He
received the M.S. degree in electrical engineering
from INSA of Lyon, Villeurbanne, France, in 2002,
the Agregation degree from the Ecole Normale Superieure de Cachan, Cachan, France, in 2003, and the
Ph.D. degree from System and Transport Laboratory,
Université de Technologie de Belfort-Montbéliard
(UTBM), Belfort, France, in 2010.
He was a Lecturer with UTBM from 2004 to 2011
and has been an Associate Professor since 2011.
His research interests include energy management,
power electronics, and hybrid systems.
Benjamin Blunier (deceased) received the M.Sc.
and Ph.D. degrees in electrical engineering from the
Université de Technologie de Belfort-Montbéliard
(UTBM), Belfort, France, in 2004 and 2007, respectively. He studied fuel cell system modeling and
particularly their air management and control for
hybrid and electric vehicles.
He was an Associate Professor with UTBM until
his death on February 23, 2012. His research interests included fuel cells systems; electric, hybrid,
and plug-in hybrid vehicles; and intelligent energy
management in smart grids and microgrids.
David Bouquain received the M.S. degree in electrical engineering from the Franche-Comté University,
Besancon, France, in 1999 and the Ph.D. degree in
electrical engineering from the Université de Technologie de Belfort-Montbéliard (UTBM), Belfort,
France, in 2008.
From 2000 to 2002, he has been an Engineer
with the Laboratory of Electrical Engineering and
Systems. He worked on a prototype of hybrid truck
for the French army. Since September 2002, he has
been Teacher and Researcher with the UTBM. He
is currently an Associate Professor with System and Transport Laboratory,
UTBM, working in the research field of energy management of electric and
hybrid vehicles and fuel cell systems.
Marcelo Godoy Simões received the B.S. and M.S.
degrees from the University of São Paulo, São
Carlos, Brazil, the Ph.D. degree from The University
of Tennessee, Knoxville, in 1985, 1990, and 1995,
respectively, and the D. Sc. degree (Liv re-Docência)
from the University of São Paulo, São Carlos,
Brazil, in 1998.
He is currently an Associate Professor with
Colorado School of Mines (CSM), Golden, where
he has been establishing research and education
activities for the development of intelligent control
for high-power-electronic applications in renewable and distributed energy
systems, where he currently serves as Director of the Center for the Advanced
Control of Energy and Power Systems. He has been involved in activities related
to control and management of smart grid applications since joining CSM.
Abdellatif Miraoui (SM’09) was born in Morocco
in 1962. He received the M.Sc. degree from Haute
Alsace University, Mulhouse, France, in 1988 and
the Ph.D. and Habilitation degrees from the University of Franche-Comté, Besancon, France, in 1992
and 1999, respectively.
He is currently the President of Cadi Ayyad
University, Marrakech, Morocco. Since 2000, he
has been a Full Professor of electrical engineering
(electrical machines and energy) with the Université de Technologie de Belfort-Montbéliard, Belfort,
France, where he was the Vice President of Research Affairs from 2008 to
2011, the Director of the Electrical Engineering Department from 2001 to 2009,
and the Head of the “Energy Conversion and Command” Research Team (38
researchers in 2007). He is a Doctor Honoris Causa of the Technical University
of Cluj-Napoca, Cluj-Napoca, Romania. He was an Editor of the International
Journal on Electrical Engineering Transportation. He is the author of more
than 80 journal and 150 international conference proceeding papers. He is also
the author of four textbooks about fuel cells. His special research interests
include fuel cell energy, energy management in transportation, and the design
and optimization of electrical propulsions/tractions.
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