1200 MHz to 2400 MHz Quadrature Modulator with
1550 MHz to 2150 MHz Frac-N PLL and Integrated VCO
ADRF6702
Data Sheet
The integrated fractional-N PLL/synthesizer generates a 2× fLO
input to the IQ modulator. The phase detector together with an
external loop filter is used to control the VCO output. The VCO
output is applied to a quadrature divider. To reduce spurious
components, a sigma-delta (Σ-Δ) modulator controls the
programmable PLL divider.
FEATURES
IQ modulator with integrated fractional-N PLL
Output frequency range: 1200 MHz to 2400 MHz
Internal LO frequency range: 1550 MHz to 2150 MHz
Output P1dB: 13.1 dBm @ 2140 MHz
Output IP3: 29.1 dBm @ 2140 MHz
Noise floor: −159.6 dBm/Hz @ 1960 MHz
Baseband bandwidth: 750 MHz (3 dB)
SPI serial interface for PLL programming
Integrated LDOs and LO buffer
Power supply: 5 V/240 mA
40-lead 6 mm × 6 mm LFCSP
The IQ modulator has wideband differential I and Q inputs,
which support baseband as well as complex IF architectures.
The single-ended modulator output is designed to drive a
50 Ω load impedance and can be disabled.
The ADRF6702 is fabricated using an advanced silicongermanium BiCMOS process. It is available in a 40-lead,
exposed-paddle, Pb-free, 6 mm × 6 mm LFCSP package.
Performance is specified from −40°C to +85°C. A lead-free
evaluation board is available.
APPLICATIONS
Cellular communications systems
GSM/EDGE, CDMA2000, W-CDMA, TD-SCDMA, LTE
Broadband wireless access systems
Satellite modems
Table 1.
Part No.
ADRF6701
GENERAL DESCRIPTION
The ADRF6702 provides a quadrature modulator and
synthesizer solution within a small 6 mm × 6 mm footprint
while requiring minimal external components.
Internal LO Range
750 MHz
1150 MHz
1550 MHz
2150 MHz
2100 MHz
2600 MHz
2500 MHz
290 MHz
ADRF6702
ADRF6703
The ADRF6702 is designed for RF outputs from 1200 MHz to
2400 MHz. The low phase noise VCO and high performance
quadrature modulator make the ADRF6702 suitable for next
generation communication systems requiring high signal
dynamic range and linearity. The integration of the IQ
modulator, PLL, and VCO provides for significant board
savings and reduces the BOM and design complexity.
ADRF6704
±3 dB RFOUT Balun Range
400 MHz
1250 MHz
1200 MHz
2400 MHz
1550 MHz
2650 MHz
2050 MHz
3000 MHz
FUNCTIONAL BLOCK DIAGRAM
VCC7
VCC6
VCC5
VCC4
VCC3
VCC2
VCC1
34
29
27
22
17
10
1
LOSEL 36
ADRF6702
LON 37
BUFFER
LOP 38
BUFFER
CLK 13
LE 14
FRACTION
REG
SPI
INTERFACE
MODULUS
2:1
MUX
INTEGER
REG
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
×2
REFIN 6
÷2
N COUNTER
21 TO 123
MUX
TEMP
SENSOR
÷4
MUXOUT 8
4
7
11 15 20 21 23 25 28 30 31 35
GND
÷2
0/90
CHARGE PUMP
250µA,
500µA (DEFAULT),
750µA,
1000µA
–
PHASE
+ FREQUENCY
DETECTOR
24
5
NC
RSET
NOTES
1. NC = NO CONNECT. DO NOT CONNECT TO THIS PIN.
DECL2
2
DECL1
18 QP
VCO
CORE
PRESCALER
÷2
9
19 QN
32 IN
33 IP
3
39
16
26
CP VTUNE ENOP RFOUT
08568-001
DATA 12
40 DECL3
DIVIDER
÷2
Figure 1.
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2011 Analog Devices, Inc. All rights reserved.
ADRF6702
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Device Programming and Register Sequencing..................... 19
Applications....................................................................................... 1
Register Summary .......................................................................... 20
General Description ......................................................................... 1
Register Description....................................................................... 21
Functional Block Diagram .............................................................. 1
Register 0—Integer Divide Control (Default: 0x0001C0) .... 21
Revision History ............................................................................... 2
Register 1—Modulus Divide Control (Default: 0x003001).. 22
Specifications..................................................................................... 3
Register 2—Fractional Divide Control (Default: 0x001802) 22
Timing Characteristics ................................................................ 6
Register 3—Σ-Δ Modulator Dither Control (Default:
0x10000B).................................................................................... 23
Absolute Maximum Ratings............................................................ 7
ESD Caution.................................................................................. 7
Pin Configuration and Function Descriptions............................. 8
Typical Performance Characteristics ........................................... 10
Theory of Operation ...................................................................... 16
Register 4—PLL Charge Pump, PFD, and Reference Path
Control (Default: 0x0AA7E4)................................................... 24
Register 5—LO Path and Modulator Control (Default:
0x0000D5) ................................................................................... 26
PLL + VCO.................................................................................. 16
Register 6—VCO Control and VCO Enable (Default:
0x1E2106).................................................................................... 27
Basic Connections for Operation............................................. 16
Register 7—External VCO Enable ........................................... 27
External LO ................................................................................. 16
Characterization Setups................................................................. 28
Loop Filter ................................................................................... 17
Evaluation Board ............................................................................ 30
DAC-to-IQ Modulator Interfacing .......................................... 18
Evaluation Board Control Software......................................... 30
Adding a Swing-Limiting Resistor ........................................... 18
Outline Dimensions ....................................................................... 35
IQ Filtering .................................................................................. 19
Ordering Guide .......................................................................... 35
Baseband Bandwidth ................................................................. 19
REVISION HISTORY
10/11—Rev. A to Rev. B
Changes Table 1 ................................................................................ 1
7/11—Rev. 0 to Rev. A
Changes to Noise Floor in Features Section and Table 1 ............ 1
Changes to Specifications Section.................................................. 2
Changes to Figure 16, Figure 17, and Figure 18 ......................... 12
Changes to Figure 28...................................................................... 14
Changes to Figure 32 and Figure 34............................................. 15
Added Figure 34, Renumbered Sequentially .............................. 15
Changes to Figure 35...................................................................... 17
Added Figure 40.............................................................................. 19
Changes to Figure 52...................................................................... 31
4/11—Revision 0: Initial Version
Rev. B | Page 2 of 36
Data Sheet
ADRF6702
SPECIFICATIONS
VS = 5 V; TA = 25°C; baseband I/Q amplitude = 1 V p-p differential sine waves in quadrature with a 500 mV dc bias; baseband I/Q
frequency (fBB) = 1 MHz; fPFD = 38.4 MHz; fREF = 153.6 MHz at +4 dBm Re:50 Ω (1 V p-p); 130 kHz loop filter, unless otherwise noted.
Table 2.
Parameter
Test Conditions/Comments
Min
OPERATING FREQUENCY RANGE
IQ modulator (±3 dB RF output range)
PLL LO range
RFOUT pin
Baseband VIQ = 1 V p-p differential
RF output divided by baseband input voltage
1200
1550
RF OUTPUT = 1850 MHz
Nominal Output Power
IQ Modulator Voltage Gain
OP1dB
Carrier Feedthrough
Sideband Suppression
Quadrature Error
I/Q Amplitude Balance
Second Harmonic
Third Harmonic
Output IP2
Output IP3
Noise Floor
RF OUTPUT = 1960 MHz
Nominal Output Power
IQ Modulator Voltage Gain
OP1dB
Carrier Feedthrough
Sideband Suppression
Quadrature Error
I/Q Amplitude Balance
Second Harmonic
Third Harmonic
Output IP2
Output IP3
Noise Floor
RF OUTPUT = 2140 MHz
Nominal Output Power
IQ Modulator Voltage Gain
OP1dB
Carrier Feedthrough
Sideband Suppression
Quadrature Error
I/Q Amplitude Balance
Second Harmonic
Third Harmonic
Output IP2
Output IP3
Noise Floor
SYNTHESIZER SPECIFICATIONS
Internal LO Range
Figure of Merit (FOM) 1
POUT − P (fLO ± (2 × fBB))
POUT − P (fLO ± (3 × fBB))
f1BB = 3.5 MHz, f2BB = 4.5 MHz, POUT ≈ −2 dBm per tone
f1BB = 3.5 MHz, f2BB = 4.5 MHz, POUT ≈ −2 dBm per tone
I/Q inputs = 0 V differential with 500 mV dc bias, 20 MHz carrier offset
RFOUT pin
Baseband VIQ = 1 V p-p differential
RF output divided by baseband input voltage
POUT − P (fLO ± (2 × fBB))
POUT − P (fLO ± (3 × fBB))
f1BB = 3.5 MHz, f2BB = 4.5 MHz, POUT ≈ −2 dBm per tone
f1BB = 3.5 MHz, f2BB = 4.5 MHz, POUT ≈ −2 dBm per tone
I/Q inputs = 0 V differential with 500 mV dc bias, 20 MHz carrier offset
RFOUT pin
Baseband VIQ = 1 V p-p differential
RF output divided by baseband input voltage
POUT − P (fLO ± (2 × fBB))
POUT − P (fLO ± (3 × fBB))
f1BB = 3.5 MHz, f2BB = 4.5 MHz, POUT ≈ −2 dBm per tone
f1BB = 3.5 MHz, f2BB = 4.5 MHz, POUT ≈ −2 dBm per tone)
I/Q inputs = 0 V differential with 500 mV dc bias, 20 MHz carrier offset
Synthesizer specifications referenced to the modulator output
Typ
Unit
MHz
MHz
4
0
13.5
−41.2
−43.7
±1
0.02
−62.2
−50.6
56
31
−158.9
dBm
dB
dBm
dBm
dBc
Degrees
dB
dBc
dBc
dBm
dBm
dBm/Hz
4.1
0.1
13.6
−40.6
−53.9
+0.7/−1.7
0.03
−74.6
−54.1
66.4
30.1
−159.6
dBm
dB
dBm
dBm
dBc
Degrees
dB
dBc
dBc
dBm
dBm
dBm/Hz
3.8
−0.2
13.1
−46.8
−44.4
±1
0.02
−71.8
−57.3
70.4
29.1
−158.1
dBm
dB
dBm
dBm
dBc
Degrees
dB
dBc
dBc
dBm
dBm
dBm/Hz
1550
2150
−220.5
Rev. B | Page 3 of 36
Max
2400
2150
MHz
dBc/Hz/Hz
ADRF6702
Data Sheet
Parameter
Test Conditions/Comments
REFERENCE CHARACTERISTICS
REFIN Input Frequency
REFIN Input Capacitance
Phase Detector Frequency
MUXOUT Output Level
REFIN, MUXOUT pins
Min
12
20
Integrated Phase Noise
Reference Spurs
PHASE NOISE (FREQUENCY =
Unit
160
MHz
pF
MHz
V
40
0.25
Low (lock detect output selected)
High (lock detect output selected)
PHASE NOISE (FREQUENCY =
1850 MHz, fPFD = 38.4 MHz)
Max
4
2.7
MUXOUT Duty Cycle
CHARGE PUMP
Charge Pump Current
Output Compliance Range
Typ
V
50
Programmable to 250 μA, 500 μA, 750 μA, 1000 μA
%
500
1
2.8
μA
V
Closed loop operation (see Figure 35 for loop filter design)
10 kHz offset
100 kHz offset
1 MHz offset
10 MHz offset
1 kHz to 10 MHz integration bandwidth
fPFD/2
fPFD
fPFD × 2
fPFD × 3
fPFD × 4
Closed loop operation (see Figure 35 for loop filter design)
−110.8
−105.8
−124.6
−150
0.27
−112
−84
−87
−93
−90
dBc/Hz
dBc/Hz
dBc/Hz
dBc/Hz
°rms
dBc
dBc
dBc
dBc
dBc
10 kHz offset
100 kHz offset
1 MHz offset
10 MHz offset
1 kHz to 10 MHz integration bandwidth
fPFD/2
fPFD
fPFD × 2
fPFD × 3
fPFD × 4
Closed loop operation (see Figure 35 for loop filter design)
−108.5
−104.2
−125.1
−149.9
0.25
−110
−83
−97
−91
−97
dBc/Hz
dBc/Hz
dBc/Hz
dBc/Hz
°rms
dBc
dBc
dBc
dBc
dBc
10 kHz offset
100 kHz offset
1 MHz offset
10 MHz offset
1 kHz to 10 MHz integration bandwidth
fPFD/2
fPFD
fPFD × 2
fPFD × 3
fPFD × 4
Measured at RFOUT, frequency = 2140 MHz
Second harmonic
Third harmonic
LOP, LON
Divide by 2 circuit in LO path enabled
Divide by 2 circuit in LO path disabled
2× LO or 1× LO mode, into a 50 Ω load, LO buffer enabled
Externally applied 2× LO, PLL disabled
Externally applied 2× LO, PLL disabled
−107.5
−102.7
−126.1
−150.4
0.25
−111
−86
−88
−91
−99
dBc/Hz
dBc/Hz
dBc/Hz
dBc/Hz
°rms
dBc
dBc
dBc
dBc
dBc
−47
−74
dBc
dBc
1960 MHz, fPFD = 38.4 MHz)
Integrated Phase Noise
Reference Spurs
PHASE NOISE (FREQUENCY =
2140 MHz, fPFD = 38.4 MHz)
Integrated Phase Noise
Reference Spurs
RF OUTPUT HARMONICS
LO INPUT/OUTPUT
Output Frequency Range
LO Output Level at 1960 MHz
LO Input Level
LO Input Impedance
Rev. B | Page 4 of 36
1550
3100
2150
4300
1
0
50
MHz
MHz
dBm
dBm
Ω
Data Sheet
ADRF6702
Parameter
Test Conditions/Comments
BASEBAND INPUTS
I and Q Input DC Bias Level
Bandwidth
IP, IN, QP, QN pins
Differential Input Impedance
Differential Input Capacitance
LOGIC INPUTS
Input High Voltage, VINH
Input Low Voltage, VINL
Input Current, IINH/IINL
Input Capacitance, CIN
TEMPERATURE SENSOR
Output Voltage
Temperature Coefficient
POWER SUPPLIES
Voltage Range
Supply Current
1
Min
Typ
Max
Unit
400
500
600
mV
POUT ≈ −7 dBm, RF flatness of IQ modulator output calibrated out
0.5 dB
3 dB
350
750
920
1
MHz
MHz
Ω
pF
CLK, DATA, LE, ENOP, LOSEL
1.4
0
3.3
0.7
0.1
5
VPTAT voltage measured at MUXOUT
TA = 25°C, RL ≥10 kΩ (LO buffer disabled)
TA = −40°C to +85°C, RL ≥10 kΩ
VCC1, VCC2, VCC3, VCC4, VCC5, VCC6, VCC7
1.64
3.9
4.75
Normal Tx mode (PLL and IQMOD enabled, LO buffer disabled)
Tx mode using external LO input (internal VCO/PLL disabled)
Tx mode with LO buffer enabled
Power-down mode
5
240
130
290
22
V
V
μA
pF
V
mV/°C
5.25
V
mA
mA
mA
μA
The figure of merit (FOM) is computed as phase noise (dBc/Hz) – 10log10(fPFD) – 20log10(fLO/fPFD). The FOM was measured across the full LO range, with fREF = 80 MHz,
fREF power = 10 dBm (500 V/μs slew rate) with a 40 MHz fPFD. The FOM was computed at 50 kHz offset.
Rev. B | Page 5 of 36
ADRF6702
Data Sheet
TIMING CHARACTERISTICS
Table 3.
Parameter
t1
t2
t3
t4
t5
t6
t7
Limit
20
10
10
25
25
10
20
Unit
ns min
ns min
ns min
ns min
ns min
ns min
ns min
Test Conditions/Comments
LE to CLK setup time
DATA to CLK setup time
DATA to CLK hold time
CLK high duration
CLK low duration
CLK to LE setup time
LE pulse width
t4
t5
CLK
t2
DATA
DB23 (MSB)
t3
DB22
DB2
(CONTROL BIT C3)
DB1
(CONTROL BIT C2)
DB0 (LSB)
(CONTROL BIT C1)
t6
t1
08568-002
LE
t7
Figure 2. Timing Diagram
Rev. B | Page 6 of 36
Data Sheet
ADRF6702
ABSOLUTE MAXIMUM RATINGS
Table 4.
Parameter
Supply Voltage (VCC1 to VCC7)
Digital I/O, CLK, DATA, LE
LOP, LON
IP, IN, QP, QN
REFIN
θJA (Exposed Paddle Soldered Down)1
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
1
Rating
5.5 V
−0.3 V to +3.6 V
18 dBm
−0.5 V to +1.5 V
−0.3 V to +3.6 V
35°C/W
150°C
−40°C to +85°C
−65°C to +150°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
Per JDEC standard JESD 51-2.
Rev. B | Page 7 of 36
ADRF6702
Data Sheet
40
39
38
37
36
35
34
33
32
31
DECL3
VTUNE
LOP
LON
LOSEL
GND
VCC7
IP
IN
GND
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
ADRF6702
TOP VIEW
(Not to Scale)
30
29
28
27
26
25
24
23
22
21
GND
VCC6
GND
VCC5
RFOUT
GND
NC
GND
VCC4
GND
NOTES
1. NC = NO CONNECT. DO NOT CONNECT TO THIS PIN.
2. THE EXPOSED PADDLE SHOULD BE SOLDERED TO A
LOW IMPEDANCE GROUND PLANE.
08568-003
GND
DATA
CLK
LE
GND
ENOP
VCC3
QP
QN
GND
11
12
13
14
15
16
17
18
19
20
VCC1 1
DECL1 2
CP 3
GND 4
RSET 5
REFIN 6
GND 7
MUXOUT 8
DECL2 9
VCC2 10
Figure 3. Pin Configuration
Table 5. Pin Function Descriptions
Pin No.
1, 10, 17, 22, 27, 29, 34
2
Mnemonic
VCC1, VCC2, VCC3,
VCC4, VCC5, VCC6,
VCC7
DECL1
3
CP
4, 7, 11, 15, 20, 21, 23,
25, 28, 30, 31, 35
24
5
GND
NC
RSET
Description
Power Supply Pins. The power supply voltage range is 4.75 V to 5.25 V. Drive all of
these pins from the same power supply voltage. Decouple each pin with 100 pF and
0.1 μF capacitors located close to the pin.
Decoupling Node for Internal 3.3 V LDO. Decouple this pin with 100 pF and 0.1 μF
capacitors located close to the pin.
Charge Pump Output Pin. Connect VTUNE to this pin through the loop filter. If
an external VCO is being used, connect the output of the loop filter to the VCO’s
voltage control pin. The PLL control loop should then be closed by routing the VCO’s
frequency output back into the ADRF6702 through the LON and LOP pins.
Ground. Connect these pins to a low impedance ground plane.
Do not connect to this pin.
Charge Pump Current. The nominal charge pump current can be set to 250 μA, 500 μA,
750 μA, or 1000 μA using DB10 and DB11 of Register 4 and by setting DB18 to 0 (CP
reference source).
In this mode, no external RSET is required. If DB18 is set to 1, the four nominal charge
pump currents (INOMINAL) can be externally tweaked according to the following
equation:
⎛ 217.4 × I CP
R SET = ⎜⎜
⎝ I NOMINAL
6
REFIN
8
MUXOUT
9
DECL2
12
DATA
⎞
⎟ − 37.8 Ω
⎟
⎠
where ICP is the base charge pump current in microamps. For further details on the
charge pump current, see the Register 4—PLL Charge Pump, PFD, and Reference Path
Control section.
Reference Input. The nominal input level is 1 V p-p. Input range is 12 MHz to 160 MHz.
This pin has high input impedance and should be ac-coupled. If REFIN is being driven
by laboratory test equipment, the pin should be externally terminated with a 50 Ω
resistor (place the ac-coupling capacitor between the pin and the resistor). When
driven from an 50 Ω RF signal generator, the recommended input level is 4 dBm.
Multiplexer Output. This output allows a digital lock detect signal, a voltage
proportional to absolute temperature (VPTAT), or a buffered, frequency-scaled
reference signal to be accessed externally. The output is selected by programming
DB21 to DB23 in Register 4.
Decoupling Node for 2.5 V LDO. Connect 100 pF, 0.1 μF, and 10 μF capacitors between this
pin and ground.
Serial Data Input. The serial data input is loaded MSB first with the three LSBs being
the control bits.
Rev. B | Page 8 of 36
Data Sheet
ADRF6702
Pin No.
13
Mnemonic
CLK
14
LE
16
18, 19, 32, 33
ENOP
QP, QN, IN, IP
26
RFOUT
36
LOSEL
37, 38
LON, LOP
39
VTUNE
40
DECL3
EP
Description
Serial Clock Input. This serial clock input is used to clock in the serial data to the
registers. The data is latched into the 24-bit shift register on the CLK rising edge.
Maximum clock frequency is 20 MHz.
Latch Enable. When the LE input pin goes high, the data stored in the shift registers is
loaded into one of the six registers, the relevant latch being selected by the first three
control bits of the 24-bit word.
Modulator Output Enable/Disable. See Table 6.
Modulator Baseband Inputs. Differential in-phase and quadrature baseband inputs.
These inputs should be dc-biased to 0.5 V.
RF Output. Single-ended, 50 Ω internally biased RF output. RFOUT must be ac-coupled
to its load.
LO Select. This digital input pin determines whether the LOP and LON pins operate as
inputs or outputs. This pin should not be left floating. LOP and LON become inputs if
the LOSEL pin is set low and the LDRV bit of Register 5 is set low. External LO drive
must be a 2× LO. In addition to setting LOSEL and LDRV low and providing an external
2× LO, the LXL bit of Register 5 (DB4) must be set to 1 to direct the external LO to the
IQ modulator. LON and LOP become outputs when LOSEL is high or if the LDRV bit of
Register 5 (DB3) is set to 1. A 1× LO or 2× LO output can be selected by setting the
LDIV bit of Register 5 (DB5) to 1 or 0 respectively (see Table 7).
Local Oscillator Input/Output. The internally generated 1× LO or 2× LO is available on
these pins. When internal LO generation is disabled, an external 1× LO or 2× LO can be
applied to these pins.
VCO Control Voltage Input. This pin is driven by the output of the loop filter. Nominal
input voltage range on this pin is 1.3 V to 2.5 V. If the external VCO mode is activated,
this pin can be left open.
Decoupling Node for VCO LDO. Connect a 100 pF capacitor and a 10 μF capacitor
between this pin and ground.
Exposed Paddle. The exposed paddle should be soldered to a low impedance
ground plane.
Table 6. Enabling RFOUT
ENOP
X1
0
1
1
Register 5 Bit DB6
0
X1
1
RFOUT
Disabled
Disabled
Enabled
X = don’t care.
Table 7. LO Port Configuration 1, 2
LON/LOP Function
LOSEL
Register 5 Bit DB5(LDIV)
Register 5 Bit DB4(LXL)
Register 5 Bit DB3 (LDRV)
Input (2× LO)
Output (Disabled)
Output (1× LO)
Output (1× LO)
Output (1× LO)
Output (2× LO)
Output (2× LO)
Output (2× LO)
0
0
0
1
1
0
1
1
X
X
0
0
0
1
1
1
1
0
0
0
0
0
0
0
0
0
1
0
1
1
0
1
1
2
X = don’t care.
LOSEL should not be left floating.
Rev. B | Page 9 of 36
ADRF6702
Data Sheet
TYPICAL PERFORMANCE CHARACTERISTICS
VS = 5 V; TA = 25°C; baseband I/Q amplitude = 1 V p-p differential sine waves in quadrature with a 500 mV dc bias; baseband I/Q
frequency (fBB) = 1 MHz; fPFD = 38.4 MHz; fREF = 153.6 MHz at +4 dBm Re:50 Ω (1 V p-p); 130 kHz loop filter, unless otherwise noted.
10
TA = –40°C
TA = +25°C
TA = +85°C
9
6
5
4
3
2
7
6
5
VS = 5V
4
3
VS = 4.75V
VS = 5.25V
2
1
0
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
0
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
08568-104
1
LO FREQUENCY (MHz)
Figure 4. Single Sideband (SSB) Output Power (POUT) vs.
LO Frequency (fLO) and Temperature; Multiple Devices Shown
Figure 7. Single Sideband (SSB) Output Power (POUT) vs.
LO Frequency (fLO) and Power Supply; Multiple Devices Shown
20
20
TA = –40°C
TA = +25°C
TA = +85°C
19
18
17
16
15
14
13
12
18
17
16
15
VS = 5V
VS = 5.25V
14
13
VS = 4.75V
12
11
10
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
10
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
08568-105
11
LO FREQUENCY (MHz)
LO FREQUENCY (MHz)
Figure 5. SSB Output 1dB Compression Point (OP1dB) vs. LO Frequency (fLO)
and Temperature; Multiple Devices Shown
SSB OUTPUT
POWER (dBm)
–20
–30
–40
–50
CARRIER FEEDTHROUGH (dBm)
0
16
–10
12
8
4
SIDEBAND
SUPPRESSION (dBc)
0
–60
–4
–70
–8
–80
–90
–100
0.1
–12
SECOND-ORDER
DISTORTION (dBc)
1
SECOND-ORDER DISTORTION (dBc),
THIRD-ORDER DISTORTION (dBc),
CARRIER FEEDTHROUGH (dBm),
SIDEBAND SUPPRESSION (dBc)
SECOND-ORDER DISTORTION (dBc),
THIRD-ORDER DISTORTION (dBc),
CARRIER FEEDTHROUGH (dBm),
SIDEBAND SUPPRESSION (dBc)
–10
20
SSB OUTPUT POWER (dBm)
THIRD-ORDER
DISTORTION (dBc)
Figure 8. SSB Output 1dB Compression Point (OP1dB) vs. LO Frequency (fLO)
and Power Supply
–20
10
BASEBAND INPUT VOLTAGE (V p-p Differential)
Figure 6. SSB Output Power, Second- and Third-Order Distortion, Carrier
Feedthrough and Sideband Suppression vs. Baseband Differential Input
Voltage (fOUT = 1960 MHz)
–20
–30
–40
SSB OUTPUT
POWER (dBm)
THIRD-ORDER
DISTORTION (dBc)
SIDEBAND
SUPPRESSION (dBc)
CARRIER
FEEDTHROUGH (dBm)
20
16
12
8
4
–50
0
–60
–4
–70
–80
–8
SECOND-ORDER
DISTORTION (dBc)
–90
–16
–100
0.1
08568-106
0
08568-108
1dB OUTPUT COMPRESSION (dBm)
19
08568-107
7
LO FREQUENCY (MHz)
1dB OUTPUT COMPRESSION (dBm)
8
SSB OUTPUT POWER (dBm)
SSB OUTPUT POWER (dBm)
8
–12
SSB OUTPUT POWER (dBm)
9
–16
1
–20
10
BASEBAND INPUT VOLTAGE (V p-p Differential)
Figure 9. SSB Output Power, Second- and Third-Order Distortion, Carrier
Feedthrough and Sideband Suppression vs. Baseband Differential Input
Voltage (fOUT = 2140 MHz)
Rev. B | Page 10 of 36
08568-109
10
Data Sheet
ADRF6702
–30
–40
–50
–60
–30
–40
–50
–60
–70
–80
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
–80
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
08568-110
–70
LO FREQUENCY (MHz)
0
TA = –40°C
TA = +25°C
TA = +85°C
TA = –40°C
TA = +25°C
TA = +85°C
–10
–20
–30
–40
–50
–60
–70
–20
–30
–40
–50
–60
–70
–80
–90
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
–90
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
LO FREQUENCY (MHz)
08568-111
–80
Figure 11. Sideband Suppression vs. LO Frequency (fLO) and Temperature;
Multiple Devices Shown
LO FREQUENCY (MHz)
Figure 14. Sideband Suppression vs. LO Frequency (fLO) and Temperature
After Nulling at 25°C; Multiple Devices Shown
–20
90
OIP2
70
60
50
40
OIP3
30
20
TA = –40°C
TA = +25°C
TA = +85°C
–30
–35
–40
THIRD-ORDER DISTORTION
–45
–50
–55
–60
–65
–70
–75
0
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
LO FREQUENCY (MHz)
Figure 12. OIP3 and OIP2 vs. LO Frequency (fLO) and Temperature
(POUT ≈ −2 dBm per Tone); Multiple Devices Shown
SECOND-ORDER DISTORTION
–80
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
08568-112
10
TA = –40°C
TA = +25°C
TA = +85°C
–25
THIRD-ORDER DISTORTION (dBc),
SECOND-ORDER DISTORTION (dBc)
80
LO FREQUENCY (MHz)
08568-115
–10
LO FREQUENCY (MHz)
Figure 13. Carrier Feedthrough vs. LO Frequency (fLO) and Temperature After
Nulling at 25°C; Multiple Devices Shown
SIDEBAND SUPPRESSION (dBc)
0
SIDEBAND SUPPRESSION (dBc)
–20
08568-113
–20
Figure 10. Carrier Feedthrough vs. LO Frequency (fLO) and Temperature;
Multiple Devices Shown
OUTPUT IP3 AND IP2 (dBm)
TA = –40°C
TA = +25°C
TA = +85°C
–10
CARRIER FEEDTHROUGH (dBm)
–10
CARRIER FEEDTHROUGH (dBm)
0
TA = –40°C
TA = +25°C
TA = +85°C
08568-114
0
Figure 15. Second- and Third-Order Distortion vs. LO Frequency (fLO) and
Temperature
Rev. B | Page 11 of 36
3.5kHz LOOP FILTER
130kHz LOOP FILTER
0.8
0.7
0.6
0.5
0.4
0.3
0.2
100k
1M
10M
100M
0
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
LO FREQUENCY (MHz)
Figure 19. Integrated Phase Noise vs. LO Frequency
–80
TA = –40°C
TA = +25°C
TA = +85°C
–85
–90
PHASE NOISE (dBc/Hz)
–95
3.5kHz LOOP FILTER
–110
–120
130kHz LOOP FILTER
–130
–140
–150
–160
1k
10k
100k
1M
10M
100M
OFFSET = 100kHz
–115
–120
–125
–130
OFFSET = 5MHz
LO FREQUENCY (MHz)
Figure 20. Phase Noise vs. LO Frequency at 1 kHz, 100 kHz, and 5 MHz Offsets
PHASE NOISE (dBc/Hz)
TA = –40°C
TA = +25°C
TA = +85°C
3.5kHz LOOP FILTER
130kHz LOOP FILTER
100k
–110
–150
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
1M
10M
100M
OFFSET FREQUENCY (Hz)
Figure 18. Phase Noise vs. Offset Frequency and Temperature,
fLO = 2140 MHz, 3.5 KHz Filter
08568-118
PHASE NOISE, LO FREQUENCY = 2140MHz (dBc/Hz)
0
10k
–105
–145
Figure 17. Phase Noise vs. Offset Frequency and Temperature,
fLO = 1960 MHz, 3.5 KHz Filter
–150
–160
1k
OFFSET = 1kHz
–140
OFFSET FREQUENCY (Hz)
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
–120
–130
–140
TA = –40°C
TA = +25°C
TA = +85°C
–100
–135
08568-117
PHASE NOISE, LO FREQUENCY = 1960MHz (dBc/Hz)
Figure 16. Phase Noise vs. Offset Frequency and Temperature,
fLO = 1850 MHz, 3.5 KHz Filter
08568-120
10k
08568-119
0.1
OFFSET FREQUENCY (Hz)
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
TA = –40°C
TA = +25°C
TA = +85°C
0.9
–80
–85
–90
–95
–100
–105
–110
–115
–120
–125
–130
TA = –40°C
TA = +25°C
TA = +85°C
OFFSET = 10kHz
OFFSET = 1MHz
–135
–140
–145
–150
–155
–160
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
LO FREQUENCY (MHz)
08568-121
–130
–140
–150
–160
1k
1.0
TA = –40°C
TA = +25°C
TA = +85°C
INTEGRATED PHASE NOISE (° rms)
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
–120
Data Sheet
08568-116
PHASE NOISE, LO FREQUENCY = 1850MHz (dBc/Hz)
ADRF6702
Figure 21. Phase Noise vs. LO Frequency at 10 kHz and 1 MHz Offsets
Rev. B | Page 12 of 36
Data Sheet
–70
TA = –40°C
TA = +25°C
TA = +85°C
–75
–80
–80
–85
–85
SPUR LEVEL (dBc)
–90
–95
–100
–105
–95
–100
–105
LO FREQUENCY (MHz)
–70
–75
1 × PFD FREQUENCY
3 × PFD FREQUENCY
LO FREQUENCY (MHz)
Figure 25. PLL Reference Spurs vs. LO Frequency (2× PFD and 4× PFD) at LO
Output
–70
TA = –40°C
TA = +25°C
TA = +85°C
–75
–80
–85
–85
SPUR LEVEL (dBc)
–80
–90
–95
–100
–105
TA = –40°C
TA = +25°C
TA = +85°C
1 × PFD FREQUENCY
3 × PFD FREQUENCY
–90
–95
–100
0.5 × PFD FREQUENCY
–105
–110
–110
–115
0.5 × PFD FREQUENCY
–120
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
LO FREQUENCY (MHz)
Figure 23. PLL Reference Spurs vs. LO Frequency (0.5× PFD, 1× PFD,
and 3× PFD) at Modulator Output
2.8
–120
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
08568-123
–115
08568-126
–115
–120
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
08568-122
–115
–120
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
Figure 22. PLL Reference Spurs vs. LO Frequency (2× PFD and 4× PFD) at
Modulator Output
SPUR LEVEL (dBc)
–90
–110
–110
LO FREQUENCY (MHz)
Figure 26. PLL Reference Spurs vs. LO Frequency (0.5× PFD, 1× PFD, and
3× PFD) at LO Output
0
TA = –40°C
TA = +25°C
TA = +85°C
2.6
–20
–40
PHASE NOISE (dBc/Hz)
2.4
2.2
2.0
1.8
1.6
–60
–100
1.2
–160
1.0
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
–180
1k
Figure 24. VTUNE vs. LO Frequency and Temperature
LO = 2140.48MHz
LO = 1841.074MHz
–120
–140
LO FREQUENCY (MHz)
LO = 1943.26MHz
–80
1.4
08568-124
VTUNE (V)
TA = –40°C
TA = +25°C
TA = +85°C
2 × PFD FREQUENCY
4 × PFD FREQUENCY
08568-125
SPUR LEVEL (dBc)
–75
2 × PFD FREQUENCY
4 × PFD FREQUENCY
10k
100k
FREQUENCY (Hz)
1M
10M
08568-127
–70
ADRF6702
Figure 27. Open-Loop VCO Phase Noise at 1841.074 MHz, 1943.26 MHz, and
2140.48 MHz
Rev. B | Page 13 of 36
1850MHz
1960MHz
2140MHz
CUMULATIVE PERCENTAGE (%)
90
80
70
60
50
40
30
20
0
–164 –163 –162 –161 –160 –159 –158 –157 –156 –155 –154
NOISE FLOOR (dBm/Hz)
08568-128
10
Figure 28. IQ Modulator Noise Floor Cumulative Distributions at 1850 MHz,
1960 MHz, and 2140 MHz
0
–10
–20
–30
–40
–50
SSB OUTPUT POWER
–60
–70
–80
LO FEEDTHROUGH
–90
–100
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
LO FREQUENCY (MHz)
08568-130
100
Data Sheet
SSB OUTPUT POWER AND LO FEEDTHROUGH (dBm)
ADRF6702
Figure 30. SSB Output Power and LO Feedthrough with RF Output Disabled
2.0
1.9
1.8
15
1.7
VPTAT (V)
20
10
5
1.6
1.5
1.4
0
1.3
–5
1.2
–15
1.0
–40
0
50
100
150
200
250
TIME (µs)
300
Figure 29. Frequency Deviation from LO Frequency at
LO = 1.97 GHz to 1.96 GHz vs. Lock Time
Rev. B | Page 14 of 36
–15
10
35
60
TEMPERATURE (°C)
Figure 31. VPTAT Voltage vs. Temperature
85
08568-131
1.1
–10
08568-129
FREQUENCY DEVIATION FROM 1960MHz (MHz)
25
Data Sheet
ADRF6702
0
–1
–2
RETURN LOSS (dB)
–3
RF OUTPUT
–4
–5
–6
–7
–8
–9
LO = 1550MHz
LO INPUT
LO = 2150MHz
–10
–11
–12
LO FREQUENCY (MHz)
Figure 32. Input Return Loss of LO Input (LON, LOP Driven Through MABA007159 1:1 Balun) and Output Return Loss of RFOUT vs. Frequency
280
270
SUPPLY CURRENT (mA)
260
250
240
TA = +85°C
TA = +25°C
230
220
210
TA = –40°C
200
180
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
LO FREQUENCY (MHz)
08568-133
190
Figure 33. Power Supply Current vs. Frequency and Temperature (PLL and
IQMOD Enabled, LO Buffer Disabled)
Rev. B | Page 15 of 36
08568-234
–14
1550 1600 1650 1700 1750 1800 1850 1900 1950 2000 2050 2100 2150
08568-032
–13
Figure 34. Smith Chart Representation of RF Output
ADRF6702
Data Sheet
THEORY OF OPERATION
The ADRF6702 integrates a high performance IQ modulator
with a state of the art fractional-N PLL. The ADRF6702 also
integrates a low noise VCO. The programmable SPI port allows
the user to control the fractional-N PLL functions and the
modulator optimization functions. This includes the capability
to operate with an externally applied LO or VCO.
The quadrature modulator core within the ADRF6702 is a part
of the next generation of industry-leading modulators from
Analog Devices, Inc. The baseband inputs are converted to
currents and then mixed to RF using high performance NPN
transistors. The mixer output currents are transformed to a
single-ended RF output using an integrated RF transformer
balun. The high performance active mixer core, coupled with
the low-loss RF transformer balun results in an exceptional
OIP3 and OP1dB, with a very low output noise floor for excellent dynamic range. The use of a passive transformer balun
rather than an active output stage leads to an improvement
in OIP3 with no sacrifice in noise floor. At 1960 MHz the
ADRF6702 typically provides an output P1dB of 13.6 dBm,
OIP3 of 30.1 dBm, and an output noise floor of −156.5 dBm/Hz.
Typical image rejection under these conditions is −44.4 dBc
with no additional I and Q gain compensation.
PLL + VCO
The fractional divide function of the PLL allows the frequency
multiplication value from REFIN to the LOP/LON outputs to
be a fractional value rather than restricted to an integer as in
traditional PLLs. In operation, this multiplication value is INT
+ (FRAC/MOD) where INT is the integer value, FRAC is the
fractional value, and MOD is the modulus value, all of which
are programmable via the SPI port. In previous fractional-N
PLL designs, the fractional multiplication was achieved by
periodically changing the fractional value in a deterministic
way. The downside of this was often spurious components close
to the fundamental signal. In the ADRF6702, a sigma delta
modulator is used to distribute the fractional value randomly,
thus significantly reducing the spurious content due to the
fractional function.
BASIC CONNECTIONS FOR OPERATION
Figure 35 shows the basic connections for operating the
ADRF6702 as they are implemented on the device’s evaluation
board. The seven power supply pins should be individually
decoupled using 100 pF and 0.1 μF capacitors located as close
as possible to the pins. A single 10 μF capacitor is also recommended. The three internal decoupling nodes (labeled DECL3,
DECL2, and DECL1) should be individually decoupled with
capacitors as shown in Figure 35.
The four I and Q inputs should be driven with a bias level of
500 mV. These inputs are generally dc-coupled to the outputs of
a dual DAC (see the DAC-to-IQ Modulator Interfacing and IQ
Filtering sections for more information).
A 1 V p-p (0.353 V rms) differential sine wave on the I and Q
inputs results in a single sideband output power of +4.1 dBm (at
1960 MHz) at the RFOUT pin (this pin should be ac-coupled as
shown in Figure 35). This corresponds to an IQ modulator
voltage gain of +0.1 dB.
The reference frequency for the PLL (typically 1 V p-p between
12 MHz and 160 MHz) should be applied to the REFIN pin,
which should be ac-coupled. If the REFIN pin is being driven
from a 50 Ω source (for example, a lab signal generator), the
pin should be terminated with 50 Ω as shown in Figure 35 (an
RF drive level of +4 dBm should be applied). Multiples or
fractions of the REFIN signal can be brought back off-chip at
the multiplexer output pin (MUXOUT). A lock-detect signal
and an analog voltage proportional to the ambient temperature
can also be brought out on this pin by setting the appropriate
bits on (DB21-DB23) in Register 4 (see the Register Description
section).
EXTERNAL LO
The internally generated local oscillator (LO) signal can be
brought off-chip as either a 1× LO or a 2× LO (via pins LOP
and LON) by asserting the LOSEL pin and making the appropriate internal register settings. The LO output must be disabled
whenever the RF output of the IQ modulator is disabled.
The LOP and LON pins can also be used to apply an external
LO. This can be used to bypass the internal PLL/VCO or if
operation using an external VCO is desired. To turn off the
PLL Register 6, Bits[20:17] must be zero.
Rev. B | Page 16 of 36
Data Sheet
ADRF6702
VCC
R43
10kΩ
(0402)
S1
VDD
C27
0.1µF
(0402)
C25
0.1µF
(0402)
C23
0.1µF
(0402)
C20
0.1µF
(0402)
C19
0.1µF
(0402)
C9
0.1µF
(0402)
C8
100pF
(0402)
C26
100pF
(0402)
C24
100pF
(0402)
C22
100pF
(0402)
C21
100pF
(0402)
C18
100pF
(0402)
C10
100pF
(0402)
VDD
34
R40
10kΩ
(0402)
VDD
VDD
27
29
VDD
22
VDD
17
LE (USB)
DATA (USB)
CLK (USB)
VDD
10
1
16
13
12
14
9
5
1
4
3
C6
100pF LOP
(0402)
37
BUFFER
MABA-007159 C5
100pF
(0402)
REF_IN
REFIN
R73
49.9Ω
(0402)
SEE TEXT
REFOUT OPEN
R16
OPEN
(0402)
FRACTION
REG
ADRF6702
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
×2
6
÷2
MUX
÷4
MUXOUT
MODULUS
TEMP
SENSOR
7
2
2:1
MUX
INTEGER
REG
18
N COUNTER
21 TO 123
R2
R37 OPEN
0Ω (0402)
(0402)
CP
TEST
POINT
(OPEN)
R38
OPEN
(0402)
C14
22pF
(0603)
RSET
CP
40
VTUNE
26
DECL3
R62
0Ω
(0402)
C13
6.8pF
(0603)
C16
100pF
(0402)
C17
0.1µF
(0402)
C42
10µF
(0603)
DECL1
C12
100pF
(0402)
C11
0.1µF
(0402)
C41
OPEN
(0603)
C2
OPEN
(0402)
C40
22pF
(0603)
33
QP
QN
R23
OPEN
(0402)
IN
IP
R3
OPEN
(0402)
QP
QN
IN
IP
RFOUT
OPEN
VTUNE
OPEN
C3
100pF
(0402)
RFOUT
R63
OPEN
(0402)
R12
0Ω
(0402)
R11
OPEN
(0402)
C43
10µF
(0603)
39
R9 10kΩ R65 10kΩ
(0402)
(0402)
R10
3kΩ
(0603)
C15
2.7nF
(1206)
19
32
3
5
24
NC
GND
÷2
0/90
CHARGE PUMP
250µA,
500µA (DEFAULT),
750µA,
1000µA
–
PHASE
+ FREQUENCY
DETECTOR
11 15 20 21 23 25 28 30 31 35
VCO
CORE
PRESCALER
÷2
8
4
SPI
INTERFACE
DIVIDER
÷2
BUFFER
38
T3
C29
100pF
(0402)
DECL2
36
C1
100pF
(0402)
08568-023
LOSEL
LON
EXT LO
C7
0.1µF
(0402)
LE
R39
10kΩ
(0402)
R47
10kΩ
(0402)
DATA
VCC
R20
0Ω
(0402)
CLK
C28
10µF
(3216)
S2
ENOP
VCC
RED
+5V
NOTES
1. NC = NO CONNECT. DO NOT CONNECT TO THIS PIN.
Figure 35. Basic Connections for Operation (Loop Filter Set to 130 kHz)
LOOP FILTER
Table 8. Recommended Loop Filter Components
The loop filter is connected between the CP and VTUNE pins.
The return for the loop filter components should be to Pin 40
(DECL3). The loop filter design in Figure 35 results in a 3 dB
loop bandwidth of 130 kHz. The ADRF6702 closed loop phase
noise was also characterized using a 2.5 kHz loop filter design.
The recommended components for both filter designs are
shown in Table 8. For assistance in designing loop filters with
other characteristics, download the most recent revision of
ADIsimPLL™ from www.analog.com/adisimpll. Operation with
an external VCO is possible. In this case, the return for the loop
filter components is ground (assuming a ground reference on
the external VCO tuning input). The output of the loop filter is
connected to the external VCO’s tuning pin. The output of the
VCO is brought back into the device on the LOP and LON pins
(using a balun if necessary).
Component
C14
R10
C15
R9
C13
R65
C40
R37
R11
R12
Rev. B | Page 17 of 36
130 kHz Loop Filter
22 pF
3 kΩ
2.7 nF
10 kΩ
6.8 pF
10 kΩ
22 pF
0Ω
Open
0Ω
2.5 kHz Loop Filter
0.1 μF
68 Ω
4.7 μF
270 Ω
47 nF
0Ω
Open
0Ω
Open
0Ω
ADRF6702
Data Sheet
AD9122
The ADRF6702 is designed to interface with minimal components
to members of the Analog Devices, Inc., family of TxDACs®. These
dual-channel differential current output DACs provide an output
current swing from 0 mA to 20 mA. The interface described in
this section can be used with any DAC that has a similar output.
An example of an interface using the AD9122 TxDAC is shown
in Figure 36. The baseband inputs of the ADRF6702 require
a dc bias of 500 mV. The average output current on each of the
outputs of the AD9122 is 10 mA. Therefore, a single 50 Ω resistor to ground from each of the DAC outputs results in an average
current of 10 mA flowing through each of the resistors, thus
producing the desired 500 mV dc bias for the inputs to the
ADRF6702.
ADRF6702
OUT1_P
IN
RBIP
50Ω
RBIN
50Ω
IP
OUT1_N
OUT2_N
08568-033
OUT2_P
QN
RBQN
50Ω
RBQP
50Ω
QP
Figure 36. Interface Between the AD9122 and ADRF6702 with 50 Ω Resistors
to Ground to Establish the 500 mV DC Bias for the ADRF6702 Baseband Inputs
The AD9122 output currents have a swing that ranges from
0 mA to 20 mA. With the 50 Ω resistors in place, the ac voltage
swing going into the ADRF6702 baseband inputs ranges from
0 V to 1 V (with the DAC running at 0 dBFS). So the resulting
drive signal from each differential pair is 2 V p-p differential
with a 500 mV dc bias.
OUT1_P
IP
RBIP
50Ω
RSL
(SEE TEXT)
RBIN
50Ω
IN
OUT1_N
OUT2_N
OUT2_P
QN
RBQN
50Ω
RBQP
50Ω
RSL
(SEE TEXT)
QP
Figure 37. AC Voltage Swing Reduction Through the Introduction
of a Shunt Resistor Between the Differential Pair
The value of this ac voltage swing limiting resistor (RSL as shown
in Figure 37) is chosen based on the desired ac voltage swing
and IQ modulator output power. Figure 38 shows the relationship between the swing-limiting resistor and the peak-to-peak
ac swing that it produces when 50 Ω bias-setting resistors are
used. A higher value of swing-limiting resistor will increase the
output power of the ADRF6702 and signal-to-noise ratio (SNR)
at the cost if higher intermodulation distortion. For most
applications, the optimum value for this resistor will be between
100 Ω and 300 Ω.
When setting the size of the swing-limiting resistor, the input
impedance of the I and Q inputs should be taken into account.
The I and Q inputs have a differential input resistance of 920 Ω.
As a result, the effective value of the swing-limiting resistance is
920 Ω in parallel with the chosen swing-limiting resistor. For
example, if a swing-limiting resistance of 200 Ω is desired
(based on Figure 37), the value of RSL should be set such that
200 Ω = (920 × RSL)/(920 + RSL)
resulting in a value for RSL of 255 Ω.
2.0
1.8
The voltage swing for a given DAC output current can be
reduced by adding a third resistor to the interface. This resistor
is placed in the shunt across each differential pair, as shown in
Figure 37. It has the effect of reducing the ac swing without
changing the dc bias already established by the 50 Ω resistors.
DIFFERENTIAL SWING (V p-p)
ADDING A SWING-LIMITING RESISTOR
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
10
100
1000
RSL (Ω)
10000
08568-035
AD9122
ADRF6702
08568-034
DAC-TO-IQ MODULATOR INTERFACING
Figure 38. Relationship Between the AC Swing-Limiting Resistor and the
Peak-to-Peak Voltage Swing with 50 Ω Bias-Setting Resistors
Rev. B | Page 18 of 36
Data Sheet
ADRF6702
BASEBAND BANDWIDTH
Figure 39 shows the frequency response of the ADRF6702’s
baseband inputs. This plot shows 0.5 dB and 3 dB bandwidths
of 350 MHz and 750 MHz respectively. Any flatness variations
across frequency at the ADRF6702 RF output have been
calibrated out of this measurement.
2
0
RESISTANCE
700
0.6
600
0.4
500
0.2
400
0
100
200
300
400
CAPACITANCE (pF)
0.8
800
0
500
BASEBAND FREQUENCY (MHz)
Figure 40. Differential Baseband Input R and Input C Equivalents (Shunt R
and Shunt C)
DEVICE PROGRAMMING AND REGISTER
SEQUENCING
The device is programmed via a 3-pin SPI port. The timing
requirements for the SPI port are shown in Table 3 and Figure 2.
Eight programmable registers, each with 24 bits, control the
operation of the device. The register functions are listed in
Table 9. The eight registers should initially be programmed
in reverse order, starting with Register 7 and finishing with
Register 0. Once all eight registers have been initially
programmed, any of the registers can be updated without any
attention to sequencing.
Software is available on the ADRF6702 product page at
www.analog.com that allows programming of the evaluation
board from a PC running Windows® XP or Windows Vista.
–2
–4
To operate correctly under Windows XP, Version 3.5 of
Microsoft .NET must be installed. To run the software on
a Windows 7 PC, XP emulation mode must be used (using
Virtual PC).
–6
–8
100
BB FREQUENCY (MHz)
1000
08568-134
BASEBAND FREQUENCY RESPONSE (dBc)
4
CAPACITANCE
08568-141
Unless a swing-limiting resistor of 100 Ω is chosen, the filter
must be designed to support different source and load
impedances. In addition, the differential input capacitance of
the I and Q inputs (1 pF) should be factored into the filter
design. Modern filter design tools allow for the simulation and
design of filters with differing source and load impedances as
well as inclusion of reactive load components.
1.0
900
RESISTANCE (Ω)
An antialiasing filter must be placed between the DAC and
modulator to filter out Nyquist images and broadband DAC
noise. The interface for setting up the biasing and ac swing
discussed in the Adding a Swing-Limiting Resistor section,
lends itself well to the introduction of such a filter. The filter
can be inserted between the dc bias setting resistors and the
ac swing-limiting resistor. Doing so establishes the input and
output impedances for the filter.
–10
10
1.2
1000
IQ FILTERING
Figure 39. Baseband Bandwidth
Rev. B | Page 19 of 36
ADRF6702
Data Sheet
REGISTER SUMMARY
Table 9. Register Functions
Register
Register 0
Register 1
Register 2
Register 3
Register 4
Register 5
Register 6
Register 7
Function
Integer divide control (for the PLL)
Modulus divide control (for the PLL)
Fractional divide control (for the PLL)
Σ-Δ modulator dither control
PLL charge pump, PFD, and reference path control
LO path and modulator control
VCO control and VCO enable
External VCO enable
Rev. B | Page 20 of 36
Data Sheet
ADRF6702
REGISTER DESCRIPTION
Integer Divide Ratio
REGISTER 0—INTEGER DIVIDE CONTROL
(DEFAULT: 0x0001C0)
The integer divide ratio bits are used to set the integer value in
Equation 2. The INT, FRAC, and MOD values make it possible
to generate output frequencies that are spaced by fractions of
the PFD frequency. The VCO frequency (fVCO) equation is
With Register 0, Bits[2:0] set to 000, the on-chip integer divide
control register is programmed as shown in Figure 41.
Divide Mode
fVCO = 2 × fPFD × (INT + (FRAC/MOD))
Divide mode determines whether fractional mode or integer
mode is used. In integer mode, the RF VCO output frequency
(fVCO) is calculated by
where:
INT is the preset integer divide ratio value (24 to 119 in
fractional mode).
MOD is the preset fractional modulus (1 to 2047).
FRAC is the preset fractional divider ratio value (0 to MOD − 1).
(1)
where:
fVCO is the output frequency of the internal VCO.
fPFD is the frequency of operation of the phase-frequency detector.
INT is the integer divide ratio value (21 to 123 in integer mode).
RESERVED
DIVIDE
MODE
DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DM
ID6
ID5
ID4
ID3
ID2
ID1
ID0
C3(0) C2(0) C1(0)
0
0
0
0
0
0
0
0
0
0
0
INTEGER DIVIDE RATIO
0
0
DM
DIVIDE MODE
0
FRACTIONAL (DEFAULT)
1
INTEGER
CONTROL BITS
DB1
ID6
ID5
ID4
ID3
ID2
ID1
ID0
INTEGER DIVIDE RATIO
0
0
1
0
1
0
1
21 (INTEGER MODE ONLY)
0
0
1
0
1
1
0
22 (INTEGER MODE ONLY)
0
0
1
0
1
1
1
23 (INTEGER MODE ONLY)
0
0
1
1
0
0
0
24
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
0
1
1
1
0
0
0
56 (DEFAULT)
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
1
1
1
0
1
1
1
119
1
1
1
1
0
0
0
120 (INTEGER MODE ONLY)
1
1
1
1
0
0
1
121 (INTEGER MODE ONLY)
1
1
1
1
0
1
0
122 (INTEGER MODE ONLY)
1
1
1
1
0
1
1
123 (INTEGER MODE ONLY)
Figure 41. Register 0—Integer Divide Control Register Map
Rev. B | Page 21 of 36
DB0
08568-014
fVCO = 2 × fPFD × (INT)
(2)
ADRF6702
Data Sheet
REGISTER 1—MODULUS DIVIDE CONTROL
(DEFAULT: 0x003001)
REGISTER 2—FRACTIONAL DIVIDE CONTROL
(DEFAULT: 0x001802)
With Register 1, Bits[2:0] set to 001, the on-chip modulus
divide control register is programmed as shown in Figure 42.
With Register 2, Bits[2:0] set to 010, the on-chip fractional
divide control register is programmed as shown in Figure 43.
Modulus Value
Fractional Value
The modulus value is the preset fractional modulus ranging
from 1 to 2047.
The FRAC value is the preset fractional modulus ranging from
0 to <MDR.
RESERVED
MODULUS VALUE
DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13
0
0
0
0
0
0
0
0
0
MD10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
MD9
MD6
MD5
MD4
MD3
MD2
MD1
MD0
C3(0) C2(0) C1(1)
MD8
MD7
DB1
DB0
MD10
MD9
MD8
MD7
MD6
MD5
MD4
MD3
MD2
MD1
MD0
MODULUS VALUE
0
0
0
0
0
0
0
0
0
0
1
1
0
0
0
0
0
0
0
0
0
1
0
2
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
1
1
0
0
0
0
0
0
0
0
0
1536 (DEFAULT)
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
1
1
1
1
1
1
1
1
1
1
1
2047
08568-015
0
CONTROL BITS
DB12 DB11 DB10
Figure 42. Register 1—Modulus Divide Control Register Map
0
0
0
0
0
0
CONTROL BITS
FRACTIONAL VALUE
0
0
0
0
FD10
DB12 DB11 DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
FD9
FD6
FD5
FD4
FD3
FD2
FD1
FD0
FD8
FD7
DB2
DB1
DB0
C3(0) C2(1) C1(0)
FD10
FD9
FD8
FD7
FD6
FD5
FD4
FD3
FD2
FD1
FD0
FRACTIONAL VALUE
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
0
1
1
0
0
0
0
0
0
0
0
768 (DEFAULT)
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
...
<MDR
FRACTIONAL VALUE MUST BE LESS THAN MODULUS.
Figure 43. Register 2—Fractional Divide Control Register Map
Rev. B | Page 22 of 36
08568-016
RESERVED
DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13
Data Sheet
ADRF6702
REGISTER 3—Σ-Δ MODULATOR DITHER CONTROL
(DEFAULT: 0x10000B)
The default value of the dither magnitude (15) should be set to a
recommended value of 1.
With Register 3, Bits[2:0] set to 011, the on-chip Σ-Δ modulator
dither control register is programmed as shown in Figure 44. The
recommended and default setting for dither enable is enabled (1).
The dither restart value can be programmed from 0 to 217 − 1,
though a value of 1 is typically recommended.
DITHER
DITHER RESTART VALUE
CONTROL BITS
ENABLE
DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0
DEN
DV16 DV15 DV14 DV13 DV12 DV11 DV10 DV9 DV8 DV7 DV6 DV5 DV4 DV3 DV2 DV1 DV0 C3(0) C2(1) C1(1)
DITH1
0
0
DITH0
0
1
DITHER MAGNITUDE
15 (DEFAULT)
7
1
0
3
1
1
1 (RECOMMENDED)
DEN
0
1
DITHER ENABLE
DISABLE
ENABLE (DEFAULT, RECOMMENDED)
DV16 DV15 DV14 DV13
DV12
DV11
DV10
DV9
DV8 DV7
DV6 DV5
DV4
DV3 DV2 DV1 DV0
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
0
...
...
1
Figure 44. Register 3—Σ-Δ Modulator Dither Control Register Map
Rev. B | Page 23 of 36
0
...
...
1
0
...
...
1
1
...
...
1
DITHER RESTART
VALUE
0x00001 (DEFAULT)
...
...
0x1FFFF
08568-017
DB23
0
DITHER
MAGNITUDE
DB22
DB21
DITH1
DITH0
ADRF6702
Data Sheet
fractional spurs. The magnitude of the phase offset is determined by the following equation:
REGISTER 4—PLL CHARGE PUMP, PFD, AND
REFERENCE PATH CONTROL (DEFAULT:
0x0AA7E4)
With Register 4, Bits[2:0] set to 100, the on-chip charge pump,
PFD, and reference path control register is programmed as
shown in Figure 45.
CP Current
The nominal charge pump current can be set to 250 μA, 500 μA,
750 μA, or 1000 μA using DB10 and DB11 of Register 4 and by
setting DB18 to 0 (CP reference source).
In this mode, no external RSET is required. If DB18 is set to 1,
the four nominal charge pump currents (INOMINAL) can be
externally tweaked according to the following equation:
⎛ 217.4 × I CP
R SET = ⎜⎜
⎝ I NOMINAL
⎞
⎟ − 37.8 Ω
⎟
⎠
(3)
where ICP is the base charge pump current in microamps.
The PFD phase offset multiplier (θPFD,OFS), which is set by
Bits[16:12] of Register 4, causes the PLL to lock with a
nominally fixed phase offset between the PFD reference signal
and the divided-down VCO signal. This phase offset is used
to linearize the PFD-to-CP transfer function and can improve
ΔΦ (deg) = 22.5
θ PFD ,OFS
I CP , MULT
(4)
The default value of the phase offset multiplier (10 × 22.5°)
should be set to a recommended value of 6 × 22.5°.
This phase offset can be either positive or negative depending
on the value of DB17 in Register 4.
The reference frequency applied to the PFD can be manipulated
using the internal reference path source. The external reference
frequency applied can be internally scaled in frequency by 2×,
1×, 0.5×, or 0.25×. This allows a broader range of reference
frequency selections while keeping the reference frequency
applied to the PFD within an acceptable range.
The device also has a MUXOUT pin that can be programmed
to output a selection of several internal signals. The default
mode is to provide a lock-detect output to allow the user to
verify when the PLL has locked to the target frequency. In
addition, several other internal signals can be passed to the
MUXOUT pin as described in Figure 35.
Rev. B | Page 24 of 36
ADRF6702
REF OUPUT
MUX SELECT
DB23 DB22
CP
INPUT REF CURRENT
REF
PATH
SOURCE
DB21 DB20 DB19
RMS2 RMS1 RMS0 RS1
RS0
PFD PHASE OFFSET
MULTIPLIER
PFD
POL
CP
CURRENT
CP
SOURCE
Data Sheet
CP
CONTROL
PFD EDGE
DB8
DB18
DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9
DB7
DB6
DB5
CPM
CPBD CPB4 CPB3 CPB2 CPB1 CPB0 CPP1 CPP0 CPS CPC1 CPC0
PE1
PE0
PFD ANTIBACKLASH
DELAY
DB4
DB3
CONTROL BITS
DB2
DB1
DB0
PAB1 PAB0 C3(1)
C2(0)
C1(0)
PAB1 PAB0 PFD ANTIBACKLASH
DELAY
0
0
0ns (DEFAULT)
1
0
0.5ns
0
1
0.75ns
1
1
0.9ns
PE1
0
1
PE0
REFERENCE PATH EDGE
SENSITIVITY
0
1
FALLING EDGE
RISING EDGE (DEFAULT)
DIVIDER PATH EDGE
SENSITIVITY
FALLING EDGE
RISING EDGE (DEFAULT)
CPC1 CPC0 CHARGE PUMP CONTROL
0
0
1
1
0
1
0
1
BOTH ON
PUMP DOWN
PUMP UP
TRISTATE (DEFAULT)
CPS
CHARGE PUMP CONTROL SOURCE
0
1
CONTROL BASED ON STATE OF DB7/DB8 (CP CONTROL)
CONTROL FROM PFD (DEFAULT)
CPP1 CPP0 CHARGE PUMP CURRENT
0
0
1
1
0
1
0
1
250µA
500µA (DEFAULT)
750µA
1000µA
CPB4 CPB3 CPB2 CPB1 CPB0 PFD PHASE OFFSET MULTIPLIER
0
0
0
0
1
1
CPM
0
1
0
0
0
1
0
1
0
0
1
0
0
1
0
0
1
1
0
1
0
1
0
0
0
1
CPBD
PFD PHASE OFFSET POLARITY
0
1
NEGATIVE
POSITIVE (DEFAULT)
0 × 22.5°/ICP,MULT
1 × 22.5°/ICP,MULT
6 × 22.5°/ICP,MULT (RECOMMENDED)
10 × 22.5°/ICP,MULT (DEFAULT)
16 × 22.5°/ICP,MULT
31 × 22.5°/ICP,MULT
CHARGE PUMP CURRENT
REFERENCE SOURCE
INTERNAL (DEFAULT)
EXTERNAL
RS1
INPUT REF
RS0 PATH SOURCE
0
1
0
1
0
0
1
1
2× REFIN
REFIN (DEFAULT)
0.5× REFIN
0.25× REFIN
RMS2 RMS1 RMS0 REF OUTPUT MUX SELECT
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
LOCK DETECT (DEFAULT)
VPTAT
REFIN (BUFFERED)
0.5× REFIN (BUFFERED)
2× REFIN (BUFFERED)
TRISTATE
RESERVED
RESERVED
08568-018
0
0
0
0
1
1
1
1
Figure 45. Register 4—PLL Charge Pump, PFD, and Reference Path Control Register Map
Rev. B | Page 25 of 36
ADRF6702
Data Sheet
Table 10. LO Port Configuration1, 2
REGISTER 5—LO PATH AND MODULATOR
CONTROL (DEFAULT: 0x0000D5)
LON/LOP
Function
Input (2× LO)
Output (Disabled)
Output (1× LO)
Output (1× LO)
Output (1× LO)
Output (2× LO)
Output (2× LO)
Output (2× LO)
With Register 5, Bits[2:0] set to 101, the LO path and modulator
control register is programmed as shown in Figure 46.
The modulator output or the complete modulator can be
disabled using the modulator bias enable and modulator
output enable addresses of Register 5.
The LO port (LOP and LON pins) can be used to apply an
external 2× LO (that is, bypass internal PLL) to the IQ
modulator. A differential LO drive of 0 dBm is recommended.
The LO port can also be used as an output where a 2× LO or
1× LO can be brought out and used to drive another mixer.
The nominal output power provided at the LO port is 3 dBm.
The mode of operation of the LO port is determined by the
status of the LOSEL pin (3.3 V logic) along with the settings
in a number of internal registers (see Table 10).
1
2
LOSEL
0
0
0
1
1
0
1
1
Register 5,
Bit DB5
(LDIV)
X
X
0
0
0
1
1
1
Register 5,
Bit DB4
(LXL)
1
0
0
0
0
0
0
0
Register 5,
Bit DB3
(LDRV)
0
0
1
0
1
1
0
1
X = don’t care.
LOSEL should not be left floating.
The internal VCO of the device can also be bypassed. In this
case, the charge pump output drives an external VCO through
the loop filter. The loop is completed by routing the VCO into
the device through the LO port.
LO
OUTPUT
MOD
RF
LO
LO
CONTROL BITS
DRIVER
BIAS
OUTPUT
OUTPUT
IN/OUT
RESERVED
ENABLE ENABLE DIVIDER CONTROL ENABLE
DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
DB7
DB6
DB5
DB4
DB3
DB2 DB1 DB0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
MBE
RFEN
LDIV
LXL
LDRV
C3(1) C2(0) C1(1)
LO OUTPUT DRIVER
LDRV ENABLE
0
1
LXL
LO INPUT/OUTPUT CONTROL
0
1
LO OUTPUT (DEFAULT)
LO INPUT
LDIV
LO OUTPUT DIVIDE MODE
0
1
DIVIDE BY 1
DIVIDE BY 2 (DEFAULT)
RFEN
RF OUTPUT ENABLE
0
1
DISABLE
ENABLE (DEFAULT)
MBE
MOD BIAS ENABLE
0
1
DISABLE
ENABLE (DEFAULT)
Figure 46. Register 5—LO Path and Modulator Control Register Map
Rev. B | Page 26 of 36
DRIVER OFF (DEFAULT)
DRIVER ON
08568-019
0
Data Sheet
ADRF6702
REGISTER 6—VCO CONTROL AND VCO ENABLE
(DEFAULT: 0x1E2106)
REGISTER 7—EXTERNAL VCO ENABLE
With Register 7, Bits[2:0] set to 111, the external VCO control
register is programmed as shown in Figure 48.
With Register 6, Bits[2:0] set to 110, the VCO control and
enable register is programmed as shown in Figure 47.
The external VCO enable bit allows the use of an external VCO
in the PLL instead of the internal VCO. This can be advantageous
in cases where the internal VCO is not capable of providing the
desired frequency or where the internal VCO’s phase noise is
higher than desired. By setting this bit (DB22) to 1, and setting
Register 6, Bits[15:10] to 0, the internal VCO is disabled, and
the output of an external VCO can be fed into the part differentially on Pin 38 and Pin 37 (LOP and LON). Because the loop
filter is already external, the output of the loop filter simply
needs to be connected to the external VCO’s tuning voltage pin.
The VCO tuning band is normally selected automatically by the
band calibration algorithm, although the user can directly select
the VCO band using Register 6.
The VCO BS SRC bit (DB9) determines whether the result of
the calibration algorithm is used to select the VCO band or if
the band selected is based on the value in VCO band select
(DB8 to DB3).
The VCO amplitude can be controlled through Register 6. The
VCO amplitude setting can be controlled between 0 and 63. The
default value of 8 should be set to a recommended value of 63.
The internal VCOs can be disabled using Register 6.
The internal charge pump can be disabled through Register 6.
By default, the charge pump is enabled.
To turn off the PLL (for example, if the ADRF6702 is being
driven by an external LO), set Register 6, Bits[20:17] to zero.
CHARGE
3.3V
VCO
VCO LDO VCO
PUMP
LDO
ENABLE ENABLE ENABLE ENABLE SWITCH
DB23 DB22 DB21
0
0
0
DB20
CPEN
DB19
L3EN
DB18
LVEN
DISABLE
ENABLE (DEFAULT)
L3EN 3.3V LDO ENABLE
0
1
DISABLE
ENABLE (DEFAULT)
LVEN
VCO LDO ENABLE
0
1
DISABLE
ENABLE (DEFAULT)
VCO
BW SW
CTRL
VCO BAND SELECT FROM SPI
CONTROL BITS
DB17
DB16 DB15 DB14 DB13 DB12 DB11 DB10
DB9
DB8 DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0
VCO EN VCO SW VC5 VC4 VC3 VC2 VC1 VC0 VBSRC VBS5 VBS4 VBS3 VBS2 VBS1 VBS0 C3(1) C2(1) C1(0)
CPEN CHARGE PUMP ENABLE
VC[5:0] VCO AMPLITUDE
VBS[5:0]
VCO BAND SELECT FROM SPI
0x00
….
0x18
….
0x2B
….
0x3F
0x00
0x01
….
0x3F
DEFAULT 0x20
0
….
8 (DEFAULT)
….
43
….
63 (RECOMMENDED)
VCO SW
VCO SWITCH CONTROL FROM SPI
0
1
REGULAR (DEFAULT)
BAND CAL
VCO EN
VCO ENABLE
0
1
DISABLE
ENABLE (DEFAULT)
VBSRC VCO BW CAL AND SW SOURCE CONTROL
0
1
BAND CAL (DEFAULT)
SPI
Figure 47. Register 6—VCO Control and VCO Enable Register Map
EXTERNAL
VCO
RESERVED
ENABLE
DB23
DB22
DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
0
XVCO
0
0
0
0
0
0
0
0
0
0
0
0
0
0
CONTROL BITS
RES
XVCO
0
1
DB7 DB6 DB5
0
0
0
EXTERNAL VCO ENABLE
INTERNAL VCO (DEFAULT)
EXTERNAL VCO
DB4 DB3 DB2 DB1 DB0
0
0 C3(1) C2(1) C1(1)
08568-021
0
1
VCO AMPLITUDE
08568-020
RESERVED
Figure 48. Register 7—External VCO Enable Register Map
Rev. B | Page 27 of 36
ADRF6702
Data Sheet
CHARACTERIZATION SETUPS
Figure 49 and Figure 50 show characterization bench setups
used to characterize the ADRF6702. The setup shown in
Figure 49 was used to do most of the testing. An automated
VEE program was used to control equipment over the IEEE
bus. The setup was used to measure SSB, OIP2, OIP3, OP1dB,
LO, and USB NULL.
For phase noise and reference spurs measurements, see the
phase noise setup on Figure 50. Phase noise was measured on
LO and modulator output.
ADRF670x TEST RACK ASSEMBLY (INTERNAL VCO CONFIGURATION)
ALL INSTRUMENTS ARE CONNECTED IN DAISY CHAIN
FASHION VIA GBIP CABLE UNLESS OTHERWISE NOTED.
E3631A POWER SUPPLY
(+6V ADJUSTED TO 5V)
+5V FOR
VPOS TO 34950
MODULE
34401A DMM (FOR SUPPLY
CURRENT MEASUREMENT)
34980A
WITH 34950 AND (×2) 34921 MODULES
PROGRAMMING
AND DC CABLE
(×6 FOR MULTISITE)
INPUT
(RFOUT)
AGILENT E4440A PSA SPECTRUM ANALYZER
10-PIN CONNECTOR
DC HEADER
9-PIN DSUB
CONNECTOR
(REGISTER
PROGRAMMING)
REF IN
6dB
KEITHLEY S46 SWITCH SYSTEM #1
(FOR RFOUT AND REFIN ON 6 SITES)
OUTPUT (REF)
RF OUT
KEITHLEY S46 SWITCH SYSTEM #2
(FOR BASEBAND INPUTS ON 6 SITES)
ADRF6702
EVAL BOARD
6dB
ROHDE AND SCHWARTZ SMT 06 SIGNAL GENERATOR
(REFIN)
BASEBAND INPUTS AT 1MHz
AEROFLEX IFR 3416 FREQUENCY GENERATOR
(WITH BASEBAND OUTPUTS AT 1MHz)
PC CONTROL
CONNECTED TO SYSTEM VIA USB TO GPIB ADAPTER
Figure 49. General Characterization Setup
Rev. B | Page 28 of 36
08568-043
BASEBAND OUTPUTS
(IN, IP, QN, QP)
Data Sheet
ADRF6702
ADRF670x PHASE NOISE STAND SETUP
ALL INSTRUMENTS ARE CONNECTED IN DAISY CHAIN FASHION
VIA GBIP CABLE UNLESS OTHERWISE NOTED.
ROHDE AND SCHWARTZ
SMA 100 SIGNAL GENERATOR
REFIN
AGILENT E5052 SIGNAL SOURCE
ANALYZER
AGILENT E4440A SPECTRUM
ANALYZER
IF OUT
KEITHLEY S46 SWITCH SYSTEM 2
(FOR IF OUT AND REFIN ON 6 SITES)
REFIN
LO OUT
BASEBAND INPUTS
(IP, IN, QP, QN)
IFR 3416 SIGNAL GENERATOR
(BASEBAND SOURCE)
KEITHLEY S46 SWITCH SYSTEM 1
(FOR BASEBAND INPUTS ON 6 SITES)
10 PIN CONNECTOR
(DC MEASUREMENT, +5V POS)
AND 9 PIN DSUB
CONNECTOR (VCO AND PLL
PROGRAMMING)
ADRF6702
EVAL BOARD
34980A MULTIFUNCTION SWITCH
(WITH 34950 AND 34921 MODULES)
INPUT DC
AGILENT 34401A DMM
(IN DC I MODE, SUPPLY CURRENT
MEASUREMENT)
PC CONTROL
CONNECTED TO SYSTEM VIA USB TO GPIB ADAPTER
Figure 50. Characterization Setup for Phase Noise and Reference Spur Measurements
Rev. B | Page 29 of 36
08568-044
AGILENT E3631A POWER
SUPPLY
ADRF6702
Data Sheet
EVALUATION BOARD
Figure 52 shows the schematic of the device’s RoHS-compliant
evaluation board. This board was designed using Rogers 4350
material to minimize losses at high frequencies. FR4 material
would also be adequate but with the slightly higher trace loss
of this material.
To operate correctly under Windows XP, Version 3.5 of
Microsoft .NET must be installed. To run the software on a
Windows 7 PC, XP emulation mode must be used (using
Virtual PC).
Whereas the on-board USB interface circuitry of the evaluation
board is powered directly from the PC, the main section of the
evaluation board requires a separate 5 V power supply.
08568-135
The evaluation board is designed to operate using the internal
VCO (default configuration) of the device or with an external
VCO. To use an external VCO, R62 and R12 should be
removed. 0 Ω resistors should be placed in R63 and R11. A
side-launched SMA connector (Johnson 142-0701-851) must
be soldered to the pad labeled VTUNE. The input of the
external VCO should be connected to the VTUNE SMA
connector and a portion of the VCO’s output should be
connected to the EXT LO SMA connector. In addition to
these hardware changes, internal register settings must also
be changed (as detailed in the Register Description section)
to enable operation with an external VCO.
Additional configuration options for the evaluation board are
described in Table 11.
The serial port of the ADRF6702 can be programmed from a
PC’s USB port (a USB cable is provided with the evaluation
board). The on-board USB interface circuitry can if desired be
bypassed by removing the 0 Ω resistors, R15, R17, and R18 (see
Figure 52) and driving the ADRF6702 serial interface through
the P3 4-pin header (P3 must be first installed, Samtec TSW104-08-G-S).
EVALUATION BOARD CONTROL SOFTWARE
USB-based programming software is available to download
from the ADRF6702 product page at www.analog.com
(Evaluation Board Software Rev 6.1.0). To install the software,
download and extract the zip file. Then run the following
installation file: ADRF6X0X_6p1p0_customer_installer.exe.
Figure 51. Control Software Opening Menu
Figure 51 shows the opening window of the software where the
user selects the device being programmed. Figure 55 shows a
screen shot of the control software’s main controls with the
default settings displayed. The text box in the bottom left corner
provides an immediate indication of whether the software is
successfully communicating with the evaluation board. If the
evaluation board is connected to the PC via the USB cable
provided and the software is successfully communicating with
the on-board USB circuitry, this text box shows the following
message: ADRF6X0X eval board connected.
Rev. B | Page 30 of 36
Data Sheet
ADRF6702
VCC
R43
10kΩ
(0402)
VDD
C27
0.1µF
(0402)
C25
0.1µF
(0402)
C23
0.1µF
(0402)
C20
0.1µF
(0402)
C19
0.1µF
(0402)
C9
0.1µF
(0402)
C8
100pF
(0402)
C26
100pF
(0402)
C24
100pF
(0402)
C22
100pF
(0402)
C21
100pF
(0402)
C18
100pF
(0402)
C10
100pF
(0402)
VDD
34
VDD
VDD
27
29
VDD
22
VDD
17
LE (USB)
DATA (USB)
CLK (USB)
VDD
10
1
16
13
12
14
9
5
1
4
3
C6
100pF LOP
(0402)
37
BUFFER
MABA-007159 C5
100pF
(0402)
REF_IN
REFIN
R73
49.9Ω
(0402)
SEE TEXT
REFOUT OPEN
R16
OPEN
(0402)
FRACTION
REG
ADRF6702
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
×2
6
÷2
MUX
÷4
MUXOUT
MODULUS
TEMP
SENSOR
7
2
2:1
MUX
INTEGER
REG
18
N COUNTER
21 TO 123
R37
0Ω
(0402)
CP
TEST
POINT
(OPEN)
R38
OPEN
(0402)
C14
22pF
(0603)
R2
OPEN
(0402)
RSET
C13
6.8pF
(0603)
C16
100pF
(0402)
C17
0.1µF
(0402)
C42
10µF
(0603)
DECL1
C12
100pF
(0402)
C11
0.1µF
(0402)
C41
OPEN
(0603)
40
VTUNE
26
DECL3
R62
0Ω
(0402)
C2
OPEN
(0402)
C40
22pF
(0603)
33
QP
QN
R23
OPEN
(0402)
IN
IP
R3
OPEN
(0402)
QP
QN
IN
IP
RFOUT
OPEN
VTUNE
OPEN
C3
100pF
(0402)
RFOUT
R63
OPEN
(0402)
R12
0Ω
(0402)
R11
OPEN
(0402)
C43
10µF
(0603)
39
CP
R9 10kΩ R65 10kΩ
(0402)
(0402)
R10
3kΩ
(0603)
C15
2.7nF
(1206)
19
32
3
5
24
NC
GND
÷2
0/90
CHARGE PUM P
250µA,
500µA (DEFAULT),
750µA,
1000µA
–
PHASE
+ FREQUENCY
DETECTOR
11 15 20 21 23 25 28 30 31 35
VCO
CORE
PRESCALER
÷2
8
4
SPI
INTERFACE
DIVIDER
÷2
BUFFER
38
T3
C29
100pF
(0402)
DECL2
36
C1
100pF
(0402)
08568-027
LOSEL
LON
EXT LO
C7
0.1µF
(0402)
LE
S1
R47
10kΩ
(0402)
DATA
R39
10kΩ
(0402)
R20
0Ω
(0402)
CLK
C28
10µF
(3216)
VCC
R40
10kΩ
(0402)
S2
ENOP
VCC
RED
+5V
NOTES
1. NC = NO CONNECT. DO NOT CONNECT TO THIS PIN.
08568-047
08568-048
Figure 52. Evaluation Board Schematic (Loop Filter Set to 130 kHz)
Figure 54. Evaluation Board Bottom Layer
Figure 53. Evaluation Board Top Layer
Rev. B | Page 31 of 36
ADRF6702
Data Sheet
Table 11. Evaluation Board Configuration Options
Component
S1, R39, R40
EXT LO, T3
REFIN SMA Connector, R73
REFOUT SMA Connector, R16
CP Test Point, R38
C13, C14, C15, C40R9, R10, R37, R65
R11, R12, R62, R63, VTUNE SMA
Connector
R2
R23, R3
P3 4-Pin Header, R15, R17, R18
Description
LO select. Switch and resistors to ground LOSEL pin. The LOSEL
pin setting in combination with internal register settings,
determines whether the LOP/LON pins function as inputs or
outputs. With the LOSEL pin grounded, register settings can
set the LOP/LON pins to be inputs or outputs.
LO input/output. An external 1× LO or 2× LO can be applied to
this single-ended input connector. Alternatively, the internal
1× or 2× LO can be brought out on this pin. The differential LO
signal on LOP and LON is converted to a single-ended signal
using a broadband 1:1 balun (Macom MABA-007159, 4.5 MHz
to 3000 MHz frequency range). The balun footprint on the
evaluation board is also designed to accommodate Johanson
baluns: 3600BL14M050 (1:1, 3.3 GHz to 3.9 GHz) and
3700BL15B050E (1:1, 3.4 GHz to 4 GHz).
Reference input. The input reference frequency for the PLL is
applied to this connector. Input resistance is set by R73 (49.9 Ω).
Multiplexer output. The REFOUT connector connects directly
to the device’s MUXOUT pin. The on-board multiplexer can
be programmed to bring out the following signals:
REFIN, 2× REFIN, REFIN/2, REFIN/4,
Temperature sensor output voltage (VPTAT),
Lock detect indicator.
Charge pump test point. The unfiltered charge pump signal
can be probed at this test point. Note that this pin should not
be probed during critical measurements such as phase noise.
Loop filter. Loop filter components.
Internal vs. external VCO. When the internal VCO is enabled,
the loop filter components connect directly to the VTUNE pin
(Pin 39) by installing a 0 Ω resistor in R62. In addition, the loop
filter components should be returned to Pin 40 (DECL3) by
installing a 0 Ω resistor in R12.
To use an external VCO, R62 should be left open. A 0 Ω resistor
should be installed in R63, and the voltage input of the VCO
should be connected to the VTUNE SMA connector. The output
of the VCO is brought back into the PLL via the LO IN/OUT SMA
connector. In addition, the loop filter components should be
returned to ground by installing a 0 Ω resistor in R11.
Loop filter return.
RSET. This pin is unused and should be left open.
Baseband input termination. Termination resistors for the
baseband filter of the DAC can be placed on R23 and R3. In
addition to terminating the baseband filters, these resistors
also scale down the baseband voltage from the DAC without
changing the bias level. These resistors are generally set in the
100 Ω to 300 Ω range.
USB circuitry bypass. The USB circuitry can be bypassed,
allowing for the serial port of the ADRF6702 to be driven
directly. P3 (Samtec TSW-104-08-G-S) must be installed, and
0 Ω resistors (R15, R17 and R18) must be removed.
Rev. B | Page 32 of 36
Default Condition/Option
Settings
T3 = Macom MABA-007159
EXT LO SMA connector = installed
FREFIN = 153.6 MHz
R73 = 49.9 Ω
REFOUT SMA connector = open
R16 = open
CP = open
R38 = open
See Table 8
R12 = 0 Ω (0402)
R11 = open (0402)
R62 = 0 Ω (0402)
R63 = open (0402)
VTUNE = open
R2 = open (0402)
R3 = R23 = open (0402)
P3 = open
R15, R17, R18 = 0 Ω (0402)
ADRF6702
08568-136
Data Sheet
Figure 55. Main Controls of the Evaluation Board Control Software
Rev. B | Page 33 of 36
Data Sheet
08568-028
ADRF6702
Figure 56. USB Interface Circuitry on the Customer Evaluation Board
Rev. B | Page 34 of 36
Data Sheet
ADRF6702
OUTLINE DIMENSIONS
6.00
BSC SQ
0.60 MAX
0.60 MAX
TOP
VIEW
0.50
BSC
5.75
BSC SQ
0.50
0.40
0.30
12° MAX
0.80 MAX
0.65 TYP
0.30
0.23
0.18
1
4.25
4.10 SQ
3.95
EXPOSED
PAD
(BOT TOM VIEW)
21
20
11
10
0.25 MIN
4.50
REF
0.05 MAX
0.02 NOM
SEATING
PLANE
40
0.20 REF
COPLANARITY
0.08
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-VJJD-2
072108-A
PIN 1
INDICATOR
1.00
0.85
0.80
PIN 1
INDICATOR
31
30
Figure 57. 40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
6 mm × 6 mm Body, Very Thin Quad
(CP-40-1)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1
ADRF6702ACPZ-R7
ADRF6702-EVALZ
1
Temperature Range (°C)
−40°C to +85°C
Package Description
40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
Evaluation Board
Z = RoHS Compliant Part.
Rev. B | Page 35 of 36
Package Option
CP-40-1
ADRF6702
Data Sheet
NOTES
©2011 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D08568-0-10/11(B)
Rev. B | Page 36 of 36