A Scalable, 2.9 mW, 1 Mb/s e-Textiles Body Area Network
Transceiver with Remotely-Powered Nodes and BiDirectional Data Communication
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Citation
Desai, Nachiket, Jerald Yoo, and Anantha P. Chandrakasan. “A
Scalable, 2.9 mW, 1 Mb/s E-Textiles Body Area Network
Transceiver With Remotely-Powered Nodes and Bi-Directional
Data Communication.” IEEE Journal of Solid-State Circuits 49,
no. 9 (September 2014): 1995–2004.
As Published
http://dx.doi.org/10.1109/JSSC.2014.2328343
Publisher
Institute of Electrical and Electronics Engineers (IEEE)
Version
Author's final manuscript
Accessed
Wed May 25 22:46:33 EDT 2016
Citable Link
http://hdl.handle.net/1721.1/98890
Terms of Use
Creative Commons Attribution-Noncommercial-Share Alike
Detailed Terms
http://creativecommons.org/licenses/by-nc-sa/4.0/
1
A Scalable, 2.9 mW, 1 Mb/s e-Textiles Body Area
Network Transceiver with Remotely-Powered Nodes
and Bi-Directional Data Communication
Nachiket Desai, Student Member, IEEE, Jerald Yoo, Member, IEEE, and Anantha P. Chandrakasan, Fellow, IEEE
Abstract—This paper presents transceivers and a wireless
power delivery system for a Body-Area Network (BAN) that
uses an e-textiles-based physical layer (PHY) capable of linking
a diverse set of sensor nodes monitoring vital signs on the
user’s body. A central base station in the network controls
power delivery and communication resource allotment for every
node using a general-purpose on-chip Node Network Interface
(NNI). The architecture of the network ensures fault-tolerance,
reconfigurability and ease of use through a dual wireless-wireline
topology. The nodes are powered at a peak end-to-end efficiency
of 1.2% and can transmit measured data at a peak rate of 1 Mb/s.
Modulation schemes for communication in both directions have
been chosen and a Medium Access and Control (MAC) protocol
has been designed and implemented on chip to reduce complexity
at the power-constrained nodes, and move it to the base station.
While transferring power to a single node at maximum efficiency,
the base station consumes 2.9 mW power and the node recovers
34 µW, of which 14 µW is used to power the network interface
circuits while the rest can be used to power signal acquisition
circuitry. Fabricated in 0.18 µm CMOS technology, the base
station and the NNI occupy 2.95 mm2 and 1.46 mm2 area
respectively.
Index Terms—Body-area networks, e-textiles, continuous
health monitoring, wireless power delivery, inductive links, integrated medium access protocol
B
I. I NTRODUCTION
ODY-area networks (BANs) allow continuous monitoring of vital signs by linking small, low-power sensors
implanted in and placed on the surface of the body and
transmitting the data they measure to healthcare providers
[1]. Current state-of-the-art biomedical front-ends [2]–[4] can
measure signals using only microwatts of power, making
it possible to aggressively minimize their volume and the
amount of energy stored locally. In order to realize their
potential afforded by these advantageous traits, it makes sense
to transfer the data using a two step process. Uncompressed
data is first sent to a local base station (like a cellphone) using
a link requiring minimal transmission effort by the sensor
node. The local base station then processes the signals and
Manuscript received August 27, 2013; revised January 16, 2014. This
work was funded by the cooperative agreement between the Masdar
Institute of Science and Technology, Abu Dhabi, UAE and the Massachusetts Institute of Technology, Cambridge, MA, USA, Reference No.
196F/002/707/102f/70/9374.
N. Desai and A. P. Chandrakasan are with the Department of Electrical
Engineering and Computer Science, Massachusetts Institute of Technology,
Cambridge, MA, 02139 USA e-mail: ndesai@mit.edu tel: 617-253-0164.
J. Yoo is with the Department of Electrical Engineering and Computer
Science, Masdar Institute of Science and Technology, Abu Dhabi, UAE.
sends them on to a healthcare provider using conventional
data networks.
Wireless communication devices forming a BAN typically
operate in the unlicensed bands, and have to contend with the
high losses in those bands in the operating environment [5],
[6]. Another scheme for communication with the local base
station is the use of body-coupled communication (BCC) [7]–
[9]. BCC transceivers avoid a number of multiple-access problems when a large number people operate BANs in close proximity. However, the path loss in BCC channels is comparable
to wireless channels. In contrast, textiles with electric routing
(e-textiles) printed on top of the cloth or woven in as part of
the yarn offer much lower path loss, and have long been a topic
of research with the aim of building wearable computers [10].
They allow remote powering of the microwatt sensors and
their associated telemetry circuits by a network base station
[11]–[13]. Building on these advantages, the proposed BAN
improves efficiency by using resonant power transfer between
coupled inductors and combines this with electric routing on
clothing to afford better aesthetics and greater comfort to the
user by avoiding wires going all the way to sensors adhered
to the body. This allows for cheap, disposable sensors that
do not need an integrated battery or radio and can be easily
attached to and removed from the user’s body depending on
his/her specific requirements. The system architecture and the
circuits used for resonant power transmission and recovery
are presented in Sections II and III respectively. Modulation
schemes for communication to and from the nodes have been
chosen to lower the power requirements at the nodes and push
complexity to the base station. These modulation schemes and
the circuits implementing them are presented in Section IV.
Medium Access and Control (MAC) protocols used in
commercially-deployed standards for wireless BANs, such
as Bluetooth LE and IEEE 802.15.4/Zigbee [14] include
significant protocol overhead at the sensor nodes and are not
compatible with microwatt-sensors. Allowing sensors to sleep
and requiring them to wake up after a preset amount of time
[15] to check if the channel is unoccupied requires an accurate,
power-hungry clock at the sensor that is not power-gated. A
MAC protocol for e-textile based BANs that relies on TimeDivision Multiple Access (TDMA) for resource allotment has
been presented in [11]. A custom MAC protocol that moves
all decision-making to the base station to reduce power consumption at the sensor nodes has been proposed and integrated
on chip. The custom MAC protocol also allows sensors to
be added and removed easily in order to keep the network
2
Inductors forming links with
sensor nodes
32 mm
Electric routing on textiles
Base Station
Fig. 1. An e-textiles shirt implemented with screen-printed spiral inductors
linking to sensor nodes (worn inside out for demonstration). The base station’s
size is limited by test and debug structures on the board.
reconfigurable and supports TDMA and Contention Access
(CA) modes for sensors with wide variation in channel access
and bandwidth requirements. Details of the implemented MAC
are presented in Section V.
II. S YSTEM A RCHITECTURE
A. e-Textiles Network
Figure 1 shows an example of the proposed network implemented on a shirt with the user holding the local base
station, whose size is limited by the test and debug structures
on the board. Spiral inductors and their associated routing that
connect them to a central base station have been screen-printed
on fabric using the process described in [16]. Restricting the
circuits on textiles to screen-printed passives avoids expensive
procedures to bond silicon dies to the routing on fabric and
keeps the costs of the textiles low. These inductors form nearfield magnetic links with similar inductors on the sensor nodes,
as shown in Figure 2. These nodes are of a band-aid patch form
factor and are worn on the body. The local base station can be
integrated in a cellphone linked to the shirt with a snap-button
interface or a secondary inductive link (with additional losses),
or onto a clip-on bluetooth-like device linked to the user’s
cellphone. This hybrid wireless/wired architecture achieves
better power efficiency compared to fully wireless- or BCCbased designs while being more comfortable to use than fully
wired systems, such as the one presented in [11].
To maximize coverage and allow different varieties of
sensors to operate in the same network, a large number of inductors must be placed on the e-textiles clothing. Additionally,
e-textiles need to be designed keeping in mind that clothes are
subject to harsher environments than most electronics used in
daily life, for example inside a washing machine. In order to
prevent network faults from arising due to a break (“opens”),
redundant paths must be included in the network routing.
Inductors are grouped to form sub-networks that can be turned
on/off independently, as shown in Figure 3. Inductors in every
sub-network are connected in parallel and the sub-network as
a whole has multiple independent paths connecting it to the
base station. This arrangement allows for a trade-off between
routing complexity when each sub-network contains only one
Fig. 2. Cross-section of inductive link between e-textiles and sensor node.
inductor and high power consumption when all inductors are
in the same sub-network. Also, by not adhering to a strict
arrangement like the (X,Y) grid in [12] or the linked list
in [17], the proposed network architecture allows for greater
freedom in covering sensors around the body based on userspecific sensor locations. An obvious arrangement would be
to form sub-networks of inductors covering one part of the
body. The only limitations would be the number of subnetworks that can be supported by the base station and the
combined inductance of the parallelly-connected coils in each
sub-network. For example, the shirt in Figure 1 can support
2-channel ECG while a cap made of similar material can
function as an EEG helmet.
Power is transferred to the sensor nodes by resonant coupling between inductors at the base station and the sensor
nodes. The network has a star topology, with all sensor nodes
talking directly to the base station. Modulation schemes for
data transmitted in both directions have been chosen to push
most of the modulation and demodulation effort away from
the sensor nodes towards the base station.
B. Base Station
A block diagram of the implemented base station for the
network-on-shirt of Figure 1 is shown in Figure 4. It can
support up to two sub-networks each containing up to two
inductors. The base station has a separate analog interface
for each of the two sub-networks it can support. Both subnetworks share the same digital back-end. The demodulated
bitstreams from both the sub-networks are combined asynchronously in the digital domain and sent on to the digital
back-end that implements the on-chip MAC and manages
power delivery to the sensor nodes in the network. A regulator
with four discrete output levels controls the amount of power
transmitted to each sub-network. A digital on/off controller
switches off a sub-network if it is devoid of active sensor
nodes and doubles up as an on-off keying (OOK) modulator
for downlink transmission. The analog interface to each subnetwork consists of the front-end oscillators, power amplifiers
and LC-tank filters for power transmission, and modulators
and demodulators for data communication.
3
SN0,0
SN0,1
SN0,P
NNI
NNI
NNI
Downlink
Sensor
Clothing
Diverse Set of
On-Body SNs
SNN,0
SNN,Q
NNI
NNI
Inductor SubNetworks on
Clothing
Uplink
Power
Body-Area Network on Clothing
Data
(with wireline interface to Base Station)
Fig. 3. Architecture of proposed BAN. Each sub-network can power and talk to a heterogeneous group of sensors. The NNIs can access a sub-network
through any one or more of its constituent inductors.
Sub-Network Analog
Interfaces
CRC
Checker
Power
Mgmt.
PWM
Network
SN List
Serial
Interface
MAC
Asynchronous
Digital Multiplexer
MAC
Command
Interpreter
Digital
Demodulator
ASK Demodulation
ASK Demodulation
Inductor
SubNetworks
VDD
AutoCommutative
Rectifier
4-Level Power
4-Level Power
Controller
Controller
PA
PA
PA
PA
PA
PA
Fabric
Inductor
Patch
VDD
Packetizer
&
Modulator
Sensor nodes are partitioned into an application-specific
biomedical data acquisition module and a general-purpose
NNI, as shown in Figure 5. The on-chip NNI module supplies
power to the sensor readout circuitry and transmits the data
measured by it. The off-chip readout circuits consist primarily
of a front-end amplifier followed by an ADC and can be any
state-of-the-art microwatt-level sensor that measures signals on
the human body [2]–[4]. An auto-commutative rectifier (ACR)
minimizes the dead zone for rectification at low-amplitude ac
inputs, and converts ac power transmitted by the base station
to dc for use by all circuits at the sensor node. A low power,
fully digital demodulator passes on instructions sent by the
base station to the digital back-end for interpretation. These
instructions directly manage the data collected by the sensor
front-end and control how and when it is packetized and
transmitted to the base station.
III. R ESONANT P OWER D ELIVERY I NTERFACE
A. Theoretical Overview
An RLC circuit model for power transfer across the inductive link is shown in Figure 6. The resonant near-field power
transfer scheme implemented is similar to the one described in
[18], except that a parallel LC-tank is employed on the primary
(base station) side. The resistor RL in Figure 6 models the ac
input resistance of the rectifier that supplies dc power to the
sensor node.
VDD
AFE
ADC
Application-Specific
Readout Circuitry
Node Network Interface
(on-chip)
Fig. 4. Block diagram of base station.
C. Sensor Nodes
CSTOR
Fig. 5. Block diagram of a typical sensor node implementation. Only the
Node Network Interface has been implemented on chip, while the applicationdependent sensing and data conversion circuits are off-chip.
I1
+
Rs
I2
+
+
R2
R1
k
L1
C1
Vs
V1
-
.
L2
Z1
sMI2
-
.
Z2
+
−
+
−
C2
RL
V2
sMI1
-
Fig. 6. Power transfer through inductively coupled resonators. The
√ mutual
inductance M and the coupling coefficient k are related as M = k L1 L2 .
The fabric inductors used have low quality factors (Q ≈
6 − 8) in the MHz range, hence the overall loaded Q of
the secondary resonator is dominated by the inductor losses
and not the load1 . Also, the coupling coefficient across the
inductive link, k, is approximately 0.1 for a separation as low
as 5 mm between the primary and secondary inductors. In
this regime of operation, the total efficiency of power transfer
1 Assuming the rectifier supplying dc power to the sensor node consumes
as much as 90 µA (p-p) current at 2.4 V (p-p) to supply 30 µA at 1.8 V dc,
this yields RL =26.7 kΩ and QL ≈ 100.
4
across the inductive link can be approximated as
2
Q2
k Q1 Q2
·
η≈
1 + k 2 Q1 Q2 QL
Class-D PA
(1)
where QL , the quality factor of load, is defined as QL ,
ω0 C2 RL and Q1 and Q2 are the intrinsic quality factors of
the primary and secondary inductors respectively. Equation (1)
can be obtained from the expressions derived in [19].
The voltage transfer function from the primary to the
secondary can be approximated as
r
√
V2 (jω0 ) ≈ k Q1 Q2 · L2
(2)
Vs (jω0 ) 1 + k 2 Q1 Q2
L1
Increasing the coupling coefficient and the Q of the resonators
improves the efficiency of power transfer and the magnitude
of the voltage transfer function, as shown by Equations (1)
and (2). From the expressions for inductance in [16], a 6
turn spiral inductor with outer diameter of 30 mm has 1.6
µH inductance. This agrees with the measured values of 1.4
µH inductance. The measured self-resonant frequency of the
inductors is 80 MHz, and the 27.12 MHz ISM band was
chosen for operation since it is sufficiently below the selfresonant frequency for the inductors to remain dominantly
inductive and the on-chip capacitors required are around 20
pF. These values lead to a link (ac-ac) efficiency of 1.7% and
voltage transfer function 0.39 at the lowest power setting of
the transmitter.2
B. Combined Oscillator-PA Transmitter
The circuit used for resonant power transfer is shown in
Figure 7. The oscillator uses a modified Colpitts’ feedback
structure, which also doubles as the output filter. The filter
is made of inductors on fabric and on-chip MIM capacitors.
A parallel LC tank is used to avoid the large voltage stress
that would arise from resonating on-chip capacitors in a series
LC configuration. A class-D power amplifier is integrated into
the oscillator feedback loop. Output power control is achieved
by reducing the active device widths of the transistors in the
PA to make them function as linear regulators in the lower
power settings. This avoids both requiring two separate tuned
networks in the MHz range and tuning those two networks to
resonate at the same frequency. The on-chip capacitors employ
two-level tuning in order to maintain operation at the desired
frequency. Coarse tuning selects the number of capacitors to
be connected in the network depending on the number of
inductors connected in parallel in the sub-network. Fine tuning
adjusts for variations in the inductance values of the fabric
inductors; a one-time calibration step can tune out up to 20%
static variation using four bits, which is sufficient based on
inductance measurements made on 24 samples.
A small-signal model of the oscillator is shown in Figure 8.
The resistance Ro is the small-signal output resistance of the
class-D PA. The loop transfer function for the oscillator has
three poles and no zeros, making the system oscillate above
a minimum dc loop gain. This dc loop gain is provided by
2 The value of R for the highest and lowest power settings of the transmitter
s
are approximately 140 Ω and 480 Ω respectively.
Vref
StaticConfigured
Cap. Banks
+
Lfabric
C
C1res
C
C2res
Fig. 7. Schematic of self-tuned combined oscillator-PA architecture.
Vo
L
gmVf
Ro
C1
Vf
C2
Fig. 8. Small signal model of oscillator with modified Colpitts’ feedback
network.
the small-signal amplification of the class-D PA, which is dcbiased around mid-rail by negative feedback. The small-signal
model is obtained by linearizing the circuit at the dc bias point.
−1
p
The oscillator oscillates at f0 = 2π LCeq
, where
Ceq = C1 ||C2 . For a desired resonant frequency and given
inductance value, the area of the on-chip capacitors is minimized when C1 = C2 . This also leads to Vo and Vf oscillating
between 0 and VDD in opposite phase, maximizing the voltage
across the fabric inductor.
C. Auto-Commutative Rectifier
AC power transmitted by the base station is rectified at the
sensor nodes by using a self-synchronous topology similar to
the one presented in [20], [21], which trades off efficiency
for the capability to generate workable dc voltages from
ac input amplitudes small enough to limit the transistors’
performance by their threshold voltages. A schematic of the
auto-commutative rectifier (ACR) used is shown in Figure 9.
The gates of all transistors are biased such that the devices
conduct strongly in the desired half-cycle and are strongly
turned off in the other. The use of additional comparators and
gate-drive circuits is avoided in order to extract net positive
power in the µW region of operation. Additionally, no charge
is stored on capacitors, which might be leaky or expensive to
integrate, in order to overcome the threshold voltages of the
transistors.
Simulation waveforms for the operation of the ACR are
shown in Figure 10. The transistors Mp1 (or Mp2 ) and Mn1
(or Mn2 ) in Figure 9 are supposed to conduct when VACp (or
VACn ) is higher than VDC+ and VACn (or VACp ) is lower than
VDC− . Reverse conduction is possible during the initial phase
5
ACp
DC Output Voltage (V)
Mn2
1
Mp1
DC-
DC+
Mn1
Mp2
ACn
Auto−Commutative Recitifer
Diode−Connected Rectifier
0.8
0.6
0.4
0.2
0
0.2
0.4
Fig. 9. Schematic of auto-commutative rectifier (ACR).
600
400
1.2
V
AC,p
V
Cc
DC+
Cc
Cc
200
100
0
−100
I
Mp1
IMn1
−300
Fig. 10. Simulation waveforms for operation of ACR.
of each half cycle and is kept lower than the forward current
by the threshold voltages of the transistors. The net dc current
supplied to the load in Figure 10 is 11 µA.
The transistors making up the ACR in Figure 9 need to
be sized to carry currents much larger than the output load
current supplied, as shown in Figure 10. However, compared
to a conventional rectifier where the same transistors are diodeconnected, the ACR supplies 2x higher output voltage than
the conventional diode-connected rectifier in the low-input
amplitude regime of operation. This is shown in Figure 11.
To generate the required dc output voltage, the four rectifiers
in Figure 9 are connected in series in their dc path while their
ac inputs are capacitively-coupled to the LC tank as shown in
Figure 12, thus avoiding the need for additional voltage boost
converters.
IV. B I -D IRECTIONAL DATA C OMMUNICATION TRX
The capability to transmit data in both directions is necessary for the star network topology with the base station acting
as the hub. The requirements for the links in both directions
are highly asymmetric: the downlink from the base station to
the sensor nodes needs to carry only small, low-duty cycle
control instructions. On the other hand, the uplink from the
sensor nodes to the base station needs to carry potentially
large amounts of uncompressed data. An added constraint is
the limited energy budget at the sensor node and the need to
push complexity to the base station.
Various schemes for transferring data to inductively-coupled
sensors have been proposed in [22]–[25], but they use mod-
Lpatch
-
Vac,n
+
Cs
-
Vac,n
Cc
+
Cs
-
Vac,n
Cc
Vac,p
+
Cs
ACR
300
Vac,p
Vac,p
ACR
Vac,p
ACR
0
Cres
Current (µA)
1
Fig. 11. Comparison of simulated output voltages generated by ACR and
diode-connected rectifier supplying dc power to a 25 kΩ load.
ACR
Voltage (mV)
800
0.6
0.8
Input AC Amplitude (V)
-
+
Cstor
Vac,n
Cc
Fig. 12. Schematic of rectifiers connected in series to generate required dc
voltage at sensor node. The schematic for the ACR block is shown in Figure 9.
ulation schemes such as FSK and OQPSK, which are more
complex than OOK-based schemes, or use multiple inductors,
or are data-only links. Since the sensor nodes have to be of
a small form factor the data link required for e-textile BANs
must use the same inductors for power and data transfer in both
directions. A pulse-width modulated (PWM) OOK scheme is
used on the downlink that can be demodulated asynchronously,
eliminating the need for clock recovery circuits at the sensor
node. On the uplink, impedance modulation is used which
reduces the modulation circuitry to a switch across the inductor
at the sensor node.
A. Downlink Communication
Binary coded messages to be transmitted to the sensor nodes
are pulse-width coded digitally using LOW and HIGH times
in a 25:75 ratio for transmitting a “1” and a 75:25 ratio for
transmitting a “0”. This pulse-width-encoded signal is used
to control a transmit on/off switch which generates the PWMOOK waveform shown in Figure 13. In order to accommodate
the startup/shutdown times of the oscillator, the data rate
chosen is 80 kb/s, much lower than the local clock frequencies
at the base station and sensor node.
Demodulation at the sensor node is done digitally using only
the local 1 MHz clock. Two incrementers count the number
of clock cycles for which the received waveform is 0 and
1, as shown in Figure 14. Since each PWM bit ends with a
negative edge, the values of the counters can be compared
against each other at the negative edge and the received bit
can be latched. Fully digital demodulation avoids static power
6
PWMEnv
CLK
N1
x
N0
x
0
1
2
3
0
1
1
2
1
3
0
1
2
TPWM = 12!s
4
1
4
D
Q
N1
PWMEnv
D Q
R
1
R
>
<
MP%
4
4
D
N0
Fig. 16. Asynchronous digital multiplexer to combine sub-network demodulator outputs. The output of the multiplexer goes to the Clock-Data Recovery
(CDR) block.
En
Negative
Edge Trigger
Fig. 14. Schematic of fully-digital digital demodulator at sensor node.
dissipation, which is significant since downlink messages have
a very low duty cycle. Additionally, the demodulator is selfreferenced, i.e. it does not need a reference for the comparator, the setting of which would require accurate calibration.
Sample waveforms for the case TPWM = 4.TOOK are shown in
Figure 15.
B. Uplink Communication
Uplink data is transmitted by short- and open-circuiting the
switch across the sensor node LC tank in Figure 5, which
changes the impedance reflected back to the base station
inductor and can be sensed as voltage V1 in Figure 6. During
uplink transmission, the sensor node primarily relies on stored
energy instead of rectifying the received ac power. The signal
received at the base station, shown in Figure 13, is an ASKmodulated waveform with non-unity modulation index. Under
the conditions outlined in Section III, the expressions for the
modulation index of the received waveform m derived in [19]
can be approximated as
m≈
To CDR
0
Q
R
CLK
Demodulator 1
Output
OUT
TOOK
Fig. 15. Waveforms for downlink demodulator. The implemented downlink
operates at TPWM = 12.TOOK .
Demodulator 0
Output
MP%
0%
x
OUT
Fig. 13. Pulse-width modulated data transmitted on downlink using OOK
(top) and impedance-modulated data with non-unity modulation index transmitted on the uplink (bottom), on a 27 MHz carrier.
3
1%
1
k 2 Q1 Q2
Rs
·
1 + k 2 Q1 Q2 Rs + Q1 ω0 L1
(3)
The ASK modulation index at the base station evaluates to approximately 0.08 for the parameter values listed in Section III.
Demodulation is done differentially by extracting and amplifying both the positive and negative envelopes of the incoming
signal shown in Figure 13. Messages arriving from different
sub-networks need to be combined to have a single clock
recovery circuit and digital back-end that handles data integrity
and network access. Combining the impedance modulated
ASK waveforms from the different sub-networks in the analog
domain results in the received voltage signal being measured
across an impedance that progressively decreases as the number of sub-networks increases. Instead, the received signal
from each sub-network is demodulated separately as shown
in Figure 4, and multiplexed in the digital domain. As it
comes before the clock recovery circuit in the receive chain,
the digital multiplexer needs to be asynchronous. Figure 16
shows the schematic of the asynchronous multiplexer. The
output of each demodulator is “0” when there is no data
being transmitted from that sub-network. The circuit shown
in Figure 16 can easily be extended to more than two subnetworks.
V. M EDIUM -ACCESS P ROTOCOL
An application-independent health-monitoring BAN needs
to be able to support different kinds of sensor nodes that have
different channel access and bandwidth requirements. These
could range from ECG and single-channel EEG recorders
that need a time-averaged data rate of up to 10 kb/s, to
temperature, blood pressure or blood oxygenation monitors
that need infrequent access and a time-averaged data rate of
only a few bits/s. In order to accommodate such a variety of
sensor nodes, a hybrid MAC protocol has been designed and
integrated on silicon along with the power and communication
TRX circuits. The hybrid MAC uses both TDMA access and
CA modes for nodes requiring frequent and infrequent access
respectively. It also ensures that all decision-making is pushed
to the base station, with the sensor nodes only following
instructions sent to them.
Due to the architecture of the network, messages on the
downlink are broadcast to all sensor nodes in a sub-network.
Downlink messages are 31b long, and contain a 4b header
that classifies the type of the message. The format of the
following 27 bits varies based on the 4b header. For example,
it could contain the addressed sensor node’s network ID in
7
Byte Count
L
100
L+16
Segment
16b CRC
Fig. 17. Uplink (sensor node to base station) packet structure.
Start
Network
Configuration
TDMA Mode
m=0
n=0
Yes
BS sends beacon to
subnet n
m++
128 bytes
64 bytes
32 bytes
16 bytes
90
80
70
60
50
40 −6
10
−5
−4
10
10
Channel Bit Error Rate
−3
10
n++
Error
?
Yes
Yes
Timeout?
TDMA request
to SN # m
Normalized Uplink Data Rate (%)
Bit Index
0
135
127
148
PLL Lock
7b SN
Preamble
Segment
Seq.
ID
n<
N
Yes
No
No
No
m<
M
No
Yes
No
SN responds
with ACK
SN classified,
added to list
Fig. 19. Attainable uplink data rate vs channel bit error rate for different
payload segment sizes.
Response
?
CA Mode
BS broadcasts
CA beacon
Yes
No
Response
?
Yes
Error
?
No
BS sends
ACK
Fig. 18. Flowchart of implemented medium-access protocol. The network
has N sub-networks with M TDMA sensor nodes combined.
a TDMA data request. The uplink packet structure for all
packets after network configuration is shown in Figure 17.
The uplink packet header contains an RX PLL lock sequence,
followed by a preamble and packet metadata. The payload
data is divided into variable-sized segments, each appended
with a 16b CRC checksum for error detection, and prevents
re-transmission of the entire packet in case of burst-mode
channel noise. Smaller segment sizes increase MAC overhead
but reduce the number of re-transmissions required. A plot
of the attainable uplink data rate vs channel bit error rate for
different possible segment sizes (Figure 19) demonstrates this.
Based on the prevailing error rate determined by the number
of re-transmissions required, the base station decides whether
the segment sizes in succeeding packets needs to increase,
decrease or remain unchanged. The network starts off with
the configuration phase and subsequently moves on to alternate
between TDMA and CA access. The sum of a TDMA and CA
period is limited to 4 seconds by the requirement to service
the sensor nodes with the highest data acquisition rates at least
during that period. A flowchart of the MAC protocol is shown
in Figure 18.
A. Network Configuration
Upon startup, the base station automatically allocates unique
identifiers to all sensor nodes connected to the network. The
base station starts out by sending a configuration beacon to
each sub-network port one at a time. Upon receiving the
configuration beacon, each sensor node backs off in time by a
16b pseudorandom value and responds with details about its
network access requirements. The base station then assigns a
network ID and classifies the sensor as a TDMA node or CA
node.
B. TDM Access Phase
Once the network is configured, it enters the TDMA phase.
During this, the base station polls through each TDMA sensor
node one by one and asks them to transmit their buffered data.
The total duration of this phase is limited by the number of
TDMA nodes multiplied with the maximum network access
time given to each node, which corresponds to 4 KB data
transferred at 1 Mb/s.
C. Contention Access Phase
After completion of the TDMA phase, the network moves
to the CA phase. The start of CA is denoted by a beacon
broadcast by the base station to all sub-networks. Since uplink
data must cross two inductive hops to be heard by other
sensor nodes, a CSMA-based protocol cannot be used for time
multiplexing. Instead, CA nodes with data to transmit back
off in time by a pseudorandom 16b value upon receiving the
beacon. The end of the 4 second period (1 TDMA phase + 1
CA phase) is again broadcast to all sub-networks and the base
station goes back to looping through all the TDMA sensors.
VI. M EASUREMENT R ESULTS
The base station and the NNI for the e-textiles network
have been fabricated in a 0.18 µm CMOS process. Die
photographs are shown in Figure 20. A brief summary of the
design characteristics are presented in Table I and performance
comparisons with other recently reported solutions in the same
application space are presented in Table II.
The power consumption of the base station during
transmit/receive-mode is 2.9 mW, which is comparable to the
narrowband RF- and BCC-based systems in Table II. The
energy per bit on the uplink is more important since the
downlink only carries small, low duty-cycle packets. At 1
Mb/s, the base station expends 2.9 nJ per uplink bit. The 14
µW maximum power consumption by the communication and
digital baseband circuits at 1 Mb/s translates to the sensor
node’s expended energy per uplink bit approaching the pJ/bit
8
2.1mm
1.6mm
2.1mm
1.8mm
(a)
(b)
Fig. 20. Die photographs of (a) base station and (b) Sensor NNI.
1.46 mm2
1
Area
2.95 mm2
VDD,min
2.0 V (PA only)
1.4 V (others)
1.2 V
fCLK
1 MHz
1 MHz
0.4
Power Consumption
2.7 mW (transmitter)
158 µW (others)
14 µW
0.2
Frequency Band
27.12 MHz ISM
Modulation Scheme
Impedance modulation (uplink, 80 kb/s)
PWM-OOK (downlink, 1 Mb/s)
Max. Sensor Count
128
Multi Access
Hybrid TDMA & CA
range seen in fully wired systems [11], [17]. The higher
power consumption at the base station, where energy is more
readily available, trades off against greater comfort due to
the wireless/wired hybrid architecture when compared against
fully wired systems.
Figure 21 shows the efficiency of power transfer and the
voltage at the output of the sensor node’s rectifier for varying
load currents. As the load current increases beyond 20 µA,
the rectifier operation is limited by the series ac-coupling
capacitor reactances and the output voltage falls rapidly. At
its lowest power setting, the power transmitter on the base
station supplying one inductor consumes 2.7 mW with the
PA operating from a 2 V supply. Owing to the weak inductive
link, the power drawn by the transmitter varies negligibly with
changes in the load current of the rectifier. The received power
peaks at 20 µA load current and the peak dc-to-dc power
transfer efficiency across the link is 1.2% with voltage transfer
function of 0.29, which are smaller by approximately 25% than
the values calculated in Section III since Equations (1) and (2)
2
0.8
1.5
0.6
1
0
1
(V)
0.18 µm CMOS
ηdc−dc (%)
Technology
1.2
Rect,out
Sensor Node
Network Interface
Base Station
3
DC−to−DC Eff.
Rectifier V
2.5
out
1.4
V
TABLE I
S UMMARY OF E - TEXTILES NETWORK CHARACTERISTICS
0.5
10
Load Current (µA)
0
100
Fig. 21. Efficiency (dc-to-dc) of power transfer across resonant inductive link
and rectifier output voltage for varying load current. The separation between
the transmitter and receiver coils in this case was 5 mm and the transmitter
PA was at its lowest power setting with 2 V supply.
ignore the effects of the source resistance Rs in Figure 6. Of
a maximum 34 µW power recovered at the sensor node, a
maximum of 14 µW is consumed by the communication and
digital baseband circuits, leaving the remaining 20 µW for
the biomedical signal acquisition front-end which is sufficient
to power state-of-the-art biomedical data-acquisition circuits
[2]–[4]. A plot of the measured efficiency and output power
normalized to its maximum value with varying misalignment
is presented in Figure 22. The base station increases transmitted power with increasing misalignment and is capable of
supplying the node’s minimum 14 µW power requirement for
misalignments up to half the radius of the primary inductor.
OOK and envelope-detected waveforms for a downlink
PWM message are shown in Figure 23. The waveforms
correspond to the initial beacon transmitted by the base station
asking sensor nodes to join the network. Waveforms pertaining
to the packet metadata from an uplink packet with simulated
ECG data are shown in Figure 24. The measured modulation
index of 0.11 agrees with the value calculated theoretically
9
TABLE II
C OMPARISON OF BODY- AREA NETWORK COMMUNICATION SYSTEMS
Wong et al.a
[6]
Roh et al.a
[7]
Yoo et al.
[12]
Lee et al.
[17]
Mercier et al.
[11]
Type
Narrowband RF
BCC
Wired+Wireless
e-textiles
Wired e-textiles
Wired e-textiles
Wired+Wireless
e-textiles
Coverage
Wide
Local
Local
Local
Wide
Wide
Network Topology
Star
Master-slave
(X,Y) grid
Bus
Star
Star
Base
Station
Power
2.3 mW
b
3.2 mW
b
5.2 mW
75 µW
180 µW
2.9 mW
Sensor
Power
2.7 mW
b
1.6 mW
b
12 µW
25 µW
20 µW
14 µW
Node
This Work
c
Data Rate
50 kb/s
1.25 Mb/s
120 kb/s
20 Mb/s
10 Mb/s
1 Mb/s (uplink),
80 kb/s (downlink)
Power Transfer
Mechanism
-
-
Inductive
Wired
Wired
Resonant inductive
Power Transfer
Efficiency
-
-
0.2%
-
96%
1.2%
a
b
c
d
d
No remote power delivery to sensor nodes
Uplink communication power only
Power consumption of communication circuits and digital baseband circuits only
Based on power consumption values of network controller and supported sensor node
1.2
DC−to−DC Eff.
Output Power 30
(µW)
20
Rect,out
0.8
0.6
0.4
0
0
20
40
60
80
100
120
140
160
180
200
100
120
140
160
180
200
Network Configuration
Beacon
P
ηdc−dc (%)
1
Voltage (V)
0.8
1.4
0.4
10
0.2
0
0
5
10
15
Center−to−Center Coil Misalignment (mm)
0
0
20
40
60
80
Time (µs)
Fig. 22. Peak efficiency (dc-to-dc) of power transfer and peak output power
for varying misalignment, measured as the center-center distance between the
coils. The diameter of the primary inductor on clothing is 32 mm.
in Equation (3). The uplink message shown consists of an
initial sequence of alternating 1s and 0s for clock recovery,
followed by the random number ACK described in Section V.
The trailing bits of the message consist of the CRC checksum
that ensures data integrity.
Figure 25 shows the network self-configuring upon startup.
The example network consists of two sub-networks, each with
one and two sensor nodes respectively. The base station sends
the beacon shown in Figure 23 to each sub-network one at a
time. Sensor nodes respond with messages similar to the one
shown in Figure 24 after a random delay to avoid collisions.3
After waiting for 1 second for all sensor nodes in the subnetwork to respond, the base station moves on to the next subnetwork. In Figure 25, both sub-networks are fully configured
3 Only the random number ACK changes between the responses sent by
different sensor nodes.
Fig. 23. Measured PWM-OOK and envelope-detected downlink waveforms
at sensor node. The packet corresponds shown corresponds to the network
configuration beacon transmitted by the base station upon startup.
at the end of 2 seconds.
VII. C ONCLUSION
In order to address the problems of high path loss that BANs
using radios and BCC face, a bi-directional communication
TRX and wireless power delivery system based on e-textiles
has been presented. The proposed e-textiles BAN has been
designed across multiple layers to support a wide variety of
sensors on the body. By delivering power wirelessly to stateof-the-art microwatt-level health monitors through resonant
coupling, the problem of long-term energy storage can be
addressed and their form factors can be reduced. Implemented
in 0.18 µm CMOS, the base station, which acts as the hub in
a star-connected network, consumes 2.9 mW per inductor it
supplies power to. The network architecture has been chosen
10
R EFERENCES
Voltage (V)
2
1.5
1
0.5
0
0
10
20
PLL Lock
Sequence
0
10
30
40
Preamble
20
30
50
60
70
80
70
80
Random
Number ACK
40
50
60
Time (µs)
Fig. 24. Measured waveforms for uplink packet header at input and output
of base station demodulator.
Fig. 25. Self-configuration of e-textiles BAN with two sub-networks. The
bottom three traces track a bit that goes HIGH once the sensor node is
configured.
to offer a good trade-off between flexibility in choosing any
subset of inductors to power while reducing complexity on the
clothing. A NNI that recovers power transmitted by the base
station and handles data modulation and demodulation has also
been implemented in 0.18 µm CMOS. It consumes 14 µW and
harvests a maximum of 34 µW, leaving the remaining 20 µW
for the biomedical front-end. Modulation schemes have been
chosen keeping in mind the asymmetric nature of resource
requirements for data flow in both directions, and such that
complexity is pushed to the base station as much as possible.
Finally, logic implementing a hybrid MAC protocol has also
been integrated on chip, with all decisions being taken by the
base station and the sensor nodes acting as slaves. The hybrid
MAC uses different network access schemes for sensor nodes
with different access and average data transfer rates.
ACKNOWLEDGMENT
The authors would like to thank members of the Semiconductor System Lab at KAIST, Daejeon, Korea for assistance
with e-textiles fabrication and MOSIS for fabrication support.
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Nachiket Desai (S’10) received the Bachelor of
Technology (B. Tech.) degree in electronics & electrical eommunication engineering from the Indian
Institute of Technology, Kharagpur, India in 2010
and the S. M. degree in electrical engineering and
computer science from the Massachusetts Institute of
Technology, Cambridge, MA, USA in 2012, where
he is currently working toward the Ph. D. degree.
Between June 2012 and August 2012, he was
at Texas Instruments, Dallas, TX, USA, designing
power converters for energy harvesting applications.
His research interests include wireless power transfer for consumer devices
and personal-area networks (PANs), circuits for near-field communication
(NFC), and power converters.
Jerald Yoo (S05-M10) received the B.S., M.S., and
Ph.D. degrees in electrical engineering from the
Korea Advanced Institute of Science and Technology
(KAIST), Daejeon, Korea, in 2002, 2007, and 2010,
respectively.
In May 2010, he joined the faculty of Microsystems Engineering, Masdar Institute of Science and
Technology, Abu Dhabi, United Arab Emirates,
where he is currently an associate professor. During
June 2010 to June 2011, he was also with Microsystems Technology Laboratory, Massachusetts
Institute of Technology (MIT), Cambridge, MA, USA, as a visiting scholar. He
developed low-energy Body Area Network (BAN) transceivers and wearable
body sensor network using Planar-Fashionable Circuit Board (P-FCB) for
continuous health monitoring system. His research focuses on low energy circuit technology for wearable bio signal sensors, wireless power transmission,
SoC design to system realization for wearable healthcare applications, and
energy-efficient biomedical circuits. He is an author of the book chapter in
Biomedical CMOS ICs (Springer, 2010).
Dr. Yoo is a co-recipient of the Asian Solid-State Circuits Conference (ASSCC) Outstanding Design Awards in 2005. Currently he serves as a technical
program committee member of Asian Solid-State Circuits Conference (ASSCC).
Anantha P. Chandrakasan (M’95-SM’01-F’04) received the B.S, M.S. and Ph.D. degrees in Electrical Engineering and Computer Sciences from the
University of California, Berkeley, in 1989, 1990,
and 1994 respectively. Since September 1994, he has
been with the Massachusetts Institute of Technology,
Cambridge, where he is currently the Joseph F. and
Nancy P. Keithley Professor of Electrical Engineering.
He was a co-recipient of several awards including
the 1993 IEEE Communications Society’s Best Tutorial Paper Award, the IEEE Electron Devices Society’s 1997 Paul Rappaport
Award for the Best Paper in an EDS publication during 1997, the 1999 DAC
Design Contest Award, the 2004 DAC/ISSCC Student Design Contest Award,
the 2007 ISSCC Beatrice Winner Award for Editorial Excellence and the
ISSCC Jack Kilby Award for Outstanding Student Paper (2007, 2008, 2009).
He received the 2009 Semiconductor Industry Association (SIA) University
Researcher Award. He is the recipient of the 2013 IEEE Donald O. Pederson
Award in Solid-State Circuits.
His research interests include micro-power digital and mixed-signal integrated circuit design, wireless microsensor system design, portable multimedia
devices, energy efficient radios and emerging technologies. He is a co-author
of Low Power Digital CMOS Design (Kluwer Academic Publishers, 1995),
Digital Integrated Circuits (Pearson Prentice-Hall, 2003, 2nd edition), and
Sub-threshold Design for Ultra-Low Power Systems (Springer 2006). He is
also a co-editor of Low Power CMOS Design (IEEE Press, 1998), Design of
High-Performance Microprocessor Circuits (IEEE Press, 2000), and Leakage
in Nanometer CMOS Technologies (Springer, 2005).
He has served as a technical program co-chair for the 1997 International
Symposium on Low Power Electronics and Design (ISLPED), VLSI Design
’98, and the 1998 IEEE Workshop on Signal Processing Systems. He
was the Signal Processing Sub-committee Chair for ISSCC 1999-2001, the
Program Vice-Chair for ISSCC 2002, the Program Chair for ISSCC 2003,
the Technology Directions Sub-committee Chair for ISSCC 2004-2009, and
the Conference Chair for ISSCC 2010-2014. He is the Conference Chair for
ISSCC 2015. He was an Associate Editor for the IEEE Journal of Solid-State
Circuits from 1998 to 2001. He served on SSCS AdCom from 2000 to 2007
and he was the meetings committee chair from 2004 to 2007. He was the
Director of the MIT Microsystems Technology Laboratories from 2006 to
2011. Since July 2011, he is the Head of the MIT EECS Department.