16-Bit, 160 MSPS 2x/4x/8x
Interpolating Dual TxDAC+® D/A Converter
AD9777
FEATURES
Versatile input data interface
Twos complement/straight binary data coding
Dual-port or single-port interleaved input data
Single 3.3 V supply operation
Power dissipation: typical 1.2 W @ 3.3 V
On-chip 1.2 V reference
80-lead thin quad flat package, exposed pad (TQFP_EP)
16-bit resolution, 160 MSPS/400 MSPS input/output
data rate
Selectable 2×/4×/8× interpolating filter
Programmable channel gain and offset adjustment
fS/4, fS/8 digital quadrature modulation capability
Direct IF transmission mode for 70 MHz + IFs
Enables image rejection architecture
Fully compatible SPI® port
Excellent ac performance
SFDR −73 dBc @ 2 MHz to 35 MHz
WCDMA ACPR 71 dB @ IF = 19.2 MHz
Internal PLL clock multiplier
Selectable internal clock divider
Versatile clock input
Differential/single-ended sine wave or
TTL/CMOS/LVPECL compatible
APPLICATIONS
Communications
Analog quadrature modulation architecture
3G, multicarrier GSM, TDMA, CDMA systems
Broadband wireless, point-to-point microwave radios
Instrumentation/ATE
FUNCTIONAL BLOCK DIAGRAM
IDAC
COS
AD9777
I AND Q
NONINTERLEAVED
OR INTERLEAVED
DATA
16
WRITE
SELECT
GAIN
DAC
OFFSET
DAC
SIN
I
LATCH
16
Q
LATCH
16
MUX
CONTROL
HALFBAND
FILTER3*
16
16
16
fDAC/2, 4, 8
16
16
SIN
16
FILTER
BYPASS
MUX
IMAGE
REJECTION/
DUAL DAC
MODE
BYPASS
MUX
I/Q DAC
GAIN/OFFSET
REGISTERS
IOFFSET
16
HALFBAND
FILTER2*
VREF
DATA
ASSEMBLER
HALFBAND
FILTER1*
COS
IDAC
/2
IOUT
(fDAC)
CLOCK OUT
/2
/2
/2
SPI INTERFACE AND
CONTROL REGISTERS
DIFFERENTIAL
CLK
PHASE DETECTOR
AND VCO
PLL CLOCK MULTIPLIER AND CLOCK DIVIDER
02706-001
* HALF-BAND FILTERS ALSO CAN BE
CONFIGURED FOR ZERO STUFFING ONLY
PRESCALER
Figure 1.
Rev. C
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
© 2006 Analog Devices, Inc. All rights reserved.
AD9777
TABLE OF CONTENTS
Features .............................................................................................. 1
Sleep/Power-Down Modes........................................................ 29
Applications....................................................................................... 1
Two Port Data Input Mode ....................................................... 29
General Description ......................................................................... 4
PLL Enabled, Two-Port Mode .................................................. 30
Product Highlights ....................................................................... 4
DATACLK Inversion.................................................................. 30
Specifications..................................................................................... 5
DATACLK Driver Strength....................................................... 30
DC Specifications ......................................................................... 5
PLL Enabled, One-Port Mode .................................................. 30
Dynamic Specifications ............................................................... 6
ONEPORTCLK Inversion......................................................... 31
Digital Specifications ................................................................... 7
ONEPORTCLK Driver Strength.............................................. 31
Digital Filter Specifications ......................................................... 8
IQ Pairing .................................................................................... 31
Absolute Maximum Ratings............................................................ 9
PLL Disabled, Two-Port Mode................................................. 31
Thermal Characteristics .............................................................. 9
PLL Disabled, One-Port Mode ................................................. 32
ESD Caution.................................................................................. 9
Digital Filter Modes ................................................................... 32
Pin Configuration and Function Descriptions........................... 10
Amplitude Modulation.............................................................. 32
Terminology .................................................................................... 12
Modulation, No Interpolation .................................................. 34
Typical Performance Characteristics ........................................... 13
Modulation, Interpolation = 2× ............................................... 35
Mode Control (via SPI Port)..................................................... 18
Modulation, Intermodulation = 4× ......................................... 36
Register Description................................................................... 20
Modulation, Intermodulation = 8× ......................................... 37
Functional Description .................................................................. 22
Zero Stuffing ............................................................................... 38
Serial Interface for Register Control ........................................ 22
Interpolating (Complex Mix Mode)........................................ 38
General Operation of the Serial Interface ............................... 22
Operations on Complex Signals............................................... 38
Instruction Byte .......................................................................... 23
Complex Modulation and Image Rejection of Baseband
Signals .......................................................................................... 39
R/W .............................................................................................. 23
N1, N0 .......................................................................................... 23
A4, A3, A2, A1, A0..................................................................... 23
Serial Interface Port Pin Descriptions ..................................... 23
MSB/LSB Transfers..................................................................... 23
Notes on Serial Port Operation ................................................ 25
DAC Operation........................................................................... 25
1R/2R Mode ................................................................................ 26
CLOCK Input Configuration ................................................... 26
Programmable PLL .................................................................... 27
Image Rejection and Sideband Suppressions of Modulated
Carriers ........................................................................................ 41
Applying the Output Configurations........................................... 46
Unbuffered Differential Output, Equivalent Circuit ............. 46
Differential Coupling Using a Transformer............................ 46
Differential Coupling Using an Op Amp................................ 47
Interfacing with the AD8345 Quadrature Modulator........... 47
Evaluation Board ............................................................................ 48
Outline Dimensions ....................................................................... 58
Ordering Guide .......................................................................... 58
Power Dissipation....................................................................... 29
Rev. C | Page 2 of 60
AD9777
REVISION HISTORY
1/06—Rev. B to Rev. C
Updated Formatting .........................................................Universal
Changes to Figure 32 .................................................................... 22
Changes to Figure 108 .................................................................. 54
Updated Outline Dimensions ..................................................... 58
Changes to Ordering Guide......................................................... 58
6/04—Data Sheet Changed from Rev. A to Rev. B.
Changes to DC Specifications ....................................................... 5
Changes to Absolute Maximum Ratings...................................... 8
Changes to DAC Operation Section........................................... 25
Changes to Figure 49, Figure 50, and Figure 51........................ 29
Changes to the PLL Enabled, One-Port Mode Section............ 30
Changes to the PLL Disabled, One-Port Mode Section........... 32
Changes to the Ordering Guide .................................................. 57
Updated the Outline Dimensions ............................................... 57
3/03—Data Sheet Changed from Rev. 0 to Rev. A.
Edits to Features .............................................................................. 1
Edits to DC Specifications ............................................................. 3
Edits to Dynamic Specifications.................................................... 4
Edits to Pin Function Descriptions............................................... 7
Edits to Table I ............................................................................... 14
Edits to Register Description—Address 02h Section ............... 15
Edits to Register Description—Address 03h Section ............... 16
Edits to Register Description—Address 07h, 0Bh Section...... 16
Edits to Equation 1........................................................................ 16
Edits to MSB/LSB Transfers Section........................................... 18
Changes to Figure 8 ...................................................................... 20
Edits to Programmable PLL Section........................................... 21
Added new Figure 14.................................................................... 22
Renumbered Figures 15 to 69...................................................... 22
Added Two-Port Data Input Mode Section............................... 23
Edits to PLL Enabled, Two-Port Mode Section ........................ 24
Edits to Figure 19 ......................................................................... 24
Edits to Figure 21 .......................................................................... 25
Edits to PLL Disabled, Two-Port Mode Section ....................... 25
Edits to Figure 22 .......................................................................... 25
Edits to Figure 23 .......................................................................... 26
Edits to Figure 26a ........................................................................ 27
Changes to Figures 53 and 54...................................................... 38
Edits to Evaluation Board Section .............................................. 39
Changes to Figures 56 to 59......................................................... 40
Replaced Figures 60 to 69 ............................................................ 42
Updated Outline Dimensions...................................................... 49
7/02—Revision 0: Initial Version
Rev. C | Page 3 of 60
AD9777
GENERAL DESCRIPTION
The AD97771 is the 16-bit member of the AD977x pin
compatible, high performance, programmable 2×/4×/8×
interpolating TxDAC+ family. The AD977x family features a
serial port interface (SPI) that provides a high level of
programmability, thus allowing for enhanced system level
options. These options include selectable 2×/4×/8× interpolation filters; fS/2, fS/4, or fS/8 digital quadrature modulation with
image rejection; a direct IF mode; programmable channel gain
and offset control; programmable internal clock divider;
straight binary or twos complement data interface; and a singleport or dual-port data interface.
The selectable 2×/4×/8× interpolation filters simplify the
requirements of the reconstruction filters while simultaneously
enhancing the TxDAC+ family’s pass-band noise/distortion
performance. The independent channel gain and offset adjust
registers allow the user to calibrate LO feedthrough and sideband
suppression errors associated with analog quadrature modulators.
The 6 dB of gain adjustment range can also be used to control the
output power level of each DAC.
The AD9777 features the ability to perform fS/2, fS/4, and fS/8
digital modulation and image rejection when combined with an
analog quadrature modulator. In this mode, the AD9777 accepts I
and Q complex data (representing a single or multicarrier waveform), generates a quadrature modulated IF signal along with its
orthogonal representation via its dual DACs, and presents these
two reconstructed orthogonal IF carriers to an analog quadrature
modulator to complete the image rejection upconversion process.
Another digital modulation mode (that is, the direct IF mode)
allows the original baseband signal representation to be frequency translated such that pairs of images fall at multiples of
one-half the DAC update rate.
The AD977x family includes a flexible clock interface accepting
differential or single-ended sine wave or digital logic inputs. An
internal PLL clock multiplier is included and generates the
necessary on-chip high frequency clocks. It can also be disabled
to allow the use of a higher performance external clock source.
An internal programmable divider simplifies clock generation
in the converter when using an external clock source. A flexible
data input interface allows for straight binary or twos complement formats and supports single-port interleaved or dual-port
data.
Dual high performance DAC outputs provide a differential
current output programmable over a 2 mA to 20 mA range. The
AD9777 is manufactured on an advanced 0.35 micron CMOS
process, operates from a single-supply of 3.1 V to 3.5 V, and
consumes 1.2 W of power.
1
Targeted at wide dynamic range, multicarrier, and
multistandard systems, the superb baseband performance of the
AD9777 is ideal for wideband CDMA, multicarrier CDMA,
multicarrier TDMA, multicarrier GSM, and high performance
systems employing high-order QAM modulation schemes. The
image rejection feature simplifies and can help to reduce the
number of signal band filters needed in a transmit signal chain.
The direct IF mode helps to eliminate a costly mixer stage for a
variety of communications systems.
PRODUCT HIGHLIGHTS
1.
The AD9777 is the 16-bit member of the AD977x pin
compatible, high performance, programmable 2×/4×/8×
interpolating TxDAC+ family.
2.
Direct IF transmission is possible for 70 MHz + IFs
through a novel digital mixing process.
3.
fS/2, fS/4, and fS/8 digital quadrature modulation and user
selectable image rejection simplify/remove cascaded SAW
filter stages.
4.
A 2×/4×/8× user selectable interpolating filter eases data
rate and output signal reconstruction filter requirements.
5.
User selectable twos complement/straight binary data
coding.
6.
User programmable channel gain control over 1 dB range
in 0.01 dB increments.
7.
User programmable channel offset control ±10% over the
FSR.
8.
Ultrahigh speed 400 MSPS DAC conversion rate.
9.
Internal clock divider provides data rate clock for easy
interfacing.
10. Flexible clock input with single-ended or differential input,
CMOS, or 1 V p-p LO sine wave input capability.
11. Low power: Complete CMOS DAC operates on 1.2 W
from a 3.1 V to 3.5 V single supply. The 20 mA full-scale
current can be reduced for lower power operation, and
several sleep functions are provided to reduce power
during idle periods.
12. On-chip voltage reference: The AD9777 includes a 1.20 V
temperature compensated band gap voltage reference.
13. An 80-lead thin quad flat package, exposed pad
(TQFP_EP).
Protected by U.S. Patent Numbers, 5,568,145; 5,689,257; and 5,703,519.
Other patents pending.
Rev. C | Page 4 of 60
AD9777
SPECIFICATIONS
DC SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, CLKVDD = 3.3 V, DVDD = 3.3 V, PLLVDD = 3.3 V, IOUTFS = 20 mA, unless otherwise noted.
Table 1.
Parameter
RESOLUTION
DC Accuracy1
Integral Nonlinearity
Differential Nonlinearity
ANALOG OUTPUT (for 1R and 2R Gain Setting Modes)
Offset Error
Gain Error (with Internal Reference)
Gain Matching
Full-Scale Output Current2
Output Compliance Range
Output Resistance
Output Capacitance
Gain, Offset Cal DACs, Monotonicity Guaranteed
REFERENCE OUTPUT
Reference Voltage
Reference Output Current3
REFERENCE INPUT
Input Compliance Range
Reference Input Resistance
Small Signal Bandwidth
TEMPERATURE COEFFICIENTS
Offset Drift
Gain Drift (with Internal Reference)
Reference Voltage Drift
POWER SUPPLY
AVDD
Voltage Range
Analog Supply Current (IAVDD)4
IAVDD in SLEEP Mode
CLKVDD (PLL OFF)
Voltage Range
Clock Supply Current (ICLKVDD)4
CLKVDD (PLL ON)
Clock Supply Current (ICLKVDD)
DVDD
Voltage Range
Digital Supply Current (IDVDD)4
Nominal Power Dissipation4
PDIS5
PDIS in PWDN
Power Supply Rejection Ratio—AVDD
OPERATING RANGE
Min
16
−6.5
−0.025
−1.0
−1
2
−1.0
Typ
Max
±6
±3
+6.5
±0.01
% of FSR
% of FSR
% of FSR
mA
V
kΩ
pF
1.26
V
nA
1.25
7
0.5
V
kΩ
MHz
0
50
±50
ppm of FSR/°C
ppm of FSR/°C
ppm/°C
±0.1
1.20
100
0.1
3.1
3.3
72.5
23.3
3.5
76
26
V
mA
mA
3.1
3.3
8.5
3.5
10.0
V
mA
23.5
3.1
−40
1
Measured at IOUTA driving a virtual ground.
Nominal full-scale current, IOUTFS, is 32× the IREF current.
3
Use an external amplifier to drive any external load.
4
100 MSPS fDAC with fOUT = 1 MHz, all supplies = 3.3 V, no interpolation, no modulation.
5
400 MSPS fDAC, fDATA = 50 MSPS, fS/2 modulation, PLL enabled.
2
Rev. C | Page 5 of 60
LSB
LSB
+0.025
+1.0
+1
20
+1.25
200
3
1.14
Unit
Bits
3.3
34
380
1.75
6.0
±0.4
mA
3.5
41
410
+85
V
mA
mW
W
mW
% of FSR/V
°C
AD9777
DYNAMIC SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, CLKVDD = 3.3 V, DVDD = 3.3 V, PLLVDD = 0 V, IOUTFS = 20 mA, Interpolation = 2×, differential
transformer-coupled output, 50 Ω doubly terminated, unless otherwise noted.
Table 2.
Parameter
DYNAMIC PERFORMANCE
Maximum DAC Output Update Rate (fDAC)
Output Settling Time (tST) (to 0.025%)
Output Rise Time (10% to 90%)1
Output Fall Time (10% to 90%)1
Output Noise (IOUTFS = 20 mA)
AC LINEARITY—BASEBAND MODE
Spurious-Free Dynamic Range (SFDR) to Nyquist (fOUT = 0 dBFS)
fDATA = 100 MSPS, fOUT = 1 MHz
fDATA = 65 MSPS, fOUT = 1 MHz
fDATA = 65 MSPS, fOUT = 15 MHz
fDATA = 78 MSPS, fOUT = 1 MHz
fDATA = 78 MSPS, fOUT = 15 MHz
fDATA = 160 MSPS, fOUT = 1 MHz
fDATA = 160 MSPS, fOUT = 15 MHz
Spurious-Free Dynamic Range within a 1 MHz Window
fOUT = 0 dBFS, fDATA = 100 MSPS, fOUT = 1 MHz
Two-Tone Intermodulation (IMD) to Nyquist (fOUT1 = fOUT2 = −6 dBFS)
fDATA = 65 MSPS, fOUT1 = 10 MHz; fOUT2 = 11 MHz
fDATA = 65 MSPS, fOUT1 = 20 MHz; fOUT2 = 21 MHz
fDATA = 78 MSPS, fOUT1 = 10 MHz; fOUT2 = 11 MHz
fDATA = 78 MSPS, fOUT1 = 20 MHz; fOUT2 = 21 MHz
fDATA = 160 MSPS, fOUT1 = 10 MHz; fOUT2 = 11 MHz
fDATA = 160 MSPS, fOUT1 = 20 MHz; fOUT2 = 21 MHz
Total Harmonic Distortion (THD)
fDATA = 100 MSPS, fOUT = 1 MHz; 0 dBFS
Signal-to-Noise Ratio (SNR)
fDATA = 78 MSPS, fOUT = 5 MHz; 0 dBFS
fDATA = 160 MSPS, fOUT = 5 MHz; 0 dBFS
Adjacent Channel Power Ratio (ACPR)
WCDMA with 3.84 MHz BW, 5 MHz Channel Spacing
IF = Baseband, fDATA = 76.8 MSPS
IF = 19.2 MHz, fDATA = 76.8 MSPS
Four-Tone Intermodulation
21 MHz, 22 MHz, 23 MHz, and 24 MHz at −12 dBFS (fDATA = MSPS, Missing Center)
AC LINEARITY—IF MODE
Four-Tone Intermodulation at IF = 200 MHz
201 MHz, 202 MHz, 203 MHz, and 204 MHz at −12 dBFS (fDATA = 160 MSPS, fDAC = 320 MHz)
1
Measured single-ended into 50 Ω load.
Rev. C | Page 6 of 60
Min
Typ
400
Max
Unit
11
0.8
0.8
50
MSPS
ns
ns
ns
pA/√Hz
71
85
85
84
85
83
85
83
dBc
dBc
dBc
dBc
dBc
dBc
dBc
73
99.1
dBc
85
78
85
78
85
84
dBc
dBc
dBc
dBc
dBc
dBc
−83
dB
79
75
dB
dB
73
73
dBc
dBc
76
dBFS
72
dBFS
−71
AD9777
DIGITAL SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, CLKVDD = 3.3 V, PLLVDD = 0 V, DVDD = 3.3 V, IOUTFS = 20 mA, unless otherwise noted.
Table 3.
Parameter
DIGITAL INPUTS
Logic 1 Voltage
Logic 0 Voltage
Logic 1 Current
Logic 0 Current
Input Capacitance
CLOCK INPUTS
Input Voltage Range
Common-Mode Voltage
Differential Voltage
SERIAL CONTROL BUS
Maximum SCLK Frequency (fSLCK)
Mimimum Clock Pulse Width High (tPWH)
Mimimum Clock Pulse Width Low (tPWL)
Maximum Clock Rise/Fall Time
Minimum Data/Chip Select Setup Time (tDS)
Minimum Data Hold Time (tDH)
Maximum Data Valid Time (tDV)
RESET Pulse Width
Inputs (SDI, SDIO, SCLK, CSB)
Logic 1 Voltage
Logic 0 Voltage
Logic 1 Current
Logic 0 Current
Input Capacitance
SDIO Output
Logic 1 Voltage
Logic 0 Voltage
Logic 1 Current
Logic 0 Current
Min
Typ
2.1
3
0
−10
−10
Max
Unit
0.9
+10
+10
V
V
µA
µA
pF
5
0
0.75
0.5
1.5
1.5
3
2.25
15
30
30
1
25
0
30
1.5
2.1
3
0
−10
−10
0.9
+10
+10
5
DRVDD − 0.6
0.4
30
30
Rev. C | Page 7 of 60
50
50
V
V
V
MHz
ns
ns
ms
ns
ns
ns
ns
V
V
µA
µA
pF
V
V
mA
mA
AD9777
DIGITAL FILTER SPECIFICATIONS
20
Table 4. Half-Band Filter No. 1 (43 Coefficients)
0
ATTENUATION (dBFS)
–20
–40
–60
–80
–100
0.5
1.0
1.5
2.0
02706-003
0
2.0
02706-004
–120
8
02706-005
Coefficient
8
0
−29
0
67
0
−134
0
244
0
−414
0
673
0
−1,079
0
1,772
0
−3,280
0
10,364
16,384
fOUT (NORMALIZED TO INPUT DATA RATE)
Figure 2. 2× Interpolating Filter Response
20
0
ATTENUATION (dBFS)
Tap
1, 43
2, 42
3, 41
4, 40
5, 39
6, 38
7, 37
8, 36
9, 35
10, 34
11, 33
12, 32
13, 31
14, 30
15, 29
16, 28
17, 27
18, 26
19, 25
20, 24
21, 23
22
–20
–40
–60
–80
Table 5. Half-Band Filter No. 2 (19 Coefficients)
Coefficient
19
0
−120
0
438
0
−1,288
0
5,047
8,192
Table 6. Half-Band Filter No. 3 (11 Coefficients)
Tap
1, 11
2, 10
3, 9
4, 8
5, 7
6
Coefficient
7
0
−53
0
302
512
–100
–120
0
0.5
1.0
1.5
fOUT (NORMALIZED TO INPUT DATA RATE)
Figure 3. 4× Interpolating Filter Response
20
0
ATTENUATION (dBFS)
Tap
1, 19
2, 18
3, 17
4, 16
5, 15
6, 14
7, 13
8, 12
9, 11
10
–20
–40
–60
–80
–100
–120
0
2
4
6
fOUT (NORMALIZED TO INPUT DATA RATE)
Figure 4. 8× Interpolating Filter Response
Rev. C | Page 8 of 60
AD9777
ABSOLUTE MAXIMUM RATINGS
Table 7.
Parameter
AVDD, DVDD, CLKVDD
AVDD, DVDD, CLKVDD
AGND, DGND, CLKGND
REFIO, FSADJ1/FSADJ2
IOUTA, IOUTB
P1B15 to P1B0, P2B15 to P2B0, RESET
DATACLK/PLL_LOCK
CLK+, CLK−
LPF
SPI_CSB, SPI_CLK, SPI_SDIO, SPI_SDO
Junction Temperature
Storage Temperature
Lead Temperature (10 sec)
With Respect To
AGND, DGND, CLKGND
AVDD, DVDD, CLKVDD
AGND, DGND, CLKGND
AGND
AGND
DGND
DGND
CLKGND
CLKGND
DGND
Min
−0.3
−4.0
−0.3
−0.3
−1.0
−0.3
−0.3
−0.3
−0.3
−0.3
−65
Max
+4.0
+4.0
+0.3
AVDD + 0.3
AVDD + 0.3
DVDD + 0.3
DVDD + 0.3
CLKVDD + 0.3
CLKVDD + 0.3
DVDD + 0.3
125
+150
300
Unit
V
V
V
V
V
V
V
V
V
V
°C
°C
°C
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only;
functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is
not implied. Exposure to absolute maximum ratings for extended periods may affect device reliability.
THERMAL CHARACTERISTICS
Thermal Resistance
80-lead thin quad flat package, exposed pad [TQFP_EP]
θJA = 23.5°C/W (With thermal pad soldered to PCB)
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. C | Page 9 of 60
AD9777
AVDD
AGND
AVDD
AGND
AVDD
AGND
AGND
IOUTB2
IOUTA2
AGND
AGND
IOUTB1
IOUTA1
AGND
AGND
AVDD
AGND
AVDD
AGND
AVDD
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
80 79 78 77 76 75 74 73 72 71 70 69 68 67 66 65 64 63 62 61
60
FSADJ1
59
FSADJ2
3
58
REFIO
CLKGND
4
57
RESET
CLK+
5
56
SPI_CSB
CLK–
6
55
SPI_CLK
CLKGND
7
54
SPI_SDIO
DATACLK/PLL_LOCK
8
53
SPI_SDO
DGND
9
52
DGND
51
DVDD
P1B15 (MSB) 11
50
P2B0 (LSB)
P1B14 12
49
P2B1
P1B13 13
48
P2B2
P1B12 14
47
P2B3
P1B11 15
46
P2B4
P1B10 16
45
P2B5
DGND 17
44
DGND
DVDD 18
43
DVDD
P1B9 19
42
P2B6
P1B8 20
41
P2B7
CLKVDD 1
LPF
2
CLKVDD
PIN 1
AD9777
TxDAC+
TOP VIEW
(Not to Scale)
DVDD 10
Figure 5. Pin Configuration
Rev. C | Page 10 of 60
02706-002
P2B8
P2B9
P2B10
P2B11
DVDD
DGND
P2B12
P2B13
ONEPORTCLK/P2B14
IQSEL/P2B15 (MSB)
P1B0 (LSB)
P1B1
P1B2
P1B3
DVDD
DGND
P1B4
P1B5
P1B6
P1B7
21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40
NC = NO CONNECT
AD9777
Table 8. Pin Function Description
Pin No.
1, 3
2
4, 7
5
6
8
Mnemonic
CLKVDD
LPF
CLKGND
CLK+
CLK−
DATACLK/PLL_LOCK
9, 17, 25,
35, 44, 52
10, 18, 26,
36, 43, 51
11 to 16, 19
to 24, 27 to
30
31
DGND
Description
Clock Supply Voltage.
PLL Loop Filter.
Clock Supply Common.
Differential Clock Input.
Differential Clock Input.
With the PLL enabled, this pin indicates the state of the PLL. A read of a Logic 1 indicates the
PLL is in the locked state. Logic 0 indicates the PLL has not achieved lock. This pin may also be
programmed to act as either an input or output (Address 02h, Bit 3) DATACLK signal running
at the input data rate.
Digital Common.
DVDD
Digital Supply Voltage.
P1B15 (MSB) to P1B0 (LSB)
Port 1 Data Inputs.
IQSEL/P2B15 (MSB)
32
ONEPORTCLK/P2B14
33, 34, 37 to
42, 45 to 50
53
P2B13 to P2B0 (LSB)
In one-port mode, IQSEL = 1 followed by a rising edge of the differential input clock latches
the data into the I channel input register. IQSEL = 0 latches the data into the Q channel input
register. In two-port mode, this pin becomes the Port 2 MSB.
With the PLL disabled and the AD9777 in one-port mode, this pin becomes a clock output
that runs at twice the input data rate of the I and Q channels. This allows the AD9777 to
accept and demux interleaved I and Q data to the I and Q input registers.
Port 2 Data Inputs.
SPI_SDO
54
SPI_SDIO
55
SPI_CLK
56
SPI_CSB
57
RESET
58
59
60
61, 63, 65,
76, 78, 80
62, 64, 66, 67,
70, 71, 74,
75, 77, 79
68, 69
72, 73
REFIO
FSADJ2
FSADJ1
AVDD
In the case where SDIO is an input, SDO acts as an output. When SDIO becomes an output,
SDO enters a High-Z state. This pin can also be used as an output for the data rate clock. For
more information, see the Two Port Data Input Mode section.
Bidirectional Data Pin. Data direction is controlled by Bit 7 of Register Address 00h. The
default setting for this bit is 0, which sets SDIO as an input.
Data input to the SPI port is registered on the rising edge of SPI_CLK. Data output on the SPI
port is registered on the falling edge.
Chip Select/SPI Data Synchronization. On momentary logic high, resets SPI port logic and
initializes instruction cycle.
Logic 1 resets all of the SPI port registers, including Address 00h, to their default values. A
software reset can also be done by writing a Logic 1 to SPI Register 00h, Bit 5. However, the
software reset has no effect on the bits in Address 00h.
Reference Output, 1.2 V Nominal.
Full-Scale Current Adjust, Q Channel.
Full-Scale Current Adjust, I Channel.
Analog Supply Voltage.
AGND
Analog Common.
IOUTB2, IOUTA2
IOUTB1, IOUTA1
Differential DAC Current Outputs, Q Channel.
Differential DAC Current Outputs, I Channel.
Rev. C | Page 11 of 60
AD9777
TERMINOLOGY
Adjacent Channel Power Ratio (ACPR)
A ratio in dBc between the measured power within a channel
relative to its adjacent channel.
Monotonicity
A DAC is monotonic if the output either increases or remains
constant as the digital input increases.
Complex Image Rejection
In a traditional two-part upconversion, two images are created
around the second IF frequency. These images are redundant and
have the effect of wasting transmitter power and system bandwidth.
By placing the real part of a second complex modulator in series
with the first complex modulator, either the upper or lower
frequency image near the second IF can be rejected.
Offset Error
The deviation of the output current from the ideal of 0 is called
offset error. For IOUTA, 0 mA output is expected when the inputs
are all 0. For IOUTB, 0 mA output is expected when all inputs are
set to 1.
Complex Modulation
The process of passing the real and imaginary components of a
signal through a complex modulator (transfer function = ejωt =
cosωt + jsinωt) and realizing real and imaginary components
on the modulator output.
Differential Nonlinearity (DNL)
DNL is the measure of the variation in analog value, normalized to
full scale, associated with a 1 LSB change in digital input code.
Gain Error
The difference between the actual and ideal output span. The
actual span is determined by the output when all inputs are set
to 1 minus the output when all inputs are set to 0.
Glitch Impulse
Asymmetrical switching times in a DAC give rise to undesired
output transients that are quantified by a glitch impulse. It is
specified as the net area of the glitch in pV-s.
Group Delay
Number of input clocks between an impulse applied at the
device input and the peak DAC output current. A half-band FIR
filter has constant group delay over its entire frequency range.
Impulse Response
Response of the device to an impulse applied to the input.
Interpolation Filter
If the digital inputs to the DAC are sampled at a multiple rate of
fDATA (interpolation rate), a digital filter can be constructed that
has a sharp transition band near fDATA/2. Images that would
typically appear around fDAC (output data rate) can be greatly
suppressed.
Linearity Error
(Also called integral nonlinearity or INL) Linearity error is
defined as the maximum deviation of the actual analog output
from the ideal output, determined by a straight line drawn from
zero to full scale.
Output Compliance Range
The range of allowable voltage at the output of a current-output
DAC. Operation beyond the maximum compliance limits may
cause either output stage saturation or breakdown, resulting in
nonlinear performance.
Pass Band
Frequency band in which any input applied therein passes
unattenuated to the DAC output.
Power Supply Rejection
The maximum change in the full-scale output as the supplies
are varied from minimum to maximum specified voltages.
Signal-to-Noise Ratio (SNR)
SNR is the ratio of the rms value of the measured output signal
to the rms sum of all other spectral components below the
Nyquist frequency, excluding the first six harmonics and dc.
The value for SNR is expressed in decibels.
Spurious-Free Dynamic Range
The difference, in dB, between the rms amplitude of the
output signal and the peak spurious signal over the specified
bandwidth.
Settling Time
The time required for the output to reach and remain within a
specified error band about its final value, measured from the
start of the output transition.
Stop-Band Rejection
The amount of attenuation of a frequency outside the pass band
applied to the DAC, relative to a full-scale signal applied at the
DAC input within the pass band.
Temperature Drift
It is specified as the maximum change from the ambient (25°C)
value to the value at either TMIN or TMAX. For offset and gain
drift, the drift is reported in ppm of full-scale range (FSR) per
°C. For reference drift, the drift is reported in ppm per °C.
Total Harmonic Distortion (THD)
THD is the ratio of the rms sum of the first six harmonic
components to the rms value of the measured fundamental. It is
expressed as a percentage or in decibels (dB).
Rev. C | Page 12 of 60
AD9777
TYPICAL PERFORMANCE CHARACTERISTICS
10
10
0
0
–10
–10
–20
–20
AMPLITUDE (dBm)
–30
–40
–50
–60
–30
–40
–50
–60
–70
–70
–80
–80
–90
0
65
130
FREQUENCY (MHz)
02706-006
–90
0
50
100
02706-009
AMPLITUDE (dBm)
T = 25°C, AVDD = 3.3 V, CLKVDD = 3.3 V, DVDD = 3.3 V, IOUTFS = 20 mA, Interpolation = 2×, differential transformer-coupled output,
50 Ω doubly terminated, unless otherwise noted.
150
FREQUENCY (MHz)
Figure 6. Single-Tone Spectrum @ fDATA = 65 MSPS with fOUT = fDATA/3
Figure 9. Single-Tone Spectrum @ fDATA = 78 MSPS with fOUT = fDATA/3
90
90
0dBFS
0dBFS
85
85
80
75
75
SFDR (dBc)
70
–12dBFS
65
–12dBFS
65
60
60
55
55
5
10
15
20
25
30
FREQUENCY (MHz)
0
5
10
15
20
25
30
02706-010
50
0
02706-007
50
–6dBFS
70
30
02706-011
SFDR (dBc)
–6dBFS
80
FREQUENCY (MHz)
Figure 7. In-Band SFDR vs. fOUT @ fDATA = 65 MSPS
Figure 10. In-Band SFDR vs. fOUT @ fDATA = 78 MSPS
90
90
85
85
0dBFS
75
75
SFDR (dBc)
80
70
–6dBFS
65
70
–6dBFS
65
60
60
55
55
50
50
0
5
10
15
20
25
FREQUENCY (MHz)
30
02706-008
SFDR (dBc)
–12dBFS
0dBFS
–12dBFS
80
Figure 8. Out-of-Band SFDR vs. fOUT @ fDATA = 65 MSPS
0
5
10
15
20
25
FREQUENCY (MHz)
Figure 11. Out-of-Band SFDR vs. fOUT @ fDATA = 78 MSPS
Rev. C | Page 13 of 60
AD9777
10
90
0dBFS
0
85
–10
–30
IMD (dBc)
AMPLITUDE (dBm)
80
–20
–40
–50
–3dBFS
75
–6dBFS
70
65
–60
60
–70
55
–80
100
200
02706-012
0
300
FREQUENCY (MHz)
0
5
10
20
25
30
Figure 15. Third-Order IMD Products vs. Two-Tone fOUT @ fDATA = 65 MSPS
Figure 12. Single-Tone Spectrum @ fDATA = 160 MSPS with fOUT = fDATA/3
90
90
0dBFS
0dBFS
85
85
80
80
75
75
IMD (dBc)
70
–6dBFS
65
–12dBFS
–6dBFS
–3dBFS
70
65
60
60
55
55
50
0
10
20
30
40
50
FREQUENCY (MHz)
02706-013
50
Figure 13. In-Band SFDR vs. fOUT @ fDATA = 160 MSPS
0
5
10
15
20
25
30
FREQUENCY (MHz)
02706-016
SFDR (dBc)
15
FREQUENCY (MHz)
02706-015
50
–90
Figure 16. Third-Order IMD Products vs. Two-Tone fOUT @ fDATA = 78 MSPS
90
90
85
85
80
80
0dBFS
IMD (dBc)
–6dBFS
–12dBFS
70
65
75
–6dBFS
70
65
0dBFS
60
60
55
55
50
0
10
20
30
40
50
FREQUENCY (MHz)
Figure 14. Out-of-Band SFDR vs. fOUT @ fDATA = 160 MSPS
0
10
20
30
40
FREQUENCY (MHz)
50
60
02706-017
50
02706-014
SFDR (dBc)
–3dBFS
75
Figure 17. Third-Order IMD Products vs. Two-Tone fOUT @ fDATA = 160 MSPS
Rev. C | Page 14 of 60
AD9777
90
90
4×
0dBFS
85
85
80
80
–6dBFS
8×
1×
70
2×
65
75
70
65
60
60
55
55
50
10
20
30
40
50
60
FREQUENCY (MHz)
50
3.1
02706-018
0
3.3
3.4
3.5
AVDD (V)
Figure 18. Third-Order IMD Products vs. Two-Tone fOUT and Interpolation
Rate, 1× fDATA = 160 MSPS, 2× fDATA = 160 MSPS,
4× fDATA = 80 MSPS, 8× fDATA = 50 MSPS
90
3.2
02706-021
75
IMD (dBc)
IMD (dBc)
–3dBFS
Figure 21. Third-Order IMD Products vs. AVDD @ fOUT = 10 MHz,
fDAC = 320 MSPS, fDATA = 160 MSPS
90
8×
85
85
80
80
PLL OFF
SNR (dB)
2×
75
70
PLL ON
65
65
60
60
55
55
50
–15
–10
–5
0
AOUT (dBFS)
50
0
50
100
150
INPUT DATA RATE (MSPS)
02706-022
70
1×
02706-019
IMD (dBc)
4×
75
Figure 22. SNR vs. Data Rate for fOUT = 5 MHz
Figure 19. Third-Order IMD Products vs. Two-Tone AOUT and Interpolation
Rate, fDATA = 50 MSPS in All Cases, 1× fDAC = 50 MSPS, 2× fDAC = 100 MSPS,
4× fDAC = 200 MSPS, 8× fDAC = 400 MSPS
90
90
78MSPS
85
0dBFS
85
80
80
–12dBFS
70
65
75
160MSPS
70
65
60
60
55
55
50
3.1
3.2
3.3
AVDD (V)
3.4
3.5
Figure 20. SFDR vs. AVDD fOUT = 10 MHz, fDAC = 320 MSPS, fDATA = 160 MSPS
Rev. C | Page 15 of 60
fDATA = 65MSPS
50
–50
0
50
TEMPERATURE (°C)
Figure 23. SFDR vs. Temperature @ fOUT = fDATA/11
100
02706-023
SFDR (dBc)
75
02706-020
SFDR (dBc)
–6dBFS
AD9777
0
0
–10
–20
–30
AMPLITUDE (dBm)
AMPLITUDE (dBm)
–20
–40
–50
–60
–70
–40
–60
–80
–80
–90
50
100
150
FREQUENCY (MHz)
02706-024
0
0
5
10
15
20
25
30
35
40
45
FREQUENCY (MHz)
02706-027
–100
–100
Figure 27. Two-Tone IMD Performance, fDATA = 90 MSPS, Interpolation = 4×
Figure 24. Single-Tone Spurious Performance, fOUT = 10 MHz,
fDATA = 150 MSPS, No Interpolation
0
0
–10
–20
–30
AMPLITUDE (dBm)
AMPLITUDE (dBm)
–20
–40
–60
–40
–50
–60
–70
–80
–80
–100
–100
10
20
30
40
50
FREQUENCY (MHz)
0
02706-025
0
50
100
150
200
250
02706-028
–90
300
FREQUENCY (MHz)
Figure 25. Two-Tone IMD Performance,
fDATA = 150 MSPS, No Interpolation
Figure 28. Single-Tone Spurious Performance, fOUT = 10 MHz,
fDATA = 80 MSPS, Interpolation = 4×
0
0
–10
–20
AMPLITUDE (dBm)
–30
–40
–50
–60
–70
–40
–60
–80
–80
–90
0
50
100
150
200
250
300
FREQUENCY (MHz)
–100
Figure 26. Single-Tone Spurious Performance, fOUT = 10 MHz,
fDATA = 150 MSPS, Interpolation = 2×
0
5
10
15
20
FREQUENCY (MHz)
Figure 29. Two-Tone IMD Performance, fOUT = 10 MHz,
fDATA = 50 MSPS, Interpolation = 8×
Rev. C | Page 16 of 60
25
02706-029
–100
02706-026
AMPLITUDE (dBm)
–20
0
–10
–20
–20
–30
–30
AMPLITUDE (dBm)
0
–10
–40
–50
–60
–70
–40
–50
–60
–70
–80
–90
–90
–100
–100
0
100
200
300
400
FREQUENCY (MHz)
Figure 30. Single-Tone Spurious Performance, fOUT = 10 MHz,
fDATA = 50 MSPS, Interpolation = 8×
0
20
40
FREQUENCY (MHz)
60
80
02706-031
–80
02706-030
AMPLITUDE (dBm)
AD9777
Figure 31. Eight-Tone IMD Performance, fDATA = 160 MSPS, Interpolation = 8x
Rev. C | Page 17 of 60
AD9777
MODE CONTROL (VIA SPI PORT)
Table 9. Mode Control via SPI Port1
Address
00h
Bit 7
SDIO
Bidirectional
0 = Input
1 = I/O
Bit 6
LSB, MSB First
0 = MSB
1 = LSB
Bit 5
Software
Reset on
Logic 1
Bit 4
Sleep Mode
Logic 1
Shuts Down
the DAC
Output
Currents
Bit 3
Power-Down
Mode Logic 1
Shuts Down All
Digital and
Analog
Functions
01h
Filter
Interpolation
Rate
(1×, 2×,
4×, 8×)
Filter
Interpolation
Rate
(1×, 2×,
4×, 8×)
Modulation
Mode
(None, fS/2,
fS/4, fS/8)
Modulation
Mode
(None, fS/2,
fS/4, fS/8)
0 = No Zero
Stuffing on
Interpolation
Filters, Logic 1
Enables Zero
Stuffing
02h
0 = Signed
Input Data
1=
Unsigned
0 = Two-Port
Mode
1 = One-Port
Mode
DATACLK
Driver
Strength
DATACLK
Invert
0 = No
Invert
1 = Invert
03h
Data Rate2
Clock
Output
0 = PLL
OFF2
1 = PLL ON
04h
05h
IDAC
Fine Gain
Adjustment
08h
09h
IDAC Offset
Adjustment
Bit 9
IDAC IOFFSET
Direction
0 = IOFFSET on
IOUTA
1 = IOFFSET on
IOUTB
QDAC
Fine Gain
Adjustment
Bit 1
PLL_LOCK
Indicator
Bit 0
0 = e−jωt
1 = e+jωt
ONEPORTCLK
Invert
0 = No Invert
1 = Invert
IQSEL
Invert
0 = No
Invert
1 = Invert
PLL Divide
(Prescaler)
Ratio
PLL
Charge
Pump
Control
DATACLK/
PLL_LOCK2
Select
0=
PLL_LOCK
1=
DATACLK
Q First
0 = I First
1 = Q First
PLL Divide
(Prescaler)
Ratio
PLL
Charge
Pump
Control
PLL
Charge
Pump
Control
0 = Automatic
Charge Pump
Control
1=
Programmable
IDAC
Fine Gain
Adjustment
IDAC
Fine Gain
Adjustment
IDAC
Fine Gain
Adjustment
IDAC Offset
Adjustment
Bit 8
IDAC Offset
Adjustment
Bit 7
QDAC
Fine Gain
Adjustment
QDAC
Fine Gain
Adjustment
IDAC Offset
Adjustment
Bit 6
IDAC
Fine Gain
Adjustment
IDAC
Coarse Gain
Adjustment
IDAC Offset
Adjustment
Bit 5
IDAC
Fine Gain
Adjustment
IDAC
Coarse Gain
Adjustment
IDAC Offset
Adjustment
Bit 4
IDAC
Fine Gain
Adjustment
IDAC
Coarse Gain
Adjustment
IDAC Offset
Adjustment
Bit 3
IDAC Offset
Adjustment
Bit 1
IDAC
Fine Gain
Adjustment
IDAC
Coarse Gain
Adjustment
IDAC Offset
Adjustment
Bit 2
IDAC Offset
Adjustment
Bit 0
QDAC
Fine Gain
Adjustment
QDAC
Fine Gain
Adjustment
QDAC
Fine Gain
Adjustment
QDAC
Fine Gain
Adjustment
QDAC
Fine Gain
Adjustment
06h
07h
Bit 2
1R/2R Mode
DAC Output
Current Set by
One or Two
External
Resistors
0 = 2R, 1 = 1R
1 = Real Mix
Mode
0 = Complex
Mix Mode
Rev. C | Page 18 of 60
AD9777
Address
0Ah
Bit 7
Bit 6
Bit 5
Bit 4
0Bh
QDAC Offset
Adjustment
Bit 9
QDAC Offset
Adjustment
Bit 8
QDAC
Offset
Adjustment
Bit 7
QDAC Offset
Adjustment
Bit 6
0Ch
QDAC IOFFSET
Direction
0 = IOFFSET on
IOUTA
1 = IOFFSET on
IOUTB
0Dh
1
2
Bit 3
QDAC
Coarse Gain
Adjustment
QDAC Offset
Adjustment
Bit 5
Bit 2
QDAC
Coarse Gain
Adjustment
QDAC Offset
Adjustment
Bit 4
Bit 1
QDAC
Coarse Gain
Adjustment
QDAC
Offset
Adjustment
Bit 3
QDAC
Offset
Adjustment
Bit 1
Bit 0
QDAC
Coarse Gain
Adjustment
QDAC
Offset
Adjustment
Bit 2
QDAC
Offset
Adjustment
Bit 0
Version
Register
Version
Register
Version
Register
Version
Register
Default values are shown in bold.
For more information, see the Two Port Data Input Mode section.
Rev. C | Page 19 of 60
AD9777
REGISTER DESCRIPTION
Bit 3: Logic 1 enables zero stuffing mode for interpolation filters.
Address 00h
Bit 2: Default (1) enables the real mix mode. The I and Q data
channels are individually modulated by fS/2, fS/4, or fS/8 after
the interpolation filters. However, no complex modulation is
done. In the complex mix mode (Logic 0), the digital
modulators on the I and Q data channels are coupled to create a
digital complex modulator. When the AD9777 is applied in
conjunction with an external quadrature modulator, rejection
can be achieved of either the higher or lower frequency image
around the second IF frequency (that is, the LO of the analog
quadrature modulator external to the AD9777) according to the
bit value of Register 01h, Bit 1.
Bit 7: Logic 0 (default) causes the SPI_SDIO pin to act as an
input during the data transfer (Phase 2) of the communications
cycle. When set to 1, SPI_SDIO can act as an input or output,
depending on Bit 7 of the instruction byte.
Bit 6: Logic 0 (default). Determines the direction (LSB/MSB
first) of the communications and data transfer communications
cycles. Refer to the MSB/LSB Transfers section for more details.
Bit 5: Writing a 1 to this bit resets the registers to their default
values and restarts the chip. The RESET bit always reads back 0.
Register Address 00h bits are not cleared by this software reset.
However, a high level at the RESET pin forces all registers,
including those in Address 00h, to their default state.
Bit 4: Sleep Mode. A Logic 1 to this bit shuts down the DAC
output currents.
Bit 3: Power-Down. Logic 1 shuts down all analog and digital
functions except for the SPI port.
Bit 2: 1R/2R Mode. The default (0) places the AD9777 in two
resistor mode. In this mode, the IREF currents for the I and Q
DAC references are set separately by the RSET resistors on
FSADJ2 and FSADJ1 (Pins 59 and 60). In 2R mode, assuming
the coarse gain setting is full scale and the fine gain setting is 0,
IFULLSCALE1 = 32 × VREF/FSADJ1 and IFULLSCALE2 = 32 ×
VREF/FSADJ2. With this bit set to 1, the reference currents for
both I and Q DACs are controlled by a single resistor on Pin 60.
IFULLSCALE in one resistor mode for both the I and Q DACs is half
of what it would be in 2R mode, assuming all other conditions
(RSET, register settings) remain unchanged. The full-scale
current of each DAC can still be set to 20 mA by choosing a
resistor of half the value of the RSET value used in 2R mode.
Bit 1: PLL_LOCK Indicator. When the PLL is enabled, reading
this bit gives the status of the PLL. A Logic 1 indicates the PLL
is locked. A Logic 0 indicates an unlocked state.
Address 01h
Bit 7, Bit 6: This is the filter interpolation rate according to the
following table.
00
1×
01
2×
10
4×
11
8×
Bit 5 and Bit 4: This is the modulation mode according to the
following table.
00
none
01
fS/2
10
fS/4
11
fS/8
Bit 1: Logic 0 (default) causes the complex modulation to be of
the form e−jωt, resulting in the rejection of the higher frequency
image when the AD9777 is used with an external quadrature
modulator. A Logic 1 causes the modulation to be of the form
e+jωt, which causes rejection of the lower frequency image.
Bit 0: In two-port mode, a Logic 0 (default) causes Pin 8 to act
as a lock indicator for the internal PLL. A Logic 1 in this register
causes Pin 8 to act as a DATACLK. For more information, see
the Two Port Data Input Mode section.
Address 02h
Bit 7: Logic 0 (default) causes data to be accepted on the inputs
as twos complement binary. Logic 1 causes data to be accepted
as straight binary.
Bit 6: Logic 0 (default) places the AD9777 in two-port mode. I
and Q data enters the AD9777 via Ports 1 and 2, respectively. A
Logic 1 places the AD9777 in one-port mode in which
interleaved I and Q data is applied to Port 1. See Table 8 for
detailed information on how to use the DATACLK/PLL_LOCK,
IQSEL, and ONEPORTCLK modes.
Bit 5: DATACLK Driver Strength. With the internal PLL
disabled and this bit set to Logic 0, it is recommended that
DATACLK be buffered. When this bit is set to Logic 1,
DATACLK acts as a stronger driver capable of driving small
capacitive loads.
Bit 4: Logic 0 (default). A value of 1 inverts DATACLK at Pin 8.
Bit 2: Logic 0 (default). A value of 1 inverts ONEPORTCLK
at Pin 32.
Bit 1: Logic 0 (default) causes IQSEL = 0 to direct input data to
the I channel, while IQSEL = 1 directs input data to the Q
channel.
Bit 0: Logic 0 (default) defines IQ pairing as IQ, IQ…, while
programming a Logic 1 causes the pair ordering to
be QI, QI…
Rev. C | Page 20 of 60
AD9777
Address 03h
Address 05h, 09h
Bit 7: This allows the data rate clock (divided down from the
DAC clock) to be output at either the DATACLK/PLL_LOCK
pin (Pin 8) or at the SPI_SDO pin (Pin 53). The default of 0 in
this register enables the data rate clock at DATACLK/
PLL_LOCK, while a 1 in this register causes the data rate clock
to be output at SPI_SDO. For more information, see the Two
Port Data Input Mode section.
Bit 7, Bit 6, Bit 5, Bit 4, Bit 3, Bit 2, Bit 1, and Bit 0: These bits
represent an 8-bit binary number (Bit 7 MSB) that defines the
fine gain adjustment of the I (05h) and Q (09h) DAC according
to Equation 1.
Bit 1, Bit 0: Setting this divide ratio to a higher number allows
the VCO in the PLL to run at a high rate (for best performance)
while the DAC input and output clocks run substantially slower.
The divider ratio is set according to the following table.
Address 06h, 0Ah
Bit 3, Bit 2, Bit 1, and Bit 0: These bits represent a 4-bit binary
number (Bit 3 MSB) that defines the coarse gain adjustment of
the I (06h) and Q (0Ah) DACs according to Equation 1.
Address 07h, 0Bh
00
÷1
Bit 7, Bit 6, Bit 5, Bit 4, Bit 3, Bit 2, Bit 1, and Bit 0: These bits
are used in conjunction with Address 08h, 0Ch, Bits 1, 0.
01
÷2
Address 08h, 0Ch
10
÷4
11
÷8
Bit 1 and Bit 0: The 10 bits from these two address pairs (07h,
08h and 0Bh, 0Ch) represent a 10-bit binary number that
defines the offset adjustment of the I and Q DACs according to
Equation 1. (07h, 0Bh–Bit 7 MSB/08h, 0Ch–Bit 0 LSB).
Address 04h
Bit 7: Logic 0 (default) disables the internal PLL. Logic 1
enables the PLL.
Address 08h, 0Ch
Bit 6: Logic 0 (default) sets the charge pump control to
automatic. In this mode, the charge pump bias current is
controlled by the divider ratio defined in Address 03h, Bits 1
and 0. Logic 1 allows the user to manually define the charge
pump bias current using Address 04h, Bits 2, 1, and 0. Adjusting
the charge pump bias current allows the user to optimize the
noise/settling performance of the PLL.
Bit 2, Bit 1, Bit 0: With the charge pump control set to manual,
these bits define the charge pump bias current according to the
following table.
000
50 µA
001
100 µA
010
200 µA
011
400 µA
111
800 µA
⎡⎛ 6 × I REF
I OUTA = ⎢⎜
⎣⎝ 8
Bit 7: This bit determines the direction of the offset of the I
(08h) and Q (0Ch) DACs. A Logic 0 applies a positive offset
current to IOUTA, while a Logic 1 applies a positive offset current
to IOUTB. The magnitude of the offset current is defined by the
bits in Addresses 07h, 0Bh, 08h, 0Ch according to Equation 1.
Equation 1 shows IOUTA and IOUTB as a function of fine gain,
coarse gain, and offset adjustment when using 2R mode. In 1R
mode, the current IREF is created by a single FSADJ resistor
(Pin 60). This current is divided equally into each channel so that
a scaling factor of one-half must be added to these equations for
full-scale currents for both DACs and the offset.
⎞⎛ COARSE + 1 ⎞ ⎛ 3 × I REF ⎞⎛ FINE ⎞⎤ ⎡⎛ 1024 ⎞⎛ DATA ⎞⎤
⎟−⎜
⎟⎥ × ⎢⎜
⎟⎜
⎟⎜
⎟⎜ 16 ⎟⎥( A)
16
⎠⎝
⎠ ⎝ 32 ⎠⎝ 256 ⎠⎦ ⎣⎝ 24 ⎠⎝ 2
⎠⎦
⎡⎛ 6 × I REF ⎞⎛ COARSE + 1 ⎞ ⎛ 3 × I REF ⎞⎛ FINE ⎞⎤ ⎡⎛ 1024 ⎞⎛ 216 − DATA − 1 ⎞⎤
⎟⎟⎥( A)
I OUTB = ⎢⎜
⎟−⎜
⎟⎜
⎟⎜
⎟⎥ × ⎢⎜
⎟⎜
16
216
⎠ ⎝ 32 ⎠⎝ 256 ⎠⎦ ⎣⎢⎝ 24 ⎠⎜⎝
⎣⎝ 8 ⎠⎝
⎠⎦⎥
⎛ OFFSET ⎞
I OFFSET = 4 × I REF ⎜
⎟( A)
⎝ 1024 ⎠
Rev. C | Page 21 of 60
(1)
AD9777
FUNCTIONAL DESCRIPTION
The AD9777 dual interpolating DAC consists of two data
channels that can be operated independently or coupled to form
a complex modulator in an image reject transmit architecture.
Each channel includes three FIR filters, making the AD9777
capable of 2×, 4×, or 8× interpolation. High speed input and
output data rates can be achieved within the following
limitations.
Interpolation Rate
(MSPS)
1×
2×
4×
8×
Input Data Rate
(MSPS)
160
160
100
50
DAC Sample Rate
(MSPS)
160
320
400
400
Both data channels contain a digital modulator capable of
mixing the data stream with an LO of fDAC/2, fDAC/4, or fDAC/8,
where fDAC is the output data rate of the DAC. A zero stuffing
feature is also included and can be used to improve pass-band
flatness for signals being attenuated by the SIN(x)/x
characteristic of the DAC output. The speed of the AD9777,
combined with its digital modulation capability, enables direct
IF conversion architectures at 70 MHz and higher.
The digital modulators on the AD9777 can be coupled to form
a complex modulator. By using this feature with an external
analog quadrature modulator, such as the Analog Devices
AD8345, an image rejection architecture can be enabled. To
optimize the image rejection capability, as well as LO feedthrough in this architecture, the AD9777 offers programmable
(via the SPI port) gain and offset adjust for each DAC.
Also included on the AD9777 are a phase-locked loop (PLL)
clock multiplier and a 1.20 V band gap voltage reference. With
the PLL enabled, a clock applied to the CLK+/CLK− inputs is
frequency multiplied internally and generates all necessary
internal synchronization clocks. Each 16-bit DAC provides two
complementary current outputs whose full-scale currents can
be determined either from a single external resistor or independently from two separate resistors (see the 1R/2R Mode
section). The AD9777 features a low jitter, differential clock
input that provides excellent noise rejection while accepting a
sine or square wave input. Separate voltage supply inputs are
provided for each functional block to ensure optimum noise
and distortion performance.
Sleep and power-down modes can be used to turn off the DAC
output current (sleep) or the entire digital and analog sections
(power-down) of the chip. A SPI-compliant serial port is used
to program the many features of the AD9777. Note that in
power-down mode, the SPI port is the only section of the chip
still active.
SDO (PIN 53)
AD9777 SPI PORT
INTERFACE
02706-032
SDIO (PIN 54)
SPI_CLK (PIN 55)
CSB (PIN 56)
Figure 32. SPI Port Interface
SERIAL INTERFACE FOR REGISTER CONTROL
The AD9777 serial port is a flexible, synchronous serial
communications port that allows easy interface to many
industry-standard microcontrollers and microprocessors. The
serial I/O is compatible with most synchronous transfer
formats, including both the Motorola SPI® and Intel® SSR
protocols. The interface allows read/write access to all registers
that configure the AD9777. Single- or multiple-byte transfers
are supported as well as MSB first or LSB first transfer formats.
The AD9777’s serial interface port can be configured as a single
pin I/O (SDIO) or two unidirectional pins for I/O (SDIO/SDO).
GENERAL OPERATION OF THE SERIAL INTERFACE
There are two phases to a communication cycle with the
AD9777. Phase 1 is the instruction cycle, which is the writing of
an instruction byte into the AD9777 coincident with the first
eight SCLK rising edges. The instruction byte provides the
AD9777 serial port controller with information regarding the
data transfer cycle, which is Phase 2 of the communication
cycle. The Phase 1 instruction byte defines whether the upcoming data transfer is read or write, the number of bytes in the
data transfer, and the starting register address for the first byte
of the data transfer. The first eight SCLK rising edges of each
communication cycle are used to write the instruction byte into
the AD9777.
A Logic 1 on the SPI_CSB pin, followed by a logic low, resets
the SPI port timing to the initial state of the instruction cycle.
This is true regardless of the present state of the internal
registers or the other signal levels present at the inputs to the
SPI port. If the SPI port is in the middle of an instruction cycle
or a data transfer cycle, none of the present data is written.
The remaining SCLK edges are for Phase 2 of the
communication cycle. Phase 2 is the actual data transfer
between the AD9777 and the system controller. Phase 2 of the
communication cycle is a transfer of one to four data bytes, as
determined by the instruction byte. Normally, using one multibyte transfer is the preferred method. However, single byte data
transfers are useful to reduce CPU overhead when register
access requires one byte only. Registers change immediately
upon writing to the last bit of each transfer byte.
Rev. C | Page 22 of 60
AD9777
INSTRUCTION BYTE
SPI_CSB (Pin 56)—Chip Select
The instruction byte contains the information shown below.
Active low input starts and gates a communication cycle. It
allows more than one device to be used on the same serial
communications lines. The SPI_SDO and SPI_SDIO pins go to
a high impedance state when this input is high. Chip select
should stay low during the entire communication cycle.
N1
0
0
1
1
N0
0
1
0
1
Description
Transfer 1 Byte
Transfer 2 Bytes
Transfer 3 Bytes
Transfer 4 Bytes
SPI_SDIO (Pin 54)—Serial Data I/O
R/W
Bit 7 of the instruction byte determines whether a read or a
write data transfer occurs after the instruction byte write. Logic
1 indicates read operation. Logic 0 indicates a write operation.
N1, N0
Bit 6 and Bit 5 of the instruction byte determine the number of
bytes to be transferred during the data transfer cycle. The bit
decodes are shown in the following table.
MSB
I7
R/W
I6
N1
I5
N0
I4
A4
I3
A3
I2
A2
I1
A1
LSB
I0
A0
A4, A3, A2, A1, A0
Bit 4, Bit 3, Bit 2, Bit 1, and Bit 0 of the instruction byte
determine which register is accessed during the data transfer
portion of the communications cycle. For multibyte transfers,
this address is the starting byte address. The remaining register
addresses are generated by the AD9777.
SERIAL INTERFACE PORT PIN DESCRIPTIONS
SPI_CLK (Pin 55)—Serial Clock
The serial clock pin is used to synchronize data to and from the
AD9777 and to run the internal state machines. SPI_CLK
maximum frequency is 15 MHz. All data input to the AD9777
is registered on the rising edge of SPI_CLK. All data is driven
out of the AD9777 on the falling edge of SCLK.
Data is always written into the AD9777 on this pin. However,
this pin can be used as a bidirectional data line. Bit 7 of Register
Address 00h controls the configuration of this pin. The default
is Logic 0, which configures the SPI_SDIO pin as
unidirectional.
SPI_SDO (Pin 53)—Serial Data Out
Data is read from this pin for protocols that use separate lines
for transmitting and receiving data. In the case where the
AD9777 operates in a single bidirectional I/O mode, this pin
does not output data and is set to a high impedance state.
MSB/LSB TRANSFERS
The AD9777 serial port can support both most significant bit
(MSB) first or least significant bit (LSB) first data formats. This
functionality is controlled by the LSB first bit in Register 0. The
default is MSB first.
When this bit is set active high, the AD9777 serial port is in LSB
first format. In LSB first mode, the instruction byte and data
bytes must be written from LSB to MSB. In LSB first mode, the
serial port internal byte address generator increments for each
byte of the multibyte communication cycle.
When this bit is set default low, the AD9777 serial port is in
MSB first format. In MSB first mode, the instruction byte and
data bytes must be written from MSB to LSB. In MSB first
mode, the serial port internal byte address generator decrements for each byte of the multibyte communication cycle.
When incrementing from 1Fh, the address generator changes to
00h. When decrementing from 00h, the address generator
changes to 1Fh.
Rev. C | Page 23 of 60
AD9777
INSTRUCTION CYCLE
DATA TRANSFER CYCLE
CS
SDIO
R/W
I6(N)
I5(N)
I4
I3
I2
I1
I0
SDO
D7N
D6N
D20
D10
D00
D7N
D6N
D20
D10
D00
02706-033
SCLK
Figure 33. Serial Register Interface Timing MSB First
INSTRUCTION CYCLE
DATA TRANSFER CYCLE
CS
I0
I1
I2
I3
I4
I5(N)
I6(N)
R/W
SDO
D00
D10
D20
D6N
D7N
D00
D10
D20
D6N
D7N
Figure 34. Serial Register Interface Timing LSB First
tSCLK
tDS
CS
tPWH
tPWL
SCLK
SDIO
tDH
INSTRUCTION BIT 7
T6
INSTRUCTION BIT
02706-035
tDS
Figure 35. Timing Diagram for Register Write to AD9777
CS
SCLK
tDV
SDIO
DATA BIT N
DATA BIT N–1
SDO
Figure 36. Timing Diagram for Register Read from AD9777
Rev. C | Page 24 of 60
02706-036
SDIO
02706-034
SCLK
AD9777
The AD9777 serial port configuration bits reside in Bit 6 and
Bit 7 of Register Address 00h. It is important to note that the
configuration changes immediately upon writing to the last bit
of the register. For multibyte transfers, writing to this register
may occur during the middle of the communication cycle. Care
must be taken to compensate for this new configuration for the
remaining bytes of the current communication cycle.
The same considerations apply to setting the reset bit in
Register Address 00h. All other registers are set to their default
values, but the software reset does not affect the bits in Register
Address 00h.
It is recommended to use only single byte transfers when
changing serial port configurations or initiating a software
reset.
The offset control defines a small current that can be added to
IOUTA or IOUTB (not both) on the IDAC and QDAC. The selection
of which IOUT this offset current is directed toward is programmable via Register 08h, Bit 7 (IDAC) and Register 0Ch, Bit 7
(QDAC). Figure 42 shows the scale of the offset current that can
be added to one of the complementary outputs on the IDAC
and QDAC. Offset control can be used for suppression of LO
leakage resulting from modulation of dc signal components. If
the AD9777 is dc-coupled to an external modulator, this feature
can be used to cancel the output offset on the AD9777 as well as
the input offset on the modulator. Figure 42 shows a typical
example of the effect that the offset control has on LO
suppression.
GAIN
CONTROL
REGISTERS
A write to Bit 1, Bit 2, and Bit 3 of Address 00h with the same
logic levels as for Bit 7, Bit 6, and Bit 5 (bit pattern: XY1001YX
binary) allows the user to reprogram a lost serial port configuration and to reset the registers to their default values. A second
write to Address 00h with reset bit low and serial port configuration as specified above (XY) reprograms the OSC IN
multiplier setting. A changed fSYSCLK frequency is stable after a
maximum of 200 fMCLK cycles (equals wake-up time).
DAC OPERATION
OFFSET
CONTROL OFFSET
DAC
REGISTERS
FINE
GAIN
DAC
FINE
GAIN
DAC
1.2VREF
IDAC
IOUTA1
IOUTB1
REFIO
COARSE
GAIN
DAC
COARSE
GAIN
DAC
0.1µF
QDAC
IOUTA2
IOUTB2
FSADJ1
OFFSET
OFFSET
CONTROL
DAC
GAIN
REGISTERS
CONTROL
REGISTERS
FSADJ2
RSET1
RSET2
02706-037
NOTES ON SERIAL PORT OPERATION
Figure 37. DAC Outputs, Reference Current Scaling, and Gain/Offset Adjust
84µA
REFIO
7kΩ
02706-038
0.7V
Figure 38. Internal Reference Equivalent Circuit
25
20
2R MODE
15
10
1R MODE
5
0
0
5
10
15
COARSE GAIN REGISTER CODE
(ASSUMING RSET1, RSET2 = 1.9kΩ)
Figure 39. Coarse Gain Effect on IFULLSCALE
Rev. C | Page 25 of 60
20
02706-039
The fine adjustment of the gain of each channel allows for improved balance of QAM modulated signals, resulting in improved
modulation accuracy and image rejection. In the Interfacing with
the AD8345 Quadrature Modulator section, the performance
data shows to what degree image rejection can be improved when
the AD9777 is used with an AD8345 quadrature modulator from
Analog Devices, Inc.
AVDD
COARSE REFERENCE CURRENT (mA)
The dual 16-bit DAC output of the AD9777, along with the
reference circuitry, gain, and offset registers, is shown in Figure
37 and Figure 38. Note that an external reference can be used by
simply overdriving the internal reference with the external
reference. Referring to the transfer functions in Equation 1, a
reference current is set by the internal 1.2 V reference, the
external RSET resistor, and the values in the coarse gain register.
The fine gain DAC subtracts a small amount from this and the
result is input to IDAC and QDAC, where it is scaled by an
amount equal to 1024/24. Figure 39 and Figure 40 show the
scaling effect of the coarse and fine adjust DACs. IDAC and
QDAC are PMOS current source arrays, segmented in a 5-4-7
configuration. The five MSB control an array of 31 current
sources. The next four bits consist of 15 current sources whose
values are all equal to 1/16 of an MSB current source. The 7
LSBs are binary weighted fractions of the middle bits’ current
sources. All current sources are switched to either IOUTA or IOUTB,
depending on the input code.
AD9777
0
–10
–0.5
OFFSET REGISTER 1 ADJUSTED
1R MODE
–1.0
2R MODE
–1.5
–2.0
–20
–30
–40
–50
–60
–2.5
OFFSET REGISTER 2
ADJUSTED, WITH OFFSET
REGISTER 1 SET
TO OPTIMIZED VALUE
–70
–3.0
200
400
600
800
–80
–1024
1000
02706-040
0
FINE GAIN REGISTER CODE
(ASSUMING RSET1, RSET2 = 1.9kΩ)
–512
–256
0
256
512
768
1024
DAC1, DAC2 (OFFSET REGISTER CODES)
Figure 42. Offset Adjust Control, Effect on LO Suppression
Figure 40. Fine Gain Effect on IFULLSCALE
In Figure 42, the negative scale represents an offset added to
IOUTB, while the positive scale represents an offset added to IOUTA
of the respective DAC. Offset Register 1 corresponds to IDAC,
while Offset Register 2 corresponds to QDAC. Figure 42
represents the AD9777 synthesizing a complex signal that is
then dc-coupled to an AD8345 quadrature modulator with an
LO of 800 MHz. The dc coupling allows the input offset of the
AD8345 to be calibrated out as well. The LO suppression at the
AD8345 output was optimized first by adjusting Offset
Register 1 in the AD9777. When an optimal point was found
(roughly Code 54), this code was held in Offset Register 1, and
Offset Register 2 was adjusted. The resulting LO suppression is
70 dBFS. These are typical numbers, and the specific code for
optimization varies from part to part.
1R/2R MODE
In 2R mode, the reference current for each channel is set
independently by the FSADJ resistor on that channel. The
AD9777 can be programmed to derive its reference current
from a single resistor on Pin 60 by putting the part into 1R
mode. The transfer functions in Equation 1 are valid for 2R
mode. In 1R mode, the current developed in the single FSADJ
resistor is split equally between the two channels. The result is
that in 1R mode, a scale factor of 1/2 must be applied to the
formulas in Equation 1. The full-scale DAC current in 1R mode
can still be set to as high as 20 mA by using the internal 1.2 V
reference and a 950 Ω resistor instead of the 1.9 kΩ resistor
typically used in 2R mode.
CLOCK INPUT CONFIGURATION
5
The clock inputs to the AD9777 can be driven differentially or
single-ended. The internal clock circuitry has supply and
ground (CLKVDD, CLKGND) separate from the other supplies
on the chip to minimize jitter from internal noise sources.
4
3
Figure 43 shows the AD9777 driven from a single-ended clock
source. The CLK+/CLK− pins form a differential input
(CLKIN) so that the statically terminated input must be dcbiased to the midswing voltage level of the clock driven input.
2R MODE
2
1R MODE
1
AD9777
RSERIES
0
200
400
600
800
COARSE GAIN REGISTER CODE
(ASSUMING RSET1, RSET2 = 1.9kΩ)
1000
CLK+
Figure 41. DAC Output Offset Current
CLKVDD
CLK–
VTHRESHOLD
0.1µF
CLKGND
02706-043
0
02706-041
OFFSET CURRENT (mA)
–768
02706-042
LO SUPPRESSION (dBFS)
FINE REFERENCE CURRENT (mA)
0
Figure 43. Single-Ended Clock Driving Clock Inputs
Rev. C | Page 26 of 60
AD9777
AD9777
0.1µF
1kΩ
CLK+
1kΩ
ECL/PECL
0.1µF
0.1µF
CLKVDD
1kΩ
CLK–
CLKGND
02706-044
1kΩ
Figure 44. Differential Clock Driving Clock Inputs
A transformer, such as the T1-1T from Mini-Circuits, can also
be used to convert a single-ended clock to differential. This
method is used on the AD9777 evaluation board so that an
external sine wave with no dc offset can be used as a differential
clock.
PECL/ECL drivers require varying termination networks, the
details of which are left out of Figure 43 and Figure 44 but can
be found in application notes such as AND8020/D from On
Semiconductor. These networks depend on the assumed
transmission line impedance and power supply voltage of the
clock driver. Optimum performance of the AD9777 is achieved
when the driver is placed very close to the AD9777 clock inputs,
thereby negating any transmission line effects such as reflections due to mismatch.
The quality of the clock and data input signals is important in
achieving optimum performance. The external clock driver
circuitry should provide the AD9777 with a low jitter clock
input that meets the minimum/maximum logic levels while
providing fast edges. Although fast clock edges help minimize
any jitter that manifests itself as phase noise on a reconstructed
waveform, the high gain bandwidth product of the AD9777’s
clock input comparator can tolerate differential sine wave inputs
as low as 0.5 V p-p with minimal degradation of the output
noise floor.
PROGRAMMABLE PLL
CLKIN can function either as an input data rate clock (PLL
enabled) or as a DAC data rate clock (PLL disabled) according
to the state of Address 02h, Bit 7 in the SPI port register. The
internal operation of the AD9777 clock circuitry in these two
modes is illustrated in Figure 45 and Figure 46.
The PLL clock multiplier and distribution circuitry produce the
necessary internal synchronized 1×, 2×, 4×, and 8× clocks for
the rising edge triggered latches, interpolation filters,
modulators, and DACs. This circuitry consists of a phase
detector, charge pump, voltage controlled oscillator (VCO),
prescaler, clock distribution, and SPI port control. The charge
pump, VCO, differential clock input buffer, phase detector,
prescaler, and clock distribution are all powered from
CLKVDD. PLL lock status is indicated by the logic signal at the
PLL_LOCK pin, as well as by the status of Bit 1, Register 00h.
To ensure optimum phase noise performance from the PLL
clock multiplier and distribution, CLKVDD should originate
from a clean analog supply. Table 10 defines the minimum
input data rates versus the interpolation and PLL divider
setting. If the input data rate drops below the defined minimum
under these conditions, VCO phase noise can increase
significantly. The VCO speed is a function of the input data
rate, the interpolation rate, and the VCO prescaler, according to
the following function:
VCO Speed ( MHz ) =
Input Data Rate ( MHz ) × Interpolation Rate × Prescaler
CLK+
CLK–
PLLVDD
PLL_LOCK
1 = LOCK
0 = NO LOCK
INTERPOLATION
FILTERS,
MODULATORS,
AND DACS
2
4
AD9777
PHASE
DETECTOR
CHARGE
PUMP
PRESCALER
VCO
LPF
8
1
CLOCK
DISTRIBUTION
CIRCUITRY
INPUT
DATA
LATCHES
INTERPOLATION
RATE
CONTROL
PLL DIVIDER
(PRESCALER)
CONTROL
INTERNAL SPI
CONTROL
REGISTERS
SPI PORT
MODULATION
RATE
CONTROL
PLL
CONTROL
(PLL ON)
Figure 45. PLL and Clock Circuitry with PLL Enabled
Rev. C | Page 27 of 60
02706-045
A configuration for differentially driving the clock inputs is
given in Figure 44. DC-blocking capacitors can be used to
couple a clock driver output whose voltage swings exceed
CLKVDD or CLKGND. If the driver voltage swings are within
the supply range of the AD9777, the dc-blocking capacitors and
bias resistors are not necessary.
AD9777
CLK+
PLL_LOCK
1 = LOCK
0 = NO LOCK
PHASE
DETECTOR
CHARGE
PUMP
8
CLOCK
DISTRIBUTION
CIRCUITRY
PRESCALER
VCO
–20
MODULATION
RATE
CONTROL
PLL
CONTROL
(PLL ON)
Figure 46. PLL and Clock Circuitry with PLL Disabled
In addition, if the zero stuffing option is enabled, the VCO doubles its
speed again. Phase noise can be slightly higher with the PLL enabled.
Figure 47 illustrates typical phase noise performance of the AD9777
with 2× interpolation and various input data rates. The signal
synthesized for the phase noise measurement was a single carrier at a
frequency of fDATA/4. The repetitive nature of this signal eliminates
quantization noise and distortion spurs as a factor in the measurement. Although the curves blend in Figure 47, the different
conditions are given for clarity in the table preceding Figure 47.
Figure 47 also contains a table detailing PLL divider settings vs.
interpolation rate and maximum and minimum fDATA rates. Note that
maximum fDATA rates of 160 MSPS are due to the maximum input
data rate of the AD9777. However, maximum rates of less than 160
MSPS and all minimum fDATA rates are due to maximum and minimum speeds of the internal PLL VCO. Figure 48 shows typical
performance of the PLL lock signal (Pin 8 or Pin 53) when the PLL is
in the process of locking.
–30
PHASE NOISE (dBFS)
PLL DIVIDER
(PRESCALER)
CONTROL
INTERNAL SPI
CONTROL
REGISTERS
SPI PORT
Div 1
Div 2
Div 2
Div 4
–10
02706-046
INTERPOLATION
RATE
CONTROL
Prescaler Ratio
0
1
INPUT
DATA
LATCHES
PLL
Disabled
Enabled
Enabled
Enabled
Enabled
–40
–50
–60
–70
–80
–90
–100
–110
0
1
2
3
4
5
FREQUENCY OFFSET (MHz)
02706-047
4
fDATA (MSPS)
125
125
100
75
50
AD9777
INTERPOLATION
FILTERS,
MODULATORS,
AND DACS
2
Table 11. Required PLL Prescaler Ration vs. fDATA
CLK–
Figure 47. Phase Noise Performance
Interpolation
Rate
1
1
1
1
2
2
2
2
4
4
4
4
8
8
8
8
Divider
Setting
1
2
4
8
1
2
4
8
1
2
4
8
1
2
4
8
Minimum
fDATA
32
16
8
4
24
12
6
3
24
12
6
3
24
12
6
3
Maximum
fDATA
160
160
112
56
160
112
56
28
100
56
28
14
50
28
14
7
02706-048
Table 10. PLL Optimization
Figure 48. PLL_LOCK Output Signal (Pin 8) in the Process of Locking
(Typical Lock Time)
It is important to note that the resistor/capacitor needed for the
PLL loop filter is internal on the AD9777. This suffices unless
the input data rate is below 10 MHz, in which case an external
series RC is required between the LPF and CLKVDD pins.
Rev. C | Page 28 of 60
AD9777
35
POWER DISSIPATION
8×
400
8×, (MOD. ON)
IDVDD (mA)
300
8×
4×
2×
200
150
100
1×
50
50
100
150
200
fDATA (MHz)
02706-049
0
0
Figure 49. IDVDD vs. fDATA vs. Interpolation Rate, PLL Disabled
20
15
1×
10
5
0
50
100
150
200
fDATA (MHz)
02706-051
0
Figure 51. ICLKVDD vs. fDATA vs. Interpolation Rate, PLL Disabled
The AD9777 provides two methods for programmable
reduction in power savings. The sleep mode, when activated,
turns off the DAC output currents but the rest of the chip
remains functioning. When coming out of sleep mode, the
AD9777 immediately returns to full operation. Power-down
mode, on the other hand, turns off all analog and digital
circuitry in the AD9777 except for the SPI port. When
returning from power-down mode, enough clock cycles must
be allowed to flush the digital filters of random data acquired
during the power-down cycle. Note that optimal performance
with the PLL enabled is achieved with the UCO in the PLL
control loop running at 450 MHz to 550 MHz.
TWO PORT DATA INPUT MODE
76.0
4×, (MOD. ON)
8×, (MOD. ON)
75.5
2×, (MOD. ON)
75.0
74.5
4×
8×
74.0
73.5
2×
73.0
1×
72.5
72.0
0
50
100
150
200
fDATA (MHz)
Figure 50. IAVDD vs. fDATA vs. Interpolation Rate, PLL Disabled
02706-050
IAVDD (mA)
2×
(Control Register 00h, Bit 3 and Bit 4)
4×, (MOD. ON)
250
4×
25
SLEEP/POWER-DOWN MODES
2×, (MOD. ON)
350
30
ICLKVDD (mA)
The AD9777 has three voltage supplies: DVDD, AVDD, and
CLKVDD. Figure 49, Figure 50, and Figure 51 show the current
required from each of these supplies when each is set to the
3.3 V nominal specified for the AD9777. Power dissipation (PD)
can easily be extracted by multiplying the given curves by 3.3.
As Figure 49 shows, IDVDD is very dependent on the input data
rate, the interpolation rate, and the activation of the internal
digital modulator. IDVDD, however, is relatively insensitive to the
modulation rate by itself. In Figure 50, IAVDD shows the same
type of sensitivity to data, interpolation rate, and the modulator
function but to a much lesser degree (<10%). In Figure 51,
ICLKVDD varies over a wide range yet is responsible for only a
small percentage of the overall AD9777 supply current
requirement.
The digital data input ports can be configured as two
independent ports or as a single (one-port mode) port. In the
two-port mode, data at the two input ports is latched into the
AD9777 on every rising edge of the data rate clock (DATACLK).
In addition, in the two-port mode, the AD9777 can be
programmed to generate an externally available DATACLK for
the purpose of data synchronization. This data rate clock can be
programmed to be available at either Pin 8 (DATACLK/
PLL_LOCK) or Pin 53 (SPI_SDO). Because Pin 8 can also
function as a PLL lock indicator when the PLL is enabled, there
are several options for configuring Pin 8 and Pin 53. The
following information describes these options.
PLL Off (Register 4, Bit 7 = 0)
Register 3, Bit 7 = 0; DATACLK out of Pin 8.
Register 3, Bit 7 = 1; DATACLK out of Pin 53.
PLL On (Register 4, Bit 7 = 1)
Register 3, Bit 7 = 0, Register 1, Bit 0 = 0; PLL lock indicator out
of Pin 8.
Rev. C | Page 29 of 60
AD9777
tOD
Register 3, Bit 7 = 1, Register 1, Bit 0 = 0; PLL lock indicator out
of Pin 53.
Register 3, Bit 7 = 0, Register 1, Bit 0 = 1; DATACLK out of Pin 8.
Register 3, Bit 7 = 1, Register 1, Bit 0 = 1; DATACLK out of Pin 53.
Test configurations showing the various clocks that are required
and generated by the AD9777 with the PLL enabled/disabled
and in the one-port/two-port modes are given in Figure 101 to
Figure 104. Jumper positions needed to operate the AD9777
evaluation board in these modes are given as well.
DATACLK
DATA AT PORTS
1 AND 2
tS
tS = 0.0ns (MAX)
tH = 2.5ns (MAX)
tH
02706-052
In one-port mode, P2B14 and P2B15 from input data port two
are redefined as IQSEL and ONEPORTCLK, respectively. The
input data in one-port mode is steered to one of the two internal data channels based on the logic level of IQSEL. A clock
signal, ONEPORTCLK, is generated by the AD9777 in this
mode for the purpose of data synchronization. ONEPORTCLK
runs at the input interleaved data rate, which is 2× the data rate
at the internal input to either channel.
CLKIN
Figure 52. Timing Requirements in Two-Port Input Mode with PLL Enabled
DATACLK DRIVER STRENGTH
(Control Register 02h, Bit 5)
With the phase-locked loop (PLL) enabled and the AD9777 in
two-port mode, the speed of CLKIN is inherently that of the
input data rate. In two-port mode, Pin 8 (DATACLK/PLL_
LOCK) can be programmed (Control Register 01h, Bit 0) to
function as either a lock indicator for the internal PLL or as a
clock running at the input data rate. When Pin 8 is used as a
clock output (DATACLK), its frequency is equal to that of
CLKIN. Data at the input ports is latched into the AD9777 on
the rising edge of the CLKIN. Figure 52 shows the delay, tOD,
inherent between the rising edge of CLKIN and the rising edge
of DATACLK, as well as the setup and hold requirements for
the data at Ports 1 and 2. The setup and hold times given in Figure
52 are the input data transitions with respect to CLKIN. Note
that in two-port mode (PLL enabled or disabled), the data rate
at the interpolation filter inputs is the same as the input data
rate at Port 1 and Port 2.
The DAC output sample rate in two-port mode is equal to the
clock input rate multiplied by the interpolation rate. If zero
stuffing is used, another factor of 2 must be included to
calculate the DAC sample rate.
DATACLK INVERSION
The DATACLK output driver strength is capable of driving
>10 mA into a 330 Ω load while providing a rise time of 3 ns.
Figure 53 shows DATACLK driving a 330 Ω resistive load at a
frequency of 50 MHz. By enabling the drive strength option
(Control Register 02h, Bit 5), the amplitude of DATACLK under
these conditions increases by approximately 200 mV.
3.0
2.5
2.0
1.5
1.0
0.5
0
DELTA APPROX. 2.8ns
–0.5
0
10
20
30
40
50
TIME (ns)
02706-053
(Control Register 02h, Bits 6 to 0 and 04h, Bits 7 to 1)
AMPLITUDE (V)
PLL ENABLED, TWO-PORT MODE
Figure 53. DATACLK Driver Capability into 330 Ω at 50 MHz
PLL ENABLED, ONE-PORT MODE
(Control Register 02h, Bits 6 to 1 and 04h, Bits 7 to 1)
(Control Register 02h, Bit 4)
By programming this bit, the DATACLK signal shown in Figure 53
can be inverted. With inversion enabled, tOD refers to the time
between the rising edge of CLKIN and the falling edge of
DATACLK. No other effect on timing occurs.
In one-port mode, the I and Q channels receive their data from
an interleaved stream at digital input Port 1. The function of
Pin 32 is defined as an output (ONEPORTCLK) that generates a
clock at the interleaved data rate, which is 2× the internal input
data rate of the I and Q channels. The frequency of CLKIN is
equal to the internal input data rate of the I and Q channels.
Rev. C | Page 30 of 60
AD9777
The selection of the data for the I or Q channel is determined by
the state of the logic level at Pin 31 (IQSEL when the AD9777 is
in one-port mode) on the rising edge of ONEPORTCLK.
Under these conditions, IQSEL = 0 latches the data into the I
channel on the clock rising edge, while IQSEL = 1 latches the
data into the Q channel. It is possible to invert the I and Q
selection by setting Control Register 02h, Bit 1 to the invert
state (Logic 1). Figure 54 illustrates the timing requirements for
the data inputs as well as the IQSEL input. Note that the 1×
interpolation rate is not available in one-port mode.
The DAC output sample rate in one-port mode is equal to
CLKIN multiplied by the interpolation rate. If zero stuffing is
used, another factor of 2 must be included to calculate the DAC
sample rate.
(Control Register 02h, Bit 0)
In one-port mode, the interleaved data is latched into the
AD9777 internal I and Q channels in pairs. The order of how
the pairs are latched internally is defined by this control register.
The following is an example of the effect this has on incoming
interleaved data.
Given the following interleaved data stream, where the data
indicates the value with respect to full scale:
I
0.5
Q
0.5
(Control Register 02h, Bit 2)
I
1
Q
1
I
0.5
Q
0.5
I
0
Q
0
I
0.5
Q
0.5
With the control register set to 0 (I first), the data appears at the
internal channel inputs in the following order in time:
I Channel
Q Channel
ONEPORTCLK INVERSION
0.5
0.5
1
1
0.5
0.5
0
0
0.5
0.5
With the control register set to 1 (Q first), the data appears at
the internal channel inputs in the following order in time:
By programming this bit, the ONEPORTCLK signal shown in
Figure 54 can be inverted. With inversion enabled, tOD refers to
the delay between the rising edge of the external clock and the
falling edge of ONEPORTCLK. The setup and hold times, tS
and tH, are with respect to the falling edge of ONEPORTCLK.
There is no other effect on timing.
I Channel
Q Channel
0.5
y
1
0.5
0.5
1
0
0.5
0.5
0
x
0.5
The values x and y represent the next I value and the previous Q
value in the series.
PLL DISABLED, TWO-PORT MODE
ONEPORTCLK DRIVER STRENGTH
With the PLL disabled, a clock at the DAC output rate must be
applied to CLKIN. Internal clock dividers in the AD9777 synthesize the DATACLK signal at Pin 8, which runs at the input
data rate and can be used to synchronize the input data. Data is
latched into input Port 1 and Port 2 of the AD9777 on the rising
edge of DATACLK. DATACLK speed is defined as the speed of
CLKIN divided by the interpolation rate. With zero stuffing enabled, this division increases by a factor of 2. Figure 55 illustrates
the delay between the rising edge of CLKIN and the rising edge
of DATACLK, as well as tS and tH in this mode.
The drive capability of ONEPORTCLK is identical to that of
DATACLK in the two-port mode. Refer to Figure 53 for
performance under load conditions.
tOD
tOD = 4.0ns (MIN)
TO 5.5ns (MAX)
CLKIN
IQ PAIRING
tS = 3.0ns (MAX)
tH = –0.5ns (MAX)
tIQS = 3.5ns (MAX)
tIQH = –1.5ns (MAX
ONEPORTCLK
The programmable modes DATACLK inversion and DATACLK
driver strength described in the PLL Enabled, Two-Port Mode
section have identical functionality with the PLL disabled.
The data rate CLK created by dividing down the DAC clock in
this mode can be programmed (via Register x03h, Bit 7) to be
output from the SPI_SDO pin, rather than the DATACLK pin.
In some applications, this may improve complex image rejection. tOD increases by 1.6 ns when SPI_SDO is used as data rate
clock out.
I AND Q INTERLEAVED
INPUT DATA AT PORT 1
tS tH
tIQS
tIQH
02706-054
IQSEL
Figure 54. Timing Requirements in One-Port
Input Mode, with the PLL Enabled
Rev. C | Page 31 of 60
AD9777
tOD
tOD
CLKIN
CLKIN
DATACLK
ONEPORTCLK
DATA AT PORTS
1 AND 2
tH
02706-055
tS
tOD = 6.5ns (MIN) TO 8.0ns (MAX)
tS = 5.0ns (MAX)
tH = –3.2ns (MAX)
I AND Q INTERLEAVED
INPUT DATA AT PORT 1
Figure 55. Timing Requirements in Two-Port Input Mode, with PLL Disabled
tS tH
PLL DISABLED, ONE-PORT MODE
IQSEL
tOD = 4.0ns (MIN)
TO 5.5ns (MAX)
tOD = 4.7ns (MAX)
tS = 3.0ns (MAX)
tH = –1.0ns (MAX)
tIQS = 3.5ns (MAX)
tIQH = –1.5ns (MAX)
(TYP SPECS)
tIQS
tIQH
02706-056
In one-port mode, data is received into the AD9777 as an
interleaved stream on Port 1. A clock signal (ONEPORTCLK),
running at the interleaved data rate, which is 2× the input data
rate of the internal I and Q channels, is available for data
synchronization at Pin 32.
Figure 56. Timing Requirements in One-Port Input Mode, with PLL Disabled
With PLL disabled, a clock at the DAC output rate must be applied
to CLKIN. Internal dividers synthesize the ONEPORTCLK signal
at Pin 32. The selection of the data for the I or Q channel is determined by the state of the logic level applied to Pin 31 (IQSEL when
the AD9777 is in one-port mode) on the rising edge of
ONEPORTCLK. Under these conditions, IQSEL = 0 latches the
data into the I channel on the clock rising edge, while IQSEL = 1
latches the data into the Q channel. It is possible to invert the I and
Q selection by setting Control Register 02h, Bit 1 to the invert state
(Logic 1). Figure 56 illustrates the timing requirements for the data
inputs as well as the IQSEL input. Note that the 1× interpolation
rate is not available in the one-port mode.
DIGITAL FILTER MODES
One-port mode is very useful when interfacing with devices,
such as the Analog Devices AD6622 or AD6623 transmit signal
processors, in which two digital data channels have been interleaved (multiplexed). The programmable modes’ ONEPORTCLK
inversion, ONEPORTCLK driver strength, and IQ pairing
described in the PLL Enabled, One-Port Mode section have
identical functionality with the PLL disable.
An online tool is available for quick and easy analysis of the
AD9777 interpolation filters in the various modes. The link can be
accessed at http://www.analog.com/Analog_Root/static/
techsupport/designtools/interactiveTools/dac/ad9777image.html.
The I and Q data paths of the AD9777 have their own
independent half-band FIR filters. Each data path consists of
three FIR filters, providing up to 8× interpolation for each
channel. The rate of interpolation is determined by the state of
Control Register 01h, Bit 7 and Bit 6. Figure 2 to Figure 4 show
the response of the digital filters when the AD9777 is set to 2×,
4×, and 8× modes. The frequency axes of these graphs have
been normalized to the input data rate of the DAC. As the
graphs show, the digital filters can provide greater than 75 dB of
out-of-band rejection.
AMPLITUDE MODULATION
Given two sine waves at the same frequency but with a 90°
phase difference, a point of view in time can be taken such that
the waveform that leads in phase is cosinusoidal and the
waveform that lags is sinusoidal. Analysis of complex variables
states that the cosine waveform can be defined as having real
positive and negative frequency components, while the sine
waveform consists of imaginary positive and negative frequency
images. This is shown graphically in the frequency domain in
Figure 57.
Rev. C | Page 32 of 60
AD9777
e–jωt/2j
The phase relationship of the modulated signals is dependent
on whether the modulating carrier is sinusoidal or cosinusoidal,
again with respect to the reference point of the viewer.
Examples of sine and cosine modulation are given in Figure 58.
SINE
DC
e–jωt/2j
Ae–jωt/2j
e–jωt/2
DC
SINUSOIDAL
MODULATION
DC
Figure 57. Real and Imaginary Components of
Sinusoidal and Cosinusoidal Waveforms
Ae–jωt/2j
Ae–jωt/2
Amplitude modulating a baseband signal with a sine or a cosine
convolves the baseband signal with the modulating carrier in
the frequency domain. Amplitude scaling of the modulated
signal reduces the positive and negative frequency images by a
factor of 2. This scaling is very important in the discussion of
the various modulation modes.
Ae–jωt/2
COSINUSOIDAL
MODULATION
DC
02706-058
COSINE
02706-057
e–jωt/2
Figure 58. Baseband Signal, Amplitude Modulated with Sine and Cosine Carriers
Rev. C | Page 33 of 60
AD9777
MODULATION, NO INTERPOLATION
With Control Register 01h, Bit 7 and Bit 6 set to 00, the
interpolation function on the AD9777 is disabled. Figure 59 to
Figure 62 show the DAC output spectral characteristics of the
AD9777 in the various modulation modes, all with the
interpolation filters disabled. The modulation frequency is
determined by the state of Control Register 01h, Bits 5 and 4.
The tall rectangles represent the digital domain spectrum of a
baseband signal of narrow bandwidth.
By comparing the digital domain spectrum to the DAC
SIN(x)/x roll-off, an estimate can be made for the characteristics required for the DAC reconstruction filter. Note also, per
the previous discussion on amplitude modulation, that the
spectral components (where modulation is set to fS/4 or fS/8) are
scaled by a factor of 2. In the situation where the modulation is
fS/2, the modulated spectral components add constructively,
and there is no scaling effect.
0
0
–20
–20
AMPLITUDE (dBFS)
–40
–60
–40
–60
–80
–80
0.2
0.4
0.6
0.8
1.0
fOUT (×fDATA)
02706-059
0
0
0.4
0.6
0.8
1.0
1.0
fOUT (×fDATA)
Figure 59. No Interpolation, Modulation Disabled
Figure 61. No Interpolation, Modulation = fDAC/4
0
–20
–20
AMPLITUDE (dBFS)
0
–40
–60
–40
–60
–80
–80
–100
–100
0
0.2
0.4
0.6
0.8
fOUT (×fDATA)
1.0
02706-060
AMPLITUDE (dBFS)
0.2
02706-061
–100
–100
02706-062
AMPLITUDE (dBFS)
The Effects of Digital Modulation on DAC Output Spectrum, Interpolation Disabled
Figure 60. No Interpolation, Modulation = fDAC/2
0
0.2
0.4
0.6
0.8
fOUT (×fDATA)
Figure 62. No Interpolation, Modulation = fDAC/8
Rev. C | Page 34 of 60
AD9777
MODULATION, INTERPOLATION = 2×
With Control Register 01h, Bit 7 and Bit 6 set to 01, the
interpolation rate of the AD9777 is 2×. Modulation is achieved
by multiplying successive samples at the interpolation filter
output by the sequence (+1, −1). Figure 63 to Figure 66
represent the spectral response of the AD9777 DAC output with
2× interpolation in the various modulation modes to a narrow
band baseband signal (again, the tall rectangles in the graphic).
The advantage of interpolation becomes clear in Figure 63 to
Figure 66, where it can be seen that the images that would
normally appear in the spectrum around the significant point is
that the interpolation filtering is done prior to the digital
modulator.
For this reason, as Figure 63 to Figure 66 show, the pass band of
the interpolation filters can be frequency shifted, giving the
equivalent of a high-pass digital filter.
Note that when using the fS/4 modulation mode, there is no
true stop band as the band edges coincide with each other. In
the fS/8 modulation mode, amplitude scaling occurs over only a
portion of the digital filter pass band due to constructive
addition over just that section of the band
0
0
–20
–20
AMPLITUDE (dBFS)
–40
–60
–40
–60
–80
–80
0.5
1.0
1.5
2.0
fOUT (×fDATA)
0
1.0
1.5
2.0
2.0
fOUT (×fDATA)
Figure 63. 2× Interpolation, Modulation = Disabled
Figure 65. 2× Interpolation, Modulation = fDAC/4
0
–20
–20
AMPLITUDE (dBFS)
0
–40
–60
–40
–60
–80
–80
–100
0
0.5
1.0
1.5
fOUT (×fDATA)
2.0
02706-064
AMPLITUDE (dBFS)
0.5
02706-065
–100
0
02706-063
–100
02706-066
AMPLITUDE (dBFS)
The Effects of Digital Modulation on DAC Output Spectrum, Interpolation = 2x
–100
Figure 64. 2× Interpolation, Modulation = fDAC/2
0
0.5
1.0
1.5
fOUT (×fDATA)
Figure 66. 2× Interpolation, Modulation = fDAC/8
Rev. C | Page 35 of 60
AD9777
MODULATION, INTERMODULATION = 4×
With Control Register 01h, Bit 7 and Bit 6 set to 10, the
interpolation rate of the AD9777 is 4×. Modulation is achieved
by multiplying successive samples at the interpolation filter
output by the sequence (0, +1, 0, −1).
Figure 67 to Figure 70 represent the spectral response of the
AD9777 DAC output with 4× interpolation in the various
modulation modes to a narrow band baseband signal.
0
0
–20
–20
AMPLITUDE (dBFS)
–40
–60
–40
–60
–80
–80
1
2
3
4
fOUT (×fDATA)
0
2
3
4
4
fOUT (×fDATA)
Figure 67. 4x Interpolation, Modulation Disabled
Figure 69. 4x Interpolation, Modulation = fDAC/4
0
–20
–20
AMPLITUDE (dBFS)
0
–40
–60
–40
–60
–80
–80
–100
–100
0
1
2
3
fOUT (×fDATA)
4
02706-068
AMPLITUDE (dBFS)
1
02706-069
–100
0
02706-067
–100
02706-070
AMPLITUDE (dBFS)
The Effects of Digital Modulation on DAC Output Spectrum, Interpolation = 4x
Figure 68. 4x Interpolation, Modulation = fDAC/2
0
1
2
3
fOUT (×fDATA)
Figure 70. 4x Interpolation, Modulation = fDAC/8
Rev. C | Page 36 of 60
AD9777
MODULATION, INTERMODULATION = 8×
With Control Register 01h, Bits 7 and 6, set to 11, the
interpolation rate of the AD9777 is 8×. Modulation is achieved
by multiplying successive samples at the interpolation filter
output by the sequence (0, +0.707, +1, +0.707, 0, –0.707, −1,
+0.707). Figure 71 to Figure 74 represent the spectral response
of the AD9777 DAC output with 8× interpolation in the various
modulation modes to a narrow band baseband signal.
Looking at Figure 59 to Figure 75, the user can see how higher
interpolation rates reduce the complexity of the reconstruction
filter needed at the DAC output. It also becomes apparent that
the ability to modulate by fS/2, fS/4, or fS/8 adds a degree of
flexibility in frequency planning
0
0
–20
–20
AMPLITUDE (dBFS)
–40
–60
–40
–60
–80
–80
1
2
3
4
fOUT (×fDATA)
02706-071
0
0
2
3
4
5
6
7
8
8
fOUT (×fDATA)
Figure 73. 8x Interpolation, Modulation = fDAC/4
Figure 71. 8× Interpolation, Modulation Disabled
0
–20
–20
AMPLITUDE (dBFS)
0
–40
–60
–40
–60
–80
–80
–100
–100
0
1
2
3
fOUT (×fDATA)
4
02706-072
AMPLITUDE (dBFS)
1
02706-073
–100
–100
02706-074
AMPLITUDE (dBFS)
The Effects of the Digital Modulation on the DAC Output Spectrum, Interpolation = 8×
0
1
2
3
4
5
6
7
fOUT (×fDATA)
Figure 74. 8x Interpolation, Modulation = fDAC/8
Figure 72. 8x Interpolation, Modulation = fDAC/2
Rev. C | Page 37 of 60
AD9777
ZERO STUFFING
(Control Register 01h, Bit 3)
As shown in Figure 75, a 0 or null in the output frequency
response of the DAC (after interpolation, modulation, and DAC
reconstruction) occurs at the final DAC sample rate (fDAC). This
is due to the inherent SIN(x)/x roll-off response in the digitalto-analog conversion. In applications where the desired frequency content is below fDAC/2, this may not be a problem. Note
that at fDAC/2 the loss due to SIN(x)/x is 4 dB. In direct RF applications, this roll-off may be problematic due to the increased
pass-band amplitude variation as well as the reduced amplitude
of the desired signal.
Consider an application where the digital data into the AD9777
represents a baseband signal around fDAC/4 with a pass band of
fDAC/10. The reconstructed signal out of the AD9777 would
experience only a 0.75 dB amplitude variation over its pass
band. However, the image of the same signal occurring at 3×
fDAC/4 suffers from a pass-band flatness variation of 3.93 dB.
This image may be the desired signal in an IF application using
one of the various modulation modes in the AD9777. This rolloff of image frequencies can be seen in Figure 59 to Figure 74,
where the effect of the interpolation and modulation rate is
apparent as well.
10
It is important to realize that the zero stuffing option by itself
does not change the location of the images but rather their
amplitude, pass-band flatness, and relative weighting. For
instance, in the previous example, the pass-band amplitude
flatness of the image at 3× fDATA/4 is now improved to 0.59 dB
while the signal level has increased slightly from −10.5 dBFS to
–8.1 dBFS.
INTERPOLATING (COMPLEX MIX MODE)
(Control Register 01h, Bit 2)
In the complex mix mode, the two digital modulators on the
AD9777 are coupled to provide a complex modulation function.
In conjunction with an external quadrature modulator, this
complex modulation can be used to realize a transmit image
rejection architecture. The complex modulation function can be
programmed for e+jωt or e−jωt to give upper or lower image
rejection. As in the real modulation mode, the modulation
frequency ω can be programmed via the SPI port for fDAC/2,
fDAC/4, and fDAC/8, where fDAC represents the DAC output rate.
OPERATIONS ON COMPLEX SIGNALS
ZERO STUFFING
ENABLED
Truly complex signals cannot be realized outside of a computer
simulation. However, two data channels, both consisting of real
data, can be defined as the real and imaginary components of a
complex signal. I (real) and Q (imaginary) data paths are often
defined this way. By using the architecture defined in Figure 76,
a system that operates on complex signals can be realized,
giving a complex (real and imaginary) output.
–10
–20
ZERO STUFFING
DISABLED
–30
–40
0
0.5
1.0
1.5
2.0
fOUT, NORMALIZED TO fDATA WITH ZERO STUFFING DISABLED (Hz)
02706-075
–50
Figure 75. Effect of Zero Stuffing on DAC’s SIN(x)/x Response
If a complex modulation function (e+jωt) is desired, the real and
imaginary components of the system correspond to the real and
imaginary components of e+jωt or cosωt and sinωt. As Figure 77
shows, the complex modulation function can be realized by
applying these components to the structure of the complex
system defined in Figure 76.
To improve upon the pass-band flatness of the desired image,
the zero stuffing mode can be enabled by setting the control
register bit to a Logic 1. This option increases the ratio of
fDAC/fDATA by a factor of 2 by doubling the DAC sample rate and
inserting a midscale sample (that is, 1000 0000 0000 0000) after
every data sample originating from the interpolation filter. This
is important as it affects the PLL divider ratio needed to keep
the VCO within its optimum speed range. Note that the zero
stuffing takes place in the digital signal chain at the output of
the digital modulator, before the DAC.
Rev. C | Page 38 of 60
a(t)
INPUT
OUTPUT
c(t) × b(t) + d × b(t)
COMPLEX FILTER
= (c + jd)
b(t)
IMAGINARY
INPUT
OUTPUT
b(t) × a(t) + c × b(t)
Figure 76. Realization of a Complex System
02706-076
0
SIN (X)/X ROLL-OFF (dBFS)
The net effect is to increase the DAC output sample rate by a
factor of 2× with the 0 in the SIN(x)/x DAC transfer function
occurring at twice the original frequency. A 6 dB loss in
amplitude at low frequencies is also evident, as can be seen in
Figure 76.
AD9777
INPUT
(REAL)
OUTPUT
(REAL)
INPUT
(IMAGINARY)
OUTPUT
INPUT
(IMAGINARY)
SINωt
90°
90°
COSωt
02706-078
INPUT
(REAL)
Figure 78. Quadrature Modulation
e–jωt = COSωt + jSINωt
02706-077
OUTPUT
(IMAGINARY)
Figure 77. Implementation of a Complex Modulator
COMPLEX MODULATION AND IMAGE REJECTION
OF BASEBAND SIGNALS
In traditional transmit applications, a two-step upconversion is
done in which a baseband signal is modulated by one carrier to
an IF (intermediate frequency) and then modulated a second
time to the transmit frequency. Although this approach has
several benefits, a major drawback is that two images are created
near the transmit frequency. Only one image is needed, the other
being an exact duplicate. Unless the unwanted image is filtered,
typically with analog components, transmit power is wasted and
the usable bandwidth available in the system is reduced.
A more efficient method of suppressing the unwanted image
can be achieved by using a complex modulator followed by a
quadrature modulator. Figure 78 is a block diagram of a
quadrature modulator. Note that it is in fact the real output half
of a complex modulator. The complete upconversion can
actually be referred to as two complex upconversion stages, the
real output of which becomes the transmitted signal.
The entire upconversion from baseband to transmit frequency
is represented graphically in Figure 79. The resulting spectrum
shown in Figure 79 represents the complex data consisting of
the baseband real and imaginary channels, now modulated onto
orthogonal (cosine and negative sine) carriers at the transmit
frequency. It is important to remember that in this application
(two baseband data channels), the image rejection is not
dependent on the data at either of the AD9777 input channels.
In fact, image rejection still occurs with either one or both of
the AD9777 input channels active. Note that by changing the
sign of the sinusoidal multiplying term in the complex
modulator, the upper sideband image could have been
suppressed while passing the lower one. This is easily done in
the AD9777 by selecting the e+jωt bit (Register 01h, Bit 1). In
purely complex terms, Figure 79 represents the two-stage
upconversion from complex baseband to carrier.
Rev. C | Page 39 of 60
AD9777
REAL CHANNEL (OUT)
A/2
A/2
–FC1
FC
–B/2J
B/2J
–FC
FC
REAL CHANNEL (IN)
A
DC
COMPLEX
MODULATOR
TO QUADRATURE
MODULATOR
IMAGINARY CHANNEL (OUT)
–A/2J
A/2J
–FC
–FC
B/2
B/2
–FC
FC
IMAGINARY CHANNEL (IN)
B
DC
A/4 + B/4J
A/4 – B/4J
A/4 + B/4J
A/4 – B/4J
–FQ2
–FQ + FC
–FQ – FC
F Q – FC
–A/4 – B/4J A/4 – B/4J
A/4 + B/4J –A/4 + B/4J
FQ
FQ + FC
OUT
REAL
QUADRATURE
MODULATOR
–FQ
IMAGINARY
FQ
REJECTED IMAGES
–FQ
A/2 – B/2J
FQ
1F = COMPLEX MODULATION FREQUENCY
C
2F = QUADRATURE MODULATION FREQUENCY
Q
Figure 79. Two-Stage Upconversion and Resulting Image Rejection
Rev. C | Page 40 of 60
02706-079
A/2 + B/2J
AD9777
COMPLEX BASEBAND
SIGNAL
A system in which multiple baseband signals are complex
modulated and then applied to the AD9777 real and imaginary
inputs, followed by a quadrature modulator, is shown in Figure 82,
which also describes the transfer function of this system and the
spectral output. Note the similarity of the transfer functions
given in Figure 82 and Figure 80. Figure 82 adds an additional
complex modulator stage for summing multiple carriers at the
AD9777 inputs. In addition, as in Figure 79, the image rejection
is not dependent on the real or imaginary baseband data on any
channel. Image rejection on a channel occurs if either the real
or imaginary data, or both, is present on the baseband channel.
1
OUTPUT = REAL
×
ej(ω1 + ω2)t
1/2
1/2
–ω1 – ω2
DC
02706-080
= REAL
ω1 + ω2
FREQUENCY
Figure 80. Two-Stage Complex Upconversion
IMAGE REJECTION AND SIDEBAND
SUPPRESSIONS OF MODULATED CARRIERS
It is important to remember that the magnitude of a complex
signal can be 1.414× the magnitude of its real or imaginary
components. Due to this 3 dB increase in signal amplitude, the
real and imaginary inputs to the AD9777 must be kept at least
3 dB below full scale when operating with the complex
modulator. Overranging in the complex modulator results in
severe distortion at the DAC output.
As shown in Figure 79, image rejection can be achieved by
applying baseband data to the AD9777 and following the
AD9777 with a quadrature modulator. To process multiple
carriers while still maintaining image reject capability, each
carrier must be complex modulated. As Figure 80 shows, single
or multiple complex modulators can be used to synthesize
complex carriers. These complex carriers are then summed and
applied to the real and imaginary inputs of the AD9777.
R(1)
COMPLEX
MODULATOR 1
BASEBAND CHANNEL 2
REAL INPUT
R(2)
COMPLEX
MODULATOR 2
IMAGINARY INPUT
BASEBAND CHANNEL N
REAL INPUT
MULTICARRIER
REAL OUTPUT =
R(1) + R(2) + . . .R(N)
(TO REAL INPUT OF AD9777)
R(1)
IMAGINARY INPUT
MULTICARRIER
IMAGINARY OUTPUT =
I(1) + I(2) + . . .I(N)
(TO IMAGINARY INPUT OF AD9777)
R(2)
R(N)
COMPLEX
MODULATOR N
R(N) = REAL OUTPUT OF N
I(N) = IMAGINARY OUTPUT OF N
02706-081
BASEBAND CHANNEL 1
REAL INPUT
R(N)
IMAGINARY INPUT
Figure 81. Synthesis of Multicarrier Complex Signal
MULTIPLE
BASEBAND
CHANNELS
IMAGINARY
MULTIPLE
COMPLEX
MODULATORS
FREQUENCY = ω1, ω2...ωN
REAL
AD9777
COMPLEX
MODULATOR
FREQUENCY = ωC
IMAGINARY
REAL
IMAGINARY
REAL
QUADRATURE
MODULATOR
FREQUENCY = ωQ
COMPLEX BASEBAND
SIGNAL
×
OUTPUT = REAL
–ω1 – ωC – ωQ
ej(ωN + ωC + ωQ)t
ω1 + ωC + ωQ
DC
REJECTED IMAGES
Figure 82. Image Rejection with Multicarrier Signals
Rev. C | Page 41 of 60
02706-082
REAL
AD9777
The complex carrier synthesized in the AD9777 digital
modulator is accomplished by creating two real digital carriers
in quadrature. Carriers in quadrature cannot be created with
the modulator running at fDAC/2. As a result, complex modulation only functions with modulation rates of fDAC/4 and fDAC/8.
Region A and Region B of Figure 83 to Figure 88 are the result
of the complex signal described previously, when complex
modulated in the AD9777 by +ejωt. Region C and Region D are
the result of the complex signal described previously, again with
positive frequency components only, modulated in the AD9777
by −ejωt. The analog quadrature modulator after the AD9777
inherently modulates by +ejωt.
Region A
Region A is a direct result of the upconversion of the complex
signal near baseband. If viewed as a complex signal, only the
images in Region A remains. The complex Signal A, consisting
of positive frequency components only in the digital domain,
has images in the positive odd Nyquist zones (1, 3, 5, and so
on), as well as images in the negative even Nyquist zones. The
appearance and rejection of images in every other Nyquist zone
becomes more apparent at the output of the quadrature
modulator. The A images appear on the real and the imaginary
outputs of the AD9777, as well as on the output of the
quadrature modulator, where the center of the spectral plot now
represents the quadrature modulator LO and the horizontal scale
now represents the frequency offset from this LO.
Region B
Region B is the image (complex conjugate) of Region A. If a
spectrum analyzer is used to view the real or imaginary DAC
outputs of the AD9777, Region B appears in the spectrum.
However, on the output of the quadrature modulator, Region B
is rejected.
Region C
Region C is most accurately described as a down conversion, as
the modulating carrier is −ejωt. If viewed as a complex signal,
only the images in Region C remains. This image appears on
the real and imaginary outputs of the AD9777, as well as on the
output of the quadrature modulator, where the center of the
spectral plot now represents the quadrature modulator LO and
the horizontal scale represents the frequency offset from this
LO.
Region D
Region D is the image (complex conjugate) of Region C. If a
spectrum analyzer is used to view the real or imaginary DAC
outputs of the AD9777, Region D appears in the spectrum.
However, on the output of the quadrature modulator, Region D
is rejected.
Figure 89 to Figure 96 show the measured response of the
AD9777 and AD8345 given the complex input signal to the
AD9777 in Figure 89. The data in these graphs was taken with a
data rate of 12.5 MSPS at the AD9777 inputs. The interpolation
rate of 4× or 8× gives a DAC output data rate of 50 MSPS or
100 MSPS. As a result, the high end of the DAC output
spectrum in these graphs is the first null point for the SIN(x)/x
roll-off, and the asymmetry of the DAC output images is
representative of the SIN(x)/x roll-off over the spectrum. The
internal PLL was enabled for these results. In addition, a
35 MHz third-order low-pass filter was used at the AD9777/
AD8345 interface to suppress DAC images.
An important point can be made by looking at Figure 91 and
Figure 93. Figure 91 represents a group of positive frequencies
modulated by complex +fDAC/4, while Figure 93 represents a
group of negative frequencies modulated by complex −fDAC/4.
When looking at the real or imaginary outputs of the AD9777,
as shown in Figure 91 and Figure 93, the results look identical.
However, the spectrum analyzer cannot show the phase
relationship of these signals. The difference in phase between
the two signals becomes apparent when they are applied to the
AD8345 quadrature modulator, with the results shown in Figure
92 and Figure 94.
Rev. C | Page 42 of 60
AD9777
0
0
–20
–20
A
B
C
D
A
B
C
–40
–40
–60
–60
–80
–80
D
–1.5
–1.0
–0.5
0
0.5
1.0
1.5
–100
–2.0
2.0
–1.5
B
–1.0
02706-083
–100
–2.0
A
(LO)
fOUT (×fDATA)
–0.5
A
0
0.5
B
1.0
C
1.5
2.0
(LO)
fOUT (×fDATA)
Figure 83. 2× Interpolation, Complex fDAC/4 Modulation
Figure 86. 2× Interpolation, Complex fDAC/8 Modulation
0
0
–20
–20
A
B
C
D
A
B
C
–40
–40
–60
–60
–80
–80
D A
–3.0
–2.0
–1.0
0
1.0
2.0
3.0
4.0
02706-084
–100
–4.0
(LO)
fOUT (×fDATA)
–100
–4.0
–3.0
B
–2.0
C D
–1.0
A
0
1.0
B
2.0
C
3.0
4.0
02706-087
D
(LO)
fOUT (×fDATA)
Figure 84. 4× Interpolation, Complex fDAC/4 Modulation
Figure 87. 4× Interpolation, Complex fDAC/8 Modulation
0
0
–20
–20
A
B
C
D
A
B
C
DA
–40
–40
–60
–60
–80
–80
–4.0
–2.0
0
2.0
4.0
6.0
8.0
(LO)
fOUT (×fDATA)
02706-085
–6.0
–100
–8.0
Figure 85. 8× Interpolation, Complex fDAC/4 Modulation
BC
–6.0
–4.0
–2.0
DA
0
BC
2.0
4.0
6.0
8.0
(LO)
fOUT (×fDATA)
Figure 88. 8× Interpolation, Complex fDAC/8 Modulation
Rev. C | Page 43 of 60
02706-088
D
–100
–8.0
CD
02706-086
D
0
0
–10
–10
–20
–20
–30
–30
AMPLITUDE (dBm)
–40
–50
–60
–70
–40
–50
–60
–70
–80
–80
–90
–90
–100
10
20
30
40
02706-089
0
–100
750
50
FREQUENCY (MHz)
760
770
780
790
800
810
820
830
840
850
FREQUENCY (MHz)
Figure 89. AD9777, Real DAC Output of Complex Input Signal Near Baseband
(Positive Frequencies Only), Interpolation = 4×, No Modulation in AD9777
02706-092
AMPLITUDE (dBm)
AD9777
Figure 92. AD9777 Complex Output from Figure 91, Now Quadrature
Modulated by AD8345 (LO = 800 MHz)
0
0
–10
–10
–20
–30
AMPLITUDE (dBm)
AMPLITUDE (dBm)
–20
–30
–40
–50
–60
–40
–50
–60
–70
–70
–80
–80
–90
–90
0
760
770
780
790
800
810
820
830
840
850
FREQUENCY (MHz)
20
30
40
50
FREQUENCY (MHz)
Figure 93. AD9777, Real DAC Output of Complex Input Signal Near
Baseband (Negative Frequencies Only), Interpolation = 4×,
Complex Modulation in AD9777 = −fDAC/4
Figure 90. AD9777 Complex Output from Figure 89, Now Quadrature
Modulated by AD8345 (LO = 800 MHz)
0
–10
–10
–20
–20
–30
–30
AMPLITUDE (dBm)
0
–40
–50
–60
–40
–50
–60
–70
–70
–80
–80
0
10
20
30
40
50
FREQUENCY (MHz)
Figure 91. AD9777, Real DAC Output of Complex Input Signal Near
Baseband (Positive Frequencies Only), Interpolation = 4×,
Complex Modulation in AD9777 = +fDAC/4
–100
750
760
770
780
790
800
810
820
830
840
850
FREQUENCY (MHz)
Figure 94. AD9777 Complex Output from Figure 93, Now Quadrature
Modulated by AD8345 (LO = 800 MHz)
Rev. C | Page 44 of 60
02706-094
–90
–90
–100
02706-091
AMPLITUDE (dBm)
10
02706-090
–100
750
02706-093
–100
AD9777
0
0
–10
–20
–30
AMPLITUDE (dBm)
–40
–60
–40
–50
–60
–70
–80
–80
0
20
40
60
80
100
FREQUENCY (MHz)
Figure 95. AD9777, Real DAC Output of Complex Input Signal Near
Baseband (Positive Frequencies Only), Interpolation = 8×,
Complex Modulation in AD9777 = +fDAC/8
–100
700
720
740
760
780
800
820
840
860
880
900
FREQUENCY (MHz)
Figure 96. AD9777 Complex Output from Figure 95, Now Quadrature
Modulated by AD8345 (LO = 800 MHz)
Rev. C | Page 45 of 60
02706-096
–90
–100
02706-095
AMPLITUDE (dBm)
–20
AD9777
APPLYING THE OUTPUT CONFIGURATIONS
A single-ended output is suitable for applications requiring a
unipolar voltage output. A positive unipolar output voltage
results if IOUTA and/or IOUTB is connected to a load resistor, RLOAD,
referred to AGND. This configuration is most suitable for a
single-supply system requiring a dc-coupled, ground referred
output voltage. Alternatively, an amplifier could be configured
as an I-V converter, thus converting IOUTA or IOUTB into a
negative unipolar voltage. This configuration provides the best
DAC dc linearity as IOUTA or IOUTB are maintained at ground or
virtual ground.
In many applications, it may be necessary to understand the
equivalent DAC output circuit. This is especially useful when
designing output filters or when driving inputs with finite input
impedances. Figure 97 illustrates the output of the AD9777 and
the equivalent circuit. A typical application where this
information may be useful is when designing an interface filter
between the AD9777 and the Analog Devices AD8345
quadrature modulator.
VOUT+
IOUTB
VOUT–
VSOURCE = 2 VP-P
ROUT = 100 Ω
Note that the output impedance of the AD9777 DAC itself is
greater than 100 kΩ and typically has no effect on the
impedance of the equivalent output circuit.
DIFFERENTIAL COUPLING USING A
TRANSFORMER
An RF transformer can be used to perform a differential-tosingle-ended signal conversion, as shown in Figure 98. A
differentially coupled transformer output provides the optimum
distortion performance for output signals whose spectral
content lies within the transformer’s pass band. An RF
transformer, such as the Mini-Circuits T1-1T, provides excellent
rejection of common-mode distortion (that is, even-order
harmonics) and noise over a wide frequency range. It also
provides electrical isolation and the ability to deliver twice the
power to the load. Transformers with different impedance ratios
may also be used for impedance matching purposes.
IOUTA
DAC
IOUTB
02706-097
VOUT
(DIFFERENTIAL)
RLOAD
The center tap on the primary side of the transformer must be
connected to AGND to provide the necessary dc current path
for both IOUTA and IOUTB. The complementary voltages appearing
at IOUTA and IOUTB (that is, VOUTA and VOUTB) swing symmetrically
around AGND and should be maintained within the specified
output compliance range of the AD9777. A differential resistor,
RDIFF, can be inserted in applications where the output of the
transformer is connected to the load, RLOAD, via a passive
reconstruction filter or cable. RDIFF is determined by the
transformer’s impedance ratio and provides the proper source
termination that results in a low VSWR. Note that approximately half the signal power dissipates across RDIFF.
RA + RB
VSOURCE =
IOUTFS × (RA + RB)
p-p
MINI-CIRCUITS
T1-1T
Figure 98. Transformer-Coupled Output Circuit
UNBUFFERED DIFFERENTIAL OUTPUT,
EQUIVALENT CIRCUIT
IOUTA
For the typical situation, where IOUTFS = 20 mA and RA and RB
both equal 50 Ω, the equivalent circuit values become
02706-098
The following sections illustrate typical output configurations
for the AD9777. Unless otherwise noted, it is assumed that
IOUTFS is set to a nominal 20 mA. For applications requiring
optimum dynamic performance, a differential output configuration is suggested. A simple differential output may be
achieved by converting IOUTA and IOUTB to a voltage output by
terminating them to AGND via equal value resistors. This type
of configuration may be useful when driving a differential
voltage input device such as a modulator. If a conversion to a
single-ended signal is desired and the application allows for ac
coupling, an RF transformer may be useful; if power gain is
required, an op amp may be used. The transformer configuration provides optimum high frequency noise and distortion
performance. The differential op amp configuration is suitable
for applications requiring dc coupling, signal gain, and/or level
shifting within the bandwidth of the chosen op amp.
Figure 97. DAC Output Equivalent Circuit
Rev. C | Page 46 of 60
AD9777
DIFFERENTIAL COUPLING USING AN OP AMP
Gain/Offset Adjust
An op amp can also be used to perform a differential-to-single
ended conversion, as shown in Figure 99. This has the added
benefit of providing signal gain as well. In Figure 99, the
AD9777 is configured with two equal load resistors, RLOAD, of
25 Ω. The differential voltage developed across IOUTA and IOUTB is
converted to a single-ended signal via the differential op amp
configuration. An optional capacitor can be installed across
IOUTA and IOUTB, forming a real pole in a low-pass filter. The
addition of this capacitor also enhances the op amp’s distortion
performance by preventing the DAC’s fast slewing output from
overloading the input of the op amp.
The matching of the DAC output to the common-mode input
of the AD8345 allows the two components to be dc-coupled,
with no level shifting necessary. The combined voltage offset of
the two parts can therefore be compensated via the AD9777
programmable offset adjust. This allows excellent LO
cancellation at the AD8345 output. The programmable gain
adjust allows for optimal image rejection as well.
AD8021
DAC
IOUTB
COPT
225Ω
25Ω
500Ω
ROPT
225Ω
02706-099
AVDD
25Ω
Figure 99. Op Amp-Coupled Output Circuit
The common-mode (and second-order distortion) rejection of
this configuration is typically determined by the resistor
matching. The op amp used must operate from a dual supply
since its output is approximately ±1.0 V. A high speed amplifier,
such as the AD8021, capable of preserving the differential
performance of the AD9777 while meeting other system level
objectives (for example, cost, power) is recommended. The op
amp’s differential gain, gain setting resistor values, and full-scale
output swing capabilities should all be considered when
optimizing this circuit. ROPT is necessary only if level shifting is
required on the op amp output. In Figure 99, AVDD, which is
the positive analog supply for both the AD9777 and the op amp,
is also used to level shift the differential output of the AD9777
to midsupply (that is, AVDD/2).
INTERFACING WITH THE AD8345 QUADRATURE
MODULATOR
The AD9777 architecture was defined to operate in a transmit
signal chain using an image reject architecture. A quadrature
modulator is also required in this application and should be
designed to meet the output characteristics of the DAC as much
as possible. The AD8345 from Analog Devices meets many of
the requirements for interfacing with the AD9777. As with any
DAC output interface, there are a number of issues that have to
be resolved. The following sections list some of the major issues.
The performance of the AD9777 and AD8345 in an image
reject transmitter, reconstructing three WCDMA carriers, can
be seen in Figure 100. The LO of the AD8345 in this application
is 800 MHz. Image rejection (50 dB) and LO feedthrough
(−78 dBFS) have been optimized with the programmable
features of the AD9777. The average output power of the digital
waveform for this test was set to −15 dBFS to account for the
peak-to-average ratio of the WCDMA signal.
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
762.5
DAC Compliance Voltage/Input Common-Mode Range
The dynamic range of the AD9777 is optimal when the DAC
outputs swing between ±1.0 V. The input common-mode range
of the AD8345, at 0.7 V, allows optimum dynamic range to be
achieved in both components.
Rev. C | Page 47 of 60
782.5
802.5
822.5
FREQUENCY (MHz)
Figure 100. AD9777/AD8345 Synthesizing a
Three-Carrier WCDMA Signal at an LO of 800 MHz
842.5
02706-100
IOUTA
AMPLITUDE (dBm)
500Ω
225Ω
The AD9777 evaluation board includes an AD8345 and
recommended interface (Figure 105 and Figure 106). On the
output of the AD9777, R9 and R10 convert the DAC output
current to a voltage. R16 can be used to do a slight commonmode shift if necessary. The (now voltage) signal is applied to a
low-pass reconstruction filter to reject DAC images. The
components installed on the AD9777 provide a 35 MHz cutoff
but can be changed to fit the application. A balun (MiniCircuits ADTL1-12) is used to cross the ground plane boundary
to the AD8345. Another balun (Mini-Circuits ETC1-1-13) is
used to couple the LO input of the AD8345. The interface
requires a low ac impedance return path from the AD8345,
therefore a single connection between the AD9777 and AD8345
ground planes is recommended.
AD9777
EVALUATION BOARD
The AD9777 evaluation board allows easy configuration of the
various modes, programmable via the SPI port. Software is
available for programming the SPI port from Windows® 95,
Windows 98, or Windows NT®/2000. The evaluation board also
contains an AD8345 quadrature modulator and support
circuitry that allows the user to optimally configure the AD9777
in an image reject transmit signal chain.
Figure 101 to Figure 104 describe how to configure the
evaluation board in the one-port and two-port input modes with
the PLL enabled and disabled. Refer to Figure 105 to Figure 114,
the schematics, and the layout for the AD9777 evaluation board
for the jumper locations described below. The AD9777 outputs
can be configured for various applications by referring to the
following instructions.
DAC Differential Outputs
Transformers T2 and T3 should be in place. Note that the lower
band of operation for these transformers is 300 kHz to 500 kHz.
Jumpers 4, 8, 13 to 17, and 28 to 30 should remain unsoldered.
The outputs are taken from S3 and S4.
Using the AD8345
Remove Transformers T2 and T3. Jumpers JP4 and 28 to 30
should remain unsoldered. Jumpers 13 to 16 should be
soldered. The desired components for the low-pass interface
filters L6, L7, C55, and C81 should be in place. The LO drive is
connected to the AD8345 via J10 and the balun T4, and the
AD8345 output is taken from J9.
DAC Single-Ended Outputs
Remove Transformers T2 and T3. Solder jumper link JP4 or
JP28 to look at the DAC1 outputs. Solder jumper link JP29 or
JP30 to look at the DAC2 outputs. Jumpers 8 and 13 to 17
should remain unsoldered. Jumpers JP35 to JP38 may be used
to ground one of the DAC outputs while the other is measured
single-ended. Optimum single-ended distortion performance is
typically achieved in this manner. The outputs are taken from
S3 and S4.
Rev. C | Page 48 of 60
AD9777
LECROY
TRIG
PULSE
INP
GENERATOR
SIGNAL GENERATOR
DATACLK
INPUT CLOCK
AWG2021
OR
DG2020
CLK+/CLK–
40-PIN RIBBON CABLE
DAC1, DB15–DB0
DAC2, DB15–DB0
AD9777
JUMPER CONFIGURATION FOR TWO PORT MODE PLL ON
SOLDERED/IN
×
UNSOLDERED/OUT
×
×
×
×
×
×
×
×
×
×
×
×
NOTES
1. TO USE PECL CLOCK DRIVER (U8), SOLDER JP41 AND JP42 AND REMOVE TRANSFORMER T1.
2. IN TWO-PORT MODE, IF DATACLK/PLL_LOCK IS PROGRAMMED TO OUTPUT PIN 8, JP25 AND
JP39 SHOULD BE SOLDERED. IF DATACLK/PLL_LOCK IS PROGRAMMED TO OUTPUT PIN 53,
JP46 AND JP47 SHOULD BE SOLDERED. SEE THE TWO PORT DATA INPUT MODE SECTION
FOR MORE INFORMATION.
02706-101
JP1 –
JP2 –
JP3 –
JP5 –
JP6 –
JP12 –
JP24 –
JP25 –
JP26 –
JP27 –
JP31 –
JP32 –
JP33 –
Figure 101. Test Configuration for AD9777 in Two-Port Mode with PLL Enabled Signal Generator Frequency = Input Data Rate,
DAC Output Data Rate = Signal Generator Frequency × Interpolation Rate
LECROY
TRIG
PULSE
INP
GENERATOR
SIGNAL GENERATOR
ONEPORTCLK
INPUT CLOCK
AWG2021
OR
DG2020
CLK+/CLK–
DAC1, DB15–DB0
DAC2, DB15–DB0
AD9777
JUMPER CONFIGURATION FOR ONE PORT MODE PLL ON
SOLDERED/IN
×
UNSOLDERED/OUT
×
×
×
×
×
×
×
×
×
×
×
×
NOTES
1. TO USE PECL CLOCK DRIVER (U8), SOLDER JP41 AND JP42 AND REMOVE TRANSFORMER T1.
02706-102
JP1 –
JP2 –
JP3 –
JP5 –
JP6 –
JP12 –
JP24 –
JP25 –
JP26 –
JP27 –
JP31 –
JP32 –
JP33 –
Figure 102. Test Configuration for AD9777 in One-Port Mode with PLL Enabled, Signal Generator Frequency = One-Half Interleaved Input Data Rate,
ONEPORTCLK = Interleaved Input Data Rate, DAC Output Data Rate = Signal Generator Frequency × Interpolation Rate
Rev. C | Page 49 of 60
AD9777
LECROY
TRIG
PULSE
INP
GENERATOR
SIGNAL GENERATOR
DATACLK
INPUT CLOCK
AWG2021
OR
DG2020
CLK+/CLK–
40-PIN RIBBON CABLE
DAC1, DB15–DB0
DAC2, DB15–DB0
AD9777
JUMPER CONFIGURATION FOR TWO PORT MODE PLL OFF
SOLDERED/IN
×
UNSOLDERED/OUT
×
×
×
×
×
×
×
×
×
×
×
×
NOTES
1. TO USE PECL CLOCK DRIVER (U8), SOLDER JP41 AND JP42 AND REMOVE TRANSFORMER T1.
2. IN TWO-PORT MODE, IF DATACLK/PLL_LOCK IS PROGRAMMED TO OUTPUT PIN 8, JP25 AND
JP39 SHOULD BE SOLDERED. IF DATACLK/PLL_LOCK IS PROGRAMMED TO OUTPUT PIN 53,
JP46 AND JP47 SHOULD BE SOLDERED. SEE THE TWO PORT DATA INPUT MODE SECTION
FOR MORE INFORMATION.
02706-103
JP1 –
JP2 –
JP3 –
JP5 –
JP6 –
JP12 –
JP24 –
JP25 –
JP26 –
JP27 –
JP31 –
JP32 –
JP33 –
Figure 103. Test Configuration for AD9777 in Two-Port Mode with PLL Disabled, DAC Output Data Rate = Signal Generator Frequency,
DATACLK = Signal Generator Frequency/Interpolation Rate
LECROY
TRIG
PULSE
INP
GENERATOR
SIGNAL GENERATOR
ONEPORTCLK
INPUT CLOCK
AWG2021
OR
DG2020
CLK+/CLK–
DAC1, DB15–DB0
DAC2, DB15–DB0
AD9777
JUMPER CONFIGURATION FOR ONE PORT MODE PLL OFF
SOLDERED/IN
×
UNSOLDERED/OUT
×
×
×
×
×
×
×
×
×
×
×
×
NOTES
1. TO USE PECL CLOCK DRIVER (U8), SOLDER JP41 AND JP42 AND REMOVE TRANSFORMER T1.
02706-104
JP1 –
JP2 –
JP3 –
JP5 –
JP6 –
JP12 –
JP24 –
JP25 –
JP26 –
JP27 –
JP31 –
JP32 –
JP33 –
Figure 104. Test Configuration for AD9777 in One-Port Mode with PLL Disabled, DAC Output Data Rate = Signal Generator Frequency,
ONEPORTCLK = Interleaved Input Data Rate = 2× Signal Generator Frequency/Interpolation Rate.
Rev. C | Page 50 of 60
Figure 105. AD8345 Circuitry on AD9777 Evaluation Board
O1P
O1N
2
+ C72
10V
10µF
02706-105
BCASE
VDDM
O2P
C54
DNP
CC0603
LC0805
CC0805
C78
0.1µF
C75
0.1µF
C73
L5 DNP
DNP
CC0805
LC0805
L4
DNP
C35
100pF CC0603
L7
DNP
LC0805
C81
DNP
1
3
1
T6
P
6
4
S
2
1
2
11 10
AD8345
3
4
RC0603
R36
51Ω
1
C77
100pF
CC0603
RC0603
7
8
P
CC0603
C80
DNP
T4
R37
DNP
RC0603
JP20
S
5
4
JP18
R26
1kΩ
VDDM
2
9
C76
100pF
3 ETC1-1-13
6
CC0603
R35
51Ω
5
U2
CC0603
C74
100pF
R34
DNP
R33
51Ω
14 13 12
4
T5
RC0603
JP19
CC0603
RC0603
RC0603
16 15
6
S
P
3
G4B
CC0805
LOIN
LC0805
G4A
G1B
G2
C55
DNP
QBBP
IBBP
CC0603
VOUT
LOIP
G3
VPS1
O2N
QBBN
IBBN
G1A
C79
DNP
RC0603
ENBL
ADTL1-12
VPS2
ADTL1-12
Rev. C | Page 51 of 60
CC0805
R32
51Ω
RC0603
L6
DNP
2
R30
DNP
2
2
J9
DGND2; 3, 4, 5
JP21
JP7
2
2
LOCAL OSC INPUT
R28 DGND2; 3, 4, 5
0Ω
J10
RC0603
RC0603
R23
0Ω
MODULATED OUTPUT
J7
J3
J6
J4
J5
J8
W12
W11
CGND
DCASE
CLKVDD_IN
AGND
DCASE
AVDD_IN
DGND
DCASE
DVDD_IN
DGND2
DCASE
VDDMIN
+ C63
16V
22µF
+ C64
16V
22µF
+ C65
16V
22µF
+ C28
16V
22µF
2
c
LC1210
L1
FERRITE
LC1210
L2
FERRITE
LC1210
L3
FERRITE
LC1210
L8
FERRITE
CC0805
CC0805
CC0805
CC0805
POWER INPUT FILTERS
DCASE
DCASE
C69
0.1µF
DCASE
JP11
C68
0.1µF
JP10
C67
0.1µF
JP9
C32
0.1µF
+ C62
16V
22µF
TP5
BLK
+ C61
16V
22µF
TP3
BLK
+ C66
16V
22µF
JP43
VDDM
JP44
TP7
BLK
TP6
RED
CLKVDD
TP4
RED
AVDD
TP2
RED
DVDD
JP45
AD9777
Figure 106. AD9777 Clock, Power Supplies, and Output Circuitry
2
3
JP12
CX2
CX1
02706-106
JP3
IQ
JP40
JP27
JP5
C29
0.1µF
JP25
RC0603
R39
1kΩ JP32
R5
49.9Ω
TP14
WHT
DVDD; 14
DGND; 7
11
DVDD; 14
DGND; 7
RC0603
R1
200Ω
DVDD
DVDD
DVDD
DVDD
c
+ C7
BCASE
10µF
6.3V
+ C8
10µF
6.3V
BCASE
+ C9
10µF
6.3V
BCASE
+ C10
10µF
6.3V
C42
0.1µF
CC0603
0.001µF
CC0603
C23
0.001µF
CC0603
C24
0.001µF
CC0603
C25
0.001µF
CC0603
C26
CLKN
CLKP
0.1µF
C11
CC0603
0.1µF
C12
BCASE
BCASE
C1 +
10µF
6.3V
R38 10kΩ
JP26
BD14
74VCX86
JP31
12
11
U4 13
U3
RC0603
JP23
74VCX86 CX3
C45
0.01µF
OPCLK
JP34
AGND; 3, 4, 5
OPCLK
S5
IQ
S6
DGND; 3, 4, 5
OPCLK_3
BD15
13
12
JP24
JP39
T1
T1-1T
JP33
ADCLK
1
5
4
JP22
6
ACLKX c CGND; 3, 4, 5
S1
CLKIN
JP2
JP1
C13
0.1µF
CC0603
R40
DVDD
5kΩ
DGND; 3, 4, 5
DATACLK
S2
c
TP15
WHT
R3
1kΩ
BD11
BD10
BD09
BD08
BD13
BD12
AD03
AD02
AD01
AD00
AD09
AD08
AD07
AD06
AD05
AD04
AD10
AD13
AD12
AD11
AD15
AD14
c
CC0603
1pF
C27
40
39
38
37
36
35
34
33
32
31
30
29
28
27
26
25
24
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
CC0805
DVDD
AD9777+TSP
CC0805
41
42
43
44
45
46
47
48
49
50
51
52
53
54
55
56
57
58
59
60
61
62
63
64
65
66
67
68
69
70
71
72
73
74
75
76
77
78
79
80
0.1µF
C36
VDDC1
VDDA6
VSSA10
LF
VDDC2
VDDA5
VSSC1
VSSA9
VDDA4
CLKP
CLKN
VSSA8
VSSA7
VSSC2
DCLK-PLLL
IOUT1P
VSSD1
IOUT1N
VDDD1
VSSA6
VSSA5
P1D15
IOUT2P
P1D14
IOUT2N
P1D13
VSSA4
P1D12
VSSA3
P1D11
VDDA3
P1D10
VSSD2
VSSA2
VDDD2
U1 VDDA2
P1D9
VSSA1
VDDA1
P1D8
FSADJ1
P1D7
FSADJ2
P1D6
REFOUT
P1D5
P1D4
RESET
VSSD3
SP-CSB
SP-CLK
VDDD3
SP-SDI
P1D3
SP-SDO
P1D2
P1D1
VSSD6
VDDD6
P1D0
P2D0
P2D15-IQSEL
P2D14-OPCLK
P2D1
P2D2
P2D13
P2D3
P2D12
P2D4
VSSD4
VDDD4
P2D5
P2D11
VSSD5
VDDD5
P2D10
P2D9
P2D6
P2D7
P2D8
CC0603
CLKVDD
CC0603
0.1µF
CC0805
C37
CC0603
BD07
BD06
C22
0.001µF
CC0603
CC0805
0.1µF
CC0603
C16
TP8
WHT
C6 +
10µF
6.3V
C5 +
10µF
6.3V
BCASE
DVDD
BCASE
JP38
JP36
J35
J37
IQ
JP46
R8
R7
1kΩ
2kΩ
0.01% 0.01%
+ C3
10µF
6.3V
DVDD
BCASE
0.1µF
+ C2
10µF
6.3V
AVDD
0.1µF
C41
AVDD
BCASE
C15
C4 +
10µF
6.3V
TP9
WHT
0.1µF
0.1µF
BCASE
C14
C58
DNP
CC0603
C17
CC0603
CC0805
0.1µF
C40
C58
DNP
CC0603
C19
0.1µF
C39
R6 0.1µF
1kΩ
BD00
C21
BD01
CC0603
BD02 0.001µF
BD03
BD04
BD05
SPCSP
SPCLK
SPSDI
SPSDO
TP10
WHT
0.1µF
TP11
WHT
CC0805
C18
C59
DNP
C57
DNP
0.1µF
CC0603
C20
0.1µF
CC0805
C38
CC0603
CC0603
RC0603
RC0603
CC0605
Rev. C | Page 52 of 60
CC0805
RC1206
CC0603
R2
1kΩ
5
6
2
1
U4
9
10
RC0603
RC0603
74VCX86
R11
51kΩ
JP47
R17
10kΩ
O1N
O1P
O2N
O2P
AGND; 3, 4, 5
OUT 2
S4
R42
49.9kΩ
RC1206
AGND; 3, 4, 5
OUT1
S3
RC0603
SPSDO
RC0603
CC0603
R43
49.9kΩ
T1-1T
JP17
R12
C70
51kΩ
0.1µF
4
3
T3
JP30
DVDD; 14
DGND; 7
8
JP14
JP15
JP29
JP16
JP13
5
6
2
1
JP8
CC0603
R16
10kΩ
4
T1-1T
T2
JP28
JP4
RC0603
RC0603
C70
0.1µF
3
R9
51kΩ
R10
51kΩ
AD9777
Rev. C | Page 53 of 60
02706-107
13
15
17
19
21
23
25
27
29
31
33
35
37
39
14
16
18
20
22
24
26
28
30
32
34
36
38
40
RC1206
R15
220Ω
11
12
RIBBON
J1
9
10
5
6
7
3
4
8
1
2
DATA-A
2
2
3
3
4
4
5
5
6
6
7
7
8
8
9
RP1
1
RP1
2
RP1
3
RP1
4
RP1
5
RP1
6
RP1
7
RP1
8
RP2
1
RP2
2
RP2
3
RP2
4
RP2
5
RP2
6
RP2
7
RP2
8
22Ω
16
22Ω
15
22Ω
14
22Ω
13
22Ω
12
22Ω
11
22Ω
10
22Ω
9
22Ω
16
22Ω
15
22Ω
14
22Ω
13
22Ω
12
22Ω
11
22Ω
10
22Ω
9
RCOM
RP5
50Ω
10
1
R1 R2 R3 R4 R5 R6 R7 R8 R9
2
3
4
5
6
7
8
9
ADCLK
OPCLK
Figure 107. AD9777 Evaluation Board Input (A Channel) and Clock Buffer Circuitry
K
2
74LCX112
U7
15
CLR
CLK
J
Q 5
1
OPCLK_2
Q 6
2
U4
OPCLK_3
3
DVDD; 14
AGND; 7
74VCX86
11
J
K
14
CLR
CLK
74LCX112
U7
13
12
10
PRE
1
3
6
C52 +
4.7µF
6.3V
DVDD
C31 +
4.7µF
6.3V
DVDD
CX3
C30 +
4.7µF
6.3V
DVDD
AGND; 8
DVDD; 16
Q 7
Q 9
DVDD; 14
AGND; 7
74VCX86
U4
8
DVDD; 14
AGND; 7
PRE
5
4
U3
6
DVDD; 14
AGND; 7
74VCX86
U3
3
DVDD; 14
AGND; 7
74VCX86
U3
74VCX86
4
DVDD
RP8
DNP
AD00
AD01
AD02
AD03
AD04
AD05
AD06
9
10
5
4
2
1
AD07
CX2
CX1
AD08
AD09
AD10
AD11
AD12
AD13
AD14
AD15
RP7
10 DNP
R1 R2 R3 R4 R5 R6 R7 R8 R9
9 10 RP6
1 2 3 4 5 6 7 8 9 10
50Ω
RCOM
RCOM
R1 R2 R3 R4 R5 R6 R7 R8 R9
R1 R2 R3 R4 R5 R6 R7 R8 R9
1
1
RCOM
ACASE
ACASE
ACASE
C53
0.1µF
C34
0.1µF
C33
0.1µF
CC0805
CC0805
CC0805
AD9777
Rev. C | Page 54 of 60
02706-108
33
35
37
39
34
36
38
40
RIBBON
J2
31
32
23
24
29
21
22
30
19
20
27
17
18
28
15
16
25
13
14
26
11
7
8
12
5
6
9
3
4
10
1
2
DATA-B
2
2
3
3
4
4
5
5
6
6
7
7
8
8
9
8
7
6
5
4
3
2
1
8
7
6
5
4
3
2
1
RP4 22Ω
RP4 22Ω
RP4 22Ω
RP4 22Ω
RP4 22Ω
RP4 22Ω
RP4 22Ω
RP4 22Ω
RP3 22Ω
RP3 22Ω
RP3 22Ω
RP3 22Ω
RP3 22Ω
RP3 22Ω
RP3 22Ω
RP3 22Ω
9
10
11
12
13
14
15
16
9
10
11
12
13
14
15
2
3
4
5
6
7
8
9
BD00
BD01
BD02
BD03
BD04
BD05
BD06
BD07
BD08
BD09
BD10
BD11
BD12
BD13
BD14
BD15
RP9
10 DNP
R1 R2 R3 R4 R5 R6 R7 R8 R9
16
RCOM
RP12
10 50Ω
1
R1 R2 R3 R4 R5 R6 R7 R8 R9
Figure 108. AD9777 Evaluation Board Input (B Channel) and SPI Port Circuitry
ACASE
DVDD
SPSDO
SPSDI
SPCLK
SPCSB
+
C43
4.7µF CC805
6.3V
9 10 RP11
1 2 3 4 5 6 7 8 9 10
RP10
50Ω
RCOM
DNP
RCOM
R1 R2 R3 R4 R5 R6 R7 R8 R9
R1 R2 R3 R4 R5 R6 R7 R8 R9
1
1
RCOM
C50
0.1µF
2
c
U5
+ C49
4.7µF
6.3V
U5
U5
U5
10
U5
74AC14
8
U5
U6
12
DGND; 7
DVDD; 14
U6
6
U6
DGND; 7
74AC14 DVDD; 14
5
DGND; 7
74AC14 DVDD; 14
3
U6
8
U6
DGND; 7
74AC14 DVDD; 14
9
DGND; 7
74AC14 DVDD; 14
11
10
13
9
DGND; 7
DVDD; 14
11
DGND; 7
DVDD; 14
12
RC0805
ACASE
DVDD
+
RC0805
C44
4.7µF
6.3V
RC0805
R24
DNP RC0805
R22
DNP
JP41
JP42
RC0805
R20
DNP
R21
DNP RC0805
RC0805
R45
9kΩ
R48
9kΩ
R50
9kΩ
CLKVDD; 8
CGND; 5
4
2
DGND; 7
DVDD; 14 74AC14
5
CC805
C48
1nF
R19
100Ω
CC805
MC100EPT22
3
6 U8
4
R18
200Ω
RC0805
DGND; 7
74AC14
DVDD; 14
3
c
1
2
CGND; 5
CLKVDD; 8
U8
13
c
RC0805
MC100EPT22
c
C47
1nF
DGND; 7
74AC14 DVDD; 14
U6
C60
0.1µF
c
7
R13
120Ω
DGND; 7
DVDD; 14
1
CC805
RC0805
R4
120Ω
R14
200Ω
DGND; 7
74AC14 DVDD; 14
1
74AC14
6
74AC14
4
74AC14
ACASE
CLKDD
ACLKX
CC805
C46
0.1µF
RC0805
CLKVDD
CLKVDD
CLKN
CLKP
CLKVDD
CC805
6
5
4
3
2
C51
0.1µF
SPI PORT
P1
1
c
c
AD9777
02706-109
AD9777
02706-110
Figure 109. AD9777 Evaluation Board Components, Top Side
Figure 110. AD9777 Evaluation Board Components, Bottom Side
Rev. C | Page 55 of 60
02706-111
AD9777
02706-112
Figure 111. AD9777 Evaluation Board Layout, Layer One (Top)
Figure 112. AD9777 Evaluation Board Layout, Layer Two (Ground Plane)
Rev. C | Page 56 of 60
02706-113
AD9777
02706-114
Figure 113. AD9777 Evaluation Board Layout, Layer Three (Power Plane)
Figure 114. AD9777 Evaluation Board Layout, Layer Four (Bottom)
Rev. C | Page 57 of 60
AD9777
OUTLINE DIMENSIONS
14.20
14.00 SQ
13.80
0.75
0.60
0.45
1.20
MAX
12.20
12.00 SQ
11.80
80
61
61
1
60
80
1
60
PIN 1
EXPOSED
PAD
TOP VIEW
(PINS DOWN)
BOTTOM VIEW
0° MIN
1.05
1.00
0.95
0.15
0.05
SEATING
PLANE
6.00
BSC SQ
0.20
0.09
7°
3.5°
0°
0.08 MAX
COPLANARITY
(PINS UP)
20
41
40
21
VIEW A
41
20
21
40
0.50 BSC
LEAD PITCH
0.27
0.22
0.17
VIEW A
ROTATED 90° CCW
COMPLIANT TO JEDEC STANDARDS MS-026-ADD-HD
Figure 115. 80-Lead Thin Quad Flat Package, Exposed Pad [TQFP_EP]
(SV-80-1)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD9777BSV
AD9777BSVRL
AD9777BSVZ1
AD9777BSZVRL1
AD9777-EB
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
80-Lead Thin Quad Flat Package, Exposed Pad [TQFP_EP]
80-Lead Thin Quad Flat Package, Exposed Pad [TQFP_EP]
80-Lead Thin Quad Flat Package, Exposed Pad [TQFP_EP]
80-Lead Thin Quad Flat Package, Exposed Pad [TQFP_EP]
Evaluation Board
Z = Pb-free part.
Rev. C | Page 58 of 60
Package Option
SV-80-1
SV-80-1
SV-80-1
SV-80-1
AD9777
NOTES
Rev. C | Page 59 of 60
AD9777
NOTES
© 2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C02706-0-1/06(C)
Rev. C | Page 60 of 60