a
Low Power
Video Op Amp with Disable
AD810
FEATURES
High Speed
80 MHz Bandwidth (3 dB, G = +1)
75 MHz Bandwidth (3 dB, G = +2)
1000 V/ms Slew Rate
50 ns Settling Time to 0.1% (VO = 10 V Step)
Ideal for Video Applications
30 MHz Bandwidth (0.1 dB, G = +2)
0.02% Differential Gain
0.048 Differential Phase
Low Noise
2.9 nV/√Hz Input Voltage Noise
13 pA/√Hz Inverting Input Current Noise
Low Power
8.0 mA Supply Current max
2.1 mA Supply Current (Power-Down Mode)
High Performance Disable Function
Turn-Off Time 100 ns
Break Before Make Guaranteed
Input to Output Isolation of 64 dB (OFF State)
Flexible Operation
Specified for 65 V and 615 V Operation
62.9 V Output Swing Into a 150 V Load (VS = 65 V)
APPLICATIONS
Professional Video Cameras
Multimedia Systems
NTSC, PAL & SECAM Compatible Systems
Video Line Driver
ADC/DAC Buffer
DC Restoration Circuits
CONNECTION DIAGRAM
8-Pin Plastic Mini-DIP (N), SOIC (R)
and Cerdip (Q) Packages
OFFSET
NULL
1
8
DISABLE
–IN
2
7
+V
S
+IN
3
6
OUTPUT
–VS
4
5
OFFSET
NULL
AD810
TOP VIEW
PRODUCT DESCRIPTION
The AD810 is a composite and HDTV compatible, current
feedback, video operational amplifier, ideal for use in systems
such as multimedia, digital tape recorders and video cameras.
The 0.1 dB flatness specification at bandwidth of 30 MHz
(G = +2) and the differential gain and phase of 0.02% and
0.04° (NTSC) make the AD810 ideal for any broadcast quality
video system. All these specifications are under load conditions
of 150 Ω (one 75 Ω back terminated cable).
The AD810 is ideal for power sensitive applications such as
video cameras, offering a low power supply current of 8.0 mA
max. The disable feature reduces the power supply current to
only 2.1 mA, while the amplifier is not in use, to conserve
power. Furthermore the AD810 is specified over a power supply
range of ± 5 V to ± 15 V.
The AD810 works well as an ADC or DAC buffer in video
systems due to its unity gain bandwidth of 80 MHz. Because the
AD810 is a transimpedance amplifier, this bandwidth can be
maintained over a wide range of gains while featuring a low
noise of 2.9 nV/√Hz for wide dynamic range applications.
0.20
–45
–90
CLOSED-LOOP GAIN – dB
1
–135
VS = ±15V
0
–180
GAIN
±5V
–225
–1
±2.5V
–270
–2
DIFFERENTIAL GAIN – %
PHASE
PHASE SHIFT – Degrees
GAIN = +2
RL = 150Ω
0.09
GAIN = +2
RF = 715Ω
RL = 150Ω
fC = 3.58MHz
100 IRE
MODULATED RAMP
0.08
0.07
0.14
0.10
0.05
GAIN
0.04
0.08
PHASE
0.03
0.06
0.02
0.04
0.01
0.02
±5V
–4
0.16
0.12
0.06
VS = ±15V
–3
0.18
DIFFERENTIAL PHASE – Degrees
0.10
0
±2.5V
–5
0
1
10
100
FREQUENCY – MHz
1000
Closed-Loop Gain and Phase vs. Frequency, G = +2,
RL = 150, RF = 715 Ω
5
6
7
8
9
10
11
12
13
14
0
15
SUPPLY VOLTAGE – ± Volts
Differential Gain and Phase vs. Supply Voltage
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
AD810–SPECIFICATIONS (@ T = +258C and V = 615 V dc, R = 150 V unless otherwise noted)
A
Parameter
DYNAMIC PERFORMANCE
3 dB Bandwidth
0.1 dB Bandwidth
Full Power Bandwidth
S
L
Conditions
VS
Min
(G = +2) RFB = 715
(G = +2) RFB = 715
(G = +1) RFB = 1000
(G = +10) RFB = 270
(G = +2) RFB = 715
(G = +2) RFB = 715
VO = 20 V p-p,
RL = 400 Ω
RL = 150 Ω
RL = 400 Ω
10 V Step, G = –1
10 V Step, G = –1
f = 3.58 MHz
f - 3.58 MHz
f = 3.58 MHz
f = 3.58 MHz
f = 10 MHz, VO = 2 V p-p
RL = 400 Ω, G = +2
±5 V
± 15 V
± 15 V
± 15 V
±5 V
± 15 V
40
55
40
50
13
15
AD810A
Typ
Max
50
75
80
65
22
30
Min
40
55
40
50
13
15
AD810S1
Typ
Max
Units
50
75
80
65
22
30
MHz
MHz
MHz
MHz
MHz
MHz
16
350
1000
50
125
0.02
0.04
0.04
0.045
MHz
V/µs
V/µs
ns
ns
%
%
Degrees
Degrees
± 15 V
±5 V
± 15 V
± 15 V
± 15 V
± 15 V
±5 V
± 15 V
±5 V
16
350
1000
50
125
0.02
0.04
0.04
0.045
± 15 V
–61
TMIN–TMAX
± 5 V, ± 15 V
± 5 V, ± 15 V
1.5
2
7
6
7.5
1.5
4
15
6
15
mV
mV
µV/°C
TMIN–TMAX
TMIN–TMAX
± 5 V, ± 15 V
± 5 V, ± 15 V
0.7
2
5
7.5
0.8
2
5
10
µA
µA
OPEN-LOOP
TRANSRESISTANCE
TMIN–TMAX
VO = ± 10 V, RL = 400 Ω
VO = ± 2.5 V, RL = 100 Ω
± 15 V
±5 V
1.0
0.3
3.5
1.2
1.0
0.2
3.5
1.0
MΩ
MΩ
OPEN-LOOP
DC VOLTAGE GAIN
TMIN–TMAX
VO = ± 10 V, RL = 400 Ω
VO = ± 2.5 V, RL = 100 Ω
± 15 V
±5 V
86
76
100
88
80
72
100
88
dB
dB
COMMON-MODE REJECTION
VOS
TMIN–TMAX
VCM = ± 12 V
VCM = ± 2.5 V
TMIN–TMAX
± 15 V
±5 V
± 5 V, ± 15 V
56
52
64
60
0.1
56
50
0.4
64
60
0.1
0.4
dB
dB
µA/V
72
0.05
0.3
72
0.05
0.3
dB
µA/V
Slew Rate2
Settling Time to 0.1%
Settling Time to 0.01%
Differential Gain
Differential Phase
Total Harmonic Distortion
INPUT OFFSET VOLTAGE
Offset Voltage Drift
INPUT BIAS CURRENT
–Input
+Input
± Input Current
0.05
0.07
0.07
0.08
0.05
0.07
0.07
0.08
–61
dBc
± 4.5 V to ± 18 V
POWER SUPPLY REJECTION
VOS
± Input Current
TMIN–TMAX
TMIN–TMAX
INPUT VOLTAGE NOISE
f = 1 kHz
± 5 V, ± 15 V
2.9
2.9
nV/√Hz
INPUT CURRENT NOISE
–IIN, f = 1 kHz
+IIN, f = 1 kHz
± 5 V, ± 15 V
± 5 V, ± 15 V
13
1.5
13
1.5
pA/√Hz
pA/√Hz
INPUT COMMON-MODE
VOLTAGE RANGE
OUTPUT CHARACTERISTICS
Output Voltage Swing3
Short-Circuit Current
Output Current
OUTPUT RESISTANCE
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
DISABLE CHARACTERISTICS4
OFF Isolation
OFF Output Impedance
RL = 150 Ω, TMIN–TMAX
RL = 400 Ω
RL = 400 Ω, TMIN–TMAX
TMIN–TMAX
65
±5 V
± 15 V
± 2.5
± 12
± 3.0
± 13
± 2.5
± 12
±3
± 13
V
V
±5 V
± 15 V
± 15 V
± 15 V
± 5 V, ± 15 V
± 2.5
± 12.5
± 12
± 2.9
± 12.9
± 2.5
± 12.5
± 12
± 2.9
± 12.9
150
60
V
V
V
mA
mA
15
Ω
10
40
2
MΩ
Ω
pF
64
(RF+ RG)i13 pF
dB
40
Open Loop (5 MHz)
+Input
–Input
+Input
60
150
60
30
15
± 15 V
± 15 V
± 15 V
f = 5 MHz, See Figure 43
See Figure 43
2.5
10
40
2
64
(RF + RG)i13 pF
–2–
2.5
REV. A
AD810
Parameter
Conditions
5
VS
Turn On Time
Turn Off Time
Disable Pin Current
ZOUT = Low, See Figure 54
ZOUT = High
Disable Pin = 0 V
Min Disable Pin Current to
Disable
TMIN–TMAX
POWER SUPPLY
Operating Range
+25°C to TMAX
TMIN
±5 V
± 15 V
170
100
50
290
± 5 V, ± 15 V
30
± 2.5
± 3.0
±5 V
± 15 V
± 5 V, ± 15 V
±5 V
± 15 V
Quiescent Current
TMIN–TMAX
Power-Down Current
AD810A
Typ
Max
Min
6.7
6.8
8.3
1.8
2.1
Min
AD810S1
Typ
Max
170
100
50
290
75
400
75
400
± 2.5
± 3.5
6.7
6.8
9
1.8
2.1
ns
ns
µA
µA
µA
30
± 18
± 18
7.5
8.0
10.0
2.3
2.8
Units
± 18
± 18
7.5
8.0
11.0
2.3
2.8
V
V
mA
mA
mA
mA
mA
NOTES
1
See Analog Devices Military Data Sheet for 883B Specifications.
2
Slew rate measurement is based on 10% to 90% rise time with the amplifier configured for a gain of –10.
3
Voltage Swing is defined as useful operating range, not the saturation range.
4
Disable guaranteed break before make.
5
Turn On Time is defined with ± 5 V supplies using complementary output CMOS to drive the disable pin.
Specifications subject to change without notice.
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V
Internal Power Dissipation2 . . . . . . . Observe Derating Curves
Output Short Circuit Duration . . . . Observe Derating Curves
Common-Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . . ± VS
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . ± 6 V
Storage Temperature Range
Plastic DIP . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +125°C
Cerdip . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C
Small Outline IC . . . . . . . . . . . . . . . . . . . –65°C to +125°C
Operating Temperature Range
AD810A . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C
AD810S . . . . . . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . +300°C
MAXIMUM POWER DISSIPATION
The maximum power that can be safely dissipated by the
AD810 is limited by the associated rise in junction temperature.
For the plastic packages, the maximum safe junction temperature is 145°C. For the cerdip package, the maximum junction
temperature is 175°C. If these maximums are exceeded momentarily, proper circuit operation will be restored as soon as the die
temperature is reduced. Leaving the device in the “overheated”
condition for an extended period can result in device burnout.
To ensure proper operation, it is important to observe the
derating curves.
2.4
2.2
TOTAL POWER
DISSIPATION – Watts
ABSOLUTE MAXIMUM RATINGS 1
NOTES
1
Stresses above those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. This is a stress rating only and functional
operation of the device at these or any other conditions above those indicated in the
operational section of this specification is not implied. Exposure to absolute
maximum raring conditions for extended periods may affect device reliability.
2
8-Pin Plastic Package: θJA = 90°C/Watt; 8-Pin Cerdip Package: θJA = 110°C/Watt;
8-Pin SOIC Package: θJA = 150°C/Watt.
8-PIN
MINI-DIP
2.0
1.8
1.6
1.4
1.2
8-PIN
CERDIP
0.8
0.6
0.4
–60
8-PIN
SOIC
–40
–20
0
20
40
60
80
100
120
140
AMBIENT TEMPERATURE – °C
ESD SUSCEPTIBILITY
ESD (electrostatic discharge) sensitive device. Electrostatic
charges as high as 4000 volts, which readily accumulate on the
human body and on test equipment, can discharge without
detection. Although the AD810 features ESD protection
circuitry, permanent damage may still occur on these devices if
they are subjected to high energy electrostatic discharges.
Therefore, proper ESD precautions are recommended to avoid
any performance degradation or loss of functionality.
Maximum Power Dissipation vs. Temperature
While the AD810 is internally short circuit protected, this may
not be sufficient to guarantee that the maximum junction
temperature is not exceeded under all conditions.
SEE TEXT
+VS
0.1µF
10kΩ
7
2
ORDERING GUIDE
1
AD810
Model
Temperature
Range
Package
Description
Package
Option
AD810AN
AD810AR
AD810AR-REEL
5962-9313201MPA
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–55°C to +125°C
8-Pin Plastic DIP
8-Pin Plastic SOIC
8-Pin Plastic SOIC
8-Pin Cerdip
N-8
R-8
R-8
Q-8
REV. A
8-PIN
MINI-DIP
1.0
3
4
5
6
0.1µF
–VS
Offset Null Configuration
–3–
20
MAGNITUDE OF THE OUTPUT VOLTAGE – ±Volts
MAGNITUDE OF THE OUTPUT VOLTAGE – ±Volts
AD810 –Typical Characteristics
20
15
NO LOAD
10
RL = 150Ω
5
15
NO LOAD
10
RL = 150Ω
5
0
0
0
5
10
15
SUPPLY VOLTAGE – ±Volts
0
20
Figure 1. Input Common-Mode Voltage Range vs.
Supply Voltage
20
10
30
9
±15V SUPPLY
SUPPLY CURRENT – mA
OUTPUT VOLTAGE – Volts p-p
10
15
SUPPLY VOLTAGE – ±Volts
Figure 2. Output Voltage Swing vs. Supply
35
25
20
15
10
VS = ±15V
8
VS = ±5V
7
6
±5V SUPPLY
5
5
0
10
100
1k
LOAD RESISTANCE – Ohms
4
–60
10k
–20
0
20
40
60
80
100
120
140
Figure 4. Supply Current vs. Junction Temperature
10
10
8
INPUT OFFSET VOLTAGE – mV
8
6
NONINVERTING INPUT
4
VS = ±5V, ±15V
2
0
–2
INVERTING INPUT
VS = ±5V, ±15V
–4
–6
6
4
VS = ±5V
2
0
VS = ±15V
–2
–4
–6
–8
–8
–10
–60
–40
JUNCTION TEMPERATURE – °C
Figure 3. Output Voltage Swing vs. Load Resistance
INPUT BIAS CURRENT – µA
5
–40
–20
0
20
40
60
80
100
120
–10
–60
140
–40
–20
0
20
40
60
80
100
120
140
JUNCTION TEMPERATURE – °C
JUNCTION TEMPERATURE – °C
Figure 6. Input Offset Voltage vs. Junction Temperature
Figure 5. Input Bias Current vs. Temperature
–4–
REV. A
Typical Characteristics– AD810
120
100
200
VS = ± 15V
OUTPUT CURRENT – mA
SHORT CIRCUIT CURRENT – mA
250
VS = ±15V
150
100
80
60
VS = ± 5V
40
VS = ±5V
50
–60
–40
–20
0
+20
+40
+60
+80
+100 +120
20
–60
+140
–40
–20
+20
+40
+60
+80
+100 +120 +140
Figure 8. Linear Output Current vs. Temperature
Figure 7. Short Circuit Current vs. Temperature
1M
10.0
GAIN = 2
VS = ±5V
OUTPUT RESISTANCE – Ω
CLOSED-LOOP OUTPUT RESISTANCE – Ω
0
JUNCTION TEMPERATURE – °C
JUNCTION TEMPERATURE – °C
RF = 715Ω
1.0
VS = ±15V
0.1
0.01
10k
100k
1M
FREQUENCY – Hz
10M
100k
10k
1k
100
100k
100M
1M
10M
100M
FREQUENCY – Hz
Figure 10. Output Resistance vs. Frequency,
Disabled State
Figure 9. Closed-Loop Output Resistance vs. Frequency
100
100
30
VS = ±15V
VS = ±5V TO ±15V
±
20
OUTPUT LEVEL FOR 3% THD
RL = 400Ω
15
10
VS = ±5V
INVERTING INPUT
CURRENT NOISE
10
10
VOLTAGE NOISE
NONINVERTING INPUT
CURRENT NOISE
5
0
100k
1M
10M
FREQUENCY – Hz
1
10
100M
100
1k
FREQUENCY – Hz
10k
1
100k
Figure 12. Input Voltage and Current Noise vs. Frequency
Figure 11. Large Signal Frequency Response
REV. A
CURRENT NOISE – pA/ Hz
VOLTAGE NOISE – nV/ Hz
OUTPUT VOLTAGE – Volts p-p
25
–5–
AD810 –Typical Characteristics
90
70
POWER SUPPLY REJECTION – dB
80
COMMON-MODE REJECTION – dB
100
80
70
60
50
40
RF = 715Ω
AV = +2
60
VS = ±15V
50
VS = ±5V
40
30
CURVES ARE FOR WORST CASE
CONDITION WHERE ONE SUPPLY
IS VARIED WHILE THE OTHER IS
HELD CONSTANT
20
10
30
20
10k
100k
1M
FREQUENCY – Hz
10M
100M
10k
Figure 13. Common-Mode Rejection vs. Frequency
100k
1M
FREQUENCY – Hz
10M
100M
Figure 14. Power Supply Rejection vs. Frequency
–40
–40
–60
HARMONIC DISTORTION – dBc
HARMONIC DISTORTION – dBc
VO = 2V p-p
RL = 100Ω
GAIN = +2
VS = ±5V
2nd HARMONIC
–80
3rd HARMONIC
VS = ±15V
–100
2nd
–120
RL = 400Ω
–80
VOUT = 20V p-p
–100
2nd HARMONIC
3rd HARMONIC
VOUT = 2V p-p
–120
2nd
3rd
100
±15V SUPPLIES
GAIN = +2
–60
3rd
1k
10k
100k
FREQUENCY – Hz
1M
–140
100
10M
1k
10k
100k
1M
10M
FREQUENCY – Hz
Figure 15. Harmonic Distortion vs. Frequency (RL = 100 Ω)
Figure 16. Harmonic Distortion vs. Frequency (RL = 400 Ω)
1200
10
6
1000
0.01%
4
0.1%
SLEW RATE – V/µs
OUTPUT SWING FROM ±V TO 0V
8
2
RF = RG = 1kΩ
0
RL = 400Ω
–2
–4
0.1%
0.01%
RL = 400Ω
800
GAIN = –10
GAIN = +10
600
GAIN = +2
400
–6
–8
–10
0
200
20
40
60
80
100 120 140
SETTLING TIME – ns
160
180
200
Figure 17. Output Swing and Error vs. Settling Time
2
4
6
8
10
12
14
SUPPLY VOLTAGE – ±Volts
16
18
Figure 18. Slew Rate vs. Supply Voltage
–6–
REV. A
Typical Characteristics, Noninverting Connection–AD810
1V
RF
20nS
100
+VS
RG
VIN 90
0.1µF
VO TO
TEKTRONIX
P6201 FET
PROBE
7
2
3
VO
RL
0.1µF
10
50Ω
0%
–VS
1V
Figure 19. Noninverting Amplifier Connection
Figure 20. Small Signal Pulse Response, Gain = +1,
RF = 1 kΩ, RL = 150 Ω, VS = ± 15 V
0
–45
CLOSED-LOOP GAIN – dB
–90
1
–135
VS = ±15V
0
–180
±5V
–1
–225
GAIN
±2.5V
–270
–2
–3
VS = ±15V
–4
±5V
±2.5V
10
100
FREQUENCY – MHz
1
VS = ±15V
–135
±5V
–180
–1
–270
VS = ±15V
–3
±5V
–4
±2.5V
10
100
FREQUENCY – MHz
1000
Figure 22. Closed-Loop Gain and Phase vs. Frequency,
G= +1, RF = 1 kΩ for ± 15 V, 910 Ω for ± 5 V and ± 2.5 V
200
G = +1
RL = 150Ω
VO = 250mV p-p
180
PEAKING ≤ 1dB
160
–3dB BANDWIDTH – MHz
–3dB BANDWIDTH – MHz
–225
GAIN
1
80
RF = 750Ω
70
60
PEAKING ≤ 0.1 dB
50
RF = 1kΩ
40
G = +1
RL = 1kΩ
VO = 250mV p-p
PEAKING ≤ 1dB
140
120
100
PEAKING ≤ 0.1dB
RF = 750Ω
80
60
RF = 1kΩ
40
30
RF = 1.5kΩ
20
RF = 1.5kΩ
20
2
4
6
8
10
12
14
16
18
2
SUPPLY VOLTAGE – ±Volts
4
6
8
10
12
14
16
18
SUPPLY VOLTAGE – ±Volts
Figure 23. Bandwidth vs. Supply Voltage,
Gain = +1, RL = 150 Ω
REV. A
±2.5V
–2
110
90
–90
0
1000
Figure 21. Closed-Loop Gain and Phase vs. Frequency,
G= +1. RF = 1 kΩ for ± 15 V, 910 Ω for ± 5 V and ± 2.5 V
100
–45
–5
–5
1
GAIN = +1
RL = 1kΩ
PHASE
CLOSED-LOOP GAIN – dB
PHASE
0
PHASE SHIFT – Degrees
GAIN = +1
RL = 150Ω
PHASE SHIFT – Degrees
HP8130
PULSE
GENERATOR
4
VO
6
AD810
VIN
Figure 24. –3 dB Bandwidth vs. Supply Voltage
G = +1, RL = 1 kΩ
–7–
AD810–Typical Characteristics, Noninverting Connection
100mV
100
90
100
VIN 90
VO
VO
10
10
0%
0%
10V
1V
Figure 26. Large Signal Pulse Response, Gain = +10,
RF = 442 Ω, RL = 400 Ω, VS = ± 15 V
Figure 25. Small Signal Pulse Response, Gain = +10,
RF = 442 Ω, RL = 150 Ω, VS = ± 15 V
GAIN = +10
RF = 270Ω
0
21
–135
20
–180
VS = ±15V
19
18
–225
±5V
GAIN
VS = ±15V
17
±5V
16
±2.5V
PHASE
–270
±2.5V
10
100
FREQUENCY – MHz
±5V
16
±2.5V
–3dB BANDWIDTH – MHz
–3dB BANDWIDTH – MHz
VS = ±15V
17
1
RL = 150Ω
VO = 250mV p-p
70
PEAKING ≤ 0.5dB
RF = 232Ω
40
RF = 442Ω
–270
±2.5V
1000
10
100
FREQUENCY – MHz
Figure 28. Closed-Loop Gain and Phase vs. Frequency,
G = +10, RL = 1 kΩ
G = +10
50
–225
±5V
18
100
60
–180
VS = ±15V
GAIN
100
80
–90
19
1000
Figure 27. Closed-Loop Gain and Phase vs. Frequency,
G = +10, RL = 150 Ω
90
–45
–135
20
15
1
RF = 270Ω
RL = 1kΩ
21
CLOSED-LOOP GAIN – dB
CLOSED-LOOP GAIN – dB
–90
0
GAIN = +10
PHASE SHIFT – Degrees
–45
RL = 150Ω
PHASE
15
50nS
1V
PHASE SHIFT– Degrees
VIN
20nS
PEAKING ≤ 0.1dB
90
G = +10
RL = 1kΩ
VO = 250m V p-p
80
PEAKING ≤ 0.5dB
70
RF = 232Ω
60
50
40
RF = 442Ω
PEAKING ≤ 0.1dB
30
30
RF = 1kΩ
20
2
4
6
8
10
12
14
SUPPLY VOLTAGE – ±Volts
20
16
18
RF = 1kΩ
2
4
6
8
10
12
14
SUPPLY VOLTAGE – ±Volts
16
18
Figure 30. –3 dB Bandwidth vs. Supply Voltage,
Gain = +10, RL = 1 kΩ
Figure 29. –3 dB Bandwidth vs. Supply Voltage,
Gain = +10, RL = 150 Ω
–8–
REV. A
Typical Characteristics, Inverting Connection– AD810
1V
20nS
RF
100
+VS
RG
VIN 90
0.1µF
VO TO
TEKTRONIX
P6201 FET
PROBE
7
2
VIN
HP8130
PULSE
GENERATOR
3
4
VO
VO
6
AD810
RL
0.1µF
10
0%
1V
–VS
Figure 31. Inverting Amplifier Connection
Figure 32. Small Signal Pulse Response, Gain = –1,
RF = 681 Ω, RL = 150 Ω, VS = ± 5 V
45
CLOSED-LOOP GAIN – dB
0
–1
GAIN
–2
0
–45
–90
VS = ±15V
±5V
–4
±2.5V
10
100
FREQUENCY – MHz
–3dB BANDWIDTH – MHz
90
GAIN
VS = ±15V
–3
±5V
–4
±2.5V
180
VO = 250mV p-p
160
PEAKING ≤ 1.0dB
RF = 500Ω
60
RF = 681Ω
PEAKING ≤ 0.1dB
40
30
140
RL = 1kΩ
VO = 250mV p-p
120
PEAKING ≤ 1.0dB
100
RF = 500Ω
80
PEAKING ≤ 0.1dB
60
RF = 649Ω
40
6
8
10
12
14
SUPPLY VOLTAGE – ±Volts
16
2
18
Figure 35. –3 dB Bandwidth vs. Supply Voltage,
Gain = –1, RL = 150 Ω
REV. A
RF = 1kΩ
20
4
1000
10
100
FREQUENCY – MHz
RF = 1kΩ
20
2
–90
G = –1
80
50
1
–45
±2.5V
–2
G = –1
RL = 150
70
0
VS = ±15V
±5V
Figure 34. Closed-Loop Gain and Phase vs. Frequency,
G = –1, RL = 1 kΩ, RF = 681 Ω for VS = ± 15 V, 620 Ω for
± 5 V and ± 2.5 V
–3dB BANDWIDTH – MHz
100
0
1000
Figure 33. Closed-Loop Gain and Phase vs. Frequency
G = –1, RL = 150 Ω, RF = 681 Ω for ± 15 V, 620 Ω for ± 5 V
and ± 2.5 V
45
–1
–5
1
90
1
±2.5V
–3
–5
VS = ±15V
±5V
135
RL = 1kΩ
PHASE SHIFT – Degrees
90
1
GAIN = –1
PHASE
CLOSED-LOOP GAIN – dB
135
RL = 150Ω
PHASE
180
PHASE SHIFT – Degrees
180
GAIN = –1
4
6
8
10
12
14
SUPPLY VOLTAGE – ±Volts
16
18
Figure 36. –3 dB Bandwidth vs. Supply Voltage,
Gain = –1, RL = 1 kΩ
–9–
AD810 –Typical Characteristics, Inverting Connection
100mV
1V
20nS
100
100
VIN
50nS
VIN
90
90
VO
VO
10
10
0%
0%
10V
1V
Figure 38. Large Signal Pulse Response, Gain = –10,
RF = 442 Ω, RL = 400 Ω, VS = ± 15 V
Figure 37. Small Signal Pulse Response, Gain = –10,
RF = 442 Ω, RL = 150 Ω, VS = ± 15 V
90
RL = 150Ω
CLOSED-LOOP GAIN – dB
135
21
45
20
0
VS = ±15V
19
±5V
GAIN
18
17
±5V
±2.5V
16
1
10
100
FREQUENCY – MHz
80
0
±5V
GAIN
±2.5V
18
±5V
16
±2.5V
1
10
100
FREQUENCY – MHz
G = –10
70
60
RF = 249Ω
RF = 442Ω
40
30
90
RF = 750Ω
4
1000
80
NO PEAKING
RL = 1kΩ
VO = 250mV p- p
70
60
RF = 249Ω
50
40
RF = 442Ω
30
2
–90
VS = ±15V
17
NO PEAKING
RL = 150Ω
VO = 250mV p- p
20
–45
100
G = –10
50
VS = ±15V
19
Figure 40. Closed-Loop Gain and Phase vs. Frequency,
G = –10, RL = 1 kΩ
–3dB BANDWIDTH – MHz
–3dB BANDWIDTH – MHz
90
45
20
1000
Figure 39. Closed-Loop Gain and Phase vs. Frequency,
G = –10, RL = 150 Ω
100
21
15
15
135
90
RL = 1kΩ
–90
±2.5V
VS = ±15V
–45
RF = 249Ω
PHASE
CLOSED-LOOP GAIN – dB
RF = 249Ω
PHASE SHIFT – Degrees
PHASE
GAIN = –10
PHASE SHIFT – Degrees
180
180
GAIN = –10
6
8
10
12
14
SUPPLY VOLTAGE – ±Volts
RF = 750Ω
20
16
2
18
Figure 41. –3 dB Bandwidth vs. Supply Voltage, G = –10,
RL = 150 Ω
4
6
8
10
12
14
SUPPLY VOLTAGE – ±Volts
16
18
Figure 42. –3 dB Bandwidth vs. Supply Voltage, G = –10,
R L = 1 kΩ
–10–
REV. A
Applications– AD810
GENERAL DESIGN CONSIDERATIONS
PRINTED CIRCUIT BOARD LAYOUT
The AD810 is a current feedback amplifier optimized for use in
high performance video and data acquisition systems. Since it
uses a current feedback architecture, its closed-loop bandwidth
depends on the value of the feedback resistor. Table I below
contains recommended resistor values for some useful closedloop gains and supply voltages. As you can see in the table, the
closed-loop bandwidth is not a strong function of gain, as it
would be for a voltage feedback amp. The recommended
resistor values will result in maximum bandwidths with less than
0.1 dB of peaking in the gain vs. frequency response.
As with all wideband amplifiers, PC board parasitics can affect
the overall closed-loop performance. Most important are stray
capacitances at the output and inverting input nodes. (An added
capacitance of 2 pF between the inverting input and ground will
add about 0.2 dB of peaking in the gain of 2 response, and
increase the bandwidth to 105 MHz.) A space (3/16" is plenty)
should be left around the signal lines to minimize coupling.
Also, signal lines connecting the feedback and gain resistors
should be short enough so that their associated inductance does
not cause high frequency gain errors. Line lengths less than 1/4"
are recommended.
The –3 dB bandwidth is also somewhat dependent on the power
supply voltage. Lowering the supplies increases the values of
internal capacitances, reducing the bandwidth. To compensate
for this, smaller values of feedback resistor are sometimes used
at lower supply voltages. The characteristic curves illustrate that
bandwidths of over 100 MHz on 30 V total and over 50 MHz
on 5 V total supplies can be achieved.
Table I. –3 dB Bandwidth vs. Closed-Loop Gain and
Resistance Values (RL = 150 V)
VS = 615 V
Closed-Loop
Gain
RFB
RG
–3 dB BW
(MHz)
+1
+2
+10
–1
–10
1 kΩ
715 Ω
270 Ω
681 Ω
249 Ω
715 Ω
30 Ω
681 Ω
24.9 Ω
80
75
65
70
65
VS = 65 V
Closed-Loop
Gain
RFB
RG
–3 dB BW
(MHz)
+1
+2
+10
–1
–10
910 Ω
715 Ω
270 Ω
620 Ω
249 Ω
715 Ω
30 Ω
620 Ω
24.9 Ω
50
50
50
55
50
Optimum flatness when driving a coax cable is possible only
when the driven cable is terminated at each end with a resistor
matching its characteristic impedance. If coax were ideal, then
the resulting flatness would not be affected by the length of the
cable. While outstanding results can be achieved using
inexpensive cables, some variation in flatness due to varying
cable lengths is to be expected.
POWER SUPPLY BYPASSING
ACHIEVING VERY FLAT GAIN RESPONSE AT
HIGH FREQUENCY
Achieving and maintaining gain flatness of better than 0.1 dB
above 10 MHz is not difficult if the recommended resistor
values are used. The following issues should be considered to
ensure consistently excellent results.
CHOICE OF FEEDBACK AND GAIN RESISTOR
Because the 3 dB bandwidth depends on the feedback resistor,
the fine scale flatness will, to some extent, vary with feedback
resistor tolerance. It is recommended that resistors with a 1%
tolerance be used if it is desired to maintain exceptional flatness
over a wide range of production lots.
REV. A
QUALITY OF COAX CABLE
Adequate power supply bypassing can be critical when
optimizing the performance of a high frequency circuit.
Inductance in the power supply leads can contribute to resonant
circuits that produce peaking in the amplifier's response. In
addition, if large current transients must be delivered to the
load, then bypass capacitors (typically greater than 1 µF) will be
required to provide the best settling time and lowest distortion.
Although the recommended 0.1 µF power supply bypass
capacitors will be sufficient in most applications, more elaborate
bypassing (such as using two paralleled capacitors) may be
required in some cases.
POWER SUPPLY OPERATING RANGE
The AD810 will operate with supplies from ± 18 V down to
about ± 2.5 V. On ± 2.5 V the low distortion output voltage
swing will be better than 1 V peak to peak. Single supply
operation can be realized with excellent results by arranging for
the input common-mode voltage to be biased at the supply
midpoint.
OFFSET NULLING
A 10 kΩ pot connected between Pins 1 and 5, with its wiper
connected to V+, can be used to trim out the inverting input
current (with about ± 20 µA of range). For closed-loop gains
above about 5, this may not be sufficient to trim the output
offset voltage to zero. Tie the pot's wiper to ground through a
large value resistor (50 kΩ for ± 5 V supplies, 150 kΩ for ± 15 V
supplies) to trim the output to zero at high closed-loop gains.
–11–
AD810
CAPACITIVE LOADS
LOAD CAPACITANCE – pF
When used with the appropriate feedback resistor, the AD810
can drive capacitive loads exceeding 1000 pF directly without
oscillation. By using the curves in Figure 45 to chose the resistor
value, less than 1 dB of peaking can easily be achieved without
sacrificing much bandwidth. Note that the curves were
generated for the case of a 10 kΩ load resistor, for smaller load
resistances, the peaking will be less than indicated by Figure 45.
Another method of compensating for large load capacitances is
to insert a resistor in series with the loop output as shown in
Figure 43. In most cases, less than 50 Ω is all that is needed to
achieve an extremely flat gain response.
VS = ±5V
100
VS = ±15V
GAIN = +2
RL = 1kΩ
10
1
Figures 44 to 46 illustrate the outstanding performance that can
be achieved when driving a 1000 pF capacitor.
1k
0
2k
3k
4k
FEEDBACK RESISTOR – Ω
RF
Figure 45. Max Load Capacitance for Less than 1 dB of
Peaking vs. Feedback Resistor
0.1µF
+VS
1000
1.0µF
RG
5V
7
2
RS (OPTIONAL)
6
AD810
3
VIN
4
RT
1.0µF
VIN 100
VO
CL
100nS
90
RL
0.1µF
–VS
VOUT
Figure 43. Circuit Options for Driving a Large
Capacitive Load
0%
CLOSED-LOOP GAIN – dB
5V
G = +2
VS = ±15V
RL= 10kΩ
CL = 1000pF
9
6
Figure 46. AD810 Driving a 1000 pF Load,
Gain = +2, RF = 750 Ω, RS = 11 Ω, RL = 10 kΩ
3
0
RF = 4.5kΩ
RS = 0
–3
DISABLE MODE
RF = 750Ω
RS = 11Ω
–6
–9
1
10
FREQUENCY – MHz
100
Figure 44. Performance Comparison of Two Methods for
Driving a Large Capacitive Load
By pulling the voltage on Pin 8 to common (0 V), the AD810
can be put into a disabled state. In this condition, the supply
current drops to less than 2.8 mA, the output becomes a high
impedance, and there is a high level of isolation from input to
output. In the case of a line driver for example, the output
impedance will be about the same as for a 1.5 kΩ resistor (the
feedback plus gain resistors) in parallel with a 13 pF capacitor
(due to the output) and the input to output isolation will be
better than 65 dB at 1 MHz.
Leaving the disable pin disconnected (floating) will leave the
AD810 operational in the enabled state.
In cases where the amplifier is driving a high impedance load,
the input to output isolation will decrease significantly if the
input signal is greater than about 1.2 V peak to peak. The
isolation can be restored back to the 65 dB level by adding a
dummy load (say 150 Ω) at the amplifier output. This will
attenuate the feedthrough signal. (This is not an issue for
multiplexer applications where the outputs of multiple AD810s
are tied together as long as at least one channel is in the ON
state.) The input impedance of the disable pin is about 35 kΩ in
parallel with a few pF. When grounded, about 50 µA flows out
–12–
REV. A
AD810
DIFFERENTIAL GAIN – %
0.09
When operated on ± 15 V supplies, the AD810 disable pin may
be driven by open drain logic such as the 74C906. In this case,
adding a 10 kΩ pull-up resistor from the disable pin to the plus
supply will decrease the enable time to about 150 ns. If there is
a nonzero voltage present on the amplifier's output at the time it
is switched to the disabled state, some additional decay time will
be required for the output voltage to relax to zero. The total
time for the output to go to zero will generally be about 250 ns
and is somewhat dependent on the load impedance.
75Ω
3
VIN
4
75Ω
0.12
0.10
0.05
GAIN
0.04
0.08
PHASE
0.03
0.06
0.02
0.04
0.01
0.02
6
7
8
9
10
11
12
13
0
15
14
+0.1
RL = 150Ω
±15V
NORMALIZED GAIN – dB
0
–0.1
±5V
±2.5
+0.1
RL= 1k
0
±15V
–0.1
75Ω
CABLE
±5V
VOUT
6
AD810
0.14
SUPPLY VOLTAGE – ± Volts
0.1µF
2
75Ω
CABLE
0.06
5
715Ω
7
0.07
0.16
0
The AD810 is designed to offer outstanding performance at
closed-loop gains of one or greater. At a gain of 2, the AD810
makes an excellent video line driver. The low differential gain
and phase errors and wide –0.1 dB bandwidth are nearly
independent of supply voltage and load (as seen in Figures 49
and 50).
+VS
0.18
GAIN = +2
RF = 715Ω
RL = 150Ω
fC = 3.58MHz
100 IRE
MODULATED RAMP
0.08
Figure 49. Differential Gain and Phase vs. Supply Voltage
OPERATION AS A VIDEO LINE DRIVER
715Ω
0.20
0.10
±2.5
75Ω
0.1µF
1M
100k
–VS
10M
FREQUENCY – Hz
100M
Figure 50. Fine-Scale Gain (Normalized) vs. Frequency
for Various Supply Voltages, Gain = +2, RF = 715 Ω
Figure 47. A Video Line Driver Operating at a Gain of +2
110
100
CLOSED-LOOP GAIN – dB
–90
1
–135
VS = ±15V
0
–180
GAIN
±5V
–225
–1
90
–3dB BANDWIDTH – MHz
–45
PHASE SHIFT – Degrees
0
GAIN = +2
RL = 150Ω
PHASE
±2.5V
–270
–2
VS = ±15V
–3
G = +2
PEAKING ≤ 1.0dB
RL = 150Ω
VO = 250mV p-p
RF = 500
80
70
60
PEAKING ≤ 0.1dB
RF = 750
50
40
RF = 1k
30
±5V
–4
20
±2.5V
–5
1
10
100
FREQUENCY – MHz
2
1000
4
6
8
10
12
14
16
18
SUPPLY VOLTAGE - ±Volts
Figure 48. Closed-Loop Gain and Phase vs. Frequency,
G = +2, RL = 150, RF = 715 Ω
REV. A
DIFFERENTIAL PHASE – Degrees
of the disable the disable pin for ± 5 V supplies. If driven by
complementary output CMOS logic (such as the 74HC04), the
disable time (until the output goes high impedance) is about
100 ns and the enable time (to low impedance output) is about
170 ns on ± 5 V supplies. The enable time can be extended to
about 750 ns by using open drain logic such as the 74HC05.
Figure 51. –3 dB Bandwidth vs. Supply Voltage,
Gain = +2, RL = 150 Ω
–13–
AD810
2:1 VIDEO MULTIPLEXER
500mV
750Ω
750Ω
The outputs of two AD810s can be wired together to form a
2:1 mux without degrading the flatness of the gain response.
Figure 54 shows a recommended configuration which results in
–0.1 dB bandwidth of 20 MHz and OFF channel isolation of
77 dB at 10 MHz on ± 5 V supplies. The time to switch between
channels is about 0.75 µs when the disable pins are driven by
open drain output logic. Adding pull-up resistors to the logic
outputs or using complementary output logic (such as the
74HC04) reduces the switching time to about 180 ns. The
switching time is only slightly affected by the signal level.
+5V
0.1µF
2
7
AD810
VINA
3
6
0.1µF
4
8
75Ω
75Ω
75Ω
CABLE
VOUT
–5V
75Ω
750Ω
750Ω
+5V
2
500nS
7
AD810
VINB
100
0.1µF
3
90
75Ω
4
6
0.1µF
8
–5V
VSW
10
74HC04
0%
5V
Figure 54. A Fast Switching 2:1 Video Mux
Figure 52. Channel Switching Time for the 2:1 Mux
0
–45
CLOSED-LOOP GAIN – dB
FEEDTHROUGH – dB
–90
0.5
–50
–60
–70
–80
0
–135
–180
–0.5
GAIN
–225
–1.0
PHASE SHIFT – Degrees
PHASE
–40
–270
–1.5
VS = ±5V
–2.0
–2.5
–3.0
1
10
FREQUENCY – MHz
100
–90
1
10
FREQUENCY – MHz
100
Figure 55. 2:1 Mux ON Channel Gain and Phase vs.
Frequency
Figure 53. 2:1 Mux OFF Channel Feedthrough vs.
Frequency
–14–
REV. A
AD810
N:1 MULTIPLEXER
1kΩ
A multiplexer of arbitrary size can be formed by combining the
desired number of AD810s together with the appropriate
selection logic. The schematic in Figure 58 shows a
recommendation for a 4:1 mux which may be useful for driving
a high impedance such as the input to a video A/D converter
(such as the AD773). The output series resistors effectively
compensate for the combined output capacitance of the OFF
channels plus the input capacitance of the A/D while
maintaining wide bandwidth. In the case illustrated, the –0.1 dB
bandwidth is about 20 MHz with no peaking. Switching time
and OFF channel isolation (for the 4:1 mux) are about 250 ns
and 60 dB at 10 MHz, respectively.
PHASE
–45
CLOSED-LOOP GAIN – dB
0.5
–90
0
–135
–180
–0.5
GAIN
–1.0
RL = 10kΩ
–2.0
CL = 10pF
2
VIN, A
75Ω
3
33Ω
6
8
4
SELECT A
0.1µF
–VS
1kΩ
+VS
0.1µF
3
75Ω
7
33Ω
AD810
VIN, B
6
8
4
SELECT B
0.1µF
–VS
VOUT
1kΩ
+VS
RL
0.1µF
2
–2.5
3
–3.0
10
FREQUENCY – MHz
100
75Ω
CL
7
33Ω
AD810
VIN, C
1
7
AD810
–225
VS = ±15V
–1.5
0.1µF
2
PHASE SHIFT – Degrees
0
+VS
6
8
4
SELECT C
0.1µF
–VS
Figure 56. 4:1 Mux ON Channel Gain and Phase vs.
Frequency
1kΩ
+VS
0.1µF
–30
FEEDTHROUGH – dB
2
3
75Ω
6
8
4
SELECT D
0.1µF
–50
–VS
Figure 58. A 4:1 Multiplexer Driving a High Impedance
–60
–70
1
10
FREQUENCY – MHz
100
Figure 57. 4:1 Mux OFF Channel Feedthrough vs.
Frequency
REV. A
33Ω
AD810
VIN, D
–40
7
–15–
AD810
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
Plastic Mini-DIP (N) Package
8
0.25
(6.35)
PIN 1
1
C1737–24–10/92
5
0.31
(7.87)
4
0.30 (7.62)
REF
0.39 (9.91) MAX
0.035 ±0.01
(0.89 ±0.25)
0.165 ±0.01
(4.19 ±0.25)
0.011 ±0.003
(0.28 ±0.08)
0.18 ±0.03
(4.57 ±0.76)
0.125
(3.18)
MIN
0.018
±0.003
(0.46 ±0.08)
0.10
(2.54)
BSC
0.033
(0.84)
NOM
15°
0°
SEATING
PLANE
Cerdip (Q) Package
0.055 (1.40) MAX
0.005 (0.13) MIN
8
5
0.310 (7.87)
0.220 (5.59)
PIN 1
1
4
0.320 (8.13)
0.290 (7.37)
0.405 (10.29) MAX
0.060 (1.52)
0.015 (0.38)
0.200
(5.08)
MAX
0.150
(3.81)
MIN
0.200 (5.08)
0.125 (3.18)
0.023 (0.58) 0.100 0.070 (1.78)
0.014 (0.36) (2.54) 0.030 (0.76)
BSC
0.015 (0.38)
0.008 (0.20)
15°
0°
SEATING
PLANE
8-Pin SOIC (R) Package
0.150 (3.81)
8
5
0.244 (6.20)
0.228 (5.79)
4
1
0.197 (5.01)
0.189 (4.80)
0.102 (2.59)
0.094 (2.39)
0.010 (0.25)
0.004 (0.10)
0.050
(1.27)
BSC
0.019 (0.48)
0.014 (0.36)
0.020 (0.051) x 45°
CHAMF
0.190 (4.82)
0.170 (4.32)
8°
0°
0.090
(2.29)
10°
0°
0.098 (0.2482)
0.075 (0.1905)
PRINTED IN U.S.A.
PIN 1
0.157 (3.99)
0.150 (3.81)
0.030 (0.76)
0.018 (0.46)
All brand or product names mentioned are trademarks or registered trademarks of their respective holders.
–16–
REV. A