10 MHz, 20 V/μs, G = 1, 10, 100, 1000 iCMOS
Programmable Gain Instrumentation Amplifier
AD8253
Data Sheet
FEATURES
FUNCTIONAL BLOCK DIAGRAM
DGND WR
Small package: 10-lead MSOP
Programmable gains: 1, 10, 100, 1000
Digital or pin-programmable gain setting
Wide supply: ±5 V to ±15 V
Excellent dc performance
High CMRR: 100 dB (minimum), G = 100
Low gain drift: 10 ppm/°C (maximum)
Low offset drift: 1.2 μV/°C (maximum), G = 1000
Excellent ac performance
Fast settling time: 780 ns to 0.001% (maximum)
High slew rate: 20 V/μs (minimum)
Low distortion: −110 dB THD at 1 kHz,10 V swing
High CMRR over frequency: 100 dB to 20 kHz (minimum)
Low noise: 10 nV/√Hz, G = 1000 (maximum)
Low power: 4 mA
2
A0
5
4
LOGIC
–IN 1
7
OUT
+IN 10
8
3
9
+VS
–VS
REF
06983-001
AD8253
Figure 1.
80
70
APPLICATIONS
60
Data acquisition
Biomedical analysis
Test and measurement
G = 1000
GAIN (dB)
50
GENERAL DESCRIPTION
The AD8253 is an instrumentation amplifier with digitally
programmable gains that has gigaohm (GΩ) input impedance,
low output noise, and low distortion, making it suitable for
interfacing with sensors and driving high sample rate analog-todigital converters (ADCs).
The AD8253 user interface consists of a parallel port that allows
users to set the gain in one of two different ways (see Figure 1
for the functional block diagram). A 2-bit word sent via a bus
can be latched using the WR input. An alternative is to use
transparent gain mode, where the state of logic levels at the gain
port determines the gain.
40
G = 100
30
20
G = 10
10
0
G=1
–10
–20
1k
10k
100k
1M
FREQUENCY (Hz)
10M
100M
006983-023
It has a high bandwidth of 10 MHz, low THD of −110 dB, and
fast settling time of 780 ns (maximum) to 0.001%. Offset drift and
gain drift are guaranteed to 1.2 μV/°C and 10 ppm/°C, respectively,
for G = 1000. In addition to its wide input common voltage range,
it boasts a high common-mode rejection of 100 dB at G = 1000
from dc to 20 kHz. The combination of precision dc performance
coupled with high speed capabilities makes the AD8253 an
excellent candidate for data acquisition. Furthermore, this
monolithic solution simplifies design and manufacturing and
boosts performance of instrumentation by maintaining a tight
match of internal resistors and amplifiers.
Rev. B
A1
6
Figure 2. Gain vs. Frequency
Table 1. Instrumentation Amplifiers by Category
General
Purpose
AD82201
AD8221
AD8222
AD82241
AD8228
1
Zero
Drift
AD82311
AD85531
AD85551
AD85561
AD85571
Mil
Grade
AD620
AD621
AD524
AD526
AD624
Low
Power
AD6271
AD6231
AD82231
High Speed
PGA
AD8250
AD8251
AD8253
Rail-to-rail output.
The AD8253 is available in a 10-lead MSOP package and is
specified over the −40°C to +85°C temperature range, making it
an excellent solution for applications where size and packing
density are important considerations.
Document Feedback
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700 ©2008–2012 Analog Devices, Inc. All rights reserved.
Technical Support
www.analog.com
AD8253
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Power Supply Regulation and Bypassing ................................ 18
Applications ....................................................................................... 1
Input Bias Current Return Path ............................................... 18
General Description ......................................................................... 1
Input Protection ......................................................................... 18
Functional Block Diagram .............................................................. 1
Reference Terminal .................................................................... 19
Revision History ............................................................................... 2
Common-Mode Input Voltage Range ..................................... 19
Specifications..................................................................................... 3
Layout .......................................................................................... 19
Timing Diagram ........................................................................... 5
RF Interference ........................................................................... 19
Absolute Maximum Ratings............................................................ 6
Driving an Analog-to-Digital Converter ................................ 20
Maximum Power Dissipation ..................................................... 6
Applications Information .............................................................. 21
ESD Caution .................................................................................. 6
Differential Output .................................................................... 21
Pin Configuration and Function Descriptions ............................. 7
Setting Gains with a Microcontroller ...................................... 21
Typical Performance Characteristics ............................................. 8
Data Acquisition ......................................................................... 22
Theory of Operation ...................................................................... 16
Outline Dimensions ....................................................................... 23
Gain Selection ............................................................................. 16
Ordering Guide .......................................................................... 23
REVISION HISTORY
10/12—Rev. A to Rev. B
Changed Digital Input Voltage Low Maximum Parameter from
1.2 V to 2.1 V and Changed Digital Input Voltage High Typical
Parameter from 1.5 V to 2.8 V ........................................................ 4
Updated Outline Dimensions ....................................................... 23
8/08—Rev. 0 to Rev. A
Changes to Ordering Guide .......................................................... 23
7/08—Revision 0: Initial Version
Rev. B | Page 2 of 24
Data Sheet
AD8253
SPECIFICATIONS
+VS = +15 V, −VS = −15 V, VREF = 0 V @ TA = 25°C, G = 1, RL = 2 kΩ, unless otherwise noted.
Table 2.
Parameter
COMMON-MODE REJECTION RATIO (CMRR)
CMRR to 60 Hz with 1 kΩ Source Imbalance
G=1
G = 10
G = 100
G = 1000
CMRR to 20 kHz1
G=1
G = 10
G = 100
G = 1000
NOISE
Voltage Noise, 1 kHz, RTI
G=1
G = 10
G = 100
G = 1000
0.1 Hz to 10 Hz, RTI
G=1
G = 10
G = 100
G = 1000
Current Noise, 1 kHz
Current Noise, 0.1 Hz to 10 Hz
VOLTAGE OFFSET
Offset RTI VOS
Over Temperature
Average TC
Offset Referred to the Input vs. Supply (PSR)
INPUT CURRENT
Input Bias Current
Over Temperature2
Average TC
Input Offset Current
Over Temperature
Average TC
DYNAMIC RESPONSE
Small-Signal −3 dB Bandwidth
G=1
G = 10
G = 100
G = 1000
Settling Time 0.01%
G=1
G = 10
G = 100
G = 1000
Conditions
Min
Typ
80
96
100
100
100
120
120
120
Max
Unit
+IN = −IN = −10 V to +10 V
dB
dB
dB
dB
+IN = −IN = −10 V to +10 V
80
96
100
100
dB
dB
dB
dB
45
12
11
10
nV/√Hz
nV/√Hz
nV/√Hz
nV/√Hz
2.5
1
0.5
0.5
μV p-p
μV p-p
μV p-p
μV p-p
pA/√Hz
pA p-p
±150 + 900/G
±210 + 900/G
±1.2 + 5/G
±5 + 25/G
μV
μV
μV/°C
μV/V
50
60
400
40
40
160
nA
nA
pA/°C
nA
nA
pA/°C
5
60
G = 1, 10, 100, 1000
T = −40°C to +85°C
T = −40°C to +85°C
VS = ±5 V to ±15 V
5
T = −40°C to +85°C
T = −40°C to +85°C
40
5
T = −40°C to +85°C
T = −40°C to +85°C
10
4
550
60
MHz
MHz
kHz
kHz
ΔOUT = 10 V step
700
680
1.5
14
Rev. B | Page 3 of 24
ns
ns
μs
μs
AD8253
Parameter
Settling Time 0.001%
G=1
G = 10
G =100
G = 1000
Slew Rate
G=1
G = 10
G = 100
G = 1000
Total Harmonic Distortion + Noise
GAIN
Gain Range
Gain Error
G=1
G = 10, 100, 1000
Gain Nonlinearity
G=1
G = 10
G = 100
G = 1000
Gain vs. Temperature
INPUT
Input Impedance
Differential
Common Mode
Input Operating Voltage Range
Over Temperature3
OUTPUT
Output Swing
Over Temperature4
Short-Circuit Current
REFERENCE INPUT
RIN
IIN
Voltage Range
Gain to Output
DIGITAL LOGIC
Digital Ground Voltage, DGND
Digital Input Voltage Low
Digital Input Voltage High
Digital Input Current
Gain Switching Time5
tSU
tHD
t WR -LOW
t WR -HIGH
Data Sheet
Conditions
ΔOUT = 10 V step
Min
Typ
Max
Unit
780
880
1.8
1.8
ns
ns
μs
μs
20
20
12
2
−110
f = 1 kHz, RL = 10 kΩ, ±10 V,
G = 1, 10 Hz to 22 kHz bandpass filter
G = 1, 10, 100, 1000
OUT = ±10 V
V/μs
V/μs
V/μs
V/μs
dB
1
OUT = −10 V to +10 V
RL = 10 kΩ, 2 kΩ, 600 Ω
RL = 10 kΩ, 2 kΩ, 600 Ω
RL = 10 kΩ, 2 kΩ, 600 Ω
RL = 10 kΩ, 2 kΩ, 600 Ω
All gains
3
1000
V/V
0.03
0.04
%
%
5
3
18
110
10
ppm
ppm
ppm
ppm
ppm/°C
GΩpF
GΩpF
V
V
4||1.25
1||5
VS = ±5 V to ±15 V
T = −40°C to +85°C
−VS + 1
−VS + 1.2
+VS − 1.5
+VS − 1.7
T = −40°C to +85°C
−13.7
−13.7
+13.6
+13.6
37
20
+IN, −IN, REF = 0
1
+VS
−VS
1 ± 0.0001
Referred to GND
Referred to GND
Referred to GND
−VS + 4.25
DGND
2.8
0
+VS − 2.7
2.1
+VS
1
325
See Figure 3 timing diagram
Rev. B | Page 4 of 24
15
30
20
15
V
V
mA
kΩ
μA
V
V/V
V
V
V
μA
ns
ns
ns
ns
ns
Data Sheet
AD8253
Parameter
POWER SUPPLY
Operating Range
Quiescent Current, +IS
Quiescent Current, −IS
Over Temperature
TEMPERATURE RANGE
Specified Performance
Conditions
Min
Typ
Max
Unit
4.6
4.5
±15
5.3
5.3
6
V
mA
mA
mA
+85
°C
±5
T = −40°C to +85°C
−40
1
See Figure 20 for CMRR vs. frequency for more information on typical performance over frequency.
Input bias current over temperature: minimum at hot and maximum at cold.
3
See Figure 30 for input voltage limit vs. supply voltage and temperature.
4
See Figure 32, Figure 33, and Figure 34 for output voltage swing vs. supply voltage and temperature for various loads.
5
Add time for the output to slew and settle to calculate the total time for a gain change.
2
TIMING DIAGRAM
tWR-HIGH
tWR-LOW
WR
tSU
tHD
06983-003
A0, A1
Figure 3. Timing Diagram for Latched Gain Mode (See the Timing for Latched Gain Mode Section)
Rev. B | Page 5 of 24
AD8253
Data Sheet
ABSOLUTE MAXIMUM RATINGS
power is the voltage between the supply pins (VS) times the
quiescent current (IS). Assuming the load (RL) is referenced to
midsupply, the total drive power is VS/2 × IOUT, some of which is
dissipated in the package and some of which is dissipated in the
load (VOUT × IOUT).
Table 3.
Rating
±17 V
See Figure 4
Indefinite1
±VS
±VS
±VS
–65°C to +125°C
–40°C to +85°C
300°C
140°C
112°C/W
140°C
The difference between the total drive power and the load
power is the drive power dissipated in the package.
PD = Quiescent Power + (Total Drive Power − Load Power)
V
V
PD  VS  I S    S  OUT
 2
RL

 VOUT 2
–

RL

In single-supply operation with RL referenced to −VS, the worst
case is VOUT = VS/2.
Assumes the load is referenced to midsupply.
2
Temperature for specified performance is −40°C to +85°C. For performance
to +125°C, see the Typical Performance Characteristics section.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
MAXIMUM POWER DISSIPATION
The maximum safe power dissipation in the AD8253 package is
limited by the associated rise in junction temperature (TJ) on
the die. The plastic encapsulating the die locally reaches the
junction temperature. At approximately 140°C, which is the
glass transition temperature, the plastic changes its properties.
Even temporarily exceeding this temperature limit can change
the stresses that the package exerts on the die, permanently
shifting the parametric performance of the AD8253. Exceeding
a junction temperature of 140°C for an extended period can
result in changes in silicon devices, potentially causing failure.
The still-air thermal properties of the package and PCB (θJA),
the ambient temperature (TA), and the total power dissipated in
the package (PD) determine the junction temperature of the die.
The junction temperature is calculated as
Airflow increases heat dissipation, effectively reducing θJA. In
addition, more metal directly in contact with the package leads
from metal traces through holes, ground, and power planes
reduces the θJA.
Figure 4 shows the maximum safe power dissipation in the
package vs. the ambient temperature on a 4-layer JEDEC
standard board.
2.00
1.75
1.50
1.25
1.00
0.75
0.50
0.25
0
–40
–20
0
20
40
60
80
100
120
AMBIENT TEMPERATURE (°C)
Figure 4. Maximum Power Dissipation vs. Ambient Temperature
ESD CAUTION
TJ  TA  PD  θ JA 
The power dissipated in the package (PD) is the sum of the
quiescent power dissipation and the power dissipated in the
package due to the load drive for all outputs. The quiescent
Rev. B | Page 6 of 24
06983-004
1
MAXIMUM POWER DISSIPATION (W)
Parameter
Supply Voltage
Power Dissipation
Output Short-Circuit Current
Common-Mode Input Voltage
Differential Input Voltage
Digital Logic Inputs
Storage Temperature Range
Operating Temperature Range2
Lead Temperature (Soldering 10 sec)
Junction Temperature
θJA (4-Layer JEDEC Standard Board)
Package Glass Transition Temperature
Data Sheet
AD8253
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
–IN 1
DGND 2
10 +IN
AD8253
9
REF
8 +VS
TOP VIEW
A0 4 (Not to Scale) 7 OUT
A1 5
6
WR
06983-005
–VS 3
Figure 5. 10-Lead MSOP (RM-10) Pin Configuration
Table 4. Pin Function Descriptions
Pin No.
1
2
3
4
5
6
7
8
9
10
Mnemonic
−IN
DGND
−VS
A0
A1
WR
OUT
+VS
REF
+IN
Description
Inverting Input Terminal. True differential input.
Digital Ground.
Negative Supply Terminal.
Gain Setting Pin (LSB).
Gain Setting Pin (MSB).
Write Enable.
Output Terminal.
Positive Supply Terminal.
Reference Voltage Terminal.
Noninverting Input Terminal. True differential input.
Rev. B | Page 7 of 24
AD8253
Data Sheet
TYPICAL PERFORMANCE CHARACTERISTICS
TA @ 25°C, +VS = +15 V, −VS = −15 V, RL = 10 kΩ, unless otherwise noted.
210
240
180
150
NUMBER OF UNITS
NUMBER OF UNITS
210
120
90
180
150
120
90
60
60
30
–60
–40
–20
0
0
06983-006
0
20
CMRR (µV/V)
–60
–40
–20
0
20
40
60
INPUT OFFSET CURRENT (nA)
Figure 6. Typical Distribution of CMRR, G = 1
06983-009
30
Figure 9. Typical Distribution of Input Offset Current
90
180
80
70
120
NOISE (nV/√Hz)
NUMBER OF UNITS
150
90
60
60
50
G=1
G = 100
40
30
G = 10
20
30
10
0
–100
100
200
INPUT OFFSET VOLTAGE, VOSI , RTI (µV)
0
G = 1000
1
10
100
1k
10k
100k
FREQUENCY (Hz)
Figure 10. Voltage Spectral Density Noise vs. Frequency
Figure 7. Typical Distribution of Offset Voltage, VOSI
250
200
150
50
2µV/DIV
0
–90
–60
–30
0
30
60
INPUT BIAS CURRENT (nA)
90
1s/DIV
Figure 11. 0.1 Hz to 10 Hz RTI Voltage Noise, G = 1
Figure 8. Typical Distribution of Input Bias Current
Rev. B | Page 8 of 24
06983-011
100
06983-008
NUMBER OF UNITS
300
06983-010
–200
06983-007
0
Data Sheet
AD8253
18
16
14
12
10
8
6
4
2
0
0.01
0.1
1
WARM-UP TIME (Minutes)
Figure 15. Change in Input Offset Voltage vs. Warm-Up Time, G = 1000
Figure 12. 0.1 Hz to 10 Hz RTI Voltage Noise, G = 1000
140
18
16
120
G = 1000
G = 100
14
100
PSRR (dB)
12
10
8
6
80
G=1
60
G = 10
40
4
10
100
1k
10k
100k
FREQUENCY (Hz)
0
10
100
1k
10k
100k
1M
06983-016
1
06983-013
0
1M
06983-017
20
2
FREQUENCY (Hz)
Figure 16. Positive PSRR vs. Frequency, RTI
Figure 13. Current Noise Spectral Density vs. Frequency
140
G = 100
120
G = 1000
PSRR (dB)
100
80
G = 10
60
40
140pA/DIV
1s/DIV
G=1
20
06983-014
NOISE (pA/√Hz)
10
06983-015
06983-012
1s/DIV
500nV/DIV
CHANGE IN INPUT OFFSET VOLTAGE (µV)
20
0
10
100
1k
10k
100k
FREQUENCY (Hz)
Figure 17. Negative PSRR vs. Frequency, RTI
Figure 14. 0.1 Hz to 10 Hz Current Noise
Rev. B | Page 9 of 24
AD8253
Data Sheet
10.5
IB–
7.5
–20
6.0
–30
4.5
IOS
–40
3.0
–50
1.5
–60
–15
–10
–5
0
5
COMMON-MODE VOLTAGE (V)
0
15
10
G = 100
60
G = 10
40
20
G=1
0
10
100
1k
10k
100k
1M
130
FREQUENCY (Hz)
Figure 21. CMRR vs. Frequency, 1 kΩ Source Imbalance
Figure 18. Input Bias Current and Offset Current vs. Common-Mode Voltage
15
30
25
10
20
5
CMRR (µV/V)
15
10
IB–
5
0
IB+
–5
IOS
–10
0
–5
–10
–60
–40
–20
0
20
40
60
80
TEMPERATURE (°C)
100
120
140
–15
–50
06983-019
INPUT BIAS CURRENT AND
OFFSET CURRENT (nA)
G = 1000
80
06983-018
–10
100
06983-021
9.0
CMRR (dB)
0
INPUT OFFSET CURRENT (nA)
IB+
06983-022
10
–30
10
30
50
70
90
110
TEMPERATURE (°C)
Figure 19. Input Bias Current and Offset Current vs. Temperature
120
–10
Figure 22. CMRR vs. Temperature, G = 1
80
G = 1000
70
G = 100
100
60
50
80
GAIN (dB)
G=1
60
G = 10
40
40
G = 100
30
20
G = 10
10
0
20
G=1
–10
0
10
100
1k
10k
FREQUENCY (Hz)
100k
1M
–20
1k
06983-020
CMRR (dB)
G = 1000
Figure 20. CMRR vs. Frequency
10k
100k
1M
FREQUENCY (Hz)
Figure 23. Gain vs. Frequency
Rev. B | Page 10 of 24
10M
100M
006983-023
INPUT BIAS CURRENT (nA)
120
12.0
20
AD8253
40
400
30
300
NONLINEARITY (10 ppm/DIV)
20
10
0
–10
–20
–6
–4
–2
0
2
4
6
8
10
–200
–400
–10
30
12
INPUT COMMON-MODE VOLTAGE (V)
16
20
10
0
–10
–20
–6
–4
–2
0
2
4
6
8
10
OUTPUT VOLTAGE (V)
Figure 25. Gain Nonlinearity, G = 10, RL = 10 kΩ, 2 kΩ, 600 Ω
8
0
–20
–40
4
6
8
10
–8
10
OUTPUT VOLTAGE (V)
Figure 26. Gain Nonlinearity, G = 100, RL = 10 kΩ, 2 kΩ, 600 Ω
VS, ±15V
–14.1V, +7.3V
+13.8V, +7.3V
0V, +3.8V
+3.8V, +1.9V
VS = ±5V
+3.8V, –1.9V
0V, –4.2V
–14.1V, –7.3V
+13.8V, –7.3V
–12
0V, –14.2V
–12
–8
–4
0
4
8
12
16
0V, +13.7V
VS ±15V
8
–14.4V, +6V
0V, +3.8V
+14.1V, +6V
4
–4.3V, +2V
+4.3V, +2V
VS = ±5V
0
–4.3V, –2V
+4.3V, –2V
–4
+14.1V, –6V
0V, –4.2V
–14.4V, –6V
–8
–12
–16
–16
06983-026
–60
2
8
OUTPUT VOLTAGE (V)
INPUT COMMON-MODE VOLTAGE (V)
20
0
6
Figure 28. Input Common-Mode Voltage Range vs. Output Voltage, G = 1
40
–2
4
0V, +13.9V
–4V, –1.9V
12
–4
2
–4
60
–6
0
–4V, +1.9V
16
–8
–2
0
80
–80
–10
–4
4
–16
–16
06983-025
–30
–8
–6
Figure 27. Gain Nonlinearity, G = 1000, RL = 10 kΩ, 2 kΩ, 600 Ω
40
–40
–10
–8
OUTPUT VOLTAGE (V)
Figure 24. Gain Nonlinearity, G = 1, RL = 10 kΩ, 2 kΩ, 600 Ω
NONLINEARITY (10ppm/DIV)
–100
06983-028
–8
OUTPUT VOLTAGE (V)
NONLINEARITY (10ppm/DIV)
0
–300
06983-024
–40
–10
100
06983-027
–30
200
0V, –14.1V
–12
–8
–4
0
4
OUTPUT VOLTAGE (V)
8
12
16
06983-029
NONLINEARITY (10ppm/DIV)
Data Sheet
Figure 29. Input Common-Mode Voltage Range vs. Output Voltage, G = 1000
Rev. B | Page 11 of 24
AD8253
Data Sheet
+VS
+85°C
–0.2
–2
+25°C
–40°C
+2
+25°C
–40°C
+1
+125°C
–VS
4
8
10
12
16
14
Figure 30. Input Voltage Limit vs. Supply Voltage, G = 1, VREF = 0 V, RL = 10 kΩ
25
+IN
–IN
–IN
–40°C
–40°C
+25°C
+85°C
+0.8
+0.6
+0.4
+125°C
–VS
4
6
8
10
12
16
14
SUPPLY VOLTAGE (±VS)
Figure 33. Output Voltage Swing vs. Supply Voltage, G =1000, RL = 10 kΩ
+25°C
10
5
+IN
+25°C
+1.0
OUTPUT VOLTAGE SWING (V)
CURRENT (mA)
+85°C
+85°C
+Vs
–5
–10
–1.0
15
FAULT
CONDITION
(OVER-DRIVEN
INPUT)
G=1000
FAULT
CONDITION
(OVER-DRIVEN
INPUT)
15
G=1000
20
0
–0.8
+0.2
SUPPLY VOLTAGE (±VS)
10
–0.6
+85°C
6
+125°C
–0.4
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
–1
06983-033
+125°C
06983-030
INPUT VOLTAGE (V)
REFERRED TO SUPPLY VOLTAGES
+VS
–Vs
–15
–40°C
5
+125°C
0
+85°C
–5
+125°C
+25°C
–10
Figure 34. Output Voltage Swing vs. Load Resistance
+VS
+VS
–0.2
–40°C
+25°C
+85°C
+125°C
–0.4
–0.4
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
+125°C
–0.6
–0.8
–1.0
–1.2
+85°C
+25°C
–40°C
+1.2
–40°C
+85°C
+25°C
+1.0
+0.8
+0.6
+125°C
+0.4
–0.8
–1.2
–1.6
–2.0
+2.0
+1.6
+1.2
+0.8
+0.4
4
6
8
10
12
SUPPLY VOLTAGE (±VS)
14
16
06983-032
+0.2
Figure 32. Output Voltage Swing vs. Supply Voltage, G = 1000, RL = 2 kΩ
Rev. B | Page 12 of 24
–VS
4
6
8
10
12
14
OUTPUT CURRENT (mA)
Figure 35. Output Voltage Swing vs. Output Current
16
06983-035
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
10k
1k
LOAD RESISTANCE (Ω)
Figure 31. Fault Current Draw vs. Input Voltage, G = 1000, RL = 10 kΩ
–VS
–40°C
06983-034
10
–15
100
06983-031
DIFFERENTIAL INPUT VOLTAGE (V)
1
100m
10m
1m
100µ
–10µ/
10µ
–100µ
–1m
–10m
–100m
–10
–25
–1
–20
Data Sheet
AD8253
5V/DIV
NO
LOAD
47pF 100pF
1392ns TO 0.01%
1712ns TO 0.001%
2µs/DIV
2µs/DIV
TIME (µs)
Figure 36. Small-Signal Pulse Response for Various Capacitive Loads, G = 1
06983-039
20mV/DIV
06983-036
0.002%/DIV
Figure 39. Large-Signal Pulse Response and Settling Time,
G = 100, RL = 10 kΩ
5V/DIV
664ns TO 0.01%
744ns TO 0.001%
0.002%/DIV
0.002%/DIV
TIME (µs)
10µs/DIV
06983-037
2µs/DIV
12.88µs TO 0.01%
16.64µs TO 0.001%
TIME (µs)
Figure 37. Large-Signal Pulse Response and Settling Time, G = 1, RL = 10 kΩ
06983-040
5V/DIV
Figure 40. Large-Signal Pulse Response and Settling Time,
G = 1000, RL = 10 kΩ
5V/DIV
656ns TO 0.01%
840ns TO 0.001%
TIME (µs)
20mV/DIV
06983-038
2µs/DIV
2µs/DIV
Figure 41. Small-Signal Response,
G = 1, RL = 2 kΩ, CL = 100
Figure 38. Large-Signal Pulse Response and Settling Time,
G = 10, RL = 10 kΩ
Rev. B | Page 13 of 24
06983-041
0.002%/DIV
AD8253
Data Sheet
1400
1200
1000
TIME (ns)
SETTLED TO 0.001%
800
600
SETTLED TO 0.01%
400
2
4
6
8
10
12
14
16
18
20
06983-045
0
20
06983-046
200
20
06983-047
2µs/DIV
06983-042
20mV/DIV
STEP SIZE (V)
Figure 42. Small-Signal Response,
G = 10, RL = 2 kΩ, CL = 100 pF
Figure 45. Settling Time vs. Step Size, G = 1, RL = 10 kΩ
1400
1200
TIME (ns)
1000
SETTLED TO 0.001%
800
SETTLED TO 0.01%
600
400
20µs/DIV
200
06983-043
20mV/DIV
0
2
4
6
8
10
12
14
16
18
STEP SIZE (V)
Figure 43. Small-Signal Response,
G = 100, RL = 2 kΩ, CL = 100 pF
Figure 46. Settling Time vs. Step Size, G = 10, RL = 10 kΩ
2000
SETTLED TO 0.001%
1800
1600
1400
TIME (ns)
SETTLED TO 0.01%
1200
1000
800
600
20mV/DIV
20µs/DIV
06983-044
400
200
0
2
4
6
8
10
12
14
16
18
STEP SIZE (V)
Figure 44. Small-Signal Response, G = 1000, RL = 2 kΩ, CL = 100 pF
Figure 47. Settling Time vs. Step Size, G = 100, RL = 10 kΩ
Rev. B | Page 14 of 24
Data Sheet
AD8253
0
20
18
–10
SETTLED TO 0.001%
–20
16
–30
–40
12
THD + N (dB)
SETTLED TO 0.01%
10
8
–50
G = 1000
–60
G = 100
–70
G = 10
–80
6
–90
4
–110
0
–120
10
2
4
6
8
10
12
14
16
18
20
06983-048
2
STEP SIZE (V)
0
–20
–30
–40
–50
G = 1000
–70
G = 100
–80
G = 10
–90
–100
G=1
100
1k
10k
100k
FREQUENCY (Hz)
1M
06983-049
–110
–120
10
1k
10k
100k
Figure 50. Total Harmonic Distortion vs. Frequency,
10 Hz to 500 kHz Band-Pass Filter, 2 kΩ Load
–10
–60
100
FREQUENCY (Hz)
Figure 48. Settling Time vs. Step Size, G = 1000, RL = 10 kΩ
THD + N (dB)
G=1
–100
Figure 49. Total Harmonic Distortion vs. Frequency,
10 Hz to 22 kHz Band-Pass Filter, 2 kΩ Load
Rev. B | Page 15 of 24
1M
06983-050
TIME (µs)
14
AD8253
Data Sheet
THEORY OF OPERATION
+VS
+VS
A0
A1
2.2kΩ
+VS
–VS
–VS
1.2kΩ
–IN
10kΩ
A1
10kΩ
–VS
+VS
DIGITAL
GAIN
CONTROL
OUT
A3
–VS
+VS
+VS
+IN
10kΩ
A2
+VS
–VS
10kΩ
REF
–VS
+VS
2.2kΩ
DGND
WR
–VS
06983-061
1.2kΩ
–VS
Figure 51. Simplified Schematic
The AD8253 is a monolithic instrumentation amplifier based
on the classic 3-op-amp topology, as shown in Figure 51. It is
fabricated on the Analog Devices, Inc., proprietary iCMOS®
process that provides precision linear performance and a robust
digital interface. A parallel interface allows users to digitally
program gains of 1, 10, 100, and 1000. Gain control is achieved
by switching resistors in an internal precision resistor array (as
shown in Figure 51).
All internal amplifiers employ distortion cancellation circuitry
and achieve high linearity and ultralow THD. Laser-trimmed
resistors allow for a maximum gain error of less than 0.03% for
G = 1 and a minimum CMRR of 100 dB for G = 1000. A pinout
optimized for high CMRR over frequency enables the AD8253
to offer a guaranteed minimum CMRR over frequency of 80 dB
at 20 kHz (G = 1). The balanced input reduces the parasitics
that in the past had adversely affected CMRR performance.
Transparent Gain Mode
The easiest way to set the gain is to program it directly via a
logic high or logic low voltage applied to A0 and A1. Figure 52
shows an example of this gain setting method, referred to throughout the data sheet as transparent gain mode. Tie WR to the
negative supply to engage transparent gain mode. In this mode,
any change in voltage applied to A0 and A1 from logic low to
logic high, or vice versa, immediately results in a gain change.
Table 5 is the truth table for transparent gain mode, and Figure 52
shows the AD8253 configured in transparent gain mode.
+15V
10μF
0.1µF
WR
A1
A0
+IN
–15V
+5V
+5V
G = 1000
AD8253
GAIN SELECTION
REF
Rev. B | Page 16 of 24
–IN
DGND
10μF
DGND
0.1µF
–15V
NOTE:
1. IN TRANSPARENT GAIN MODE, WR IS TIED TO −VS.
THE VOLTAGE LEVELS ON A0 AND A1 DETERMINE
THE GAIN. IN THIS EXAMPLE, BOTH A0 AND A1 ARE
SET TO LOGIC HIGH, RESULTING IN A GAIN OF 1000.
06983-051
This section describes how to configure the AD8253 for basic
operation. Logic low and logic high voltage limits are listed in
the Specifications section. Typically, logic low is 0 V and logic
high is 5 V; both voltages are measured with respect to DGND.
Refer to the specifications table (Table 2) for the permissible
voltage range of DGND. The gain of the AD8253 can be set
using two methods: transparent gain mode and latched gain
mode. Regardless of the mode, pull-up or pull-down resistors
should be used to provide a well-defined voltage at the A0 and
A1 pins.
Figure 52. Transparent Gain Mode, A0 and A1 = High, G = 1000
Data Sheet
AD8253
Table 5. Truth Table Logic Levels for Transparent Gain Mode
Table 6. Truth Table Logic Levels for Latched Gain Mode
WR
WR
A1
Low
Low
High
High
−VS
−VS
−VS
−VS
A0
Low
High
Low
High
Gain
1
10
100
1000
Latched Gain Mode
Some applications have multiple programmable devices such
as multiplexers or other programmable gain instrumentation
amplifiers on the same PCB. In such cases, devices can share a
data bus. The gain of the AD8253 can be set using WR as a latch,
allowing other devices to share A0 and A1. Figure 53 shows a
schematic using this method, known as latched gain mode. The
AD8253 is in this mode when WR is held at logic high or logic
low, typically 5 V and 0 V, respectively. The voltages on A0 and A1
are read on the downward edge of the WR signal as it transitions
from logic high to logic low. This latches in the logic levels on
A0 and A1, resulting in a gain change. See the truth table listing
in Table 6 for more on these gain changes.
+15V
WR
10μF
0.1µF
A1
A1
A0
+IN
+5V
0V
+5V
0V
WR
+
A0
G = PREVIOUS
STATE
+5V
0V
G = 1000
AD8253
REF
–
–IN
DGND
0.1µF
–15V
NOTE:
1. ON THE DOWNWARD EDGE OF WR, AS IT TRANSITIONS
FROM LOGIC HIGH TO LOGIC LOW, THE VOLTAGES ON A0
AND A1 ARE READ AND LATCHED IN, RESULTING IN A
GAIN CHANGE. IN THIS EXAMPLE, THE GAIN SWITCHES TO G = 1000.
1
A0
Low
High
Low
High
X1
X1
X1
Gain
Change to 1
Change to 10
Change to 100
Change to 1000
No change
No change
No change
X = don’t care.
On power-up, the AD8253 defaults to a gain of 1 when in
latched gain mode. In contrast, if the AD8253 is configured in
transparent gain mode, it starts at the gain indicated by the
voltage levels on A0 and A1 on power-up.
Timing for Latched Gain Mode
In latched gain mode, logic levels at A0 and A1 must be held for
a minimum setup time, tSU, before the downward edge of WR
latches in the gain. Similarly, they must be held for a minimum
hold time, tHD, after the downward edge of WR to ensure that
the gain is latched in correctly. After tHD, A0 and A1 may change
logic levels, but the gain does not change until the next downward
edge of WR. The minimum duration that WR can be held high
is t WR -HIGH, and t WR -LOW is the minimum duration that WR can
be held low. Digital timing specifications are listed in Table 2.
The time required for a gain change is dominated by the settling
time of the amplifier. A timing diagram is shown in Figure 54.
When sharing a data bus with other devices, logic levels applied
to those devices can potentially feed through to the output of
the AD8253. Feedthrough can be minimized by decreasing the
edge rate of the logic signals. Furthermore, careful layout of the
PCB also reduces coupling between the digital and analog
portions of the board.
DGND
06983-052
10μF
High to Low
High to Low
High to Low
High to Low
Low to Low
Low to High
High to High
A1
Low
Low
High
High
X1
X1
X1
Figure 53. Latched Gain Mode, G = 1000
tWR-HIGH
tWR-LOW
WR
tSU
tHD
06983-053
A0, A1
Figure 54. Timing Diagram for Latched Gain Mode
Rev. B | Page 17 of 24
AD8253
Data Sheet
INCORRECT
POWER SUPPLY REGULATION AND BYPASSING
The AD8253 has high PSRR. However, for optimal performance,
a stable dc voltage should be used to power the instrumentation
amplifier. Noise on the supply pins can adversely affect performance. As in all linear circuits, bypass capacitors must be
used to decouple the amplifier.
REF
–VS
–VS
TRANSFORMER
TRANSFORMER
+VS
+VS
10µF
WR
A1
A0
AD8253
AD8253
REF
VOUT
10MΩ
LOAD
–VS
REF
–IN
–VS
THERMOCOUPLE
DGND
THERMOCOUPLE
+VS
10µF
–VS
+VS
C
06983-054
0.1µF
REF
C
1
fHIGH-PASS = 2πRC
AD8253
Figure 55. Supply Decoupling, REF, and Output Referred to Ground
C
REF
INPUT BIAS CURRENT RETURN PATH
The AD8253 input bias current must have a return path to its
local analog ground. When the source, such as a thermocouple,
cannot provide a return current path, one should be created
(see Figure 56).
R
AD8253
C
REF
R
–VS
–VS
CAPACITIVELY COUPLED
CAPACITIVELY COUPLED
06983-055
AD8253
DGND
AD8253
REF
+VS
+IN
+VS
AD8253
Place a 0.1 μF capacitor close to each supply pin. A 10 μF tantalum
capacitor can be used farther away from the part (see Figure 55)
and, in most cases, it can be shared by other precision integrated
circuits.
0.1µF
CORRECT
+VS
Figure 56. Creating an IBIAS Path
INPUT PROTECTION
All terminals of the AD8253 are protected against ESD. An external
resistor should be used in series with each of the inputs to limit
current for voltages greater than 0.5 V beyond either supply rail.
In such a case, the AD8253 safely handles a continuous 6 mA
current at room temperature. For applications where the AD8253
encounters extreme overload voltages, external series resistors
and low leakage diode clamps such as BAV199Ls, FJH1100s, or
SP720s should be used.
Rev. B | Page 18 of 24
Data Sheet
AD8253
REFERENCE TERMINAL
Coupling Noise
The reference terminal, REF, is at one end of a 10 kΩ resistor
(see Figure 51). The instrumentation amplifier output is
referenced to the voltage on the REF terminal; this is useful
when the output signal needs to be offset to voltages other than
its local analog ground. For example, a voltage source can be
tied to the REF pin to level shift the output so that the AD8253
can interface with a single-supply ADC. The allowable reference
voltage range is a function of the gain, common-mode input,
and supply voltages. The REF pin should not exceed either +VS
or −VS by more than 0.5 V.
To prevent coupling noise onto the AD8253, follow these
guidelines:
For best performance, especially in cases where the output is
not measured with respect to the REF terminal, source impedance to the REF terminal should be kept low because parasitic
resistance can adversely affect CMRR and gain accuracy.

INCORRECT
CORRECT
AD8253


Do not run digital lines under the device.
Run the analog ground plane under the AD8253.
Shield fast-switching signals with digital ground to avoid
radiating noise to other sections of the board, and never
run them near analog signal paths.
Avoid crossover of digital and analog signals.
Connect digital and analog ground at one point only
(typically under the ADC).
Power supply lines should use large traces to ensure a low
impedance path. Decoupling is necessary; follow the
guidelines listed in the Power Supply Regulation and
Bypassing section.
Common-Mode Rejection
The AD8253 has high CMRR over frequency, giving it greater
immunity to disturbances, such as line noise and its associated
harmonics, in contrast to typical in amps whose CMRR falls off
around 200 Hz. They often need common-mode filters at the
inputs to compensate for this shortcoming. The AD8253 is able
to reject CMRR over a greater frequency range, reducing the
need for input common-mode filtering.
AD8253
VREF
VREF
+
06983-056
OP1177
–



Figure 57. Driving the Reference Pin
COMMON-MODE INPUT VOLTAGE RANGE
The 3-op-amp architecture of the AD8253 applies gain and then
removes the common-mode voltage. Therefore, internal nodes
in the AD8253 experience a combination of both the gained
signal and the common-mode signal. This combined signal can
be limited by the voltage supplies even when the individual
input and output signals are not. Figure 28 and Figure 29 show
the allowable common-mode input voltage ranges for various
output voltages, supply voltages, and gains.
LAYOUT
Grounding
In mixed-signal circuits, low level analog signals need to be
isolated from the noisy digital environment. Designing with the
AD8253 is no exception. Its supply voltages are referenced to an
analog ground. Its digital circuit is referenced to a digital ground.
Although it is convenient to tie both grounds to a single ground
plane, the current traveling through the ground wires and PC
board can cause an error. Therefore, use separate analog and
digital ground planes. Only at one point, star ground, should
analog and digital ground meet.
Careful board layout maximizes system performance. To maintain
high CMRR over frequency, lay out the input traces symmetrically.
Ensure that the traces maintain resistive and capacitive balance;
this holds for additional PCB metal layers under the input pins
and traces. Source resistance and capacitance should be placed
as close to the inputs as possible. Should a trace cross the inputs
(from another layer), it should be routed perpendicular to the
input traces.
RF INTERFERENCE
RF rectification is often a problem when amplifiers are used in
applications where there are strong RF signals. The disturbance
can appear as a small dc offset voltage. High frequency signals
can be filtered with a low-pass RC network placed at the input
of the instrumentation amplifier, as shown in Figure 58. The
filter limits the input signal bandwidth according to the following
relationship:
FilterFreq DIFF 
FilterFreq CM 
where CD ≥ 10 CC.
The output voltage of the AD8253 develops with respect to the
potential on the reference terminal. Take care to tie REF to the
appropriate local analog ground or to connect it to a voltage that
is referenced to the local analog ground.
Rev. B | Page 19 of 24
1
2 π R( 2C D  C C )
1
2 π RC C
AD8253
Data Sheet
+15V
10µF
CC
R
+IN
VOUT
AD8253
CD
R
REF
–IN
CC
0.1µF
–15V
06983-057
10µF
+15V
Figure 58. RFI Suppression
10μF
Values of R and CC should be chosen to minimize RFI.
Mismatch between the R × CC at the positive input and the
R × CC at negative input degrades the CMRR of the AD8253.
By using a value of CD that is 10 times larger than the value of
CC, the effect of the mismatch is reduced and performance is
improved.
DRIVING AN ANALOG-TO-DIGITAL CONVERTER
0.1µF
WR
A1
+12V
–12V
A0
+IN
0.1μF
0.1μF
49.9Ω
AD8253
REF
AD7612
1nF
+5V
–IN
ADR435
DGND
10μF
An instrumentation amplifier is often used in front of an analogto-digital converter to provide CMRR. Usually, instrumentation
amplifiers require a buffer to drive an ADC. However, the low
output noise, low distortion, and low settle time of the AD8253
make it an excellent ADC driver.
Rev. B | Page 20 of 24
DGND
0.1µF
06983-058
0.1µF
In this example, a 1 nF capacitor and a 49.9 Ω resistor create an
antialiasing filter for the AD7612. The 1 nF capacitor also serves
to store and deliver necessary charge to the switched capacitor
input of the ADC. The 49.9 Ω series resistor reduces the burden
of the 1 nF load from the amplifier and isolates it from the kickback
current injected from the switched capacitor input of the AD7612.
Selecting too small a resistor improves the correlation between
the voltage at the output of the AD8253 and the voltage at the
input of the AD7612 but may destabilize the AD8253. A tradeoff must be made between selecting a resistor small enough to
maintain accuracy and large enough to maintain stability.
–15V
Figure 59. Driving an ADC
Data Sheet
AD8253
APPLICATIONS INFORMATION
DIFFERENTIAL OUTPUT
SETTING GAINS WITH A MICROCONTROLLER
+15V
In certain applications, it is necessary to create a differential
signal. High resolution analog-to-digital converters often require a
differential input. In other cases, transmission over a long distance
can require differential signals for better immunity to interference.
10μF
0.1µF
WR
A1
A0
+IN
Figure 61 shows how to configure the AD8253 to output a
differential signal. An op amp, the AD8675, is used in an
inverting topology to create a differential voltage. VREF sets the
output midpoint according to the equation shown in the figure.
Errors from the op amp are common to both outputs and are
thus common mode. Likewise, errors from using mismatched
resistors cause a common-mode dc offset error. Such errors are
rejected in differential signal processing by differential input
ADCs or instrumentation amplifiers.
MICROCONTROLLER
+
AD8253
REF
–
–IN
DGND
DGND
0.1µF
06983-059
10μF
–15V
Figure 60. Programming Gain Using a Microcontroller
When using this circuit to drive a differential ADC, VREF can be
set using a resistor divider from the ADC reference to make the
output ratiometric with the ADC.
+15V
0.1μF
AMPLITUDE
WR
+5V
A1
A0
+IN
–5V
AMPLITUDE
+
VOUTA = VIN + VREF
2
AD8253
VIN
G=1
–
0.1μF
+2.5V
0V
–2.5V
REF
TIME
4.99kΩ
DGND
–
–15V
4.99kΩ
–15V
56pF
+
AD8675 +15V
VREF
0V
AMPLITUDE
10μF
0.1µF
–15V
0.1µF
10μF
DGND
VOUTB = –VIN + VREF
2
Figure 61. Differential Output with Level Shift
Rev. B | Page 21 of 24
+2.5V
0V
–2.5V
TIME
06983-060
+15V
Data Sheet
DATA ACQUISITION
AMPLITUDE (dB)
The AD8253 makes an excellent instrumentation amplifier
for use in data acquisition systems. Its wide bandwidth, low
distortion, low settling time, and low noise enable it to
condition signals in front of a variety of 16-bit ADCs.
Figure 63 shows the AD825x as part of a total data acquisition
system. The quick slew rate of the AD8253 allows it to condition
rapidly changing signals from the multiplexed inputs. An FPGA
controls the AD7612, AD8253, and ADG1209. In addition,
mechanical switches and jumpers allow users to pin strap the
gains when in transparent gain mode.
This system achieved −116 dB of THD at 1 kHz and a signal-tonoise ratio of 91 dB during testing, as shown in Figure 62.
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
–120
–130
–140
–150
–160
–170
0
5
10
15
20
25
30
35
40
45
50
FREQUENCY (kHz)
Figure 62. FFT of the AD825x in a Total Data Acquisition System
Using the AD8253 1 kHz Signal
JMP
+12V +
+12V
0.1µF
10µF
+CH2
+CH3
+CH4
–CH4
–CH3
–CH2
–CH1
806Ω
806Ω
806Ω
806Ω
806Ω
806Ω
806Ω
4 S1A
+5V
2kΩ
10µF
2
EN
DGND
DGND
JMP
+5V
DGND
5 S2A
2kΩ
2
6 S3A
ALTERA
EPF6010ATC144-3
DGND 6
7 S4A
0Ω
DA 8
0Ω
CC +IN
10
ADG1209
10 S4B
11 S3B
DB 9
12 S2B
GND 15
A0
13
–VS
S1B A1
VSS 16
0Ω
0Ω
CD
–IN
CC
1
+
5
WR
A1 4
A0
AD8253 REF
–
+VS
–VS
9
3
VOUT
+IN
0Ω 49.9Ω
AD7612
1nF
C4
0.1µF
C3
0.1µF
3
7
ADR435
8
1
DGND
+12V –12V
JMP
0.1µF
–12V
+5V
2kΩ
DGND
JMP
+5V
R8
2kΩ
06983-067
+CH1
VDD
JMP
–12V
GND
14
806Ω
+
DGND
Figure 63. Schematic of ADG1209, AD8253, and AD7612 Used with the AD825x in a Total Data Acquisition System
Rev. B | Page 22 of 24
06983-062
AD8253
Data Sheet
AD8253
OUTLINE DIMENSIONS
3.10
3.00
2.90
3.10
3.00
2.90
10
5.15
4.90
4.65
6
1
5
PIN 1
IDENTIFIER
0.50 BSC
0.95
0.85
0.75
15° MAX
1.10 MAX
0.30
0.15
6°
0°
0.70
0.55
0.40
0.23
0.13
091709-A
0.15
0.05
COPLANARITY
0.10
COMPLIANT TO JEDEC STANDARDS MO-187-BA
Figure 64. 10-Lead Mini Small Outline Package [MSOP]
(RM-10)
Dimensions shown in millimeters
ORDERING GUIDE
Model1
AD8253ARMZ
AD8253ARMZ-RL
AD8253ARMZ-R7
AD8253-EVALZ
1
Temperature Range
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Package Description
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
Evaluation Board
Z = RoHS Compliant Part.
Rev. B | Page 23 of 24
Package Option
RM-10
RM-10
RM-10
Branding
Y0K
Y0K
Y0K
AD8253
Data Sheet
NOTES
©2008–2012 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06983-0-10/12(B)
Rev. B | Page 24 of 24