AN-851
APPLICATION NOTE
One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106, U.S.A. • Tel: 781.329.4700 • Fax: 781.461.3113 • www.analog.com
A WiMax Double Downconversion IF Sampling Receiver Design
by Eric Newman and Cecile Masse
INTRODUCTION
This application note describes an intermediate frequency (IF)
sampling receiver intended for use in the wireless communication services (WCS) band from 2.3 GHz to 2.36 GHz and the
unlicensed ISM band from 2.4 GHz to 2.48 GHz. The receiver
is designed for broadband Orthogonal Frequency Division
Multiplexing (OFDM) systems, as described according to IEEE
802.16 standards documentation. The design approach and
implementation procedures are presented to allow designers to
easily modify the receiver chain to address other bands such as
Wireless Broadband (WiBro) and other cellular standards such
as TDS-CDMA. The presented design is capable of addressing
channel bandwidths as wide as 10 MHz using the depicted
surface acoustic wave (SAW) filter; however, larger bandwidths
can be addressed using wider bandwidth channel-select filters
and increased sampling rates.
BACKGROUND
The 802.16 standard describes an adjacent channel rejection
requirement for the receiver. The following adjacent and
nonadjacent channel interferer to desired channel power ratios
must be achieved while maintaining a 10−6 bit error rate (BER)
while the desired signal is no more than 3 dB above the
specified reference sensitivity.
Table 1. Adjacent and Nonadjacent Channel Rejection
Requirements as Described in the 802.16-2004 Standard
Modulation/Coding
16-QAM-¾
64-QAM-¾
Adjacent Channel
Rejection (dB)
−11
−4
Nonadjacent
Channel
Rejection (dB)
−30
−23
ARCHITECTURE
OFDM is the modulation scheme used in the next emerging
WiMax standard. OFDM utilizes multiple subcarriers of various IQ
modulations to achieve a high aggregate data rate that has inherent
immunity to multipath propagation. The baseband information is
spread among subcarriers so that little information is lost if
multiple propagation paths result in destructive interference and
heavily attenuate a portion of the transmitted spectrum. The
variability in the subcarrier modulation schemes allows for an
adaptive signaling approach that propagates lower data rates at the
greatest distances, while the highest data rates can be utilized when
the received signal-to-noise ratio (SNR) is high.
The variety of subcarrier modulation schemes and coding
results in different SNR requirements at the receiver. The
reference sensitivity of a WiMax receiver is defined in the IEEE
Std. 802.16-2004 to be −91 dBm for a 1.5 MHz channel using ½
coding rate QPSK, and −65 dBm for a 20 MHz channel using ¾
coding rate 64-QAM. This results in a receiver NF requirement
of 7 dB with 5 dB of implementation margin.
The 802.16 standard defines a maximum input power level of
−30 dBm for successful detection, with a maximum tolerated
power level of 0 dBm. While the base station or subscriber
station is not expected to decode a 0 dBm input level successfully, the equipment must be able to withstand the large 0 dBm
input without damage.
Figure 1 depicts a classic double downconversion IF sampling
receiver. IF sampling architectures are quite appropriate when
dealing with large signal bandwidths as employed in WiMax, or
even multicarrier systems. By using multiple down conversions,
it is possible to employ several channel select filters that help to
improve the selectivity of the receiver and improve immunity to
blocking signals that would otherwise degrade receiver sensitivity. The double downconversion also allows the use of a high
enough first IF where the image frequency band falls out of the
passband of the front-end, band-select RF filter.
The receiver architecture described in this application note is
based on a 14-bit ADC. A 12-bit ADC could be used to address
the 802.16 receiver requirements, although it is recommended
to use a 14-bit ADC for single-down conversion or for multicarrier architectures in order to compensate for the less efficient
selectivity and avoid the saturation of the ADC in presence of
high interferer levels. In order to design a receiver capable of
successfully addressing the multiple data rate options available,
it is necessary to select IF center frequencies carefully and
ensure that appropriate SAW filter selections are available.
Rev. 0 | Page 1 of 8
AN-851
ADF4360-8
~100Ω
BAND
SELECT
ATF541M4
ATF541M4
IMAGE
REJECT
LNA1
AV = –1.5dB
AV = +14.5dB
NF = 0.9dB
IIP3 = 18dBm
SAWTEK
855898
LO1
LCR
LPF
LO2
LNA2
AV = –2.5dB
AD8370
~1kΩ
LCR
AAF
IF DGA
ADL5350
AV = –8dB
AV = –7.5dB
NF = 7.5dB
IIP3 = 24dBm
AV = +14.5dB
NF = 0.9dB
IIP3 = 18dBm
AD8344
AV = +10dB
AV = –1dB
AV = +11.5dB
A
=
–15dB
TO
+34dB
V
NF = 8dB
NF = 8dB (AT MAX GAIN)
IIP3 = 12dBm
OIP3 = 35dBm
AD9246 FULL SCALE =
–3dBV rms
ADC
en = 21nV/ Hz
IIP3 = 33dBV rms
AV = 0dB
NF = 20.5dB
IIP3 = 33dBm
06147-001
ADF4153
Figure 1. Double Downconversion IF Sampling Receiver Chain
Figure 2 shows the spurious trajectories for a downconverting
mixer using low-side LO injection. In order to cover the
2.3 GHz to 2.4 GHz bands while minimizing the number of
spurious mixer components that can interfere with the desired
signal, inspection of the spurious trajectories indicates that the
first IF should be within a range of 210 MHz to 400 MHz. The
grey shaded area indicates the fractional bandwidth bounded by
fRF/fLO and fIF/fLO for a 374 MHz IF using a low-side LO (LO1 =
1926 MHz to 2026 MHz). It is possible to substitute other IF
center frequencies, but caution must be taken to ensure that the
spurious responses of the first mixer do not cause in-channel
interference. For this design, a first IF of 374 MHz is selected.
There are several commercially available SAW filters of various
bandwidths from Sawtek, as well as other manufacturers, that
address this center frequency.
0.5
2L
-
-2R
3L
0.4
0.3
1.7
1.8
R
1.0
3R
4L-
0.2
0.1
0
1.0
Gain (dB)
65.5
58.5
48.5
37
23.5
1.1
1.2
1.3
1.4
1.5
fRF
fLO
1.6
1.9
2.0
06147-002
fIF
fLO
0.6
R
-2
4L
3R
5L-
0.7
After establishing the target IF frequencies it is possible to
consider the cascaded performance. Table 2 depicts the
anticipated dynamic performance of the double downconversion receiver from the output of the band-select filter down to
the ADC interface.
Table 2. Anticipated Dynamic Performance of Receiver
Signal Chain depicted in Figure 1
L
R-
2R
3R
-5L 3L
0.8
3R
-4L
0.9
2R
-2L
4R-4
L
3R3L
1.0
Next it is necessary to consider the final IF. The second down
converting mixer has the benefit of a narrower input bandwidth, typically no more than 20 MHz. This allows several
possible second IFs to be considered, with ranges from 10 MHz
to 70 MHz and higher IF frequency bands centered at 107 MHz
and 140 MHz. A 70 MHz IF is selected in order to avoid higher
order LO harmonics that can leak back to the antenna receive
port and fall in the desired receive band. Additionally, the
70 MHz IF allows for a wide selection of commonplace SAW
filters to be applied, or alternatively, practically lumped element
LC passive filters.
Figure 2. Spurious Trajectories for a Down Converting Mixer Using Low-Side
LO Injection (Note that the grey box indicates the fractional bandwidth
boundaries for a 374 MHz IF using low-side LO injection for the 2.3 GHz to
2.4 GHz WiBro frequency band.)
.
Rev. 0 | Page 2 of 8
IIP3 (dBm)
−32.8
−26
−14.5
−7.7
−5.6
NF (dB)
2.6
2.7
3.5
4.6
9.7
AN-851
The first LNA stage follows the front-end, band-select filter. The
insertion loss of the filter and the noise figure of the first stage
dominate the overall cascaded sensitivity of the receiver. As a
result, it is critical to achieve very low noise figure in the first
LNA stage. The ATF541M4 GaAs E-mode pHEMT was selected
from Agilent Technologies for its low noise and high OIP3. The
circuit implementation for the first and second stage LNAs is
presented in Figure 3, along with the measured gain and NF
performance in Figure 4.
5V
10nF
1nF
294Ω
50Ω
1.21kΩ
8.2pF
32.4Ω
5
20
4
15
3
10
2
5
1
0
2100
2200
2300
2400
2500
0
2600
FREQUENCY (MHz)
NF (dB)
25
06147-004
The first component in the receiver chain is a band-select filter.
Several manufacturers, such as Anatech Electronics, Inc., K & L
Microwave Inc., and Digital Communications Inc., provide
cavity- and ceramic-based filters that address the WCS and
unlicensed ISM frequency bands utilized for 2.3 GHz to
2.5 GHz WiMax deployments. The cavity-based filters are
capable of low insertion loss of less than 2 dB with up to 60 dB
of stop-band rejection at only 25 MHz offset from the center of
the pass band. The lower cost ceramic-based filters offer closer
to ~50 dB of stop-band rejection at 150 MHz offsets. The choice
of band select filter strongly depends on the image rejection
performance of the receiver and the expected magnitude of the
interfering signals present in the vicinity of the band of
operation. For this demonstration, a Digital Communications
Inc. 10-section band-pass cavity filter is selected at a center
frequency of 2350 MHz.
GAIN (dB)
IMPLEMENTATION
Figure 4. ATF541M4 Gain and NF, Input Return Loss is Measured to be −8 dB
with an IIP3 of 18 dBm
In order to help improve the image rejection of the front-end
design and to minimize the presented broadband noise to the
first mixer, a simple high-pass filter was used between the first
and second LNA. The rejection requirement is not very critical
considering the first band-select filter provides >60 dB rejection
of unwanted signals appearing at the image frequency. Using a
374 MHz IF with low-side LO injection, the fundamental image
band is from 1552 MHz to 1652 MHz for a 2.3 GHz to 2.4 GHz
input frequency range. A simple 3-pole lumped element filter
was designed between the first and second LNA stages. The
filter provides better than 20 dB of image band rejection and
less than 2.5 dB of insertion loss in the desired pass band. The
measured frequency response and circuit implementation is
depicted in Figure 5.
0
8.2pF
–5
5.6nH
–10
1.5nH
8.2pF
–15
Figure 3. Basic LNA Implementation Schematic
S21 (dB)
3.3pF
06147-003
0.6pF
–20
–25
–30
4.3nH
–35
–40
2.2nH
2.2pF
2.2nH
–45
–50
0
500
1000 1500 2000 2500 3000 3500 4000 4500 5000
FREQUENCY (MHz)
06147-005
1.5nH
Figure 5. Simple Lumped Element Image Reject Filter Implementation and
Measured Response
Rev. 0 | Page 3 of 8
AN-851
12
30
IIP3
9
6
IIP3, IP1dB (dBm)
20
The additional AD8353 gain stage is needed to amplify the
output power of the PLL/VCO synthesizer circuit to ~4 dBm
drive level at the LOIN pin. There is additional filtering applied
to the LO input to help reduce harmonic content coming from
the LO synthesizer. The filter is a simple 3-pole, low-pass filter
constructed using lumped element components. Without the
filter the second harmonic of the LO would cause some additional high frequency noise power to be downconverted to the
desired IF, degrading the reference sensitivity of the receiver.
The measured dynamic performance of the ADL5350, including the external LO buffer and filter networks, is presented in
Figure 6. The mixer provides better than 24 dBm input IIP3
with ~8 dB conversion loss and single-sideband NF.
3
15
IP1dB
10
0
–3
5
GAIN
0
–6
CONVERSION GAIN (dB)
25
–9
–5
–10
2200
2250
2300
2350
2400
2450
–12
2500
06147-007
After the second LNA stage, the signal is downconverted to a
fixed high IF of 374 MHz using the ADL5350 mixer. The
ADL5350 is a single-ended passive mixer with an onboard LO
buffer amplifier. The mixer relies on off-chip filtering networks to
isolate the RF and IF ports. The subcircuit depicted in Figure 7
includes the ADL5350 mixer, external filter networks, and an
external LO buffer to provide sufficient drive into the LOIN pin.
RF FREQUENCY (MHz)
Figure 6. Measured Performance of ADL5350 Mixer
During the downconversion, the phase noise of the PLL is
imposed onto each subcarrier of the OFDM modulated signal
through convolution. To minimize the impact on the receiver
sensitivity level, the first agile PLL for the generation of LO1 is a
fractional-N PLL. It is designed with the ADF4153 synthesizer.
The closed loop bandwidth is about 30 kHz and the estimated
phase jitter is 0.3 degrees.
ADL5350 EVALUATION BOARD
1
RFIF
2
GND2
3
LOIN
4
GND1
GND1
5
3nH
100pF
2.2pF
2.7nH
3V
100pF
2.2pF
1nF
1nF
RFIF
8
GC
7
VPOS
6
18pF
6pF
9.5nH
15nH
374MHz
IF OUT
0.7pF
3V
2.1nH
100pF
4.7µF
3V
VP
VDD
0.47µF
100nF
1nF
3dB PAD
18Ω
100pF
AD8353
EVALUATION BOARD
VCO190-1960T
18Ω
1.5kΩ
100pF
CP
600Ω
REFIN
ADF4153
MUXOUT
18Ω
100pF
820pF 40nF
2.7nF
51Ω
RFINA
CLK
DATA
LE
RFINB
RSET
10MHz
XTAL
REF
SPI
CONTROL
BUS
100pF
ADF4153 EVALUATION BOARD
Figure 7. First Mix-Down Stage
Rev. 0 | Page 4 of 8
06147-006
RF INPUT
FROM LNA2
SAWTEK
855898
1.5nH
ADL5350
1pF
0.67nH
AN-851
The first IF is then passed through a 374 MHz SAW filter for
channel selection. The excellent stop-band rejection of the
Sawtek 855898 SAW filter allows the receiver to have
exceptional selectivity, improving immunity to adjacent
interfering signals. The filter is matched to 50 Ω using the
external LC component indicated in Figure 7. The filters
frequency response characteristic is presented in Figure 8.
The first filtered IF signal is then downconverted to the second
IF of 70 MHz using the AD8344 active mixer. A low-side LO
injection scheme was used to ensure optimal spurious response
rejection and to achieve greater gain in the mixer. In general,
the AD8344 offers slightly higher conversion gain when using a
low-side LO. The 304 MHz LO is provided by the ADF4360-8.
This is an integrated PLL + VCO providing good cost and
board space savings over discrete solutions. Two external
inductors that are the tank inductors of the on-chip VCO set
the center frequency. The differential outputs of the ADF4360-8
are combined in a balun. Using a 304 MHz LO the AD8344
offers a conversion gain of ~11 dB, with 12 dBm IIP3 and 8 dB
SSB NF. The 70 MHz output is then passed through a fourth
order low-pass filter to help reject LO feedthrough and higher
frequency mixer spurii.
0
–10
–20
–40
–50
–60
–70
–80
–90
–100
300
325
350
375
400
425
450
FREQUENCY (MHz)
06147-008
Figure 8. Sawtek 855898 Frequency Response
AD8344
EVALUATION BOARD
1nF
5V
2.43kΩ
1nF
5V
12
11
10
9
VPDC PWDN EXRB COMM
13
COMM
14
RFCM
COMM
1nF
8
100pF
374MHz
INPUT
15
AD8344
RFIN
IFOP 7
4:1
1nF
220nH
39pF
240nH
70MHz
IF OUT
18pF
IFOM 6
100pF
5V
16
1nF
VPMX
COMM
5
VPLO LOCM LOIN COMM
1
1nF
2
3
1nF
VDD
ADF4360-8
EVALUATION BOARD
4
1nF
ADF4360-8
4.3kΩ
VTUNE
REFIN
CP
CLK
DATA
LE
1.5kΩ
150pF
680nF
390pF
SPI
CONTROL
BUS
43.8nH
VDD
L1
240nH
1:4
10MHz
XTAL
REF
L2
RFOUTA
470Ω
RSET
100pF
RFOUTB
4.7kΩ
06147-009
RESPONSE (dB)
–30
Figure 9. The AD8344 Active Mixer Driven by the ADF4360-8 Through an External 1:4 Impedance Ratio Balun
Rev. 0 | Page 5 of 8
AN-851
The final 70 MHz IF signal is then passed through the AD8370
digitally controlled variable gain amplifier before being IF
sampled by the AD9246. The AD8370 provides a high output
IIP3 and greater than 40 dB gain adjustment range. This allows
the overall receiver conversion gain to adjust and tailor the
cascaded input dynamic range to accommodate widely varying
input signal powers.
to raise the presented loading impedance at the desired IF
frequency. The filter network provides an impedance
transformation from 100 Ω to 600 Ω and has the effect of
stepping up the voltage by ~8 dB. The step-up transformation
needs to be accounted for when analyzing the receiver line-up.
For more information regarding the design approach used to
yield the driver/ADC interface, please refer to AN-827, A
Resonant Approach to Interfacing Amplifiers to SwitchedCapacitor ADCs. The key details of the AD8370 and AD9246
network interface are captured in Figure 11. The simulated filter
response is plotted against the measured response through the
ADC in Figure 10.
The selected ADC provides excellent spurious free dynamic
range out to greater than 200 MHz IF frequencies while only
consuming ~250 mW. The high analog input bandwidth of
650 MHz allows the AD9246 to be applied to higher IF
frequencies. In this demonstration a 70 MHz IF was selected
using an 80 MSPS sampling clock, placing the IF in the second
Nyquist zone of the ADC.
0
–5
–10
In order to prevent degradation of the ADC’s sensitivity level it
was necessary to employ an anti-aliasing filter. The anti-aliasing
filter helps to reject higher frequency spurious signals such as
LO leakage from degrading the perceived noise floor of the
ADC. Additionally the anti-aliasing filter helps to reject
wideband noise generated by the driving amplifier stages that
would otherwise alias additional noise into the desired Nyquist
band. Elliptical low-pass architecture is selected to help provide
better rejection of strong spurious components in the higher
Nyquist zones. A resonant parallel tank network is formed by
the 72 nH bias inductors in combination with the total
presented input capacitance. The resonant tank network helps
RESPONSE (dB)
MEASURED
–15
–20
SIMULATED
–25
–30
–35
–45
20
40
60
80
100
120
140
160
FREQUENCY (MHz)
Figure 10. Simulated and Measured Response of the Anti-Aliasing Filter into
the AD9246 ADC
AD9246
VCOM
39pF
SERIAL CONTROL INTERFACE
1nF
1nF
16
15
14
13
12
ICOM
DATA
CLCK
LTCH
11
10
47nH
56nH
9
VCCO OCOM OPLO
72nH
1:4
AD8370
15pF
22pF
18pF
AD9246-80
72nH
INHI
ICOM
VCCI
1
2
3
PWUP VOCM VCCO OCOM
4
5
6
OPHI
7
8
1nF
1nF
1nF
0.1µF
47nH
56nH
0.1µF
39pF
5V
Figure 11. Circuit Interface Between the VGA and ADC
Rev. 0 | Page 6 of 8
06147-010
70MHz
INPUT
1nF
INLO
06147-011
–40
AN-851
SUMMARY OF COMPLETE RECEIVER PERFORMANCE
While attempting to verify the overall cascaded performance of
the full receiver, it is necessary to capture the sampled data from
the AD9246 ADC using the Analog Devices high speed ADC
FIFO USB evaluation kit. The additional first-in-first-out
(FIFO) daughtercard serves as a data buffer to allow capture of
long data output strings generated by the ADC at the sampling
rate. The FIFO can then transmit the captured data to the PC at
a lower data rate that can be supported via a standard USB
interface. The ADC evaluation hardware is controlled by
Analog Devices ADC Analyzer™ software. The ADC Analyzer
allows for time domain and frequency domain analysis. The
single-tone and two-tone distortion performance is readily
captured using ADC Analyzer, and the results are presented in
Figure 12 and Figure 13. Note the discontinuity in Figure 13
around 48 dB of signal gain. This is where the AD8370 VGA
transitions from a low gain mode to a high gain mode. The
discontinuity does not result in any notable degradation of the
BER performance of the full receiver.
Suitable BER test equipment for an 802.16 OFDM waveform
was not available at the time of the evaluation. However, error
vector magnitude (EVM) analysis allows for a fair estimate of
the receiver performance and available dynamic range. In order
to perform EVM analysis on OFDM 802.16 test signals it was
necessary to use a FIFO card with an extended memory depth
of at least 65 kb. This allows capture of a complete RF burst.
The captured integer vectors range from 0 to 65,536 (216) with a
midscale value of 32,768. Note that the AD9246 is a 14-bit
ADC, and that the data output words are 16-bit representation.
The captured integer vectors were post-processed in the
Advanced Design System (ADS) 2006A from Agilent. The ADS
program divides the 16-bit represented IF vector by four to
yield a 214 integer representation. The vector is then shifted and
scaled in magnitude to yield a zero-mean value waveform with
±1 V peak magnitude. The signal is then demodulated using an
ideal IQ demodulator. The IQ vectors are then decimated in
time to yield baseband vectors at ¼ the original sampling rate.
–50
The processed IQ vectors are then fed into the 89600 Vector
Signal Analyzer software from Agilent to reveal the EVM
performance. The measured EVM performance vs. input power
is depicted in Figure 14 for a 64-QAM OFDM signal with a
10 MHz modulation bandwidth and ¾ coding rate. The EVM is
plotted with and without the ADF4153 and ADF4360 PLLs to
provide a comparison of the performance degradation due to
phase noise.
–60
–65
–70
HD2
–75
HD3
–80
0
–85
–5
–90
30
35
40
45
50
55
60
65
SIGNAL GAIN (dB)
06147-012
–10
–15
0
7
–5
6
EVM (dB)
Figure 12. Second and Third Harmonic Distortion for the Full Receiver
Cascade Measured at −1 dBFS Input Levels at the AD9246 Input
–20
–25
–30
–35
WITH ADF4153 AND
ADF4360-8 PLLs
5
–15
4
–20
3
–25
2
–30
1
–35
30
35
40
45
50
55
60
65
–45
SSB NF (dB)
–10
0
70
SIGNAL GAIN (dB)
–50
–80
USING EXTERNAL GENERATORS FOR LO SIGNALS
–70
–60
–50
–40
INPUT POWER (dBm)
–30
–20
–10
Figure 14. EVM Performance Measured Through the Full Receiver for 64-QAM
¾ Rate Coding, 10 MHz OFDM Signaling
06147-013
IIP3 (dBm)
–40
06147-014
HARMONIC DISTORTION (dBc)
–55
Figure 13. Two-Tone IIP3 and Single-Sideband NF for the Full Receiver
Cascade Measured vs. Receiver Conversion Gain
Rev. 0 | Page 7 of 8
AN-851
In order to access the selectivity of the receiver, it is necessary to
measure the relative response for single-tone inputs swept over
a reasonable input frequency range. Figure 15 presents the
selectivity of the full receiver in the absence of the front-end
band-select filter. The frequency selective nature of the double
downconversion receiver provides very high immunity to
nearby interferers. With the addition of the front-end bandselect filter, greater than 60 dB rejection can be achieved at
adjacent RF frequency bands.
10
NORMALIZED RESPONSE (dB)
0
–10
A complete summary of the receiver’s performance is provided
in Table 3. The design provides better than −25 dB EVM performance for input signals greater than −74 dBm, with more
than 60 dB image rejection and excellent adjacent and nonadjacent channel rejection.
–20
–30
–40
Table 3. Performance Summary (measured with 64-QAM ¾
rate coding OFDM 10 MHz bandwidth input signal)
–50
–70
2280
2300
2320
2340
2360
2380
2400
2420
2440
RF INPUT FREQUENCY (MHz)
06147-015
–60
Figure 15. Receiver Selectivity vs. Input Frequency, Local Oscillators Tuned for
2350 MHz Input Signal (fLO1 = 1976 MHz, fLO2 = 304 MHz)
–20
SPECTRAL MAGNITUDE (dBm)
–30
–40
–50
–60
Parameter
Gain Range
EVM from −74 dBm to −20 dBm Input Power
IIP3 at Mid-Gain (Av = 48 dB)
NF at Mid-Gain (Av = 48 dB)
Image Rejection
Adjacent Channel Rejection
Power Dissipation (from LNA to ADC,
including PLLs)
Measured
Performance
30 dB to 67 dB
<−25 dB
>−8 dBm
<4 dB
>60 dB
>58 dB
~2.2 Watts
REFERENCES
–70
Newman, Eric and Reeder, Rob. 2006. “A Resonant Approach to
Interfacing Amplifiers to Switched-Capacitor ADCs.”
Application Note AN-827. Analog Devices, Inc. (January)
–80
–90
–100
–110
IEEE Std 802.16-2004. IEEE Standard for Local and
Metropolitan Area Networks - Part 16: Air Interface for Fixed
Broadband Wireless Access Systems. Institute of Electrical and
Electronics Engineers, Inc. (June).
0
5
10
15
20
25
30
35
FREQUENCY (MHz)
40
06147-016
–120
–130
The typical spectral characteristics of the full receiver is
depicted in Figure 16. By using the ADC Analyzer software, it
is easy to see the instantaneous spectrum and capture time
domain data for later signal processing. Whereas the IF is at
70 MHz with an 80 MSPS sampling clock, the signal appears as
a 10 MHz signal in the first Nyquist zone. The ADC Analyzer
software makes it possible to quickly analyze and debug
spurious clock and LO signal components that can otherwise
compromise in-band receiver performance. The multiple IF
filter stages and LO filters helped to minimize spurious clutter
to negligible levels. Note the weak spur at 32 MHz. This is the
LO to IF leakage of the first mixer.
Figure 16. FFT Plot of ADC Spectrum at Mid-Gain for a 64-QAM ¾ Rate
Coded, 10 MHz Bandwidth OFDM Input at 2.342 GHz (Note that the signal
appears as a 10 MHz IF but is actually a 70 MHz input in the second Nyquist
zone of the converter.)
©2006 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
AN06147-0-6/06(0)
Rev. 0 | Page 8 of 8