DC to DC Power Conversion Module for the All-Electric Ship
by
Weston L. Gray
B.S., Electrical Engineering
University of Akron, 1999
Submitted to the Department of Mechanical Engineering and the Department of Electrical
Engineering and Computer Science in Partial Fulfillment of the Requirements for the Degrees of
Naval Engineer
and
Master of Science in Electrical Engineering and Computer Science
at the
ARCHIVES
SSACHUSETTS INSTITUTE
OF TECHNOLOGY
Massachusetts Institute of Technology
June 2011
JUL 2 9 2011
© 2011 Weston L. Gray
All rights reserved
LIE
The author hereby grants to MIT permission to reproduce and to distribute publicly paper and electronic
copies of this thesis document in whole or in part in any medium now known or hereafter created.
Signature of Author...........................................................
Center
Certified
By ....
.
..
Weston L. Gray
r Ocean Engineyring, Department of Mechanical Engineering
. .r.
....................................
..........
Chryssostomidis
oChrys
P e
Professor of Mechanical Engineering and Ocean Engineering
L. Kirtley Jr.
CertifiedJames
Professor of Electrical Engingr ng and Com
A ccepted By .............................................
t
Science
. ....
av
al
Chairman, Committee on Graduate Students
Department of Mechanical Engineering
Accepted By...
............................
/
C/
Leslie A. Kolodziejski
Chairman, Committee on Graduate Students
Department of Electrical Engineering and Computer Science
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Page 2
DC to DC Power Conversion Module for the All-Electric Ship
by
Weston L. Gray
B.S., Electrical Engineering
University of Akron, 1999
Submitted to the Department of Mechanical Engineering and the Department of
Electrical Engineering and Computer Science in Partial Fulfillment of the Requirements for
the Degrees of
Engineer in Naval Architecture and Marine Engineering
and
Master of Science in Electrical Engineering and Computer Science
ABSTRACT
The MIT end to end electric ship model is being developed to study competing
electric ship designs. This project produced a model of a Power Conversion Module (PCM)4, DC-to-DC converter which interfaces with the MIT model. The focus was on the Medium
Voltage DC (MVDC) architecture, and therefore, the PCM-4 converts a MVDC bus voltage of
3.3, 6.5 or 10 kVDC to 1 kVDC. The design describes the transient and steady-state
behavior, and investigates the naval architecture characteristics.
A modular architecture, similar to SatCon Applied Technology's Modular
Expandable Power Converters, was selected as the best balance for the wide variation in
loads experienced. The model consists of a standard module that can be paralleled
internally to provide for a wide range of system power requirements. Naval architecture
parameters, such as weight, volume, efficiency, and heat load, were compiled into a
parametric format allowing a reasonable approximation of actual weight and volume as a
function of rating and efficiency and heat load as a function of loading. All of the
parameters were evaluated for dependence on the MVDC bus voltage.
Verification of the model was pursued through comparison to available simulations
of similar power electronics to ensure that the model provided reasonable time response
and shape. Finally, the model met all requirements with the exception of efficiency which
was slightly lower than the requirement although several ideas were presented to improve
efficiency.
Page 3
TABLE OF CONTENTS
ABST RACT ........................................................................................................................................
3
TAB LE OF CO N TEN T S .................................................................................................................
4
LIST OF TA BLES .............................................................................................................................
6
...........................................................................................................................
LIST OF FIG URESS
7
LIST OF A BBR EV IA TIO NS ......................................................................................................
9
Chapter 1 Introduction ......................................................................................................................
10
1.1 overview ......................................................................................................................................
10
1.2 background ................................................................................................................................
11
1.3 Project Goals..............................................................................................................................
12
Chapter 2 Model Specification ....................................................................................................
15
2.1 System Specifications.............................................................................................................
15
2.2 Basic m odel structure .........................................................................................................
16
2.3 Pow er Conversion Module Control..............................................................................
18
Chapter 3 Model Design ....................................................................................................................
20
3.1 Base Ship Service Electrical Distribution System Layout ...............
20
3.2 PCM -4 functional block Diagram ...................................................................................
22
3.3 Pow er Converter......................................................................................................................
23
3.3.1 Converter Module Size ..............................................................................................
24
3.3.2 Pow er Sw itch ....................................................................................................................
25
3.3.3 Converter Layout..........................................................................................................
37
3.3.4 Converter Waveform s ..............................................................................................
38
3.3.5 Converter Efficiency ....................................................................................................
44
3.3.6 Filters ...................................................................................................................................
53
3.3.7 Capacitor Selection ......................................................................................................
61
Page 4
3.3.8 W ire Selection...................................................................................................................
62
3.3.9 Inductor D esign................................................................................................................
63
3.3.10 High Frequency Transform er ..............................................................................
66
3.3.11 Converter Sum m ary .................................................................................................
72
3.4 Load Sharing..............................................................................................................................72
3.4.1 Droop ...................................................................................................................................
73
3.4.2 V oltage Regulator.......................................................................................................
74
3.4.3 Load Sharing Verification.........................................................................................
76
3.5 Power Control Module.......................................................................................................
79
3.5.1 power Control Module shipwide control and sensing..................................
79
3.5.2 PCO N m odule PCM -4 Control................................................................................
80
3.6 PCO N Module.............................................................................................................................
80
3.7 Model Verification ...................................................................................................................
82
Chapter 4 Naval Architecture Param etric Extraction.........................................................
83
4.1 Efficiency and Heat Load .................................................................................................
83
4.2 Weight..........................................................................................................................................
85
4.3 Volum e .........................................................................................................................................
86
4.4 Naval A rchitecture Sum m ary.........................................................................................
88
Chapter 5 Conclusions .......................................................................................................................
89
5.1 Recom m endations for future w ork ..............................................................................
89
BIBLIO GRA PHY ...........................................................................................................................
91
Appendix A: Model Detail.........................................................................................
Page 5
A-1
LIST OF TABLES
TABLE 1 EFFICIENCY AND POW ER QUALITY REQUIREMENTS [4] ....................................................................
....... 15
TABLE 2 POWER CONVERTER VARIANTS ................................................................................................................................
17
TABLE 3 SWITCH PROPERTIES ..............................................................................................................................................
37
TABLE 4 TRANSFORMER INPUT W AVEFORM ..........................................................................................................................
39
TABLES CONVERTER SWITCHING MODES ...............................................................................................................................
40
TABLE 6 UPDATED SWITCHING SEQUENCE.............................................................................................................................42
TABLE 7 EFFICIENCY FOR 3.3 KV CONVERTER .........................................................................................................................
45
TABLE 8 EFFICIENCY FOR 6.6 KV CONVERTER .........................................................................................................................
46
TABLE 9 EFFICIENCY FOR 10KV CONVERTER ...........................................................................................................................
46
TABLE 10 SWITCH SUMMARIES ...........................................................................................................................................
51
TABLE 11 3.3KV POW ER CONVERTER EFFICIENCY...................................................................................................................52
TABLE 12 6.5 KV POW ER CONVERTER EFFICIENCY ..................................................................................................................
52
TABLE 13 10KV POW ER CONVERTER EFFICIENCY....................................................................................................................53
TABLE 14 CAPACITOR PROPERTIES.......................................................................................................................................
61
TABLE 15 W IRE SIZE FOR PRIMARY AND SECONDARY...............................................................................................................63
TABLE 16 INDUCTOR EQUATION SYMBOLS .............................................................................................................................
64
TABLE 17 INDUCTOR DESIGN SUMMARY...............................................................................................................................66
TABLE 18 TURNS RATIOS ....................................................................................................................................................
71
TABLE 19 EFFICIENCY AND HEAT LOAD FOR A200 KW PCM AT VARIOUS LOADS AND VOLTAGES......................................................84
TABLE 20 W EIGHT............................................................................................................................................................86
TABLE 21 CONVERTER VOLUME HOUSED IN ENCLOSURE.......................................................................................................
Page 6
87
LIST OF FIGURES
FIGURE 1 FUTURE (POTENTIAL) IFTP IN-ZONE ARCHITECTURE [1].............................
................
...................................... 11
FIGURE 2 SATCON APPLIED TECHNOLOGY DISTRIBUTED POWER SYSTEMS [3] ..........................................................................
13
FIGURE 3 BASIC FORWARD CONVERTE RFROM [11] ................................................................................................................
18
FIGURE 4 BASELINE ELECTRICAL DISTRIBUTION SYSTEM............................................................................................................21
FIGURE 5 BASELINE ZONAL ELECTRICAL DISTRIBUTION SYSTEM................................................................................................21
FIGURE 6 PCM -4 M ODEL BLOCK DIAGRAM ..........................................................................................................................
22
FIGURE 7 SIMULINK MODEL OF CONVERTER M ODULE..............................................................................................................23
FIGURE 8 3.3KV IGBT FORWARD VOLTAGE DROP [12]............................................................................................................27
FIGURE 9 3.3KV IGBT TURN ON LOSS [12]...........................................................................................................................28
FIGURE 10 3.3KV IGBT TURN OFF LOSS [12 ........................................................................................................................
29
FIGURE 11 3.3KV IGBT REVERSE RECOVERY LOSS [12] ..........................................................................................................
30
FIGURE 12 3.3KV IG BT SW ITCHING TIMES [12] ....................................................................................................................
31
FIGURE 13 6.5KV IGBT FORWARD VOLTAGE DROP [13]..........................................................................................................32
FIGURE 14 6.5KV IGBT TURN-ON LOSS [13].........................................................................................................................33
FIGURE 15 6.5KV IGBT TURN-OFF LOSS [13]........................................................................................................................34
FIGURE 16 6.5KV IGBT REVERSE RECOVERY LOSS [13]............................................................................................................35
FIGURE 17 6.5KV iGBT SW ITCHING TIMES [13] ....................................................................................................................
36
FIGURE 18 SECOND ITERATION OF CONVERTER LAYOUT............................................................................................................37
FIGURE 19 BASE TRANSFORMER INPUT W AVEFORM ...............................................................................................................
FIGURE 20 VOLTAGE CONTROL CIRCUIT WITH ZVS DELAY FACTOR (F)
...........................................
39
41
FIGURE 21 SECOND ITERATION OF VOLTAGE CONTROLLER........................................................................................................43
FIGURE 22 FINAL VOLTAGE CONTROL CIRCUIT .......................................................................................................................
44
FIGURE 23 3300V SW ITCH FORWARD VOLTAGE DROP [17] .....................................................................................................
48
FIGURE 24 3300V SW ITCHING ENERGY [17] .........................................................................................................................
49
FIGURE 25 6500V FORWARD VOLTAGE DROP [16].................................................................................................................49
FIGURE 26 6500V SWITCH TURN-ON LOSS [16] ....................................................................................................................
50
FIGURE 27 POW EREx 6500V TURN-OFF LOSS [16].................................................................................................................50
FIGURE 28 POW EREx 6500V REVERSE RECOVERY [16] ...........................................................................................................
51
FIGURE 29 INITIAL TEST OF OUTPUT FILTER ............................................................................................................................
55
FIGURE 30 SHUNT CIRCUIT .................................................................................................................................................
57
FIGURE 31 SHUNT CONTROL CIRCUIT ...................................................................................................................................
58
FIGURE 32 FULL LOAD STEADY STATE RIPPLE..........................................................................................................................59
FIGURE 33 TRANSIENT RESPONSE 33% TO 100% AT 0.04 SECONDS ..........................................................................................
60
FIGURE 34 TRANSIENT RESPONSE 0% TO 50% AT 0.04 SECONDS ............................................................................................
60
Page 7
FIGURE 35 LIFE EXPECTANCY M ULTIPLIER FOR GENERAL ATOMICS TYPE C CAPACITOR [20]........................................................
62
FIGURE 36 SANDW ICH W INDINGS SHOW ING PRIMARY AND SECONDARY [6] ..............................................................................
67
FIGURE 37 SIMULATION W ITH 3000V SECONDARY VOLTAGE .................................................................................................
68
FIGURE 38 SIMULATION W ITH 2000V SECONDARY VOLTAGE .................................................................................................
69
FIGURE 39 SIMULATION W ITH 1500V SECONDARY VOLTAGE .................................................................................................
69
FIGURE 40 SIMULATION W ITH 4000V ON THE SECONDARY SIDE .............................................................................................
70
FIGURE 41 GENERATORS IN PARALLEL OPERATION...................................................................................................................73
FIGURE 42 VOLTAGE REGULATOR ........................................................................................................................................
75
FIGURE 43 REGULATOR FUNCTIONAL TEST.............................................................................................................................76
FIGURE 44 PARALLEL MODULE OPERATIONAL BLOCK DIAGRAM ...............................................................................................
77
FIGURE 45 PARALLEL OPERATION AT HALF LOAD .....................................................................................................................
78
FIGURE 46 PARALLEL OPERATION AT FULL LOAD......................................................................................................................78
FIGUR E 47 BASIC PCON MODULE .......................................................................................................................................
Page 8
81
LIST OF ABBREVIATIONS
BJT
Bipolar power Junction Transistors
D
Duty Ratio
GTO
Gate Turn-Off Thyristors
HFAC
High Frequency AC
IGBT
Insulated Gate Bipolar Transistors
IPS
Integrated Propulsion System
MOSFET
Metal Oxide Semiconductor Field Effect Transistors
MVDC
Medium Voltage DC
NGIPS
Next Generation Integrated Power System
PCM
Power Conversion Module
PGM
Power Generation Module
PCON
Power Control Module
SSCM
Ship Service Converter Module
SSIM
Ship Service Inverter Modules
ZCS
Zero Current Switching
ZVS
Zero Voltage Switching
Page 9
CHAPTER 1 INTRODUCTION
1.1 OVERVIEW
The early stage ship design requires analysis and comparison of many different ship
variants. The ships that are designed and then built based on these early studies are a huge
capital investment for the country. Many ship designs have a life of 50 or more years by the
time the initial design, the construction, and then the service life of the last constructed
ship is taken into account. All of these factors add up to the conclusion, the right decisions
need to be made early in the design of a ship.
The early stage designer is usually working his designs to meet a set of
requirements which are outside his control. Additionally, they have to produce affordable
designs that are the convergence of maximum durability, mission capability and
survivability all at a minimum cost. Ship designs are the definition of a systems
engineering problem with many designs rivaling the complex integrations of the world's
most complicated systems.
Much of the early stage design is completed by wise individuals with multiple
decades of experience. They have developed artful techniques and gut feelings that
produce very good results. The purpose of the "MIT End-to-End All-Electric Ship" model is
to supplement these talented individuals by providing a tool which can show higher level
simulations of a ship system with a minimum amount of set-up time.
The design tool was set up to allow these early-stage designers to make good
choices with better information. The end goal was to provide system wide modeling of a
new ship design.
This project designed a model of one of modules in the ship's electrical distribution
system. This module, a Power Conversion Module (PCM)-4, converts the main bus voltage,
3kV to 10kV, to 1kV DC to supply loads and other power conversions modules throughout
the ship. The model provides transient responses for evaluation. The Naval architect also
needs data on the efficiency, heat load, weight and volume of the component. The physicsPage 10
based model was used to develop a set of parametric relationships to answer these
questions.
1.2 BACKGROUND
The Navy has produced a roadmap of its view of technology development in the area
of electric ships utilizing an integrated power system. Published in the Navy's Next
Generation Integrated Power System (NGIPS) roadmap [1], the roadmap was used
extensively to produce the high-level characteristics of the PCM-4. The roadmap also
projects a more compact integrated propulsion system (IPS) layout. This next IPS layout
was utilized in this model, and the operation and major components are outlined below.
Figure 1 shows a notional electrical distribution system architecture.
load
oad
load
Emerge ncy -oad
xia CBT
V
C1
a-
MVAC
HFAC
->
>
1
1000
VDC
MVAC
Cod
od
FAC
dH
HVDC
or
0
gL loa
HVDC
load
Emerency _oad
-
J0dJVi
-or
1000
d3iJ3:-1 lydidt
VDC
via PCM-4
via PCM-4
n interupib
an
in
uple
P CM-2A
Un inter-utble
U -nem ll
load
load
lod
load
Variable eed
aiabl
*Otag
Spenial Freq uerty
Load
Figure 1 Future (potential) IFTP in-zone architecture [1]
The PCM-4 is used to convert the main generation voltage to 1000 VDC which is
supplied to the PCM-1A. In this specific application, the PCM-4 will also contain the control
center which will coordinate the power available and the amount of loading that is allowed
to run downstream. The NGIPS refers to this unit as the Power Control Module (PCON).
Page 11
"The PCON module consists of the software necessary to coordinate the behavior of the
other modules [2]."
The PCM-1A is made up of a variety of Ship Service Converter Modules (SSCMs) and
Ship Service Inverter Modules (SSIMs) which supply power to the individual DC and AC
loads. These loads can be powered from more than one PCM for reliability via
auctioneering diodes or a bus transfer switch. The individual modules can be paralleled
together to supply larger loads. Additionally, the modules would ideally be available in a
number of different power ratings to allow for the maximum efficiency and minimum cost
to power any additional load added to the ship.
The PCM-2A is a smaller Converter that could even power and be located with a
single load. The PCM-2A would replace large motor drives and be able to output variable
frequency AC as well as supply smaller uninterruptable loads with redundant power. The
redundancy would be achieved by powering the PCM-2A from both buses in a given zone.
However, the PCM-2A would not be a redundant component; although, redundant modules
could be included in a PCM-2A for the most critical loads.
In general, this next step in IPS architecture would minimize cabling by only
requiring the main distribution system lines to cross zones to power each zone's PCM-4.
For certain loads, it would also reduce the number of power stages required to supply
power to a load, specifically DC loads. Finally, this architecture allows the reduction of the
number of motor drives throughout the ship by supplying a common method to drive large
motors which could easily be integrated into the electrical distribution system's control
network.
1.3 PROJECT GOALS
There are nearly as many power conversion designs as there are power supply
designers. Nevertheless, there are a limited number of different power converter
structures which can then be implemented with a variety of components and different
options. For this project, the concern is not only the power converter design, but also the
PCM-4 design which must be able to take power from a variety of input voltages and
output, efficiently, to a variety of load sizes.
Page 12
The PCM-4 for this design must include variations for 3kV to 10kV input with power
rating variable between 1 and 5MW. The PCM-4 was divided up between the controller
and the modules. The PCM contains one controller which performs several critical
functions. These functions include maintaining the voltage setting of the individual power
modules to maintain 1000V DC output. In addition, the controller would ensure smooth
operation of the ship's electrical distribution system.
Main Propulson
Zonal
Distribution
ZONE 1
ZONE 2
ZONE N
800 VDC - 4SO VAC
nyerter
w
PMRDC
DC- DCCONVERTER
DC- DCCONVERTER
DC-DCCONVERTIR
0 350 - 800 Vdc
4SVac 30
45
100 Vd
DC- ACINVERTER
1000 vde 1
DC- DOCONVERTE
DC- ACINVERTER
DC- ACIN
,
350 - 800 Vdc
1000 Vde
DC - DCCONERTR
WC
- Dr.
8ON1R"E
1000-800 VDC
4160 Vaw
3+
Converter
Figur e 2 SatCon Applied Technology Distributed Power Systems [3]
Early on the power conversion strategy from SatCon Applied Technology was
selected. These PCMs use small modules which can be connected together to supply larger
loads. Using these ideas, the PCM-4 could be made up of a small number of modules where
the number var ies depending on the total load the PCM-4 is expected to supply.
A key co mponent of any power converter is the solid state switch. The power
converter envis ioned here is a switching type of power supply using Silicon based switches
.
The switches m ust be able to survive the maximum current and voltage difference they will
experience in t he converter design. Additionally, the switches must be kept cool enough to
avoid damagin gthe switching structure. Finally, the switches must be set up to allow the
switching to oc cur when there is no voltage and current on or through the switch.
Page 13
Depending on the specific component used, there are various other losses which must be
minimized.
From a technology point of view, the other components of the PCM, inductors,
transformer, computer control, are all very well established. The technology of the
switches is the component that is still being improved. This continuing improvement offers
a chance for future capability in terms of voltage limits, current capacity, and efficiency
improvements, but it also supplies a host of unknowns which must be dealt with in the
modeling process.
Page 14
CHAPTER 2 MODEL SPECIFICATION
2.1 SYSTEM SPECIFICATIONS
The model will have to conform to the same transient limits and other specifications
as a shipboard PCM. This will allow the model to provide useful results when coupled with
the other systems of the all-electric ship model. The specifications to be met are outlined in
this section along with the basis used to determine these specifications.
The Navy's NGIPS Roadmap indicates that the PCM-4 and PCM-1 will likely be
replaced by a PCM-1A which performs both tasks. With this in mind, this project will
attempt to make the PCM-4 modules such that they can be easily combined with a future
PCM-1 to maintain a consistent architecture with the Navy.
While Navy Specifications for their systems are not available in open literature, a
reference was found in the SBIR Program. The Navy is requesting design work on power
conversion devices that meet the requirements shown in Table 1. From the description,
these requirements are perhaps a bit conservative, but they should be representative of
what will be expected of future naval combatants. Only the threshold values were used in
the table below to simplify the design process.
Parameter
Threshold
SS Voltage Regulation
Transient Voltage Regulation
+/- 3.5%
+8.5%/-16.5%
Conversion Efficiency
Conversion Efficiency
Conversion Efficiency
Steady State Ripple Voltage
75%
96%
96.5%
2%
Condition
0 to 50%, 33 to 100% and 100 to
0% at 70 MW/sec
20% rated load
35% rated load
40 to 100% rated load
Table 1 Efficiency and Power Quality requirements [4]
The controller and converter circuitry work together to produce an easily
controllable output voltage. In addition, the output from the PCM-4 will be used to drive
another power conversion stage which will produce a particular voltage required by a load.
Page 15
2.2 BASIC MODEL STRUCTURE
Selecting the structure of the model is a much more complex question than simply
selecting the best of several options. Like the rest of ship design, the converter design is a
complex systems engineering problem. This implies that many of the parameters for the
converter depend on each other, and an optimal converter for one use is a poor choice for
another use. This selection is being made for a converter operated in the all-electric ship as
explained above. Efficiency, reliability, size and weight are all important considerations
which must be traded off against each other to select the best converter structure.
On a circuit level, there are more specific concerns that must be optimized. The size
of the energy storage components should be minimized. Additionally, minimizing the
stresses applied to the switching components will allow smaller and/or cheaper
components to be used and/or will increase the operational life of the converter.
Losses must also be minimized to increase the converter's efficiency. Switching
losses, conduction losses, and field losses must all be analyzed and minimized.
After the model has been completed, details such as the power capacity of each
module and component selection will be determined. Some level of optimization will be
performed for the output power level based on number of units for given maximum power
and efficiency. Finally, the components will be selected and/or designed. The actual
component parameters will be input into the model and used to validate all component
assumptions made during the design phase.
The different types of power converters have their relative advantages [5]. The
Direct Converter has significantly lower switch stresses than the indirect converter
topology. Additionally, the Direct Converter requires about half or less of the energy
storage that an indirect converter requires during reasonable operation. Therefore, the
direct topology will be utilized in this model.
Page 16
Bus voltage
1 module
modules
Multiple
Multiple modules
3.3 kV
Al
B1
6.5 kV
A2
B2
10 kV
A3
B3
Table 2 Power Converter Variants
As shown in Table 2, the conversion ratios will be from 0.1 (10kV input) to 0.3 (3kV
input). While 0.3 might produce reasonable switch stresses, the reduction in voltage from
10kV produces too much switch stress for a single stage design. One possible method to
reduce the switch stresses is to add a transformer to the direct converter. This type of
converter is called a forward converter. In addition to reducing the switch stresses, the
transformer also provides electrical isolation between the input and output. This feature
could be significant for the high voltage system, especially during a fault induced transient.
The basic forward converter is shown in Figure 3 below. The operation of this
forward converter really only draws power from the source for half of the cycle. When the
switch is on, the source supplies current to the transformer which charges up the filter
elements on the secondary side. For the second half of the cycle, the filter elements solely
supply the load. This type of converter works well, but it requires a mechanism to reset the
transformer on the primary side. Resetting the transformer refers to reducing the flux to
zero during each cycle to prevent a buildup of flux and saturation of the core. Additionally,
the basic forward converter also requires relatively large filter components to maintain a
constant voltage on the secondary side, especially at low switching frequencies.
Page 17
Figure 3 Basic Forward Converter from [11]
To minimize both of these limitations, a different approach was tried. The high
power switches have a relatively low switching frequency; this characteristic will tend to
increase the size of the energy storage components. As a result, full wave rectification was
chosen for the secondary side. To utilize a full wave rectifier, it was realized that the
transformer flux must not only be reset, but it must be reversed. This was accomplished on
the primary side of the transformer by modifying the inverter. Using a different switching
configuration, it was hoped that the transformer primary could be reversed yielding an AC
voltage on the secondary which would be suitable for full wave rectification.
Based on the above considerations and research, the Forward Converter was
selected for converter topology. This topology should minimize the switch stresses
allowing a larger input voltage and longer component life. Also, the lower voltage switches
are easier to develop if they are not already commercially available. The forward converter
also limits the amount of stored energy required which will minimize the size and weight of
the required inductors and capacitors in the filtering of the converter.
2.3 POWER CONVERSION MODULE CONTROL
The desire to enable the ship to reconfigure itself in response to damage or system
failures requires a networked power system. In addition to the reconfiguration options,
the networked controllers enable the different electrical distribution components to
respond very quickly.
The reconfiguration options included in this design are the ability to select a
different power source. This feature allows a PCM-4 to draw power from either the port or
Page 18
starboard bus. The switch is an electrically controlled bus transfer switch. With the size of
the filters required in the power supply, the assumption is that the switching of source
voltage could be done with loads energized.
A quick controller allows the system to respond many times faster than the cycle
speed of the machines at the sources or loads. For example, if a power generation module
(PGM) powering the PCM-4 was tripped on a fault, the PCM could instantly determine to
switch its source to another bus. An additional option is the PCM-4, working with the other
PCM-4's, could determine that the loading is too much for the remaining PGM, and the
PCM-4 could initiate load shedding through the PCM-1 and 2 modules. A fast controller and
efficient control codes could enable these changes to occur before the bus voltage has had a
chance to drop appreciably or a circuit breaker is tripped.
The PCM-4 power conversion modules must operate in parallel to supply the total
PCM-4 loads. In addition, the PCM-4 power conversion modules will have to share loads
with other PCM-4's. These features will have to be included in the control system.
Page 19
CHAPTER 3 MODEL DESIGN
To meet the specifications determined in chapter 2, the converter will have to be
carefully designed. The input and output filters will be designed and tuned to result in a
high efficiency while meeting the input and output requirements. Additionally, the
switching losses must be reduced to allow the converter to function at a higher efficiency.
Finally, the components required to perform these functions must be selected and/or
designed. It is recognized that some of the components may not exist, especially for the
higher bus voltage, but projected values will be used for components not commercially
available.
3.1 BASE SHIP SERVICE ELECTRICAL DISTRIBUTION SYSTEM LAYOUT
Using the general guidance developed in Chapters 1 and 2, a baseline electrical
distribution system layout was developed as shown in Figure 4 and Figure 5. This baseline
attempts to demonstrate the reliability features and general configuration that is
envisioned to operate around the PCM-4 designed herein. A simple 3-zone ship is shown
for emphasis only.
Page 20
MVDC
Figure 4 Baseline Electrical Distribution System
MVDC
P
PCM-1
1000 VDC
PCM - 4
PCM-1
MVDC
1000 VDC
Figure 5 Baseline Zonal Electrical Distribution System
Page 21
The baseline configuration has a port and starboard MVDC bus which feeds the
PCM-4's. This provides reliability in that all the PCM-4's can be supplied with power
independent of which set of generators in online. Likewise, the PCM-4's output of 1000
VDC is fed to a port and starboard ship service bus. Typically, the port and starboard ship
service buses will be fed by half of the PCM-4's to increase redundancy. Inside each zone,
the PCM-1's will be supplied by the opposite bus. Individual loads will then be supplied
from the PCM-1 or PCM-2's with vital and redundant loads powered from multiple sources.
The net effect of these reliability considerations is an electrical distribution system
that can get electrical power from any available source to any available load through any
available PCM configuration.
It should be noted that there are multiple configurations that the PCMs can be
arranged in. For example, the PCM-4 could be combined with the PCM-1 to form a PCM-1A.
There could be 2 PCM-1A's per zone to produce a redundant system with, perhaps, a
different survivability characteristic.
3.2 PCM-4 FUNCTIONAL BLOCK DIAGRAM
Simulink is the primary modeling tool utilized for this project. As such, the PCM-4 is
made up of subsystems: PCON and converter modules. The converter modules are further
broken down later. The overall PCM is shown below in Figure 6. More converter modules
can be added in parallel to supply larger loads.
Enuous
po we rgul
Figure 6 PCM-4 Model Block Diagram
Page 22
The minimum control signal is a voltage control signal from PCON that sets the
operating voltage of the modules. Sensing lines were also required to determine the actual
operating voltage of the PCM. These minimum signals are expanded in later sections as the
system design is expanded.
3.3 POWER CONVERTER
The first piece to explore is the Power Converter stage. The Power Converter stage
is made up of a switch, transformer, and diode as the major components. The power
converter stage produces an average output voltage at 1000 VDC. The input and output
filters work with the Power Converter to produce a voltage with a limited amount of ripple,
2% of steady state voltage. The converter is shown in Figure 7 below.
D1
N:1
Continuous
D2
out gnd
D3
powergui
D4
Switch Control
Figure 7 Simulink model of Converter Module
The forward converter consists of a direct converter and a transformer. The direct
converter produces an AC voltage with four switches. Therefore, even though the input
and output of the converter are DC voltages, the switched AC voltage allows the use of a
transformer in the middle of the converter, which transforms the voltage to the level
Page 23
required to produce the desired output voltage at the ideal duty ratio (D). The ideal duty
ratio value depends on the switch used and the circuit, but in general the ideal duty ratio
reduces stress on the switch and maximizes efficiency. Typically a 50% duty ratio or D of
0.5 is a good tradeoff.
The type of converter shown above is sometimes referred to as a double-ended
isolated forward converter. In addition to the advantages mentioned above, this converter
also reduces the voltage ratings required for the switches. Based on the current level of
switch technology, it is still expected that the higher MVDC bus voltages will require circuit
modification to achieve a realizable switch voltage rating.
3.3.1 CONVERTER MODULE SIZE
The PCM-4 high level architecture decided on earlier was a Converter made up of a
control module and the needed number of converter modules. To complete the design of
the PCM-4, the size of the individual converter modules had to be determined. This size
was bounded on both the lower and upper ends by physical considerations inherent in the
ship and the electrical distribution system
The size of the converter modules was determined to be large enough that the
design for paralleling the multiple modules was not overly complex. Too many modules
operating in parallel could have led to unexpected interactions. Additionally, it was
believed that having too many modules would decrease the overall efficiency of the PCM-4.
On the other hand, having too few modules would also impact efficiency by ensuring
that an individual module would rarely be operated at its ideal loading. The requirement to
have an installed spare module would also become cumbersome if the modules were too
large.
Initially, an in-depth optimization was planned for the module size. However, it
quickly became apparent that the number of variables from zone loading to ship type made
the prospect of optimizing the module size for any one configuration unproductive. As a
result, the possibility of multiple module sizes was introduced, but a 200 kW module size
was selected as the initial offering. This size was intended as a compromise between the
Page 24
previously mentioned parameters as well as trying, in advance, to limit the size of the
filtering components.
3.3.2 POWER SWITCH
The type of switch selected for this design affected all of the other components and
even influenced the converter layout. As a result, after exploring the basic transformer
frequency ranges, the switch selection was the next step. The proper switch selection was
critical to the efficiency and durability of the converter.
There are several types of switches to choose from. Common designs include
variations of Gate Turn-Off Thyristors (GTOs), Insulated Gate Bipolar Transistors (IGBTs),
Metal Oxide Semiconductor Field Effect Transistors (MOSFETs), and Bipolar power
Junction Transistors (BJTs). Each of these devices has advantages and disadvantages that
were explored to select the best device for the PCM-4. Silicon-Carbide (SiC) devices were
not seriously considered due to the lack of established devices. As larger SiC devices are
produced and tested, their advantages should be reviewed for inclusion in this type of
converter.
GTOs can handle high currents and high voltages, but they require commutation or
high turn off currents [7]. As a result, they were placed at the bottom of the component
selection.
Barkhordarian [8] provides a comparison of BJT and MOSFET devices as follows.
BJTs are current driven devices, and as a result, as much as 20% of the collector current is
required to keep the device on which drives up the cost and complexity of the switching
circuit. In addition, BJTs have a relatively slow switching period and are difficult to operate
in parallel. MOSFETs by comparison are faster switching and can be paralleled easily due
to the forward voltage decrease with temperature increases which helps to evenly
distribute the current to each parallel device. However, the forward voltage drop at high
voltages becomes worse than BJTs which limits the effectiveness of MOSFETs.
IGBTs are a combination of a FET and BJT. As a result, IGBTs are voltage controlled
devices that do not require a snubber circuit. IGBTs can also be combined in parallel to
provide added current capability [7]. The IGBT also has the low on resistance of a BJT and
Page 25
higher current capability than a MOSFET [9]. As a result, IGBTs have become "the
switching element of choice" in high voltage applications. Although the IGBTs have a lower
frequency range than MOSFETs, recent advances have increased the frequency range of
IGBTs while minimizing switching losses. As a result of these characteristics, the IGBT was
the first choice for a switch for the PCM-4 model.
To provide a safety margin for survivability, a factor of 2 was used for the switch
rating. For example, if 6 kV was expected across the switch during normal operation, the
switch or switches must be rated at 12 kV. Not only does this provide an inherent
capability to survive voltage transients, but it also improves the switch life.
The highest rating IGBTs commercially produced were rated at 6500 VDC. Of
course, this voltage rating would not support the 10 kV model variant and provide the
safety factor of 2. As a result, two options were considered. The first option was to assume
that eventually a higher voltage rating IGBT would be produced. The second option was to
use the IGBTs in a scheme that would allow an effective voltage splitting between multiple
IGBTs.
The first option appeared to be low risk. Based on the literature review, a 10 kV
switch appears to be the next iteration of the IGBT design. The second option evaluated
allowed for operating IGBTs in series. Ju Won Baek [10] presents a simple and efficient
series connection which was used as a starting design for series switch connection. He uses
an RC circuit to balance the voltage across 2 switches and allow for a higher operating
voltage.
The variants shown in Table 2 will be analyzed individually starting with variant Al
which is a single module operated at 3.3kV input voltage. In order to achieve the 2 times
rated voltage, at least a 3kV switch must be used since 2 switches are in series. As
estimated in Equation (1) below [5], the switch current is approximately 128 A.
P=
2
pf
-
Pf=
I, =
P
Vppf
2(200kW)
=2P
128A
-3300V(0.9S)
(1)
The Hitachi 3.3kV switch referenced for this paper has a current rating of 1500A
providing a comfortable current margin. The higher current and voltage rating will also
Page 26
allow the converter module to absorb a rather large transient without damage to the
switches. As shown below in Figure 8, the forward voltage drop is approximately 1.3V at a
worst case 125"C. Figure 9 indicates that the turn on loss will be approximately 0.4 J/pulse,
and Figure 10 indicates an approximately 0.35 J/Pulse turn off loss. Finally, the reverse
recovery losses will be approximately 0.35 J/Pulse as shown in Figure 11. The turn on time
is approximately 1.9[tsec with a turn off time of about 5.8 psec as shown in Figure 12.
TYPICAL|
3000
-
VGE=OV
2500
Tc=25 'C
2000
LL
3
1500
-o
- ---
12
0
1000
Tol
-- TG
25 C50
150C
C_
500
0
0
1
2
3
4
Forward Voltage, VF(V)
Figure 8 3.3kV IGBT forward voltage drop [12]
Page 27
TYPICAL
4.0
-[EConditional
Ls=10-nH
3.5
-va-t
Eon(full)
av
Eon(fu--0
R13=2.79
(10C)
CGE=330nF
Tc=125"O
3.0
Eon(1 0%)
2.5
We-
I
aS!
0
(n 2.0
(P2
0
'
0
0
a- 1.5
PR-
t
Eon(0)
-7
Eon(full) =
0
200
400
600
800
1000
7t3
4
12
dt
(exC)-
(lex VcE)- dt
1200
1400
Collector Current, IC(A)
Figure 9 3.3kV IGBT Turn on loss [12]
Page 28
1600
T YPICALI
4.0
[Conditions]
Vow= IIIOV
3.5
v-*15v
-CGE=330nIF
d0V
Ven
_
TU=t25te
= PICA vee
.of(tA)
3.0
a
E-off(full)=((k - VCE
Q.
Eoff(full)
S2.5
Eoff(ful)
(150 c)
CL
U
S2.0
0
C
Eoff(10%
1.5
1-
F :z
0
200
400
600
800
1000
1200
1400
Collector Current, Ic(A)
Figure 10 3.3kV IGBT Turn off loss [12]
Page 29
1600
TYPICAL
3.0
[Conditions)
_eaI lH
VeeGt 650V
VG=*1 5V
2.5
RG=2.7Q
CGE=330nF
Te125%C
UD
Err(full)
-(150 C)
Err(full)
0
-
Err(1 0%)
1.0
0.5
t11
110
Err(10%.)=
cl
Err(full)=
(1,x VCE)-d
xVe) .dt
--
-
0.0
0
200
400
600
800
1000
1200
1400
1600
Forward Current, IF(A)
Figure 11 3.3kV IGBT Reverse Recovery Loss [12]
Page 30
TYPICA L
6.0
1
1
Lon dition
- -
- -
-
- -
-
-
-
-
-
-
-
-
-
~ALa100fnM
-
c=1UUIIV
-
-
VG-*I5V
G=2.70
5.0
0-E- 330oF
4
Ta=12
T
toff
4.0
~~
-
0
~
ton
--
E
1.0
trr
-
0.0
0
200
400
600
800
1000
1200
1400
1600
Collector Current, IC(A)
Figure 12 3.3kV IGBT switching times [12]
A larger switch was required for variant A2. In order to achieve the 2 times rated
voltage, at least a 6kV switch must be used since, again, 2 switches are in series. As
estimated in Equation (2) below, the peak switch current is approximately 65 A.
I
=
2
2P
P
Vppf
_
2(200kW)
-
65A
(2)
6500V(0.95)
The Hitachi 6.5kV switch referenced for this paper has a current rating of 750 A
providing a comfortable margin. As with the 3.3 kV switch, it is believed that using the
larger switch at a lower than rated power increases the operational life and reliability. As
shown below in Figure 13, the forward voltage drop is approximately 1.4V at 125"C. Figure
14 indicates that the turn on loss will be approximately 0.8 J/pulse, and Figure 15 indicates
an approximately 0.6 J/Pulse turn off loss. Finally, the reverse recovery losses will be
Page 31
approximately 0.55 J/Pulse as shown in Figure 16. The turn-on time is approximately 2.3
ptsec with a turn off time of about 5.8 iisec as shown in Figure 17.
TYPICAL
I I
1400
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Tc=-1251C
1200
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2
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5
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Forward Voltage, VF(V)
Figure 13 6.5kV IGBT forward voltage drop [13]
0
+ -+ + + - -+ +
ill
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Page 32
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VGE=±15V,
Rg on'g
2Q
Vcc-3600V, L4200nH, Tj-I 25 C
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500
750
1000
Collector Current , Ic (A)
1250
Turn-on Loss vs. Collector Current
Figure 14 6.5kV IGBT turn-on loss [13]
Page 33
1500
TYPICAL
12.0
-F-Ft
L
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--
-----
LL
'
L1
J- J -- J
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11.0
L
.
Vcc
3600VL-2OOnH.
125'C
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±15Vgffh2
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--
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1-
---
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-+
L L -J.J.J e - + i
750
1000
1250
Colector Current , Ic (A)
Turn-off Loss vs.Collector Current
Figure 15 6.5kV IGBT turn-off loss [13]
Page 34
+
1500
~ -I~[-
LMLL . J. J .1J
TYPICAL
I
5.0
,,
-r T VGE ±15VRgon, 32Q
- T Vcc 36COVL-2OOnHTj 125C
F-T ltduA ve luid
FT
I If
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4.0
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li
liii
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iIi i
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0.0)
0
250
500
750
1000
1250
1500
IIIA)
Figure 16 6.5kV IGBT reverse recovery loss [13]
Forward Current,
Page 35
TYPICA
10.0
L
V L
i-
II
9.0
15V Rgo -.
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500
750
CollectorCirmat
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1000
(A)
1250
1500
Figure 17 6.5kV IGBT switching times [13]
In order to proceed with the design, the decision was made to assume that a 10kV
switch will become available. Since no data is available, the parameters were extrapolated
based on the differences from the 3.3kV to 6.5kV switches. This should provide a
reasonable starting point for efficiency calculations. Future design work could investigate
the methods of combining switches in series if the 10kV switches are not yet producible.
The table below summarizes the switch parameters from above as well as the
estimates of the 10kV switch parameters. These values will be used later in the analysis to
determine the efficiency of the converter. Equation (3) below determines the value of peak
current as 42A.
Page 36
P2=
pf2 f
|
*
P
Vppf
f
Load
Design I SwitchI Current
Al
A2
A3
3.3 kV
6.5 kV
10.0 kV
2(200kW)=
42A
(3)
-10000V(0.95)
Forward
Voltage Drop
128 A
65 A
42A
1.3 V
1.4 V
1.4 V
Turn-on
Loss
Turn-off
Loss
Reverse
Recovery
Turn-on
Time
Turn-off
Time
0.40J/pulse 0.35J/pulse 0.35J/pulse 1.9 usec 5.8 usec
0.80J/pulse 0.60J/pulse 0.55J/pulse 2.3 usec 5.8 usec
1.10 J/pulse 0.75J/pulse 0.70J/pulse 2.6usec 5.8usec
Table 3 Switch Properties
3.3.3 CONVERTER LAYOUT
The second iteration of this converter layout is shown below in Figure 18. This
circuit includes a Voltage Control block and a Voltage regulator block. In addition, because
a relatively low switching frequency is expected, a block called, Shunt, was added as a
placeholder for some active filtering components. This system provided the starting point
for the design that follows.
Figure 18 Second iteration of converter layout
As the design progressed, it was expected that the Voltage Control block would be
used to transform the control signal (duty ratio, D) into a control signal to actuate the
Page 37
switches which take MVDC and produce a switched AC voltage at the input to the
transformer.
The voltage regulator block was intended to modify the output voltage as required
to allow multiple modules to be operated in parallel. In addition, the voltage regulator
would also help to minimize transients through providing the control signal to the voltage
controller as well as controlling the active filtering components in the Shunt block.
A maximum steady-state and transient input voltage variation of 3.5% of the MVDC
bus voltage will be accepted while still producing 1000 VDC output voltage within the
acceptable band. The final consideration for the converter layout will be the combining of
switches in series. This will be done to ensure a safety factor of 2 above the normal
operating voltage.
3.3.4 CONVERTER WAVEFORMS
Some considerable energy was applied to developing an optimal waveform. As
mentioned above, the input waveform to the transformer must reverse polarity to allow
full wave rectification and decrease the size of the filtering components as well as eliminate
the clamping circuit. Additionally, a waveform was chosen which would allow the output
voltage to be increased or decreased without changing the switching frequency or input
amplitude.
A square sine-wave input to the transformer would allow the maximum energy
transfer with the smallest energy storage. However, some allowance must be made for
input voltage variation. In addition, a small period of zero input voltage will help the
transformer transition to a reverse current. An initial waveform incorporating these
factors is shown in Figure 19. The waveform variables are described in Table 4.
Page 38
t (sec)
OT
Figure 19 Base Transformer Input Waveform
Variable
Description
T
OT
D
F
Period of one cycle
Off Time between positive and negative half-cycle
Duty ratio between 0 and 1 with a base value of 0.5
Delay Factor for ZVS and ZCS initiation
(smaller factor is more delay)
Table 4 Transformer Input Waveform
The transformer operation and design can be simplified if the input waveform
approximates a sinusoidal waveform. While a multiple level waveform could be used to
more closely approximate a sinusoid, it was decided that a simpler approach could be used
for this application. The area under the square wave was set equal to the area of a sinusoid.
This has the added advantage of producing the same Volt-sec, or flux, in the transformer as
the sinusoid per cycle. Equation (4) shows the calculation of the area to be placed under
the square wave and the value of the variables.
f1
sin(" t) dt = 4 f 4 sin (
t) dt
(4)
=
The area under the unit sinusoid is 2T/w. To mimic the same flux distribution, the
square wave in Figure 19, the square wave must have a total area of 2/7r per cycle. This
calculation is shown in Equation (5) below. This value of OT will be selected as the base
value for the space between the square wave halves. When the frequency of the switching
circuit is selected, the value of OT will be a constant value until a control signal, D, is
applied to raise or lower the voltage.
(T -
20T)
* 1 =
2T
iF
=
OT
=
f
2
T
T
21r
=
2
-
ir
T
(5)
The first run through the converter operation was made with an attempt to
minimize the stresses on the switches. It is assumed that additional circuitry would have to
Page 39
be added to achieve Zero Voltage Switching (ZVS) or Zero Current Switching (ZCS);
however, efforts were made to reduce the switching stresses on average. The switching
sequence shown in Table 5 has 4 unique modes of operation which are designed to
produce ZVS switching for switches 2 and 4. This will help to minimize the size of the
module as well as increase converter efficiency. The delay factor, F, was adjusted to
provide the desired delay between the operation of the two switches in series.
Mode
Switch
Time
Comment
1
S4--C
S1-C
S1--O
S4-+O
S2->C
S3-C
S3-O
OT(1-D)*F
OT(1-D)
0.5T-OT(1-D)
0.5T-OT(1-D)*F
O.5T+OT(1-D)*F
0.5T+OT(1-D)
T-OT(1-D)
ZVS
S2-+O
T-OT(1-D)*F
ZVS
2
3
4
ZVS
ZVS
Table 5 Converter switching modes
The converter control is realized using time delays as shown in Figure 20. A pulse
circuit was used as a clock signal. The frequency of the clock was determined by the
switching frequency of the switch. When the positive pulse is initiated, the delay timers are
activated. As shown in Table 5, there are 8 timers. Each of the timers is used as an input
signal which either shuts or opens the inverter switches S1-S4. The operating frequency,
or clock, of the system is not adjusted after the system is constructed. This voltage control
circuit is shown below in Figure 20.
Page 40
Generator
INV5
Figure 20 Voltage Control Circuit with ZVS Delay Factor (F)
The control signal, D, from PCON is used to adjust the voltage of the inverter. As can
be seen mathematically in Table 5, when D is increased, the total area under the volt-sec
curve in Figure 19 will increase. This is equivalent to more flux being generated in the
transformer and more power being transferred to the secondary side. As a result, more
energy is stored in the secondary filtering components, and the average voltage increases.
The converter is designed to operate in a steady state with D equal to 0.5. As the converter
responds to transients on either the input or output side, D is adjusted to minimize the
change in voltage on the output.
As the design evolved, the delay factor was abandoned as a means to control the
switches to achieve one switch operation under ZVS. It was acknowledged that a path must
be provided to allow continuity of current in the windings of the transformer. As a result,
two switches are always left shut. The switches will first shut to provide a path from the
source to the transformer, and then the switches will operate to provide a path for the
transformer primary current to coast down to zero. The revised converter switching is
shown below in Table 6.
Page 41
SwitchComment
Time
S1
S2
S3
0 0 C
0
C 0 0
OT(1-D)
C C 0
O.5T-OT(1-D)
0 C C
O.5T+0T(1-D)
0 0 C
T-OT(1-D)
Table 6 Updated
S4
Carryover State
C
C
O
0
C__
Switching Sequence
Table 6 above takes into account the continuity of transformer primary current. In
addition, the number of switching states was increased to equalize the switching stresses.
The first task was accomplished by always leaving 2 switches closed across the
transformer primary. These closed switches provide a current path for primary current to
continue to flow during the OT periods while the input is not connected to the transformer.
To balance out the heat loading and current stress on the switches, S1 and S3 would be
shut one half cycle and then S2 and S4 during the other half cycle. This also requires that
the delay factor (F) be removed and the switching order be modified from Table 5 as
shown in Table 6.
Eliminating the delay factor (F) provided some benefit but also added some
complications. The benefit of eliminating F was that now the duty ratio (D) could be varied
from 0 to 1 without additional circuitry being added. However, now, all four switches will
be switched under full voltage. To minimize the switching losses, ZVS/ZCS needs to be
addressed 4 times vice 2 times, adding to the circuitry, cost, and weight.
These changes required the Voltage Control circuit to be redesigned. In addition to
the above discussion, this redesign was brought on by early testing which showed that D
might have to vary significantly in order to help minimize transients. The second iteration
of the voltage control circuit is shown below in Figure 21.
Page 42
(1 -D)2
Figure 21 Second Iteration of Voltage Controller
After some initial testing of the operation, another variation was analyzed. If the
transformer current was allowed to coast down on the secondary side, the primary
switches would not have to be configured to provide an off-state conduction path. This
approach also allows the duty ratio to be adjusted throughout a prescribed range that
covers 0 to nearly 1. The upper limit is still based on giving the transformer time to reverse
fields, but the voltage control circuit does not inherently limit these values.
Page 43
Figure 22 Final Voltage Control Circuit
Figure 22 above was also modified to allow greater flexibility in the control of PCM
voltage. In addition to simplifying the circuit, timers were added to allow short cycling its
operation. For example, if the requested duty ratio changes during the current cycle, the
new duty ratio will modify the control signals in real time. The advantage is that the
relatively slow switching frequencies can effectively be increased without increasing the
actual frequency and therefore switching losses. The circuit's extra features were designed
to allow a switch to be opened or shut at any time during its normal half-cycle. However, to
limit the increases in switching losses, once the switch pairs are opened during a half cycle,
they cannot be shut again until their next operation period. This allows the switch to be
opened at any time, even if the requested duty ratio was 90% when the switch was shut.
3.3.5 CONVERTER EFFICIENCY
After the switching schema was determined, a comparison analysis was performed
to optimize the switching frequency and switch losses with the size of the transformer and
filtering components. An initial calculation was performed in Table 7, Table 8 and Table 9
below. This calculation was based on the manufacturer's data presented in the previous
section and summarized in Table 3. The primary losses accounted for included the
conduction loss calculated from the forward voltage drop and conduction current.
Additionally, the turn-on and turn-off losses and reverse recovery losses were determined
Page 44
from the manufacturer's data. These are typically the primary losses of concern. These
losses were calculated based on the 2 series switches.
The diode losses were based on a DYNEX rectifier diode with a voltage drop of
approximately 0.8V with about 200A of average current. The losses are relatively small,
but they are given their own category to differentiate them from the other losses involved.
3.3 kV 500 Hz
750 Hz
Power
200 kW 200 kW
power factor
0.95
0.95
Conduction Current
128A
128A
Forward Drop
1.3V
1.3V
D
0.50
0.50
F
1.00
1.00
T
2.OE-03 1.3E-03
Turn-on loss
0.4J/pul 0.4J/pul
Turn-off loss
0.4 J/pul 0.4 J/pul
Reverse recoveries 0.4J/pul 0.4 J/pul
Non-ZVS On Trans
2000
3000
Non-ZVS Off Trans
2000
3000
Reverse recoveries
2000
3000
Cond. Time/cycle
1.OE-03 6.7E-04
Conduction diss.
0.2 kW
0.2 kW
Switching Diss.
02.4 kW 03.6 kW
Transformer Loss
04.0 kW 04.0 kW
Filter Loss
02.0 kW 02.0 kW
Diode Loss
00.3 kW 00.3 kW
Power Loss
08.9 kW 10.1 kW
Switching Efficiency
95.6%
95.0%
1000Hz
200 kW
0.95
128A
1.3V
0.50
1.00
1.OE-03
0.4J/pul
0.4J/pul
0.4J/pul
4000
4000
4000
5.OE-04
0.2 kW
04.8 kW
04.0 kW
02.0 kW
00.3 kW
11.3 kW
94.4%
1500 Hz2000Hz
200 kW 200 kW
0.95
0.95
128A
128A
1.3V
1.3V
0.50
0.50
1.00
1.00
6.7E-04 5.OE-04
0.4 J/pul 0.4 J/pul
0.4 J/pul 0.4 J/pul
0.4J/pul 0.4J/pul
6000
8000
6000
8000
6000
8000
3.3E-04 2.5E-04
0.2 kW
0.2 kW
07.2 kW 09.6 kW
04.0 kW 04.0 kW
02.0 kW 02.0 kW
00.3 kW 00.3 kW
13.7 kW 16.1 kW
93.2%
92.0%
2500 Hz
200 kW
0.95
128A
1.3V
0.50
1.00
4.OE-04
0.4J/pul
0.4J/pul
0.4 J/pul
10000
10000
10000
2.OE-04
0.2 kW
12.0 kW
04.0 kW
02.0 kW
00.3 kW
18.5 kW
90.8%
Table 7 Efficiency for 3.3 kV converter
Page 45
3000Hz_
13500-Hz
200 kW 200 kW
0.95
0.95
128A
128A
1.3V
1.3V
0.50
0.50
1.00
1.00
3.3E-04 2.9E-04
0.4J/pul 0.4J/pul
0.4J/pul 0.4J/pul
0.4 J/pul 0.4J/pul
12000
14000
12000
14000
12000
14000
1.7E-04 1.4E-04
0.2 kW
0.2 kW
14.4 kW 16.8 kW
04.0 kW 04.0 kW
02.0 kW 02.0 kW
00.3 kW 00.3 kW
20.9 kW 23.3 kW
89.6%
88.4%
40
z
200 kW
0.95
128A
1.3V
0.50
1.00
2.5E-04
0.4 J/pul
0.4 J/pul
0.4J/pul
16000
16000
16000
1.3E-04
0.2 kW
19.2 kW
04.0 kW
02.0 kW
00.3 kW
25.7 kW
87.2%
6.5 kV 500 Hz
Power
power factor
Conduction Current
Forward Drop
D
F
T
Turn-on loss
Turn-off loss
Reverse recoveries
Non-ZVS On Trans
Non-ZVS Off Trans
Reverse recoveries
Cond. Time/cycle
Conduction diss.
Switching Diss.
Transformer Loss
Filter Loss
Diode Loss
Power Loss
Switching Efficiency
J
750 Hz
200 kW 200 kW
0.95
0.95
065 A
065 A
1.4V
1.4V
0.50
0.50
1.00
1.00
2.OE-03 1.3E-03
0.8 J/pul 0.8 J/pul
0.6 J/pul 0.6J/pul
0.6 J/pul 0.6 J/pul
2000
3000
2000
3000
2000
3000
1.OE-03 6.7E-04
0.09 kW 0.09 kW
04.0 kW 06.0 kW
04.0 kW 04.0 kW
02.0 kW 02.0 kW
00.3 kW 00.3 kW
10.4 kW 12.4 kW
94.8%
93.8%
1000 Hz
1500 Hz
2000 Hz
2500 Hz
200 kW 200 kW 200 kW 200 kW
0.95
0.95
0.95
0.95
065 A
065 A
065 A
065 A
1.4V
1.4V
1.4V
1.4V
0.50
0.50
0.50
0.50
1.00
1.00
1.00
1.00
1.OE-03 6.7E-04 5.OE-04 4.OE-04
0.8 J/pul 0.8 J/pul 0.8 J/pul 0.8 J/pul
0.6J/pul 0.6 J/pul 0.6J/pul 0.6J/pul
0.6J/pul 0.6J/pul 0.6 J/pul 0.6 J/pul
4000
6000
8000
10000
4000
6000
8000
10000
4000
6000
8000
10000
5.OE-04 3.3E-04 2.5E-04 2.OE-04
0.09 kW 0.09 kW 0.09 kW 0.09 kW
08.0 kW 12.0 kW 16.0 kW 20.0 kW
04.0 kW 04.0 kW 04.0 kW 04.0 kW
02.0 kW 02.0 kW 02.0 kW 02.0 kW
00.3 kW 00.3 kW 00.3 kW 00.3 kW
14.4 kW 18.4 kW 22.4 kW 26.4 kW
92.8%
90.8%
88.8%
86.8%
3000 Hz
3500 Hz 14000 Hz
200 kW 200 kW
0.95
0.95
065 A
065 A
1.4V
1.4V
0.50
0.50
1.00
1.00
3.3E-04 2.9E-04
0.8 J/pul 0.8 J/pul
0.6 J/pul 0.6J/pul
0.6 J/pul 0.6 J/pul
12000
14000
12000
14000
12000
14000
1.7E-04 1.4E-04
0.09 kW 0.09 kW
24.0 kW 28.0 kW
04.0 kW 04.0 kW
02.0 kW 02.0 kW
00.3 kW 00.3 kW
30.4 kW 34.4 kW
84.8%
82.8%
200 kW
0.95
065 A
1.4V
0.50
1.00
2.5E-04
0.8 J/pul
0.6J/pul
0.6 J/pul
3000 Hz
200 kW
0.95
042 A
1.4V
0.50
1.00
3.3E-04
1.1J/pul
0.8 J/pul
0.7 J/pul
12000
12000
12000
1.7E-04
0.06 kW
31.2 kW
04.0 kW
02.0 kW
00.3 kW
37.6 kW
81.2%
4000Hz
200 kW
0.95
042 A
1.4V
0.50
1.00
2.5E-04
1.1J/pul
0.8J/pul
0.7 J/pul
16000
16000
16000
1.3E-04
0.06 kW
41.6 kW
04.0 kW
02.0 kW
00.3 kW
48.0 kW
76.0%
16000
16000
16000
1.3E-04
0.09 kW
32.0 kW
04.0 kW
02.0 kW
00.3 kW
38.4 kW
80.8%
Table 8 Efficiency for 6.6 kV converter
10.0 kV
Powe r
power factor
Conduction Current
Forward Drop
D
F
T
Turn-on loss
Turn-off loss
Reverse recoveries
Non-ZVS On Trans
Non-ZVS Off Trans
Reverse recoveries
Cond. Time/cycle
Conduction diss.
Switching Diss.
Transformer Loss
Filter Loss
Diode Loss
Power Loss
Switching Efficiency
500 Hz
200 kW
0.95
042 A
1.4V
0.50
1.00
2.OE-03
1.1J/pul
0.8 J/pul
0.7 J/pul
2000
2000
2000
1.OE-03
0.06 kW
05.2 kW
04.0 kW
02.0 kW
00.3 kW
11.6 kW
94.2%
750 Hz 1_1000 Hz
200 kW 200 kW
0.95
0.95
042 A
042 A
1.4V
1.4V
0.50
0.50
1.00
1.00
1.3E-03 1.OE-03
1.1J/pul 1.1J/pul
0.8 J/pul 0.8 J/pul
0.7 J/pul 0.7 J/pul
3000
4000
3000
4000
3000
4000
6.7E-04 5.OE-04
0.06 kW 0.06 kW
07.8 kW 10.4 kW
04.0 kW 04.0 kW
02.0 kW 02.0 kW
00.3 kW 00.3 kW
14.2 kW 16.8 kW
92.9%
91.6%
1500 Hz
200 kW
0.95
042 A
1.4V
0.50
1.00
6.7E-04
1.1J/pul
0.8 J/pul
0.7 J/pul
6000
6000
6000
3.3E-04
0.06 kW
15.6 kW
04.0 kW
02.0 kW
00.3 kW
22.0 kW
89.0%
2000 Hz 12500 Hz
200 kW 200 kW
0.95
0.95
042 A
042 A
1.4V
1.4V
0.50
0.50
1.00
1.00
5.OE-04 4.OE-04
1.1J/pul 1.1J/pul
0.8 J/pul 0.8 J/pul
0.7 J/pul 0.7 J/pul
8000
10000
8000
10000
8000
10000
2.5E-04 2.OE-04
0.06 kW 0.06 kW
20.8 kW 26.0 kW
04.0 kW 04.0 kW
02.0 kW 02.0 kW
00.3 kW 00.3 kW
27.2 kW 32.4 kW
86.4%
83.8%
Table 9 Efficiency for 10kV converter
Page 46
3500 Hz
200 kW
0.95
042 A
1.4V
0.50
1.00
2.9E-04
1.1J/pul
0.8J/pul
0.7 J/pul
14000
14000
14000
1.4E-04
0.06 kW
36.4 kW
04.0 kW
02.0 kW
00.3 kW
42.8 kW
78.6%
These calculations yielded an approximate efficiency of the switching circuit
between 76% and 95.6% depending on the voltage and frequency variations. This is
significantly below the requirement of 96.5% at full load for most configurations. The
transformer efficiency of approximately 98% and an additional 1% loss in the filtering
components was assumed for this calculation.
Although the switches are less efficient at high frequencies, a higher frequency will
allow a smaller transformer and filtering components. Therefore, ZVS and ZCS techniques
should be investigated in a future study to determine their suitability in this application. A
typical ship's service electrical loading is approximately 5-10 MW. A 3% efficiency increase
would amount to as much as 150-300 kW in power savings as well as heat load on the
ship's cooling systems. Based on the results above, the optimal switching frequency for
these switches was 1500 to 2000 Hz. Above this frequency, the efficiency of the switch
decreases rapidly without additional circuitry to limit the voltage and current at switching.
After analyzing the efficiency results above, two factors became obvious. The
conduction losses were approximately 0.3% of the switching losses. As a result, the type of
switch used was investigated to determine if there were any better matches. The ideal
switch for this application does not need to have a high current rating, and in fact, a lower
current rating might allow the switch to be made such that the switching losses are lower.
Also, because the conduction losses are so low, a switch's conduction losses are not really a
controlling factor. MOSFETs are typically faster switching but have higher forward voltage
drops than IGBTs. Initially, this last factor led to the selection of the IGBT. However, at this
stage, MOSFETs were investigated more closely to determine if they could reduce the
switching losses at the expense of increasing conduction losses.
Additionally, the initial assumption was that using an IGBT that was overrated
would significantly increase component durability. True as this may be, it was later
realized that this could also increase the losses significantly more than expected. This also
led to lower rated IGBTs being evaluated. The overall goal of the switch re-evaluation was
to increase the efficiency of the converter to the required level.
Page 47
A review of several power switch manufacturers such as Toshiba and Powerex did
not reveal any MOSFET technologies that allowed even 3300 V switching. As a result,
MOSFETs were once again removed from consideration. However, two lower rated IGBTs
were located. One was a Powerex module, and the other was a Mitsubishi module. With a
much lower rating of 400A and 200A for 3300V and 6500V respectively, these IGBTs more
closely matched the required values.
These switches were still overrated, so a factor was applied to the key
characteristics. This factor was intended to approximate the change in these parameters
for a switch that was rated for 2 times the current requirement. The key graphs for the
new switches are shown in Figures 23 through 28 below. The values are summarized in
Table 10.
800
Q
z
w
600
400
0
0
W
200
0
0L
0
2
4
6
8
10
COLLECTOR-EMITTER VOLTAGE VCE (V)
Figure 23 3300V Switch forward voltage drop [17]
Page 48
2.0
Vcc = 1650V, VGE= +15V,
RG = 7.52, Tj = 125*C,
Inductive load
0L
1.5
0
z
0
1.0
--
Eon
_
z
0.5
-
__
__
Eoff
_
C/)
Erec
-
0
0
200
600
400
800
CURRENT (A)
Figure 24 3300V switching energy [17]
2000 r..
.
.
.
-
4
8
12
16
1600
1200
800
400
20
COLLECTOR-EMITTER VOLTAGE, VCE(sat, (VOLTS)
Figure 25 6500V forward voltage drop [16]
Page 49
4.0
,
3.5
Vcc = 3600V
VGE=±15V
G (f) =
3.0
-
72 0
RG(on) = 301
=200nH
2.5
L
2 .0
T1= 125*C
--
-
- - - -
1.5
1 .0
-_-
- - --
-
---
0.5
0
0
100
200
300
400
500
COLLECTOR CURRENT, IC,(AMPERES)
Figure 26 6500V Switch turn-on loss [16]
4.0
I
3.5
Vcc = 3600V
VGE = ±15V
RG(ofUj = 7212
3.0
-
RGoen' = 30L2
2.5
Ls
= 200nH
T = 125*C
2.0
- - -0
1.5
- -
,0 - -
-
-
-
-
1.0
0.5
0
100
200
300
400
COLLECTOR CURRENT, IC,(AMPERES)
Figure 27 Powerex 6500V turn-off loss [16]
Page 50
500
1.2
Vcc =3600V
VGE=±15V
1.0
q
RG(on) = 300
w
:0.8-
L= 200nH
T = 125*C---
w0
Lp0.6
CE)
cc
a: 0
0
100
200
300
400
500
EMITTER CURRENT, 'E, (AMPERES)
Figure 28 Powerex 6500V reverse recovery [16]
Design Switch
Al
A2
A3
3.3 kV
6.5 kV
10.0 kV
Load
Current
Forward
Voltage Drop
128 A
65 A
42A
3.0 V
3.0 V
3.0 V
Turn-on
Loss
Turn-off
Loss
Reverse
Recovery
0.20J/pulse 0.25J/pulse 0.12J/pulse
0.50J/pulse 0.40J/pulse 0.35J/pulse
0.65J/pulse 0.70J/pulse 0.60J/pulse
Table 10 Switch summaries
As can be seen from a comparison of Table 10 and Table 3, the switching losses are
much lower for the lower rating switches. However, the forward voltage drop is
significantly higher. The higher voltage intuitively makes sense because the lower current
rating switch has a smaller conduction surface area. The tables below summarize the 3.3,
6.5, and 10kV power converter efficiencies using the smaller switches.
Page 51
3.3 kV 500 Hz
Power
power factor
Conduction Current
Forward Drop
D
F
T
Turn-on loss
Turn-off loss
Reverse recoveries
Non-ZVS On Trans
Non-ZVS Off Trans
Reverse recoveries
Cond. Time/cycle
Conduction diss.
Switching Diss.
Transformer Loss
Filter Loss
Diode Loss
Power Loss
Switching Efficiency
200 kW
0.95
128 A
3.OV
0.50
1.00
2.OE-03
0.2 J/pul
0.3 J/pul
0.1J/pul
2000
2000
2000
1.OE-03
0.4 kW
01.1kW
04.0 kW
02.0 kW
00.3 kW
07.8 kW
96.1%
750 Hz
1000 Hz
1500 Hz
2000 Hz
200 kW
0.95
128A
3.0 V
0.50
1.00
1.3E-03
0.2 J/pul
0.3 J/pul
0.1J/pul
3000
3000
3000
6.7E-04
0.4 kW
01.7 kW
04.0 kW
02.0 kW
00.3 kW
08.4 kW
95.8%
200 kW
0.95
128 A
3.0 V
0.50
1.00
1.OE-03
0.2 J/pul
0.3 J/pul
0.1J/pul
4000
4000
4000
5.OE-04
0.4 kW
02.3 kW
04.0 kW
02.0 kW
00.3 kW
09.0 kW
95.5%
200 kW 200 kW
0.95
0.95
128 A
128A
3.0 V
3.OV
0.50
0.50
1.00
1.00
6.7E-04 5.OE-04
0.2 J/pul 0.2 J/pul
0.3 J/pul 0.3 J/pul
0.1J/pul 0.1J/pul
6000
8000
6000
8000
6000
8000
3.3E-04 2.5E-04
0.4 kW
0.4 kW
03.4 kW 04.6 kW
04.0 kW 04.0 kW
02.0 kW 02.0 kW
00.3 kW 00.3 kW
10.1 kW 11.3 kW
94.9%
94.4%
2500 Hz
3000 Hz
3500Hz
4000 Hz
200 kW 200 kW 200 kW 200 kW
0.95
0.95
0.95
0.95
128 A
128A
128 A
128 A
3.0 V
3.0 V
3.0 V
3.0 V
0.50
0.50
0.50
0.50
1.00
1.00
1.00
1.00
4.OE-04 3.3E-04 2.9E-04 2.5E-04
0.2 J/pul 0.2 J/pul 0.2 J/pul 0.2 J/pul
0.3 J/pul 0.3 J/pul 0.3 J/pul 0.3 J/pul
0.1J/pul 0.1J/pul 0.1J/pul 0.1J/pul
10000
10000
10000
2.OE-04
0.4 kW
05.7 kW
04.0 kW
02.0 kW
00.3 kW
12.4 kW
93.8%
12000
12000
12000
1.7E-04
0.4 kW
06.8 kW
04.0 kW
02.0 kW
00.3 kW
13.5 kW
93.2%
14000
14000
14000
1.4E-04
0.4 kW
08.0 kW
04.0 kW
02.0 kW
00.3 kW
14.7 kW
92.7%
16000
16000
16000
1.3E-04
0.4 kW
09.1kW
04.0 kW
02.0 kW
00.3 kW
15.8 kW
92.1%
Table 11 3.3kV Power Converter Efficiency
6.5 kV 500 Hz
Powe r
200 kW
powerfactor
0.95
Conduction Current
065 A
Forward Drop
3.OV
D
0.50
F
1.00
T
2.OE-03
Turn-on loss
0.5 J/pul
Turn-off loss
0.4J/pul
Reverse recoveries 0.4J/pul
Non-ZVS On Trans
2000
Non-ZVS Off Trans
2000
Reverse recoveries
2000
Cond. Time/cycle
1.OE-03
Conduction diss.
0.19 kW
Switching Diss.
02.5 kW
Transformer Loss
04.0 kW
Filter Loss
02.0 kW
Diode Loss
00.3 kW
Power Loss
09.0 kW
Switching Efficiency
95.5%
750 Hz 1 1000 Hz 1 1500 Hz 12000 Hz 12500 Hz 3000 Hz 3500 Hz 14000 Hz]
200 kW
0.95
065 A
3.0 V
0.50
1.00
1.3E-03
0.5 J/pul
0.4J/pul
0.4J/pul
3000
3000
3000
6.7E-04
0.19 kW
03.8 kW
04.0 kW
02.0 kW
00.3 kW
10.3 kW
94.9%
200 kW
0.95
065 A
3.0 V
0.50
1.00
1.OE-03
0.5J/pul
0.4J/pul
0.4J/pul
4000
4000
4000
5.OE-04
0.19 kW
05.0 kW
04.0 kW
02.0 kW
00.3 kW
11.5 kW
94.2%
200 kW
0.95
065 A
3.0 V
0.50
1.00
6.7E-04
0.5 J/pul
0.4 J/pul
0.4 J/pul
6000
6000
6000
3.3E-04
0.19 kW
07.5 kW
04.0 kW
02.0 kW
00.3 kW
14.0 kW
93.0%
200 kW
0.95
065 A
3.0 V
0.50
1.00
5.OE-04
0.5 J/pul
0.4 J/pul
0.4 J/pul
8000
8000
8000
2.5E-04
0.19 kW
10.0 kW
04.0 kW
02.0 kW
00.3 kW
16.5 kW
91.7%
200 kW
0.95
065 A
3.0 V
0.50
1.00
4.OE-04
0.5J/pul
0.4J/pul
0.4J/pul
10000
10000
10000
2.OE-04
0.19 kW
12.5 kW
04.0 kW
02.0 kW
00.3 kW
19.0 kW
90.5%
Table 12 6.5 kV Power Converter Efficiency
Page 52
200 kW
0.95
065 A
3.0 V
0.50
1.00
3.3E-04
0.5J/pul
0.4J/pul
0.4J/pul
12000
12000
12000
1.7E-04
0.19 kW
15.0 kW
04.0 kW
02.0 kW
00.3 kW
21.5 kW
89.2%
200 kW
0.95
065 A
3.0 V
0.50
1.00
2.9E-04
0.5 J/pul
0.4J/pul
0.4J/pul
14000
14000
14000
1.4E-04
0.19 kW
17.5 kW
04.0 kW
02.0 kW
00.3 kW
24.0 kW
88.0%
200 kW
0.95
065 A
3.0 V
0.50
1.00
2.5E-04
0.5 J/pul
0.4J/pul
0.4J/pul
16000
16000
16000
1.3E-04
0.19 kW
20.0 kW
04.0 kW
02.0 kW
00.3 kW
26.5 kW
86.7%
10.0 kV
Power
power factor
Conduction Current
Forward Drop
D
F
T
Turn-on loss
Turn-off loss
Reverse recoveries
Non-ZVS On Trans
Non-ZVS Off Trans
Reverse recoveries
Cond. Time/cycle
Conduction diss.
Switching Diss.
Transformer Loss
Filter Loss
Diode Loss
Power Loss
Switching Efficiency
500 Hz
J
200 kW
0.95
042 A
3.OV
0.50
1.00
2.OE-03
0.7J/pul
0.7 J/pul
0.6 J/pul
2000
2000
2000
1.0E-03
0.13 kW
03.9 kW
04.0 kW
02.0 kW
00.3 kW
10.3 kW
94.8%
[ 1500 Hz
750 Hz
1000 Hz
200 kW
0.95
042 A
3.OV
0.50
1.00
1.3E-03
0.7J/pul
0.7 J/pul
0.6J/pul
3000
3000
3000
6.7E-04
0.13 kW
05.9 kW
04.0 kW
02.0 kW
00.3 kW
12.3 kW
93.9%
200 kW 200 kW 200 kW
0.95
0.95
0.95
042 A
042 A
042 A
3.0 V
3.0 V
3.0 V
0.50
0.50
0.50
1.00
1.00
1.00
1.OE-03 6.7E-04 5.OE-04
0.7J/pul 0.7 J/pul 0.7 J/pul
0.7J/pul 0.7 J/pul 0.7 J/pul
0.6J/pul 0.6J/pul 0.6 J/pul
4000
6000
8000
4000
6000
8000
4000
6000
8000
5.OE-04 3.3E-04 2.5E-04
0.13 kW 0.13 kW 0.13 kW
07.8 kW 11.7 kW 15.6 kW
04.0 kW 04.0 kW 04.0 kW
02.0 kW 02.0 kW 02.0 kW
00.3 kW 00.3 kW 00.3 kW
14.2 kW 18.1 kW 22.0 kW
92.9%
90.9%
89.0%
2000 Hz 1_2500 Hz_[3000 Hz
200 kW
0.95
042 A
3.0 V
0.50
1.00
4.0E-04
0.7J/pul
0.7 J/pul
0.6 J/pul
10000
10000
10000
2.OE-04
0.13 kW
19.5 kW
04.0 kW
02.0 kW
00.3 kW
25.9 kW
87.0%
J3500Hz
200 kW
0.95
042 A
3.0 V
0.50
1.00
3.3E-04
0.7 J/pul
0.7J/pul
0.6J/pul
12000
12000
12000
1.7E-04
0.13 kW
23.4 kW
04.0 kW
02.0 kW
00.3 kW
29.8 kW
85.1%
200 kW
0.95
042 A
3.0 V
0.50
1.00
2.9E-04
0.7 J/pul
0.7 J/pul
0.6J/pul
14000
14000
14000
1.4E-04
0.13 kW
27.3 kW
04.0 kW
02.0 kW
00.3 kW
33.7 kW
83.1%
4000Hz
200 kW
0.95
042 A
3.0 V
0.50
1.00
2.5E-04
0.7 J/pul
0.7 J/pul
0.6J/pul
16000
16000
16000
1.3E-04
0.13 kW
31.2 kW
04.0 kW
02.0 kW
00.3 kW
37.6 kW
81.2%
Table 13 10kV Power Converter Efficiency
These revised selections yield efficiencies of 94.4%, 91.7%, and 89% for the 3.3kV,
6.5kV, and 10kV power converters respectively. As a result, it appears unlikely that this
type of power converter layout would be able to achieve 96.5% efficiency without adding
circuitry to minimize the switching losses even further.
3.3.6 FILTERS
After a working circuit was produced, the filters were designed and tested. The
initial input and output filter were assumed to be second order LC filters. It was
rationalized that a capacitor at the input and output would be required to maintain the
voltage at the desired values. For space and reliability considerations, only a small
capacitor was considered for this application. The use of small filtering components was
also a very big concern for the transient response and required additional circuitry to limit
the transients to the allowable values.
The voltage ripple for this circuit was analyzed across half a cycle. The positive and
negative half cycles are identical as far as the input and output filters are concerned. As a
Page 53
result of the waveform developed for this circuit, the apparent frequency to the input and
output filters was two times the actual switching frequency. This helped, as postulated
previously, to decrease the size of the filtering components by minimizing the length of
time between pulses to the output filter and load.
Voltage and current ripple were used to determine the initial size of the inductors
and capacitors in the filter circuits. For a first order approximation, all current ripple in the
inductor was assumed to go into the capacitor [5]. The inductance, Loi, was calculated to
be at least 31 mH to maintain the output current ripple less than 4A and thus keep the
output voltage ripple to less than 20V for the largest load of 5Q. Equation (6) outlines the
method used to arrive at the inductance, Loi. The larger duty ratio (smallest input voltage)
was used for the set up below because this would give the larger variation in inductor
current and capacitor voltage.
di
VL = L-dt
1
diL = -VLdt
L
==>
1
(V
L (Di
A=
1
- DV 2 )T
1f
-
AiL
=*
DT
V2 )dt
L
-V
2 )T
-
AiL
___
_______
=; L >
T(1-D)V 2
=
_
-
1
-(V
L
1
-
V 2 )(D)T
31mH
(6)
Using this same methodology, the minimum value of the capacitance was
determined. The maximum load of 200kW was also used for the voltage ripple calculation
to attain the worst case result. Equation (7) shows the calculation used to determine the
starting value for the capacitance.
dv
IC = C->
dt
Av
C
1
dve
C
(D1, - DI )T
1
LDT
=
C
- 12 )dt -> Ave 5 - (I
C
ve =-j(I1
C 0
)T
=(-DC
T(1-D)1 2
AVC
Page 54
= 1. 25 mF
-2)(D)T
(7)
250
lll
2000
lout
Overshoot
1500
1000
500
0
0.1
0.2
0.3
04
05
lime (sec)
0.6
0.7
0.8
0.9
Figure 29 Initial test of output filter
The testing screen adopted for this project shows the output voltage versus time,
but it also shows a variety of control signals and other measurements as well. Starting
from the top in Figure 29, the graph shows the output current going to the load shifted by
2500 to separate the graphs. Next is a digital indicator (ON or OFF) of whether the circuit
has reached steady state as defined by the logic of the shunt control circuit. The curve at
2000 is the slope of the output voltage. Below that is the digital control signal for the shunt
resistor to limit overshoot. The next curve offset at 1500 is the ON or OFF state for the
switch to discharge the capacitor during load increases. At 1250 is the ON or OFF state for
the switch which charges the capacitor through a 50 resistor. The dashed line is the shunt
capacitor voltage which tracks the output voltage. The green line at 1000 is the desired
voltage from the regulator. Finally, the red signal at the bottom is the rectified voltage at
the output side of the rectifier.
The output filter components from above were tested to determine the validity of
the approximations. As can be seen from Figure 29, the transient response is slow and
includes significant overshoot. In addition, due to the very low switching frequency, the
Page 55
converter is not very controllable. In a typical low power converter, the switching
frequency would be an order of magnitude greater or more. This would allow more fidelity
in the control of the switched output. As a result, effort was made on shunt components to
attempt to utilize smaller capacitors and inductors while increasing the speed of the
transient response and controllability. Figure 30 shows the resulting shunt circuit.
The shunt circuit is designed to minimize transient effects on the output voltage
while minimizing the size of the filtering components and increasing response speed. The
capacitor is initially charged during the module start-up. As the converter is run in steady
state, the capacitor voltage will be checked periodically to ensure it remains at the
regulated voltage. During a large load increase, the capacitor can be switched into the
circuit and supply its energy to the load to maintain the output voltage while the output
filter is charged to its new required value. During a large load decrease, the capacitor can
also be charged slightly to help limit the increase in output voltage. The shunt circuit also
contains a 5 Q resistor. This resistor can be switched in to limit the maximum increase in
output voltage during transients from high to low loads. The resistor dissipates the stored
energy in the output filter preventing exceeding the output voltage limits.
Page 56
Charge
QDis-Charge
CP
C D O
~Over-shoot
-
A a
+ Load
+input
S5
S6
S7
Ro
Vcap
Co2
Rol
I measurement
-
A
input
-Load
Figure 30 Shunt circuit
Both of these shunt branches are designed to only be used during transients. As a
result, it is expected that they will have a minimum effect on the overall efficiency of the
converter. Additionally, through trial and error, this circuit allowed a drastic decrease in
the size of the output filter inductor and capacitor; although, it required the addition of
three switches, a capacitor and two resistors.
The Shunt Control circuit is shown in Figure 31. This provides the logic sequences
to control the shunt circuit to limit large transients to within the required specifications.
Page 57
Figure 31 Shunt Control Circuit
The output filter was adjusted to a value which resulted in an average steady state
ripple of approximately 20V. This resulted in an inductor of 16mH and a capacitor of
100,pF. Another factor that allowed the inductance and capacitance to be much lower than
predicted was the variable duty ratio, D. The approximation assumed a constant D,
although the actual duty ratio varies dynamically to control the output voltage. Although
these components are not small for the voltages under consideration, the components are
available. The steady state ripple voltage is distorted by the adjustments of the PCON as
shown in Figure 32.
Page 58
1040-
10200
1000_
Vout
Vrec
Vreg
Charge
Vcap
Slope
Discharge
ss
lout
Oershoot
980
960-
940-
920-
9008800.046
0.048
0.05
0.052
0.054
Time(sec)
0.056
0.058
0.06
Figure 32 Full Load Steady state ripple
A 5 Q resistor was used for both of the resistors to ensure that the maximum output
current would be limited to full load. The resistor was based on a Power Wire Wound
Resistor from MF Power Resistor Ltd. A 20kW resistor was chosen to reduce the overall
size of the resistor. The manufacturer stated that a 10 times rated power is acceptable for
short periods less than 5 seconds. For the operation of this circuit, the resistor will be used
for very short periods of time (much less than 1 second) and only during transients.
The shunt circuit was then adjusted to limit the total voltage swing during transients
as discussed above. The minimum capacitor value was determined to be 430ptF. A variety
of transients were run, and a summary is shown below in Figure 33 and Figure 34. The
transients were selected based on the limits given in Table 1. The first set of transients
started with a 0% to 33% transient followed by the 33% to 100% transient and finally a
100% to 0% transient. The second transient set, started at 0.03sec, was a 0 to 50%
transient. At 0.05sec, the load was increased to 100%, and then at 0.07sec, the load was
decreased to 0%. During all of these transients, the output voltage was maintained
between the limits of +8.5% and -16.5%.
Page 59
lout
SS
Slope
Owrshoot
Discharge
Charge
Vreg
Vout
Vcap
Vreg
NNW
1000-
III
/ I
500
0
0.01
0.04
Time (sec)
0.03
0.02
0.05
0.06
0.08
0.07
Figure 33 Transient response 33% to 100% at 0.04 seconds
lout
G9 SS
Slope
-Overshoot
Discharge
Charge
Vreg
Vout
Vcap
Vreg
2500;M1
-
~
mAD
2000 1-
1000 H-
)0
0
0.01
002
003
0.04
lime (sec)
0.05
0.06
Figure 34 Transient response 0% to 50% at 0.04 seconds
Page 60
007
008
The input filter was not designed because there was limited information about the
electrical sources that would be seen in the systems. Instead, it was assumed that the same
size capacitor would be used on the input as the output. For a specific supply system, an
optimized input filter could be designed, but it is not expected to significantly affect the
efficiency or operation of the converter.
3.3.7 CAPACITOR SELECTION
The capacitors chosen for the converter come from the General Atomics Electronic
Systems [20]. Three capacitors are required for this circuit: 1 input filter, 1 output filter,
and 1 shunt circuit. The capacitor values and pertinent data are summarized in the table
below.
Weight
H
1W
IL
Component Value
74.9 kg
1OOuF 203.2mm 355.6mm 612.1mm
Cil
74.9 kg
100 uF 203.2 mm 355.6 mm 612.1 mm
Col
Cshunt
430uF 184.2mm 355.6mm 693.4 mm
Table 14 Capacitor Properties
77.2 kg
The capacitors selected are of a mixed dielectric type using either polyester or
polypropylene film in combination with paper in the dielectric. This type of dielectric is
designed for higher repetition rates based on the manufacturer's literature. Figure 35
shows the operational life of the capacitor versus the charge voltage [20]. On the
secondary side of the power converter, we are running at less than 10% of the rated
voltage which should yield a very large number of cycles. The figure shows greater than
104 times the 3x10 3 cycles the capacitors are rated for. Additionally, the capacitors are not
cycled more than a small amount for the ripple voltage. While no more complex analysis
was run on these capacitors, this amount of information satisfied the author that these
capacitor specifications were representative of the actual type of capacitor that would be
used in this type of power supply.
Page 61
1ol'
CHARGE VOLTAGE VS. LIFE
S
2
103
10
2
15
.2
20% 30 40 So 60 70 80 90 100 110 120
Charge Voltage % of Rated Voltage
Figure 35 Life Expectancy Multiplier for General Atomics Type Ccapacitor [20]
3.3.8 WIRE SELECTION
Due to the high voltages involved in this design, special care was taken to ensure
that the wire for the inductors and capacitors was chosen correctly. Most wire is
commonly rated to 600V. The voltages in this application are approximately 3kV to 10kV
on the primary side and 1kV on the secondary side of the transformer. Also, to get a good
estimate of the properties of the inductors and transformer, the size wire must be known to
account for the size and weight of the windings and also core material.
The 600V rating for wire is based on the insulation rating. Higher voltages have a
chance of arcing through the insulation creating a hazard and causing damage to the
equipment. Thicker materials provide better insulation, but the windings would then grow
in size as well. A common means to insulate wire in magnetic windings is applying a
special varnish to the wires. By searching available products, an insulation value of 3000
V/mil was used. This is for dry windings. If the winding is wetted, the insulation value
drops to 1500 V/mil. These values came from Red Insulating Varnish 4228 on MG
Page 62
Chemicals website. For this level of design analysis, these numbers were deemed
sufficient.
The conduction of the conductor is also critical for wire size. The primary effects
analyzed were the skin depth of the conductor and the area required to carry current on
the primary side and secondary side. The Skin effect causes the current to flow through an
area of the conductor near the surface. As frequency increases, the area becomes smaller
and smaller as the current flows only through an area nearer and nearer to the surface.
Equation 8 was used to determine the skin depth, 6 eU [5]:
Cu
=
(8)
2
,
(LIcuOcu
where w is the frequency, peU is the permeability of copper (1.256x10- 6 H/m), and ~cu is the
conductivity of copper (5.69x10 7 s/n). This yields a skin depth of 1.5mm for the copper
wire. Table 15 summarizes the size of the wire on the primary side of the transformer for
the three converter variations as well as the secondary side which is common to all three.
ZVotlage
Primary
Primary
Primary
Secondary
ICurrent
3.3 kV
6.5 kV
10.0 kV
1.0 kV
128A
65A
42 A
200 A
Area Req Cond.
0.256cm2
0.130cm2
0.084 cm2
0.400 cm2
Diam
Final Diam [Insulation
2.9mm
2.0mm
1.6mm
3.6mm
2.9mm
2.0mm
1.6mm
3.6mmj
0.03mm
0.06mm
0.09mm
0.01mm
Iwire
2.89mm
2.09mm
1.73mm
3.58mm
jStandard Wire Diam
9 AWG
11AWG
13AWG
7AWG
2.94 mm
2.15mm
1.82mm
3.59mm
Table 15 Wire size for Primary and Secondary
As shown above, the only wire that showed an issue with the skin depth was the
secondary side windings in the transformer and inductor filter. The electricity does not
flow through the center portion of the conductor, so a tenth of a millimeter added to the
conductor diameter alleviated this concern. The wire sizes used for these calculations were
based on dry insulation as it is the expected choice for this type of installation.
Additionally, the required wire size was rounded up to the nearest standard wire size
which provides a small margin in addition to simplifying the construction.
3.3.9 INDUCTOR DESIGN
This section details the design and selection of the filter inductors. The design of
these components was completed for several reasons. First, having an actual design of the
component would allow at least a first order estimation of the parasitic losses of the
Page 63
component to increase the fidelity of the model. Second, to provide Naval Architecture
parameters, the weight and volume for the components is required. An accurate weight
estimate could only be found by referencing an actual component. Last, designing an actual
component provided verification that the selected component parameters could actually be
obtained.
These inductors, as with most inductors, are custom designed for this application.
The inductor design procedure from Fundamentals of Power Electronics [15] was
referenced heavily to get a rough estimate of the inductor properties. To initiate the
inductor design, we start by determining the size of wire which will be required for the
windings. The calculations in the previous section show the wire sizes required for the
primary and secondary side for the different variations.
Iron was chosen as the core material and has a maximum allowable flux density of
1.2T. By calculating the saturation flux in the core by Equation (9), the minimum number
of turns on the inductor can then be found to avoid saturating the core. The area of the
toroid was calculated and the dimensions were adjusted so that the required number of
turns of wire could be wound. No toroids could be found of this size which prevented the
use of AL which is essentially a manufacturers specification for the inductance per turn. As
a result, the core parameters were chosen as shown in the equations below. Table 16 lists
the name and units for each variable used in the inductor calculations.
Ac
Aw
Ka
Core cross-sectional area [cm 2]
Wire conductor area [cm 2]
Maximum core flux density [T]
Magnetic Core figure of merrit
Ku
Winding fill factor
Bmax
Imax
L
MLT
p
R
WA
K >
Peak winding current [A]
Inductance [H]
Mean-length per turn [cm]
Wire resistivity [Q/cm]
Winding resistance [Q]
Window area [cm 2]
Table 16 Inductor equation symbols
108 [cm 5 ]
(9)
Page 64
Kg =
AW
l1
*
=
"104
2
1
A-
AL
A
[M]
(1
(12)
[unpH]
Bmax
L = ALN
N
(10)
[cmS]
-
A. !
R =
2 10- 9
(13)
[H]
Umax
10 4
(14)
KuWA
[Cm2]
(15)
pNMLT
[l
(16)
Bmax Ac
Aw,actual
The summary of this iterative design process is shown in Table 17. The filter
inductors are rather large at 16mH each, but a toroidal core with an air gap was initially
tried. The equations above were stepped through. The core properties and the size of the
wire required were calculated first. Then, the resistance of the winding was estimated.
This resistance was used in Equation (9) above to give an estimate for the core parameter,
Kg. Next a winding fill factor was chosen. In this case, 75% was used because of the
relatively thin varnish insulation chosen. These parameters yielded a required core
parameter. By adjusting the size of the core and the window area, the actual core
parameter was adjusted to be approximately equal to the required value.
Using the other relationships given in the equations, the gap length was calculated.
This allowed a calculation of the inductance. At this stage, the primary inductor
parameters were fairly accurately estimated. The weight and volume of the core were
calculated as well as the actual resistance of the winding.
Page 65
Voltage
Power
Current (max avg)
Wire size (diameter)
Resistivity
Core Flux Density
Inductance
Winding Resistance
Winding Fill Factor
3.2E-5 Ohm/cm
5.OE-5 Ohm/cm
8.OE-5 Ohm/cm
1.2T
16 mH
0.2 Ohm
0.75
1.2T
16 mH
0.2 Ohm
0.75
1.2 T
16 mH
0.2 Ohm
0.75
Kg required
13 756 cm5
05 644 cm5
03 788 cm5
Height
Width
MLT
Cross Sectional Area
Window Area (WA)
9.7 cm
7.0 cm
33.4 cm
67.9 cm2
100 cm2
8.0 cm
5.0 cm
26.0 cm
40.0 cm2
100 cm2
13 804 cm5
06 154 cm5
Kg of the core
Length of Gap
AL
Turns
Thickest Wire
Required Window Diam
DiameterofToroid
InductorDiameter
Inductor Height
InductorVolume
Core Weight
3.3 kV
6.5 kV
200 kW
200 kW
61 A
31 A
2.94 mm 2.15 mm
10.0 kV
200 kW
20 A
1.82 mm
1.0 kV
200 kW
200 A
3.59 mm
2.OE-5 Ohm/cm
1.2 T
16 mH
0.3 Ohm
0.75
62 894 cm5
7.5 cm
14.8 cm
4.0 cm
12.5 cm
23.0 cm
54.6 cm
30.0 cm2 185.0 cm2
100 cm2 100 cm2
03 913 cm5
62 683 cm5
Actual Winding Resistance
0.8 cm
0.3 cm
0.2 cm
3.0 cm
1130uH
1521uH
2025uH
770uH
119 turns 103 turns 89turns 144 turns
0.45 cm
0.48 cm
0.52 cm
0.41 cm
5.64 cm
5.64 cm
5.64 cm
5.64 cm
19.6cm
15.6cm
13.6cm
30.6cm
20.2cm
16.1cm
14.0cm
31.2cm
10.8 cm
8.9 cm
8.3 cm
15.9 cm
0.0139 m3 0.0072 m3 0.0051 m3 0.0485 m3
24.10 kg 11.56 kg
7.68 kg 99.24 kg
0.13 Ohm 0.13 Ohm 0.16 Ohm 0.16 Ohm
Per Unit Wire Weight
5.9E-4 kg/cm 3.7 E-4 kg/cm 2.3 E-4 kg/cm 9.4E-4 kg/cm
WindingWeight
Total Weight
2.3 kg
26.4 kg
1.0 kg
12.5 kg
0.5 kg
8.2 kg
7.4 kg
106.6 kg
Table 17 Inductor Design Summary
3.3.10 HIGH FREQUENCY TRANSFORMER
Determining the practical frequency range of the transformer is the first priority of
the power converter design. The design process will be iterative, but it is still important to
start as near as possible to the ideal design point. The higher switching frequency of the
converter will impact not only the size of the filter components, but it will also affect the
size of the transformer. Both of these sets of components will decrease in size as the
switching frequency increases. However, there are limits on the frequency such as the
efficiency of the switch and even the maximum frequency capability of the switch.
Page 66
It was initially expected that the transformer would be the limiting component on
the switching frequency of the converter along with the efficiency tradeoffs. However, F. K.
Wong showed that the development of new core materials and winding techniques have
increased the switching frequency of high frequency transformers [6]. The data shows that
operation through the megahertz range is possible although maybe not for the size range
required in this project. This discovery moved the frequency limit to the power switch for
the upper switching frequency.
The eddy current losses of the core material transformer are not the only
consideration as frequency increases. The windings also become a significant source of
losses due to the skin effect and proximity effect. The skin effect can be minimized by using
litz wire. Litz wire is made up of many strands vice a single conductor. The increased
surface area tends to reduce the resistance increase due to the higher frequency. Wong
presents a common approach called sandwich winding, shown in Figure 36, which can
reduce the proximity effect. The principle is to eliminate large areas with a uniform
magnetic field which can cause eddy-current losses in the conductors. Proximity losses are
typically more significant than skin effect losses for high frequency transformers.
PHs
PSPPsPP
Figure 36 Sandwich windings showing primary and secondary [6]
After doing a quick review on the depth of information available for transformer
design principles, it was decided that a more basic approach would be appropriate for this
project. An entire thesis could be written on transformer design, but in this case, only a
good approximation of the transformer's size, weight, and efficiency were required. As a
result, a generic transformer design method was sought that would fit into the overall time
constraints of this project.
Page 67
The turns ratio of the transformer will be adjusted so that all of the voltage drop
occurs across the transformer. This results in the direct converter having an equivalent
conversion ratio of unity which reduces the switch stresses [5]. In this case, the
transformer ratio was adjusted to determine not only the minimum switch stresses but
also to allow maximum controllability. As can be seen from Figures 36-38, as the
transformer lowers the secondary voltage, the controller cannot adequately minimize the
load increase transients. As a result of these tests, a run with 4000 V on the secondary side
was made. As can be seen in Figure 40, the transient response is a little better, but the
steady state ripple is now made worse. This is likely due to the amount of energy that is
required to be stored in the filter to maintain the output voltage with a very low duty ratio.
fi
Vout
Vrec
Vreg
Charge
2500
-_-_Vcap
Discharge
s
2000-
ut
1500
1000
500
0
0.01
0.02
0.03
0.04
Time (sec)
0.05
0.06
Figure 37 Simulation with 3000V secondary voltage
Page 68
0.07
0.08
Vout
Vrec
Vreg
Charge
Vcap
Slope
Discharge
C'S
lout
Overshoot
,/""- ; -
I
-
.
4"
bI hF
DHk
II
!]
0V
0
0.03
10.02
0.04
lime (sec)
0.05
0.06
Figure 38 Simulation with 2000V secondary voltage
19)
Vout
Vrec
Vreg
9311IIIIINAM==
Charge
Vcap
Slope
Discharge
SS
lout
Owrs hoot
500
0
F01
002
003
004
ime (sec)
005
006
Figure 39 Simulation with 1500V secondary voltage
Page 69
007
008
Mlm
Vout
Vrec
Vreg
Charge
-- Vcap
Slope
Discharge
m
oSS
lout
- -Owrshoot
1000
01IL
0
IL
0.01
0.02
0.03
0.04
0.05
0.06
0.07
0.08
Time (sec)
Figure 40 Simulation with 4000V on the secondary side
As a result of the above experiments, 3000V was chosen for the peak secondary
voltage. This adds further weight to the earlier supposition that the 3.3kV converter may
not need a transformer since this value is close to the 3kV selected here. However, for this
project, the transformers are being maintained as a means of electrical isolation. In
practice for a 3.3kV system, the transformer might not be used to increase the efficiency by
nearly 2%. To decrease the voltage per the above discussion, Table 18 below shows the
required turns ratios.
It should be kept in mind that a secondary voltage below 3kV might be
advantageous if the switch stresses need to be reduced. The peak currents will be higher
with the lower duty ratio required for a 3kV secondary. From the above experiments, it
appears that a secondary voltage down to approximately 2kV will still produce an adequate
transient response.
Page 70
Voltage ITurns ratio for 3000 V secondary
1.1:1
3.3 kV
6.5:3
10:3
6.5 kV
10.0 kV
Table 18 Turns ratios
Several procedures were investigated for the design of the transformer. These
procedures came from a Texas Instruments technical publication and Fundamentals of
Power Electronics. However, after extensive experimentation, neither of these procedures
was adequate to design a transformer of this size and a relatively low frequency of 2 kHz.
On the other hand, typical power transformers were mostly designed to operate at 60Hz
with a few for primarily aircraft operations at 400 Hz. Due to the wide breadth of subjects
covered in this project, it was deemed that sufficient time was not available to attempt a
transformer design from first principles.
With these constraints, there were two thoughts on how to proceed. The first
thought was to attempt to define a few constraints for the transformer design such as
ensuring that the core did not saturate. However, the transformer must also be efficient,
and there was not time to calculate the efficiency and refine the transformer design.
The second approach required a re-examination of the overall purpose of the
project. This project is intended to allow simulations of the system and to provide
parametric data for an early stage ship-design tool. An early stage ship design has a fair
margin of uncertainty, and it was decided that an estimation of the efficiency, weight, and
volume would be close enough to satisfy these needs. As a result, the transformer weight
and volume were assumed to be the same as the low voltage inductor at 0.0485 m 3 and 110
kg. It is expected that a custom-designed transformer would be slightly smaller and lighter.
The efficiency of the transformer consists of some base core losses which are always
present as long as the transformer windings are energized. Additionally, as the
transformer power is increased, the losses in the windings increase. An amount of 0.5% of
transformer rating was taken as the constant loss with an additional 1.5% of transformer
power as the variable loss. This equates to 2% at full load yielding 98% efficiency.
Page 71
3.3.11 CONVERTER SUMMARY
The converter design worked very well and met all of the requirements from the
SBIR referenced earlier except for efficiency. While the efficiency of this design is lower
than desired, it is still functional. Additionally, this converter is only envisioned for use on
the ship service side of the electrical distribution system which is a small fraction of the
ship's total power. As a result, the inefficiency may be acceptable, especially as an early
stage design comparison.
As the design above progressed, it was noticed that the 3.3kV variant could have
been designed without the transformer at an increased efficiency. Not only would this have
saved the -2% loss of the transformer, but fewer switches and diodes would have been
required. This likely would have resulted in an efficiency of 97% or higher for this
converter variant. In addition, it would be worth investigating whether the 6.5kV version
could be simplified as well. However, for the 6.5kV variant, it is much harder to estimate
whether it would return a higher efficiency due to the much higher peak currents.
As stated earlier, there are many different types of converter topologies, and this
topology might not be the ideal method for this application. However, it is believed that the
physical attributes and performance of this power converter would represent other power
converter designs. This should allow the use of this one topology as a tool for analyzing the
viability of different ship electrical distribution systems regardless of the exact
configuration that might eventually be built for a future ship. It is expected that there will
be some error, but that these errors will not be significant enough to sway a designer's
choice in ship configuration selection.
3.4 LOAD SHARING
The adopted PCM-4 design requires that the system be reconfigurable for many
different size loads. The system concept from SatCon provides a versatile system which
can be used in many different ship concepts without redesign and with enhanced reliability
due to the redundant sub-modules. However, now the circuits must function so that not
only do the PCM-4's have to be able to operate in parallel for some configuration, but also,
the sub-modules have to operate in parallel within a single PCM-4. This section explains
Page 72
the methodology used to share the loads, the voltage regulator, which realizes the load
sharing methodology, and the load sharing operational verification.
3.4.1 DROOP
A common method of sharing loads is a linear droop. For a DC application, a voltage
versus real power, or more accurately, current, was chosen. The idea is loading is shared
equally as a percentage of rating between multiple machines in parallel. If the load
changes, each machine in parallel has a proportional change in its load. This leads to a
predictable and usually small change in voltage without large fluctuations or other
undesirable consequences. The figure below shows two generators in parallel.
V
VNL
VoP
Module 2
Module 1
VFL
P [kW]
<
>
-
Load
Figure 41 Generators in parallel operation
The case shown above is for two modules of unequal power rating. The load is the
total amount shown by the scope of the arrows around the load. By observing the
geometry of the problem, one could imagine that if the no-load voltage, VNL' of module 1 is
increased, module 1 will take on a larger portion of the load and module 2's load will
decrease. In addition, the operating voltage would increase. The VNL of the module would
be the typical parameter adjusted by a controller or an operator depending on the
Page 73
operation. It can be imagined that by adjusting the VNL on multiple machines operating in
parallel, the distribution of the load could be set as desired.
Droop=
vNLVNL
X
(17)
%
Voltage droop is the amount of voltage decrease with a change in real power as
shown in Equation (17). A voltage droop of 5% was chosen for this application. This
appeared to be a large enough value that the generators would be able to stably share the
load. In addition, 5% was small enough to limit the amount of output voltage variation.
3.4.2 VOLTAGE REGULATOR
The voltage regulator's purpose was to maintain the output voltage along the linear
droop line as load changes. The voltage regulator takes an input, SS V from Figure 42, and
uses this value as the no-load voltage. SS V comes from the PCON module which allows the
voltage regulator's no-load voltage to be adjusted, but the voltage regulator's operation is
independent of where no-load voltage is set.
Implementation of the voltage regulator in Simulink required the use of voltage and
current sensing blocks. A PID controller was used to translate the difference in regulated
voltage and output voltage into a control signal, duty ratio D. The voltage sensing block
was used to measure output voltage. The current sensing block measured output current
which was multiplied by a factor of 'A. The factor of 'A changed the current into a voltage
where the full load current of 200A then equaled 50V which is 5% of the operating voltage,
providing the 5% voltage droop desired.
Page 74
Vout
To Workspace
L-],-
Vload
Vm1
Vload -
+
lout
4PID
D
lout
PID
Charge
Vreg oVreg
Dis-Charge
Vout
To Workspace3
Charge
Dis-Charge
CD+-
Gain
Over-shoot
Vcap
Over-shoot
SS V
Shunt Controller
Vcap_
To Workspace1
Vm2
Figure 42 Voltage Regulator
By setting the no-load voltage to a constant 100OVDC, the voltage regulator was
tested with several loads over the operating range. As can be seen in Figure 43, the
regulator produces a very good droop curve. This graph was made by plotting only the
regulator voltage from the main troubleshooting graph. The load was changed from 0 to
30% to 50% to 100% load. The plot is regulator voltage versus time, but the transients
were initiated at the corresponding percentage of the simulation time to yield a correlated
plot. The line shows where the set-points would be for all loading conditions.
Page 75
995
Actual
990
Regulated
Voltage Test
985
Points
I 980
C))
4-'
o 975
970
965
960
-
Regulated
Voltage
955
950
Predicted
-
0
0
0.001 0.002 0.003 0.004 0.005 0.006 0.007 0.008 0.009
30
50
0.01
100
Time[s] / Power[%]
Figure 43 Regulator functional test
3.4.3 LOAD SHARING VERIFICATION
The final test for the voltage regulator involved operating while sharing load. The
initial set up to share loads between modules requires that the modules be connected to
the same load. Additionally, each PCM-4 has only one PCON which controls all of the
modules. All of the modules were driven by the same clock to control the timing of the
power switches.
The verification started with two modules connected to the load module. The
individual modules contain their own voltage controllers and voltage regulators which
control the converter operation as described above. The clock signal was moved external
to the converter so that one clock could be used for multiple modules. Finally, the no-load
voltage control signal was routed to both modules as shown in Figure 44.
Page 76
Continuous
-___SS
po we rg u i
V
1000VDC +
100OVDC
PCON
0 SS V
Conni
Clock
1VDC
CLK
MVDC+
MVDC
I_ -
_
__
i
100OVDC-
Can2
MVDCPower Converter
Load Str cture
SS V
100OVDC
+
CLK
MVDC+
100OVDC
MVDCPower Converter1
Figure 44 Parallel module operational block diagram
The circuit was run using the required transients. Initially, half loads were used to
test that the converters shared power using 15%, 25% and 50% and the transients 0% to
50%, 30% to 100%, and 100% to 0%. Under half load conditions, Figure 45 demonstrates
that the converter is stable. In fact, perhaps as expected, the smaller transients did not
even require the use of the shunt circuit except for the start-up transient.
Page 77
Vout
Vrec
Vreg
Charge
Vcap
Slope
Discharge
2500
0
SS
[out
Overshoot
-I-
1000 |
500
Jill
01
0.05
0.04
Time (sec)
0 06
0.07
0.OE
Figure 45 Parallel operation at half load
Vout
Vrec
Vreg
Charge
Vcap
2500v.
am
-
c~
-
U
m
mSlope
Discharge
C
2000
SS
lout
Overshoot
1500 -
1000 [-
500
0
0.09
0
Time (sec)
Figure 46 Parallel operation at full load
Page 78
0.1
The operation under full loads as shown in Figure 46 was also stable. These tests
were used to verify the operation of the converter for loads using more than one module.
As the tests showed, the converter modules operated stably in parallel. However,
another possible improvement to the overall design could come from driving the paralleled
modules by an offset clock signal. If the offset is done correctly, this could result in the
ripple from the two modules cancelling out at the load. As a result, it might be possible to
reduce the filter component sizes.
Another method that might result in a higher efficiency is to have a variable
frequency drive for the power conversion switches. For example, during periods of
constant and small loads the frequency could be lowered resulting in less switching losses.
The output voltage ripple would increase during these periods, but the circuit would have
to be designed to limit the output voltage ripple to an acceptable value. When combined
with operating two modules in parallel, which would be the most likely operation, the
output ripple could conceivably be limited at a much reduced frequency at lower loading
levels.
3.5 POWER CONTROL MODULE
The PCON module is the control center of the PCM-4 and even the ship's service
electrical distribution system. As such, the PCON system must have inputs from across the
ship to know the current state of electrical power loading and generation. The PCON
module must also be able to control the PCM-4 to maintain 1000 VDC output and minimize
the effects of transients on the MVDC distribution bus.
3.5.1 POWER CONTROL MODULE SHIPWIDE CONTROL AND SENSING
The PCON must get information from across the ship's electrical power system.
Then the PCON must use these inputs to determine the correct actions necessary to
minimize the loss of power to ship service loads and especially vital loads. To accomplish
this tasking, the PCON must also control the PCM-1A and PCM-2A modules which are fed
from the PCM-4 output. Using a predetermined hierarchy, these output signals could force
loads to be de-energized starting with the lowest importance load and moving up. The end
Page 79
result is a power control system which is capable of quickly managing the loads for power
generation module (PGM) fluctuations, propulsion motor power demands, and casualties.
3.5.2 PCON MODULE PCM-4 CONTROL
The PCON is responsible for maintaining power to as many ship service loads as
possible. This requires the regulation of PCM-4 1000 VDC output voltage as well as the
coordination of the PCM-4 modules. Each of these modules can be activated as required by
the expected loading, or deactivated when loading levels are low. Additionally, the
activated modules must share load.
The primary purpose of the PCON is to control the PCM-4. The PCON module has a
single output to the PCM-4 power modules. This output signal is in the form of a control
signal D or duty ratio. When D is increased, the PCM-4 will increase the output voltage and
vice versa. A small logic circuit is included with each power module to translate the control
signal D into the proper switch timing to increase or decrease the voltage as required.
The electrical distribution system will have enough excess capacity to allow any ship
service load to be started without overloading the distribution system. When a large load
is started, the PCON will determine whether an additional sub-module should be started to
maintain adequate excess capacity.
The modules will have a voltage droop to share load. This voltage droop will be
initiated within each module to allow PCON to provide a minimum number of control
signals.
3.6 PCON MODULE
Initially, it was planned to produce a control system which would take and evaluate
the ship-wide input described above to control the ship service electrical distribution
system. However, the scope of the project was narrowed to focus more on the power
converter. As a result, the control module was designed with two goals in mind. These
goals were basic functionality and expandability.
As shown in Figure 47 below, the PCON module designed for this application was
simplified compared to the previous discussion. Essentially a proportional controller, the
Page 80
PCON's primary purpose was to maintain the desired operating voltage. This task was
accomplished by measuring the difference between actual voltage and desired voltage. The
difference was multiplied by 0.5 to limit the rate of change before modifying the converter
control signal.
Vavg
To Workspace
+
Saturation
.5Mean
C+
Gain
Vm1
Mean Value
1
z
Unit Delay
0100VDCC3
100VC3
Figure 47 Basic PCON module
The control signal in this case essentially adjusts the no-load voltage. The no-load
voltage would be equivalent to what an operator would adjust on a manually controlled
generator. In this situation, the controller adjusts no-load voltage to maintain a constant
1000 VDC output.
The saturation output filter prevents too high or low of a no-load voltage from being
produced. Therefore, if the no-load voltage is adjusted during a transient, the requested
voltage cannot be above 1085V or below 835V which is the range of variation for a
transient.
To avoid interaction with the voltage regulator, the sampling rate for the PCON was
set to 1/5 of the switching frequency. This allows the PCON to gradually fine tune the
voltage after the initial transient has died down. The interaction was shown to be minimal
during transient simulations. If interaction was shown between the controller and voltage
regulator, an interlock could be placed in the controller to prevent changes to the no-load
voltage during a transient.
One final area for future work on PCON is the ability to bring additional modules
online and shut off unused modules. By using the properties of the voltage regulator and
adjusting no-load voltage, PCON would be able to parallel a module by matching its no-load
Page 81
voltage to the operating voltage before connecting the new module to the bus. Then, the
no-load voltage would be increased gradually until it matched the operating voltage. The
process would work in reverse to shut down an unneeded module. This would eliminate
any transient associated with starting or stopping modules.
3.7 MODEL VERIFICATION
Ideally, as mentioned earlier, it would be best to compare the model to an actual
system, or failing that, compare the model to a physical model to ensure adequate
similarity. Unfortunately, no similar models were identified. However, although the size
and voltage of this converter are not common, the general type of converter is common. As
shown in the above description and analysis, many transients were run. These transients
show a strong correlation to the expected output.
At every step in the design of the converter, great care was taken to try and ensure
that the model behaved as realistically as possible. This included an in-depth analysis of
the model parameters to ensure that the expected values were in fact received. This review
caught several problems such as an uncontrolled sample time. The sample time of one of
the blocks was set at an arbitrarily high value, and the rest of the circuit did not limit its
effects. As a result, the effective switching frequency of the circuit was increased several
fold. The end effect drastically limited the magnitude of the output variations for both
steady state and transient. Of course, this effect was not real, and the switching frequency
was limited after this was noticed. This is just one of the examples of the errors that were
found.
The simulation runs presented earlier show expected and explainable results.
When the load was varied, the resulting transients were consistent with this type of power
converter. Despite the care taken to get to this point and the simulations run, it would still
be desirable to compare the output of this power converter to a physical model should one
be made available.
Page 82
CHAPTER 4 NAVAL ARCHITECTURE PARAMETRIC EXTRACTION
This portion of the project is focused on providing estimates of the key naval
architecture parameters to aid in ship design configuration analysis. These values are used
to compare ship variants and produce a gross approximation of a ship's performance in
reliability and survivability. While the provided values are approximations based on an
early stage design, it is believed that they are accurate enough to allow true comparisons
and, therefore, provide useful analysis.
The main parameters of concern are efficiency and heat load, weight, volume, and
cost. While this design can provide useful data on the first three parameters, it is not
refined enough to offer any cost information. Therefore, efficiency, weight, and volume will
be determined in this section. The data is presented in an easily scalable format to allow
application to a wide variety of ship designs.
4.1 EFFICIENCY AND HEAT LOAD
The efficiencies shown in the table below were calculated based on the explanations
in Section 3. The losses are shown in two groups based on whether they are on the
primary or secondary side of the transformer.
The secondary side of the transformer is the same for all three PCM voltage
variations. The Transformer losses are made up of a constant and a proportional loss as
shown in Equation (18). The rated power (Prated) and operating power (P) are used for this
approximation. The filtering losses are proportional to the load. The Diode losses are also
approximated as proportional to the load based on the low switching frequency.
Transformer Loss = 0.005
X
Prated +
.015 P
(18)
The primary side losses depend on the switch ratings required for each
configuration. The switch conduction loss is proportional to load, but the switching losses
are primarily a function of the switching frequency. These losses are calculated in Section
3.3.4
Page 83
The heat load was estimated to be equal to the losses from the converter. This
provides a slightly conservative estimate of the cooling requirements.
Power
Transformer Loss
Filtering Losses
Diode Losses
20 kW
1.3 kW
0.2 kW
0.03 kW
40 kW
1.6 kW
0.4 kW
0.06 kW
60 kW
1.9 kW
0.6 kW
0.10 kW
80 kW
2.2 kW
0.8 kW
0.13 kW
100 kW 120 kW 140 kW
2.5 kW 2.8 kW
3.1kW
1.0 kW
1.2 kW
1.4 kW
0.16 kW 0.19 kW 0.22 kW
0.04 kW
4.6 kW
6.7 kW
83%
6.7 kW
0.05 kW
4.6 kW
7.3 kW
88%
7.3 kW
0.07 kW
4.6 kW
7.8 kW
90%
7.8 kW
0.09 kW 0.11kW
4.6 kW
4.6 kW
8.4 kW
8.9 kW
92%
93%
8.4 kW
8.9 kW
0.13 kW
4.6 kW
9.5 kW
93%
9.5 kW
0.01 kW 0.02 kW
10.0 kW 10.0 kW
11.5 kW 12.1 kW
42%
70%
11.5kW 12.1kW
0.03 kW
10.0 kW
12.6 kW
79%
12.6kW
0.04 kW
10.0 kW
13.2 kW
84%
13.2 kW
0.01 kW 0.01 kW
15.6kW 15.6kW
17.1 kW 17.7 kW
14%
56%
17.1kW 17.7 kW
0.02 kW
15.6kW
18.2 kW
70%
18.2 kW
0.02 kW
15.6kW
18.8 kW
77%
18.8 kW
3.3kV
160 kW
3.4 kW
1.6 kW
0.26 kW
180 kW 200 kW
3.7 kW
4.0 kW
1.8 kW
2.0 kW
0.29 kW 0.32 kW
0.15 kW
4.6 kW
10.0 kW
94%
10.0 kW
0.16 kW 0.18 kW
4.6 kW 4.6 kW
10.6 kW 11.1kW
94%
94%
10.6 kW 11.1kW
0.05 kW 0.06 kW
10.0 kW 10.0 kW
13.7 kW 14.2 kW
86%
88%
13.7kW 14.2kW
0.06 kW 0.07 kW
10.0 kW 10.0 kW
14.8 kW 15.3 kW
89%
90%
14.8kW 15.3kW
0.08 kW 0.09 kW
10.0 kW 10.0 kW
15.9 kW 16.4 kW
91%
92%
15.9kW 16.4kW
0.03 kW 0.04kW
15.6kW 15.6kW
19.3 kW 19.8 kW
81%
83%
19.3 kW 19.8 kW
0.04 kW
15.6kW
20.4 kW
85%
20.4 kW
0.05 kW
15.6kW
21.4 kW
88%
21.4 kW
I
Switch conduction
Switching Losses
Total Losses
Efficiency
HeatLoad
0.02 kW
4.6 kW
6.2 kW
69%
6.2 kW
I
6.6kV
Switch Conduction
Switching Losses
Total Losses
Efficiency
HeatLoad
10kV
Switch conduction
SwitchingLosses
Total Losses
Efficiency
HeatLoad
0.05 kW
15.6kW
20.9 kW
87%
20.9 kW
0.06 kW
15.6kW
22.0 kW
89%
22.0kW
Table 19 Efficiency and Heat Load for a200 kW PCM at various loads and voltages
When selecting the values to use from the table above, it is important to consider
whether the module will be operating in parallel with other modules. The start-up time for
the modules is relatively short at about 0.025 seconds, but this is still more than enough
time to cause a voltage transient. The start-up transient could be significantly reduced by
keeping the capacitors on an idle module charged, but it is still likely that there would be a
large voltage transient in just charging the inductor.
Operating two modules in parallel will result in lower efficiencies because each
module would be under a lower load, but it would insure that the failure of a single module
would not cause a voltage transient. When choosing the configuration for a particular ship
zone, it is recommended that if a momentary interruption in power is significant, then the
modules should be operated at such a load so that there is at least one module's worth (200
kW) of excess capacity. If the PCMs are connected such that power is shared between more
than one PCM, this methodology could be extended across all of the modules between the
multiple PCMs.
Page 84
4.2 WEIGHT
The weights were calculated based on the main components and the structural
cabinets to house the equipment. Additionally, an additional weight percentage was added
to account for items which were not included in the detail of this design. These items
include wiring, shock absorption equipment, cooling tubes, etc. In future iterations of the
design, this factor should be adjusted to increase the accuracy of the estimate.
To use the weight table below, it is necessary to include one controller and at least
one enclosure with the PCM. Two converter modules will fit in each enclosure with the
exception of the first enclosure which also includes the controller. For example, if a 400
kW module with no redundancy is required, then two enclosures must be selected to house
the controller and two converter modules.
The enclosure cabinet weight was based on a commercial electrical enclosure from
Austin Electrical Enclosures. This enclosure was approximately 0.45 m 3 and weighed
108kg. Extrapolating these characteristics to get an approximate weight for the 1.4 m 3
enclosure required in this example yields a weight of approximately 336kg. Additionally,
25% was added to the weight to approximate the requirements of mounting the cabinet
shipboard.
A weight factor was also added to the components which make up the converter
module. The 20% weight factor was used to account for the connections and mounting
brackets and hardware that would be required. As the design is refined, this 20% increase
in weight could be tuned.
Page 85
Weight
Controller
Enclosure
200kW Module
Input Capacitor
Output Capacitor
Shunt Capacitor
Shunt Resistors
Output Inductor
Transformer
Rectifier Diodes
Weight Factor
3300 V
Switches
Sub-Total
6500 V
Switches
Sub-Total
10000 V
Switches
Sub-Total
200.00 kg
420.00 kg
74.90 kg
74.90 kg
77.20 kg
10.00 kg
106.60 kg
106.60 kg
0.22 kg
20%
6.00 kg
547.70 kg
6.00 kg
547.70 kg
7.00 kg
548.90 kg
Table 20 Weight
The application of the weight estimation to the PCM design begins with adding the
weight of the controller to the weight of the enclosure. The weight of each rating power
converter is also shown in the table above. As with the other parameters, two modules fit
in one enclosure with the exception of the enclosure with the controller which only holds
one converter module.
4.3 VOLUME
The volumes were calculated in a similar manner to the weights. The volume of the
control module was based on the SatCon control module for their AC to DC power
conversion equipment.
A table was developed listing the sizes of each of the main components in the
converter module, and then these volumes were summed up. A packing factor of 40% was
used to estimate the volume required for the module. The 40% was based on visual
Page 86
observations of the SatCon power converter pictures as well as similar types of power
cabinets.
There are two parts to the power converter that are required to arrive at an
operational module. The first required piece is the controller. One controller is required
for each PCM-4. The controller includes any switches required to connect the PCM to the
supply and source lines.
The next required part is a module for the rated voltage required. The sub-total
listed in each variations list is the size of one module of 200kW. This value is multiplied by
the number of modules required to achieve the required power. The enclosure size
envisioned for the converter is 1220x610x1906 mm. This enclosure can house a control
module and a converter module or two converter modules.
[Depth
Width
Height
Volume
Controller
200kW Modu e
Input Capacitor
Output Capacitor
Shunt Capacitor
1220mm
610mm
953mm 0. 7089 m3
203 mm
203 mm
185 mm
356 mm
356 mm
356 mm
612 mm 0.0442 m3
612 mm 0.0442 m3
694 mm 0.0457 m3
Shunt Resistors
1030 mm
300 mm
Output Inductor
Transforme r
Recifer ioes
312 mm
312 mm
168 mm
312 mm
312 mm
42 mm
350 mm 0. 1082 m 3
159 mm 0.0155 m3
159 mm 0.0155 m3
Packing Factor
3300 V
Switches
Sub-Total
5500 V
Switches
10000 V
Switches
15 mm 0.0001 m3
40%
130 mm
140 mm
4mm 0. 0007 m3
0.6853 m3
140 mm
Sub-Total0.
73 mm
50 mm 10.0005 m3
6847 m 3
150 mm
90 mm
60 mm 0.0008 m3
.65m3
Suib-Tota
Table 21 Converter Volume housed in enclosure
Based on the concept of this PCM, at least one enclosure is required with a volume of
1.4 m 3 . Up to two modules can fit in the enclosure, but a controller is required, so the first
enclosure can only hold one converter module and the controller.
Page 87
4.4 NAVAL ARCHITECTURE SUMMARY
The recommended method to use for selecting a PCM's characteristics from the
above data is to first determine the power required.
Then, the level of redundancy required should be determined. For example, based
on the electric plant configuration, determine whether there will be PCMs operating in
parallel. Also, determine the number of redundant modules required. It should be
assumed that the control module is a high-reliability unit and the most likely failure would
be in the converter module. This information will allow the determination of the number
of PCMs and the number of converter modules per PCM.
Once the number of modules for a given PCM has been determined, the operation of
the modules should be decided. The main choice in this design is whether a redundant
module should be operated in parallel with the others. For example, a redundant module
would be adding a
3rd
module if the total load was under 400kW but above 200kW. This
module would then be available to supply part of the load if one of the other modules failed.
If the module must be able to pick up load instantly, then it should be operated in parallel
which would reduce the loading on all three modules and affect their efficiency. However,
if a momentary delay is acceptable, then the third module could be off until it is needed.
This would not affect the efficiency of the two operating units.
The cooling heat load can be estimated from the number of converter modules
operating and the load on each module. This heat load must be removed by some kind of
cooling plant. Most likely, water cooling would be utilized for shipboard applications
where volume is at a premium.
Now that the number of PCMs and modules has been selected, the number of
enclosures can be determined. Each enclosure has a volume and weight as explained
above, and the enclosure can hold either two modules or a module and a controller. Using
the volume of the enclosures and the weight of the enclosures and converter or controller
modules, the size and weight of the PCMs can be determined.
Combining each of these factors allows an initial estimate of the impact of the
electrical distribution system on the ship's design.
Page 88
CHAPTER 5 CONCLUSIONS
This project designed a DC to DC power converter for use in medium-voltage DC
(MVDC) based ship electrical distribution system architecture. In addition, the naval
architecture parameters of weight, volume, and heat load were estimated.
While the DC to DC converter design was relatively straight forward, the low
switching frequency proved to be a challenge. A shunt, active filtering circuit was designed
to limit the converter transients. Also, the control circuits for the converter were designed
to produce the desired operation while allowing simple control signals such as a duty ratio
from 0 to 1 and the desired voltage. The simple control signals will allow future control
system designs to easily control the converter module without extensive modification.
The Naval architecture parameters described above allow the development of more
accurate ship models. It is expected that future naval combatants will use MVDC
distributions systems, and this model will allow more detailed exploration of those designs.
5.1 RECOMMENDATIONS FOR FUTURE WORK
While as much rigor as possible was applied to this design, it is still an early-stage
design. There are many more iterations that would have to be completed to arrive at a
complete design. Several questions arose during the preceding design, but none of the
issues were expected to invalidate the design. Barring these concerns, the converter design
yields a perhaps innovate way to actively filter the output while minimizing the size of the
filters and increasing the speed of the response.
In this case, the early stage design definition is meant to indicate that there are ideal
components in the final simulation model. Some of the ideal components were used to
speed up the simulation while others were used to simplify the analysis. By using many
ideal components in this early stage of the design, it was easier to determine operational
and performance issues with the circuit. The outputs were also cleaner. These factors
allowed the operation of the circuit to be understood without as much "noise" from
leakages and parasitic effects that would alter the characteristics of the circuits. The next
iteration of the design could work through the issues of adding these second and third
Page 89
order effects. It is expected that the basic circuit will function after this second iteration
although perhaps efficiency will drop slightly.
One of the primary concerns is the transformer. A transformer design was not
completed. A high-frequency transformer design would be another excellent place to
concentrate future research. The efficiency of the actual transformer is expected to be
close to the values used in this project, but the volume and weight might be quite different.
The active filtering circuit presented here appears to have the desired effect.
Another way to address the converter's ability to respond to a transient would be to use a
variable frequency drive circuit. The frequency would be increased during a transient to
ensure a quick response and minimal overshoot. Another improvement that might yield
significant results is operating two converter modules in parallel with their switching
frequencies out of phase by 180 degrees. It was surmised that the ripple from the two
modules would cancel allowing a smaller filter. By combining these two effects, it seems
likely the size and weight of the converter could be decreased. In addition, if the ripple was
minimized, the variable frequency might be decreased below 2kHz in steady state
particularly at low loads. This could significantly increase the efficiency of the converter
especially when lightly loaded.
A final issue concerns the switch stresses due to low duty ratio. The lower duty
ratio was required to allow the low frequency converter to respond effectively to load
increases. However, the lower duty ratio meant higher peak currents for the components
in the converter. More analysis could be completed in this area to ensure the component
ratings are sufficient.
This project produced a first iteration design of a power converter. The issues and
concerns raised here would require a more detailed design should this converter be
constructed. Overall, the converter met the requirements with the exception of the
efficiency.
Page 90
BIBLIOGRAPHY
[1] Doerry, Norbert. "Next Generation Integrated Power System (NGIPS) Technology
Development Roadmap". Washington Navy Yard, DC, Naval Sea Systems Command,
2007.
[2] CAPT Doerry, Norbert. Functions of PCON. NGIPS PCON Workshop. Annapolis, MD, Jun
2009.
[3] SatCon Applied Technology. Distributed Power Systems Flier. Boston, MA.
[4] Tannenbaum, Joseph, Dana Delisle, and Joseph Borraccini. High Efficiency and High
Power Quality Electrical Conversion. SBIR Program Topic Submission.
[5] Kassakian, John G., Martin F. Schlecht, and George C.Verghese. Principlesof Power
Electronics. Reading, MA: Addison-Wesley Publishing Company, 1992.
[6] Wong, Fu Keung, B. Eng, and M. Phil. High Frequency Transformer for Switching Mode
Power Supplies. School of Microelectronic Engineering, Griffith University, Brisbane,
Australia, 2004.
[7] Johnson, C.Mark. "Current state-of-the-art and future prospects for power
semiconductor devices in power transmission and distribution applications".
International Journal of Electronics Vol. 90, 2003.
[8] Barkhordarian, Vref. Power MOSFET Basics. International Rectifier, El Segundo, CA.
[9] Laud, Satyavrat. "IGBT evolution enhances switched-mode power conversion".
Electronic Products and Technology Nov-Dec 2007.
[10]
Baek, Ju Won, Dong-Wook Yoo, and Heung-Geun Kim. "High-Voltage Switch using
series-connected IGBTs with simple auxiliary circuit". IEEE transactions on Industry
Applications vol. 37 no. 6, Nov-Dec 2001.
[11]
Designing Single-Switch, Resonant-Reset, Forward Converters. MAXIM Application
Note 3983, Mar 2007.
[12]
Hitachi. Specification sheet - Silicon N-Channel IGBT 3300V E2 version. Spec No.
IGBT-SP-08002 R3.
Page 91
[13]
Hitachi. Specification Sheet - Silicon N-Channel IGBT 6500V E2 version. Spec No.
IGBT-SP-09008 R5.
[14]
Borage, Mangesh, Sunil Tiwari, and S. Kotaiah. MOSFET - Assisted Soft-Switching of
IGBTs: A Re-Examination. Nov 2003.
[15]
Eriickson, Robert W. Fundamentals of Power Electronics. University of Colorado,
Boulder. 2004
[16]
POWEREX. Specification sheet Single IGBTMOD HVIBT Module 200 Amperes/6500
Volts. CM200HG-130H.
[17]
Mitsubishi. Mitsubishi HVIGBT Modules. High Power Switching Use Insulated Type.
CM400DY-66H.
[18]
Dynex. Rectifier Diode. DS502ST. June 2002.
[19]
Marden, M. Milosevic, P. Prempraneerach, J.L. Kirtley, G.Karniadakis, C.
Chryssostomidis. "An End-to-End Simulator for the All-Electric Ship MVDC Integrated
Power System," in Proceedings of the Grand Challenges in Modeling and Simulation
(GCMS10), Ottawa, Canada, 2010.
[20]
Series CHigh Voltage Energy Storage Capacitors. General Atomics website.
http://www.ga-esi.com/EP/capacitors/series-c-high-voltage-capacitors.php
[21]
Limpeacher, Rudy. "Novel Converters for Electric Ship Propulsion System and
Shipboard Power Distribution." Science Applications International Corporation.
[22]
Jovcic, D. "Step-up DC-DC converter for megawatt size applications." IET Power
Electronics. Nov 2008.
Page 92
Appendix A - Model Detail
The purpose of this appendix is to provide all the necessary information for a third
party to reconstruct this model and produce verifiable results. This appendix shows the
details of the model blocks required to implement this design in Simulink. The figures and
explanations below show the parameters of each component in the simulation model.
The model was created using many components from SimPower Systems. This
module must be added in addition to the Simulink software in MATLAB. The screen below
shows the solvers settings. The model might work with other solver settings, but these
were the settings which produced reasonable results for this project.
.t64
Data .mport/Export
Cptlrrization
Diagnostics
Sample Time
Data Vaidity
Type Conversion
Connectivity
Compatibility
MrdAeRafarpn-Ing
Saving
Stateflow
--Hardware Implementation
Node Referencinc
L Simulation Target
Symbols
Custom Code
Real-ime Workshop
Report
i-Comments
Symhnls
Start time: 0.0
mi!ape
SimMechanics
Stop time: .001
Solver options
Type:
Solwr:
Jod
Max step size:
auto
Relative tolerance:
le-5
Min step size:
auto
Absolute tolerance:
auto
IrtUal step size:
auto
Shape preservation: rq
Solver reset rrethod:
Fs
n
Number of corsecutive min steps:
Solver
/
V
1
laablan method
Tasidng
and sample time options
uto
Tasidng mode for perodic sampe tines:
Fl Automatically handle rate transitlon for data transfer
l Higher prio y value indcates higher tast priority
Zeross
Custom Ccde
Debug
Interface
Si
AS
Simulation time
Select:
Options
Zero-crossing control: Use local selng
Algorithm:
Time tolerance:
Signal threshold: auto
10*128*eps
Number of consecutive zero crossings:
in1p
11
1000
The following MATLAB code was used to run the simulation. While the code is not
explicitly required to run the model, it does contain the values for some constants in the
simulation blocks.
clear all;
sf = 0.9;
%safety factor for over and undershoot
A-1
shunting default
0.9
%SET to the MVDC bus voltage
MVDC = 3000;
%Used only if the load structure is used
Rloadl = 1000; %starting load resistance
Rload2 = 10; %transition to load
Rload3 = 5; %transistionn 2
Rload4 = 5; %transistion 3
%The PID gains below are default = 1
kp = 1; %Proportional gain
ki =
kd =
1;
1;
%Integral Gain
%Derivative Gain
ul = 0.95; %Upper limit of PID output for duty ratio
Ttrans = .02; %Time when the first transient will occur if using load
structure
Ttrans2 = .04; %Time of second transient
Ttrans3 = .06; %Time of third transient
time = .1; %simulation time
close all;
f = 2e3; %frequency in Hz default 2kHz
sim('Model 6500rev3',time)
points = length(Vout); %number of data points
t = (0:time/points:time*(1-1/points))'; %time for x axis
scrsz = get(0,'ScreenSize');
figure('Position', [5 5 scrsz(3)-75 scrsz(4)]) %[left bottom width height]
hold on
subplot (1,1,1)
plot(t,Iout+2500,'mo', t,SS*100+2250,'ko',t,(Slope+4e6)/2000,'y',...
t,Overshoot*100+1750,'k',t,Discharge*100+1500,'m',t,Charge*100+1250,...
'c',t,Vreg,'g',t,Vout,'b',t,Vcap,'k--',t,Vrec/5,'r')
axis([0 time 0 2750])
xlabel('Time (sec)')
ylabel('V')
legend('Iout', 'SS','Slope','Overshoot','Discharge','Charge', 'Vreg',...
'Vout', 'Vcap','Vreg')
%Determines the Steady State Ripple
vt = 2/f;%time for 2 cycles
pa = round(vt*points/time);%points added to get at least 1 full cycle
m=Vout(round(3*points/4):round(3*points/4)+pa);
Vout_min = min(Vout(round(points/4):points))
Voutmax = max(Vout)
SSripple = max(m)-min(m)
SSV=mean(m)
V_reg = mean(Vreg)
The diagram below is the structure for the power conversion module. The series of
diagrams and tables below show the settings for each block to ensure that the model is
reproducible without having access to the electronic content. The voltage source was set to
6500V for this example. If a different input voltage is desired, change the source voltage
and also adjust the transformer accordingly. Unless otherwise shown throughout this
appendix, the sampling time is inherited.
A-2
g
lockParaetI
Soste
Pulse type determines the computational technique used.
Time-based is recommended for use with a variable step solver, while
Sample-based is recommended for use with a fixed step solver or withi
a discrete portion of a model using a variable step sover
Parameters
Pulse type: Trigbased
Time (t):
U|4ggpm ime
Amplitude:
Period (secs):
Block Parameters: MVDC
DC voltage Source (mask) (link)
.1
Ideal DC voltage source.
Pulse Width (%of period):
so
Parameters
Phase delay (secs):
Amplitude (V):
0
6500
C Interpret vector parameters as 1-D
EIK Cnl
Measurements N_
_e
S
E
Cance
L
App
1000VDC +
SS V
Value
1000VDC
Unit Delay
PCON
A-3
-
Functir
lck Parameters; Saturation
Saturation
------------ ----- ,-------------
Limit input signal to the upper and lower saturation vdlues.
Function
z
Block
Param
Unit Delay
Upper imit:
Sample and hold with one sample period delay.
1060
Main
State Attributes
Lower limit:
Initial conditions:
1000
1000
Treat as gain when linearizing
Input processing:
'1Enable zero-crossing detection
Elements as channels (sample based)
Sample time (1 for inerited):
Sample time (-1 for inherited):
O7k7
Cancel
17 Helpo
OK
pl
Function Block Paameters Mean Value
Mean value (mask) (link)
This block computes the mean value of the input signal over a running
window with specified averaging period.
Parameters
Averaging period (s):
2/f|
ance
Help
pp
A-4
Cancel
Ip
Ap
Conn1
S6
Timer2
RL1
RL2
RL4
RL3
Conn2
Load Structure
The switches are initially open except for the bottom two switches S1 and S3.
A-5
Source Slock Parameters: Timer
Tiner (mbsk) (kink)
Generates a signal changing at
OK
Cancel
H
Sorce Block Parameters: Timer2
Timer (mask) (link)
Generates a signal changing at specified times.
if a signal value is not specified at time zero, the output
is kept at 0 until the first specified transition time.
Parameters
Time (s):
[Ttrans3]
SAmplitude:
OK
cancei
Help
A-6
Power Converter
Block Parameters: S1
Block ParmnetersLinear T
m
Ideal Switch (mask) (link)
Unear Transformer (mask) (link)
Switch controlled by a gate signal in parallel with a series RC snubber
circuit.
In on-state the Switch model has an internal resistance (Ron).
In off-state this internal resistance is infinite
The internal resistance must be greater than zero.
The switch model is on-state when the gate signal (g) isset to 1.
Implements a three windings linear transformer.
Parameters
Units [Pu
Internal resistance Ron (Ohms):
Nominal power and frequency [Pn(VA) fn(Hz)]:
0.01
Parameters
[250e3 2000]
Initial state (0 for 'open', 1 for 'closed'):
Winding 1 parameters [Vi(Vrms) R1(pu) L1(pu)]:
0
[MVOC 0.002 0.08]
Snubber resistance Rs (Ohms):
Winding 2 parameters [V2(Vrms) R2(pu) L2(pu)]:
1e5
(3000 0.002 0.08]
Snubber capacitance Cs (F):
LJ Three windings transformer
inf
]
Click the Apply or the OK button after a change to the Units popup
to confirm the conversion of parameters.
Winding 3 parameters [V3(Vrms) R3(pu) L3(pu)]:
31503 0.002 0.08]
Show measurement port
Magnetization resistance and inductance [Rm(pu) Lm(pu)]:
(500 500 1
Measurements [None
K
Applnpy
A-7
Cance
Help
Apply
BckParameters: D1
Implements a diode in parallel with a series RC snubber circuit.
In on-state the Diode model has an internal resistance (Ron) and
inductance (Lon).
For most applications the internal inductance should be set to zero.
The Diode impedance is infinite in off-state mode.
Parameters
Resistance Ron (Ohms)
0 01
Inductance Lon (J
0
Forward voltage Vf(V
0 8
Intiall current c (A)
Snubber resistance Rs (Ohms)
500
Snubber capacitance Cs (F)
Show measurement port
O
LCancel
Help
Ap
A-8
+input
+ Load
S5
S6
Ro
Vcap
Cshunt
~u*T
I
Rol
I
I measurement
o
A
input
SHUNT
Voltage Control
A-9
-Load
1RFunctn Block
am rDelay
Function Block Parameters: Delay2
Discrete variable transport delay (mask) (link)
Apply a delay to the first input (In)signal.
The second input (D)specifies the delay time.
Discrete variable transport delay (mask) (link)
Apply adelay to the first input (In)signal.
The second input (D)specifies the delay time.
Parameters
Parameters
Maximum delay (s):
1/40
Maximum delay (s)!
Initial input:
Initial input:
1/40
0
Sample time:
Sample time;
te-6
Cacl
Help
LApy
OK
A-10
Cancel
Help
Apply
11Functon
Block Pake I
Functin Block Pararnetem I
I-
Function Block PaaameterzM~
3 Dtsaete
Monoabe Rip-lop (mask) (lnk)
~
Block ParRieteri~41
U Function
Flip-Rop (mask) (kink)
Discrete
Monotable
After being triggered by the specified edge (Rising, Felng or EthIer),
this block outputs a pulse (TRUE signal) for the tne specMed by
parameter -Puise duraon.
if the spedfied edge isdeteded while the output is TRUE, the
monogable is not retriggered
FaMng or Either),
After being triggered by the specled edge (Ris
this block outpts a pulse (TRUE signa) for the time specified by
paramete 'Pulse duramon".
I the specified edge is deteded while the output is TRUE, the
nwotable Isnot retriggered
The input value (0(1) at time step preceding t = 0 Isspecified by the
parameter "input at t = -Ts.
The input value (011) at Ome step preceding t = 0 isspecified by the
parameter Input at t = -T.
Parameters
Parameters
Edge detection Wing
Edge
Pulse duration (s):
Pulse duraon (s):
detection
v1
-
1/(2-f)
1/(2-f)
Input at t = -Ts:.
0
Input at t = -Ts:
0
sample time Ts:
Sample time Ts:
1/(100-f)
LI(100-f)
Ix
Apply
Voltage Regulator
A-11
7caoZ LIWp
App y
Discrete PID Control er (mask) (link)
This block implements a discrete PID controller.
Parameters
Proportional gain (Kp):
kp
Integral gain (Ki):
ki
Derivative gain (Kd):
kd
Time constant for derivative (s):
1
Output limits: [ Upper Lower]
[ul 0]
Output initial value:
.4
Sample time:
1/(100ef)
_____
7
Cane
1 271
A-12
Apply
Shunt Controller
A-13