A Spaceborne Microwave Radiom eter Design For
Ocean Wind Remote Sei ising
by
Michael W. Miller
Submitted to the Department of Electrical Engiuieering and Computer Science in partial fulfillment of the requi rements for the
degrees of
Bachelor of Science in Electrical and Computer 5cience and Engineering and Master of Engineering in Electrical Engineering and
Computer Science
MASSACHUSETTS INSTITUTE OF TECIHNOLOGY
May 15, 1996
@ Michael W. Miller, MCMXCVI. All Rigl tts Reserved.
The Author hereby grants to MIT permission to r -produce and distribute publicly paper and electronic copies of thi, Sthesis document
in whole or part, and to grant others the rig ht to do so.
.................................
Author .............. ...............................................
Department of Electrical Engineering aid Computer Science
May 15, 1996
C ertified by .....
..................................................
Jin A. Kong
1
•
,
Professor
Thesis Supervisor
Accepted by .......
I
........................................
F.R. Morgenthaler
v dChairman, Departme al Committe c,ni Graduate Students
9arker Eng
MASSACHUSETTS INSTITU TE
IF:
1rr--ýOLOGY
996
It.4
JUN 111996
IRRARIFC
A Spaceborne Microwave Radiomete r Design For
Ocean Wind Remote Sensi ng
by
Michael W. Miller
Submitted to the Department of Electrical Engioleering and Computer Science in partial fulfillment of the requ rements for the
degrees of
Bachelor of Science in Electrical and Computer dience and Engineering and Master of Engineering in Electrical Engineering and
Computer Science
Abstract
In this thesis, the design of a spaceborne passive radar is pre ented for the remote sensing
of wind vector at the surface of the ocean. Its bases are the 10.35 and 37 GHz radiometers
built at the Jet Propulsion Lab at Caltech in Pasadena, Califctnia which have been tested
in airborne configurations aboard the NASA DC-8. The radioneter detects orthogonal
linear polarization brightness temperatures from the surface ;f the ocean, then extrapolates a modified form of Stokes vector. The quantities in this vector are then used to determine ocean surface wind direction. The spaceborne radiometer is a descendant of the
airborne radiometers built at JPL, using the same basic desig4 concept, but with the addition and deletion of subsystems to make it more efficient and smaller. The waveguide
switch network to calculate the modified Stokes vector was r -designed in microstrip, the
serial operation of the airborne radiometers was replaced wit 1parallel detection ability,
and the noise diode was rearranged to provide for a different calibration scheme. After the
overall design was completed, particular design and construc ion of the crucial microstrip
adding circuit was begun. Due to time constraints of the inte nship period during which
this work took place, only this piece of the radiometer was built and tested. The center frequency of this subsystem was 13.8 GHz so a test radiometer could be constructed with relative ease using parts already on hand. Its operation was veri~ed and it was determined
that the relative phases of the sum and difference signals were within the plus-or-minus 14
degree limit over a 500 MHz bandwidth. The magnitudes of Qhe signals were degraded
due to a faulty connector design, but this was corrected in a redesign of the subsystem at
an 18 GHz center frequency.
Thesis Supervisor: Jin A. Kong
+l;tT
. D
fp
essor
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Table of Contents
1 Introduction.................................................................... ........... ...........
10
1.1 Passive Remote Sensing ..................................................... 10
1.2 Current Methods ........................................................... 10
1.3 Previous W ork ......................................... .......... ...................................... 11
1.4 Current Work .........................................
12
2 Remote Sensing Theory ................................................
14
.................................. 14
2.1 Introduction .............................................
2.2 Modified Stokes Vector ............................
.......... 16
3 W ind Radiom eter Design..................................................
.................................... 18
3.1 Looking At Other Designs ...............................................................
18
3.2 Design Considerations ....................................................................... 20
........................................ 24
3.3 D esign Topography............................................
3.4 M icrostrip D esign ............................................................................... 30
4 Construction And Testing ............................................................ 34
4.1 Microstrip Circuit Construction .........................................
....... 34
4.2 Testing the Microstrip Circuits ................................................ 36
5 C onclusions.........................................................
................................................. 40
40
5.1 Test Results ................................................................................................
5.2 Trials And Tribulations..............................................................................41
5.3 Further Study ................................................................................................
42
Appendix A Spaceborne Polarimetric Radiometer Types At 18 GHz........................44
... 46
Appendix B Footprint / Swath Coverage Calculations ...................................
48
...............................
....
Appendix C Calibration Measurement Procedure .................
........ ....................................... 48
C .1 O verview ....................................................
C .2 N oise D iode Calibration .......................................... ....................................... 48
C.3 Radiometer System Temperature Calibration ................................................. 49
... 52
Appendix D 6-Channel Radiometer Block Diagram ....................................
Appendix E 13.8 GHz Band-Pass Filter Simulation ............................................... 54
..... 54
E.1 HP Microwave Design Simulator Layout................ ...............
E.2 Simulation Frequency Response...........................i.................................55
Appendix F 13.8 GHz Adding Circuit Simulation ...................................
..... 56
F.1 Adding Circuit Block Diagram.............................................56
F.2 M D S Layout................................................... ............................................ 57
F.3 MDS Simulation Results ..................................... ................
58
B ibliography .........................................................
..................................................
60
4
List of Figures
Figure 2.1: Polarization Vectors .....................................................................
...
15
Figure 3.1: Russian Gyrator Radiometer block diagram. Taken from [7] .................. 19
Figure 3.2: How footprint pixels make up the circular scan ........................................ 22
Figure 3.3: Calibration Line ............................................
....................................... 23
Figure 3.4: Noise Diode Placement Analysis. Note the largei uncertainty in the calibration
line for Scheme II when the Dicke switch is before the noisc diode ........................... 27
Figure 3.5: Gain calculation model ................................................................................
28
Figure 3.6: The bandpass filter mask (AutoCAD), shown actual size........................ 31
Figure 3.7: The modified 90-degree hybrid coupler without nagnitude factors on the outpu ts .................................................................................................................................
32
Figure 4.1: Microstrip adding circuit, actual size. .................
3.......................................
Figure 4.2: Transition from coaxial line to microstrip; the important ground contact, point
A , is also show n [17]. .....................................................................................................
36
Figure C.1: Simplified Radiometer Block Diagram .....................................................
48
List of Tables
Table A. 1: Spaceborne Radiometer Comparisons 44
Table A.2: Spaceborne Radiometer Comparisons (continued) 44
Chapter 1
Introduction
1.1 Passive Remote Sensing
The knowledge of ocean surface winds is important for uncerstanding and predicting the
interactions between the sea and the atmosphere. The near surface ocean wind is the
momentum flux driving the circulation of the ocean, and its accurate prediction is an
important component of reliable weather forecasts. Currentl3, the United States is employing two main methods of detecting the surface ocean wind speed and direction. Each is
costly to set up and maintain. This thesis will be concerned iith a more cost effective and
reliable approach to making higher resolution measurements through passive remote sensing in the microwave region.
-1
,%
A"
A
- A'1At
- -
1.2 CurrentMVetnods
The more primitive of the two is through the use
. Scattered on the surface
of the oceans, at not nearly a global scale, are
cord the wind speed and
direction through mechanical means. They then transmit their data to a processing center
where global forecasts are attempted. The other method, which came into maturity in the
last decade, is to sense ocean wind speed and direction through the use of spaceborne scatterometers. "The principle of scatterometry for ocean wind •neasurement is based on the
empirical relationship between near surface wind and norma ized ocean radar backscatter
cross section," [1]. There have been several spaceborne scatt rometer experiments verifying the correlation between ocean radar backscatter and wind speed, as well as a variation
of the backscatter with the relative azimuth angle between the wind direction and the radar
look direction. These include the scatterometers SEASAT and ERS-1. Note that this
method is active, and therefore requires substantial power o generate useful measure-
ments over a prolonged period of time.
1.3 Previous Work
There has recently been a large body of research published regarding passive remote sensing of ocean features. It has been shown that the most approiriate method for wind speed I
wind direction analysis is a full polarization radiometer looking at incidence angles
between 45 and 65 degrees from earth orbit [2]. An import4nt stepping stone to this finding, however, was a radiometer launched through the Detfnse Meteorological Satellite
Program in 1987.
This satellite was launched with the capability to monitor the oceans on a global scale
in the microwave region at 19.35 GHz and 37.0 GHz by m asuring linear polarization of
surface emission. The instrument on board this satellite, the Special Sensor Microwave
Imager (SSM/I), was a total power radiometer used to measure orthogonal polarizations of
the emission from ocean and land surfaces. The SSM/I had goals in addition to the remote
sensing of ocean wind direction and thus had several other frequency radiometers on
board. It was shown previous to this launching that using ort1ogonal polarizations at 19.35
and 37.0 GHz, knowledge of ocean surface wind speed was possible.
The SSM/I was designed so that, while in orbit, calibration could be performed once
each scan period of 1.9 seconds, with the radiometer gain temaining constant over this
interval. This was done through the use of a reflector mounted in the scan path of the radiometer which would reflect cold space into the antenna once ach scan. This frequent calibration kept the gain drift to a minimum, and thus aided in the stabilization of the
radiometer sensitivity. For the 19 and 37 GHz channels, the s nsitivity was specified at 0.8
K and 0.6 K, respectively, while the actual values measure
spacecraft were slightly lower [3].
during the lifetime of the
The algorithm used by the SSM/I to determine ocean surface wind speed measurements was "based on the 'D--Matrix' approach which seek a linear relationship between
measured SSM/I brightness temperatures and environment 1 parameters," [4]. This algorithm is rather complex and uses matrix coefficients based n particular seasons and latitude bands that the measurements were taken from. The algorithm was validated at the
University of Massachusetts by comparing derived wind ;peeds with references taken
from buoys maintained by the National Oceanic and Atmospheric Administration
(NOAA). "Results indicate that for approximately 85% •f the time, the D--matrix-retrieved winds will have an accuracy better than the Defen e Meteorological Space Program (DMSP) goal of 2 m/s. For the remaining 15% of th4 time,... retrieval accuracies
will be worse than 2 m/s," [4].
Within the past several years, it has been theoretically shown that "the new technique
for remote sensing of the near--ocean surface wind is based n the brightness temperature
anisotropy -- the dependence of the microwave emission oi azimuthal angle," [5]. This
technique is different from that originally used in the analysis of the SSM/I data. However,
recently a study by Wentz [6] using this data has confirmed the importance of wind direction as a factor that affects microwave brightness temperature. This fact has been supported by several other studies [2, 6, 7, 8, 9], as well. It is witi this technique in mind that
the current work on ocean surface wind remote sensing will he discussed
1.4 Current Work
Using the dependence of microwave brightness temperature f the sea surface at azimuth
angles relative to the wind direction has been confirmed as !a viable method by several
experimental and theoretical studies. Experiments conducted in the Space Research Institute (Moscow, Russia), the Environmental Technology Lab atory (Boulder, Colorado),
and the Jet Propulsion Laboratory (Pasadena, California), de onstrate the effectiveness of
12
the method.
The aircraft radiometer experiments conducted by the ussians at the Space Research
Institute measured brightness temperatures of the horizontal and vertical polarizations at
incidence angles near nadir. They found a few degrees Kel in difference in the up/down
wind direction verses the cross--wind direction in the three measured Stokes parameters
[2]. However, one should note that the experiment was "c nducted at near normal incidence, which though scientifically significant, is not useful f om the swath coverage standpoint," [1]. This is in regards to the coverage needed if remOte sensing from a spaceborne
platform on a global scale is the goal.
At the Jet Propulsion Lab, where the research performed in this paper took place, several aircraft flights have been taken to show the usefulnesg of 19 and 37 GHz linearly
polarized radiometers to ocean wind remote sensing. The res4lts of several of these flights,
in [2, 9], demonstrate that at incidence angles in the range fr m 45 to 65 degrees there are
as large as a few Kelvin signals in all Stokes parameters.
"It was found that the wind direction signals ave a broad
frequency spectrum from 19 to 37 GHz. The p and downwind asymmetry of sea surface brightness ha a small
increasing trend versus incidence angle, while the up and
crosswind asymmetry may have a dramatic v iation with
incidence angle. The observed magnitudes of imuthal
wind direction signals in the vertical and horizontal polarizations,... are easily measurable with present Onicrowave
radiometers. In addition, the third and fourth Stokes parameters, U and V, which have an odd symmetry with respect to
the wind direction,... are also measurable with a single
antenna plus microwave switch network desig . The results
indicate that spaceborne passive microwave ra iometers
have a strong potential for ocean wind remote ensing," [9].
Chapter 2
Remote Sensing Theory
2.1 Introduction
The basis of the previous discussion on sensing ocean wind vector stems from the fact that
electromagnetic waves emitted from natural media due to ra dom thermal motion of electric charges are in general partially polarized. The way tha wind over the ocean causes
small ripples in the surface serves to intensify this parti
polarization of the emitted
microwaves. Because these ripples are anisotropic in azim th direction, the polarization
vector of emitted microwaves will vary with that direction relative to the wind vector.
In order to completely classify the partial polarization oý the emitted microwaves, Sir
George Stokes in 1852 introduced four parameters I, Q, U,
d V These Stokes parameters
deal with the correlations between the magnitudes of two o hogonal components, Eh and
Ev, of the thermally radiated electric field present in a partiall polarized wave [10]. "Isignifies the total radiated power, while Q characterizes the pola ization difference. The third
parameter U and the fourth parameter V represent the real' and imaginary parts of the
mutual correlation between Ev and Eh," as can be s(
StokesVector =
2.1 [2].
I Tvt+ Th
=
Tv-Th = c
(2.1)
U Tp-Tm
Y TI - Tr
Since radiometers for earth remote sensing usually measure the brightness temperature
of the surface rather than electric energy density, this notatio has been used in equation
2.1. Here, TV and Th are the temperatures of the vertical and horizontal polarizations, Tp
and Tm are the brightness temperatures of the +45and -45 de ree linear polarizations, and
T1 and Tr represent the left and right-hand circular polarization brightness temperatures as
illustrated in figure 2.1. The angular brackets denote the •nsemble average of the argument, and c is a proportional constant relating the brightn ss temperature to the electric
energy density. [9].
rr
Iv
T1
Th
'r
111
Figure 2.1: Polarization Vector,
Typically, the approach for Stokes parameter measuremetit is to make power measurements at each of the six different polarizations. This can bec me quite cumbersome, especially when a different linearly or circularly polarized antenna is used for each
measurement. These power measurements are then used with the following identities to
calculate the third and fourth Stokes parameters.
2
(2.2)
21m (EhEv*) = ElcI2 - ErcI2
(2.3)
2Re (EhEV*) = Ep 2 -IEm
where
EP
Eh + Ev
(2.4)
Em
h- E
(2.5)
EIc
E h + iE v
(2.6)
F2ErcE-
Eh - iEv
'i
(2.7)
It has been shown, however, that all polarization measur ments can be carried out by a
single antenna and a microwave switch network to coherently combine the vertically and
horizontally polarized electric fields [9]. Equations 2.2 through 2.7 illustrate the fact that
to determine Ep and Em one merely needs to determine the coherent sum and difference of
vertical and horizontal polarization field components. Anc to calculate Elc and Erc the
same process is followed, except for a 90-degree phase shift in the vertical component
before the addition or subtraction.
2.2 Modified Stokes Vector
In the experimental work of the researchers at JPL, it has be n shown that the results from
the I and Q Stokes vectors are very similar and that the resuls from the Th and Tv signals
are individually more scientifically interesting. Therefore, a modified form of the Stokes
vector has been used in their research, in which the I and
Q components
are replaced with
the horizontal and vertically polarized brightness temperatur s.
TTl rEatj2)
Th
UI
Th
c (Eh
Tp-Tm
2Re(EVEh*)
LT-Tr
L21m(EvEh*)
(2.8)
For the design of the radiometer in this paper, the above nodified form has been used
exclusively. Clearly, then, from the brightness temperatures btained from the radiometer
(Tv and Th), we can easily calculate the elements of the mod fied Stokes vector using the
following equations:
U = Tp- Tm
(2.9)
V = Tic - Trc
(2.10)
Chapter 3
Wind Radiometer Design
3.1 Looking At Other Designs
Because the research in this paper was conducted through the MIT VI-A internship program at JPL, the radiometers developed there were natur ly the basis for this design.
However, the designs by researchers mentioned earlier in this paper were considered for
their stability, bandwidth, power, noise, and size.
The radiometer developed by the Russians at the Space Research Institute in Moscow
made measurements from nadir viewing angle, which are not significant from a swath
coverage standpoint, as pointed out earlier. However, an innovative design was used to
determine the I, Q, and U elements of the Stokes vector. Ir particular, they use a ferrite
gyrator based on the Faraday effect, which rotates the polarization plane 45-degrees based
on a control current. This gyrator is placed after an omnipýlarizational antenna (conical
horn) but before an orthogonal-mode transducer to establish both horizontal and vertical
linearly polarized signals and plus-and-minus 45-degree linear signals [7], without needing a complicated adding circuit. The orthomode transducer ( MT) is a waveguide device
that takes a signal from a conical horn (or other circular wa eguide) and separates it into
its two orthogonal linear components. The Russians then switch between these four measurements and a reference load in the familiar Dicke radiometer scheme (see figure 3.1).
The main drawback to this design is the restricted bandwi th due to the ferrite gyrator.
As an example of the bandwidth for a Faraday device such 4s the gyrator, the data sheet
for a typical gyrator was looked up in the Alpha Advanced
oducts Division catalog of
millimeter wave components and subsystems. There, the s
cification was an instanta-
neous bandwidth of approximately 1%of the center frequenc . Thus, for the Russian con
Reference
Load
Figure 3.1: Russian Gyrator Radiometer block diag am. Taken from [7].
figuration, at a wavelength of 1.6 cm (18 GHz), the expected bandwidth would be about
180 MHz. This number was too low for the RMS noise (calcilations to be detailed below)
to be at an acceptable level.
A radiometer developed by A.J. Gasiewski at Georgia T ch presented another design
option. Here, a dual orthogonal linearly polarized radiomete is used along with a digital
cross-correlator to determine Th, T v, and the U Stokes vect r parameter. In this method,
the researchers have demonstrated a technique to electronicaliy (rather than mechanically)
rotate the polarization basis of a dual orthogonally-polarize4 radiometer [11]. This technique, called "Electronic Polarization Basis Rotation" (EBPR), allows the feedhorn polarization basis to be rotated by a simple matrix operation in soft are to any desired angle. In
addition to this innovation, the Georgia Tech researchers al o implemented a detection
scheme using digital correlators. Then, they used a conventio al hot and cold load calibration scheme for absolute calibration of each of the polariz tion channels. This method
forced the use of polarized calibration loads, which lead to a bulky setup with highlyabsorbing microwave blackbodies and a polarization-splitting wire grid.
4 weaknesses of the above two
The table in Appendix A shows the relative strengths an
designs alongside the statistics for the radiometer designed i this paper. The discussion of
the elements of the 6-channel design presented in this paperl follow.
3.2 Design Considerations
The radiometers built at JPL to remotely sense ocean wini vector were designed to be
used from an airborne platform. In their tests, the DC-8' would make circular flights
around a known ocean buoy at varying angles of incidence, ýneasuring the different polarization brightness temperatures. Because of the speed of thyse circles, the radiometer had
plenty of time to make the measurements and to average several of them in order to get
consistent numbers. Also, the size of the radiometers and tle power to be used were not
large constraints on the designs because the aircraft was q ite large. When designing a
radiometer for a spaceborne platform, these issues come to the forefront.
3.2.1 Footprint Calculations
First, when a radiometer orbits the earth on a satellite, the area that it "listens" to on
the surface, known as the footprint, is determined by severo factors. These include the
satellite'saltitude, pointing angle, and the antenna beamwidt. For the study in this paper,
typical values for a satellite to obtain global coverage were dsed for the spacecraft height
and speed. Because the previous studies by Yueh et. al. [24 9] indicated that 45 to 65degree incidence angles gave very favorable results, a 65-degree incidence angle was chosen. This choice also facilitates the ability of a spaceborne radiometer to achieve global
coverage. In order to obtain the resolution needed for NOAA to make use of the measurements provided by this radiometer, a footprint width of about 50 km is needed.
At this particular incidence angle, the pointing angle of t e radiometer antenna (with
respect to nadir) is 53.6 degrees at a spacecraft height of 800
. Also, in order to get the
required 50 km footprint width, the diameter of the circular a tenna needs to be about half
a meter. Therefore, at 18 GHz, the 3 dB beamwidth will be ibout 1.8 degrees. See Appendix B for the calculations of these values.
3.2.2 Radiometric Sensitivity
A radiometer is a highly sensitive receiver designed to measure thermal electromagnetic emission by different media. In order to make the radicmeter as sensitive as possible,
the power delivered from the antenna to the receiver needs tp be maximized. This quantity
is given by the equation: P=kTAB. Here, the power delivered is based on the radiometer
bandwidth, B, and the antenna temperature, TA, where k is the Boltzmann constant. The
function of the radiometer is to measure the antenna tempe ature. However, this quantity
represents the average value of a fluctuating noiselike signal.; Thus, the radiometer transfer
function (relating TA to the output voltage) and the precisi n with which TA can be estimated are of interest. The latter value, often referred to as the radiometer sensitivity, AT,
is a key quantity characterizing the performance of a microw.ave radiometer [12].
This sensitivity is defined as the smallest change in TA th4t can be detected by the radiometer output. Its value is governed by the radiometric bandwidth, the antenna temperature, the noise temperature of the receiver, and the postdet ction integration time, as in
equation 3.1:
AT =
2 (TA + TRN)
[12].
(3.1)
Here, TRN is the equivalent noise temperature of the rece ver, which can be calculated
from a cascading of the noise temperatures and gains of th• subsystems making up the
microwave receiver. The integration time, t, is the time available for the radiometer to
integrate measurements over one pixel. In this case, it is de ermined by the spacecraft's
altitude, speed, and footprint characteristics. In equation 3.1 B is the radiometric bandwidth. Clearly, maximizing the bandwidth and the integration time will increase the sensi-
tivity. However, since the integration time is set by the orbit of the spacecraft, maximizing
the bandwidth becomes the main criteria in laying out the design of the radiometer. The
radiometer is to scan in a circle, taking measurements at each pixel long enough to cover
the entire circular scan, as can be seen in figure 3.2.
Pixel
cular Pcan
Figure 3.2: How footprint pixels make up the circular scan.
For the data associated with this design, the antenna tdmperature summed with the
equivalent receiver noise temperature is referred to as Tsy s. Rough calculations of the
expected contributions to the noise of each component of the design show that Tsys will be
about 450 Kelvin. This leads to a sensitivity of 0.17 K, if the bandwidth is 500 MHz. Now,
considering the Russian design above, if the bandwidth were '180 MHz, and the noise temperature was the same, the sensitivity would be 0.3 K.
Equation 3.1 above applies to a radiometer set up in a Dicke switching scheme. In this
scheme, as in the Russian Gyrator Radiometer, the total power radiometer measurements
are switched between the antenna and a known reference load. In this way, gain fluctuations (which are the limiting factor to achieving high radiorietric resolutions) are minimized. In addition to the switching at the beginning of the signal chain, a synchronous
demodulator is used at the detector which sums the two ieasurements with opposite
polarity, to increase the received signal from the integrator i(which follows the demodulator) [12].
3.2.3 Calibration
In the above section, the radiometric sensitivity was disdussed as one important aspect
of a radiometric measurement process. This process is also zharacterized by the measurement accuracy. Conceptually, the transfer function of the radiometer receiver is established by measuring the output voltage as a function of the noise temperature of a noise
source connected to the receiver input terminals in place of the antenna. Since the radiometers this design is based on use square-law detectors, the ouiput voltage is linearly related
to the noise temperature of the input source. Therefore, it is sufficient to measure the output voltage corresponding to each of two input noise temper4tures to establish the calibration line (See figure 3.3). This calibration line is then used for converting the output
voltage measured by the receiver to antenna temperature values (provided the antenna is
connected to the receiver).
Vout
Tal (K)
Figure 3.3: Calibration Line
The noise source used in calibration is a separate entity from the reference load used in
Dicke switching. In previous spaceborne passive radiomet rs [3] a hot and cold load
would be presented to the antenna feedhorn once per scan, thus allowing a calibration to
occur on each rotation of the antenna assembly. In this desi n, calibration will occur by
taking four measurements, one of the reference load (also us d for Dicke switching), two
of the ambient antenna temperature (one with the noise diod coupled into the signal and
one without), and one with the radiometer off (to obtain the1zero offset measurement). We
can therefore establish the equivalent noise temperature o1 the receiver once each scan,
and thus construct the calibration line to increase the accutracy of the antenna measurements. See Appendix C for the calibration measurement procedure of both the noise diode
itself and for the radiometer temperature.
3.3 Design Topography
One of the main drawbacks of the radiometers developed at jPL for a spaceborne platform
is their serial nature. Each measurement is made in turn, the horizontal brightness temperature, the vertical brightness temperature, etc. There isn't enough time to take each of
these measurements when the spacecraft the radiometer is rnounted on is travelling at 6.7
km/s around the earth. Therefore, a major design change for the radiometer presented here
is a parallel measurement operation. Here, the OMT splits the partially polarized signal
into its orthogonal linear components and then processes each to calculate the needed
polarization brightness temperatures in parallel. This can be seen from the 6-channel radiometer block diagram in Appendix D. The signal flows froi the antenna into the OMT
where it is split up. Each of these signals then flow throug1 a directional coupler where
they can have noise injected into them depending on the state of the switch in front of the
noise diode. They then each go through a Dicke switch and on to a pair of matched amplifiers. These amplifiers serve to add enough gain to the signal so the losses due to the
microstrip adding circuit are not significant. Each signal then passes through a bandpass
filter and then into the adding circuit. Where in the JPL airbor#e radiometers, the functions
to add or subtract the signals were implemented in waveguide with the "magic-tee" (180degree hybrids), the 6-channel radiometer implements this functionality in microstrip
hardware. In this way, all the power division, phase shifts, smins, and differences can be
carried out in one small piece of hardware concurrently. Afte this subsystem, the signals
are each re-amplified before arriving at the square-law dete4tors.
In the following sections, each item in the proposed spaceborne radiometer will be
detailed. In addition, the calculations of the gains required for the matched amplifiers and
the pre-detection amplifiers will be given. In these calcul tions, estimations have been
made regarding the losses of the individual elements of the radiometer.
3.3.1 OMT
The orthogonal-mode transducer (OMT) is a waveguide Idevice that allows the separation of a signal in circular waveguide into its orthogonal liner components. It is a key element in the 6-channel radiometer, for it allows the use of one antenna horn to get several
different polarized signals. Otherwise, two separate rectan ular-horn antennas, mounted
perpendicular to each other, would be used to get the two orthogonal linearly-polarized
signals from the scene.
3.3.2 Dicke Switches
The functionality of a Dicke switch, as given in section 3.2.2, is to minimize the gain
fluctuations inherent in any highly sensitive receiver by switching between the antenna
and a known reference load. It is known that the bulk of these fluctuations lie below 1 Hz,
and that almost none exist above 1 KHz [12]. Thus, typical switching rates are chosen so
that the period of one cycle is between 1 and 20 ms. If the switch rate is 150 Hz (6.7 ms
per measurement), given the integration time of 55 ms, there can be 4 measurements of the
scene and 4 measurements of the reference load. In the WINDRAD radiometers developed at JPL, the Dicke rate was 125 Hz [9]; this was witho xt the strict requirements on
integration time for a spaceborne platform.
3.3.3 Noise Source
The noise diodes included in the radiometer design are primarily used for the calibration of the radiometer system temperature. This measurem nt is taken once each scan
cycle so that when an antenna measurement is taken, the difference in the magnitudes of
the antenna plus noise temperature and the reference load telnperature is decreased. In this
way, there will be less error in the synchronous demodulator that is used with the Dicke
switch.
One of the variables in the design of the radiometer d4alt with the placement of the
noise diodes. The options were to place it before or after th. Dicke switch. It was argued
that placing the Dicke switch as close to the antenna (before the noise diode) was the correct goal, because there would be less loss before it. However, in doing that, the loss and
error in the ferrite switch cannot be accounted for in the antenna/noise switching. So from
an absolute calibration standpoint, placing the noise diode irI front of the Dicke switch is a
more reasonable idea (see Figure 3.4). Of course, the drawback to this scheme is that in
the calibration of the system temperature done once per scan, there needs to be one extra
measurement. This would be to cancel the Tamb term arising from not being able to see the
noise diode with the Dicke switch to the reference (see calibration equations in Appendix
C).
3.3.4 Matched Amplifiers
The next elements of the design are a pair of low noise a plifiers matched in gain and
phase to stringent requirements. The purpose of these amplifiprs is to provide enough gain
to the signal so that the losses incurred in the adding circuit can be neglected. These
amplifiers were going to be used in both a prototype layout circuit at 13.8 GHz and an aircraft prototype at 18 GHz. Therefore, the specification was for the gain and phase matching to cover 1 GHz bandwidths around each center frequencyý The reason for the stringent
requirements on the gain and phase matching is so that the two orthogonally polarized sig
nals are as equal as possible going into the adding circuit. F r, in the adding circuit their
gains and more importantly, phases, must be equal so that th 180-degree hybrids (which
take two signals and outputs them added at 0 ahd 180' relati e phase) operate properly.
SCHEME I
Radiometer
Vout
N
Noiseout
Diode
I
I
SCHEME II
IT
Vout
loo
Radiometer
Vout
Tref
Noise
Diode
I
J
K
Figure 3.4: Noise Diode Placement Analysis. Note the larg r uncertainty in the calibration line for Scheme II when the Dicke switch is before the noise diode.
In order to determine what gain is needed for the matched amplifiers, a model of one
of the radiometer channels can be used with estimated losses for each group of elements.
The various noise temperatures for each can be worked out and then the amplifier can be
specified to contribute 5% of the system noise temperature. Figure 3.5 shows the model
used for these calculations.
According to this model,
T 1 T T ( 1- )L 1 T2 L1 L 2
Trec =TZ(ol-h)+ll+
+
l
+0 ,
(3.2)
where theislast
zero
term
because the second stage gain, G,is assumed to be very high
where the last term iszero because the second stage gain, G2,;is assumed to be very high
Figure 3.5: Gain calculation model.
(on the order of 45 dB). An estimation of these parameters yields the following:
L 1 = 1.25, To = 290, and L2 = 1.97. After communicating with the contractor slated
to build the matched amplifiers, the noise figure was appro imated at 6 dB for a gain on
the order of 20 dB, which was expected. Plugging in this n ise figure for the temperature
of the first stage amplifier using this relation, T1 = To (F 1 - 1) , and assuming the second
stage amplifiers have a noise figure of 3 dB, gives
Trec = 1138 +
re
178 + 714
G
(3.3)
So, if the amplifier were to have enough gain so the losses ii the second half of the radiometer contributed 5% of the system noise temperature, G, could be solved to be 12 dB.
Because there should be some margin for error in guessing Idsses and estimating the noise
figure of the post-detection amplifiers, the matched amplifiers were specified to have 25
dB of gain. This is well above the minimum set by the above calculations.
3.3.5 Microstrip Circuit
After being amplified, each signal then passes into the ipicrostrip circuit subsystem.
This system realizes the band-pass filters, power dividers, phase shifter, and 180-degree
hybrids. First, the signals are routed through the band pass filters so as to reduce effects of
reflections at unwanted frequencies. These filters have a 1GH• 3-dB bandwidth around the
center frequency. The signals then each flow into a three-wiy power divider. This is to
provide each of the two hybrids with that signal and to provide the given signal as an out-
put. The 180-degree hybrid is a device which takes two signils and provides their sum at 0
and 180-degrees out of phase, thus giving the sum and differtence. One of the hybrids takes
as inputs the two orthogonal signals, unmodified, and presents the + and - 45-degree linearly polarized signals as outputs. The other hybrid takes the horizontal input unmodified
and the vertical input after an extra phase shift of 90-degrees. In this way the outputs are
the left and right-hand circularly polarized signals, of the foim of equations 2.6 and 2.7.
3.3.6 Pre-Detection Amplification
The outputs of the microstrip circuit are then amplified obce more so that the power of
the signals presented to the square-law detectors will be moie than -27 dBm. This number
is from the specifications for the detectors used in the 13.8 GHz prototype circuit. In order
to calculate what this gain needs to be, the most lossy channe will be modelled in terms of
its temperatures and losses, much like section 3.3.4 above As in that section, the loss
before the matched amplifiers is 1.26, the gain of the matcled amplifiers is 316, and the
loss before the pre-detection amplifiers is 1.95. The lumped 4lements from Figure 3.5 will
be used in the following calculations.
P = k(Tant+ Trec) B = (1.38x10
23 )
(1100) (500x06) = 7.59x10l
) = 6.02x10 1 2
6)(3.)
After L:P = (7.59X1012)(
-12
AfterGI:P = (6.02x10F12 ) (316) = 1.90x10
6
(3.4)
(3.5)
9
(3.6)
9.76x1010
(3.7)
= (9.76x0 ol0 ) (G2 )
(3.8)
After L2 :P = (1.90x10F9)( ~)=
After G2: P = -27dBm = 1.99x10
12
And thus G2 = 2053 = 33dB. So if the pre-detection
plifiers used have over 33
dB gain, there will be more than -27 dBm power delivered to he detectors. The amplifiers
ordered for the prototype had 44 dB gain, and thus were well ver the minimum required.
3.4 Microstrip Design
One of the more complex redesign issues of the JPL airborne radiometers was how to
implement the adding circuit. It was determined that it w.s possible for the waveguide
switch network and its associated functions to be implemeoted in microstrip circuitry. In
this way, the size and weight of this element of the radiometers will both be reduced dramatically. Here, the microwaves are guided in a dielectric sandwiched between a ground
conductor and an overhead strip of copper, usually on the order of tens of mils thick (thousandths of an inch). It is constructed much like a printed cOircuit board, in that the lines
which are to constrain the microwaves are laid out in a CAD package and a photoresist
mask is made from this drawing. The mask is then used in an etching process to eat away
the copper that does not make up the lines where the signals will flow. Care must be taken
in the launching of the microwaves from a coaxial line into the microstrip circuit, for mismatches in dielectrics and air gaps can easily degrade the performance significantly.
3.4.1 Bandpass Filters
The bandpass filters to go in between the matched amplifiers and the adding circuit
were easily designed in microstrip. In this way, the size and weight of the radiometer is
further reduced. From earlier, the specification for the filters is to have a 3 dB bandwidth
of 1GHz around the center frequencies. The slope of the attenuation curve outside the
pass-band is not a critical factor in the performance of the circuit, so it was specified to
have about 20 dB attenuation at one-and-a-half 3 dB bandjidth's from the center frequency.
The band-pass filters were designed with a center frequency of 13.8 GHz to use in the
first prototype circuit. They were designed in a parallel coupled-line fashion, with values
calculated from [13], as in figure 3.6. It was determined that a third order structure would
MO-
Figure 3.6: The bandpass filter mask (AutoCAD), shown actual size.
provide the attenuation that was specified, with the thickness of the dielectric at 25 mils
and a dielectric constant of E = 2.2. The formulas from [Ib] provided the even and odd
mode impedances for each of the coupled line sections. Hewlett Packard's Microwave
Design Simulator (MDS) was then used on a Sun Sparc 10 toicalculate the physical parameters for the filter sections given the electrical characteristics and the substrate definition.
MDS then verified that the filter layout calculated from [13] led to the performance
expected. After a shortening of the ends of each of the couplod line sections to account for
fringe capacitance, the filter simulation performed exactly as expected. See Appendix E
for the simulation layout and results.
3.4.2 Power Dividers
After being filtered the orthogonal linearly polarized signals are routed into the adding
circuit proper. First, each goes through a three-way power diVider so that it can be sent to
both hybrids and straight to an output. These dividers were designed relatively simply in
microstrip by making an impedance transformation for a quarter-wavelength section and
then splitting the line three ways back to the original impedance. Because the signals are
being fed to the adding circuit from 50-ohm coaxial line i the microstrip circuit was
designed to have 50-ohm impedance lines. Thus, for a quarter-wavelength before the
power divider,
29 50 - ()
3.4.3 Hybrids
the
strip would
go
from 50 to 29-ohm
impedance,
since
The crucial elements of the microstrip circuit are the actual adders themselves, for here
is where the bandwidth of the circuit is limited the most. The first implementation to be
considered were the traditional "rat-race" hybrids, which take two signals and provide
their sum and difference over a bandwidth of about 10% of the center frequency [14].
However, their layout was such that in order to use the same inputs for two hybrids concurrently, the input lines would have to cross somehow. As doing this is rather unfeasible,
another method was sought.
It was noted that the 90-degree hybrid is rather easy to design in microstrip, and has a
performance similar to the rat-race design [14, 15]. Because the signals need to be onehalf wavelength out of phase at the output of the hybrids, a few extra delays added to a 90degree hybrid can make it perform like a 180-degree hybrid. If two extra one-quarter
wavelength delays are added, on opposite inputs and outputs (as in figure 3.7), it will provide the sum and difference of a pair of signals. These delays are realized as extra 90degree lengths of microstrip line.
Inputs
90
o
Outputs
-(A-B)
B
A+B)
Figure 3.7: The modified 90-degree hybrid coupler without magnitude factors on the outputs.
As with the band-pass filters, these coupler designs were laid out and simulated on
MDS. The substrate they were designed for had a dielectric constant of 2.2, so that any
errors in the lengths of the lines would translate into smaller phase errors. This is because
the higher the dielectric constant in a waveguide, the shorter the length for a given electrical distance [14]. In order to keep the width-to-length aspect ratio of a 90 degree transmission line low, a thinner dielectric than the band-pass filters was used. In this case, a
thickness of 5 mils was chosen so the length of a quarter-wavelength section would be
about 155 mils with a width of about 15 mils. As in figure 3.7, the series lines of the
hybrid coupler appear wider because they have lower impedance (35.4 ohms) than the rest
of the coupler (50 ohms) [14].
3.4.4 Layout And Simulation
The microstrip power dividers and hybrid couplers were laid out on the same piece of
duroid with the 5 mil thickness and dielectric of 2.2. MDS was used in two modes for the
simulation: first, just the layout with the electrical characteristics was used, then a better
simulation was run in which the physical parameters are used. The first simulation is basically to make sure the numbers for the elements are in the right ballpark, while the second
is usually the more accurate one. In the second simulation, the actual geometry of the layout becomes important, for here is where the microstrip discontinuities are factored into
the performance of the circuit. These include 90-degree bends, tee junctions, and steps in
width.
The results of the simulation proved that over a 1 GHz bandwidth around the center
frequency, the relative phases of the six outputs varied by a maximum of about 16 degrees
from the expected value (see Appendix F for the layout and results of the MDS simulation). The magnitudes of the signals were as expected, for the simulation was run as
though the entrances and exits from the microstrip circuit were perfectly matched.
Chapter 4
Construction And Testing
4.1 Microstrip Circuit Construction
After the design of the radiometer was completed, construction was begun on the most
complicated part of the circuit first, the microstrip networks. The goal was to complete as
much of the construction and testing of the radiometer as possible before the completion
of the internship period at the Jet Propulsion Lab in Pasadena, California. Unfortunately,
there was only time to completely build and test the 13.8 GHz prototype circuits. Plans
were drawn up and simulations left for the JPL researchers to continue the construction of
an 18 GHz spaceborne radiometer.
4.1.1 Laying out the Mask
The first step after the simulations were complete was to actually draw those transmission lines to actual size in a CAD package. In this case, AutoCAD was used for both the
layout of the bandpass filter and the adding circuit. Using the lengths of the various transmission lines from the MDS simulations, the CAD layout became a simple matter. The
only stumbling block occurred during the drawing of the adding circuit. The problem was
achieving the correct phase ratios at the inputs to the hybrid couplers. Because there had to
be an extra 90-degree phase shift to get the circularly polarized components, the adding
circuit could not be totally symmetric. For this reason, transmission lines into one hybrid
had to be lengthened and shortened to compensate for each other and for the extra phase
shift needed (see figure 4.1 for the adding circuit drawing). After completion of the CAD
layout, the file was sent to a local contractor to create a positive and a negative photoresist
mask for the etching procedure.
Input Tv Output Tv
Output Tp
L
Output
~:::-T M
Output TL
Output TR
Input TH Output TH
Figure 4.1: Microstrip adding circuit, actual size.
4.1.2 Etching
After securing the appropriate duroid and receiving the photoresist mask, the two were
taken to the etching lab located at the JPL facilities. There, the mask was used to cover the
areas on the top of the microstrip that were not to be etched away while it was immersed in
a chemical bath. The pieces of duroid were returned and appeared to match up well with
the CAD drawings used to create them.
4.1.3 Mounting and Connectors
Once the duroid had been etched, the next step was to decide how to launch the microwaves into the circuit from the 50-ohm coaxial lines they will be coming from. Due to
inexperience with this issue, the mounting hardware designed first was faulty. It was
decided to bond each circuit to a base plate of aluminum with conducting epoxy, and then
to place that into a shielded box with coax-to-microstrip connectors on it. An example
connection from [17] can be seen in figure 4.2. The design error made was the choice of
coax-to-microstrip connectors used, and not the overall scheme. In the case of the filters,
the error was not that large, and so below, the focus will be on the adding circuit problems.
First, the connectors used were SMA "tab" connectors, with a female coaxial jack on
one side and a small gold plated tab sticking out of the cylindrical dielectric material of the
coax on the other. Due to time considerations outlined earlier, the only suitable connector
available was one in which the radius of the dielectric surrourding the tab was about ten
Launch Irogon
ctor
nmrostrlp
I-
~FlU
0
I
1M 11/~IhI
I
WEE
''
rYr~lYrlll
rmYY~ld~lEW
~u~eA
p
rr:n
&
Ag
Fixture Base
Figure 4.2: Transition from coaxial line to microstrip; the important ground contact, point
A, is also shown [17].
times the thickness of the dielectric in the microstrip circuit. So, in matching them up,
there was a decision to be made whether or not to align the conductors (the microstrip
transmission line and the tab) or to align the grounds (the ground plane of the microstrip
and the coax shielding). The alignment will have a grave impact on the performance of the
circuit because in each case, there will be some portion of the incident wave's energy
reflected by either an air gap, or by what looks to the wave like a short.
Upon consultation with colleagues (all of which, incidentally, had no experience with
coax to microstrip transitions) it was decided to align the ground conductors. Thus, there
would be mismatches due to an air gap where the tab was bent down to the surface of the
microstrip. It was thought that even with the losses caused by the mismatches, the operation of the circuit could be verified. Only after the true reflection coefficients were realized
in the testing was a search undertaken to find a vendor of specialty connectors for thin
microstrip.
4.2 Testing the Microstrip Circuits
For each piece of hardware, a suitable enclosure with SMA connectors for coaxial input
and output was constructed for the testing. A Hewlett Packard I P8510 Network Analyzer,
which injects electromagnetic waves into a device and measures its response, was used for
the testing. The analyzer is capable of carrying out a variety of measurements, including
VSWR, loss, and numerous phase details. For the circuits rnder test here, the important
measurements were loss and phase delay.
4.2.1 Band-Pass Filters
The filters were designed and built on the same substrate, next to each other, so they
could be used in parallel for each input channel of the radiometer. However, because of the
presence of just one network analyzer, each channel was tested in turn. On the first design
pass, the filters appeared to have the correct rolloff and the' right size ripples in the pass
band, but the center frequency was skewed. Unfortunately, in the CAD stage of laying out
the circuit, the wrong amount was trimmed from the ends of the parallel lines to negate the
effects of fringe capacitance. Once this was realized, the filters were simply changed in the
AutoCAD drawing so that new masks could be made. Due to time constraints, the updated
filters could not be built at 13.8 GHz, and thus were redesigned for the 18 GHz implementation to be continued after the absence of this researcher.
4.2.2 Adding Circuit
The adding circuit was tested with 50 ohm terminations on the SMA connectors not
being used with the analyzer. In this way, unwanted reflections could be minimized. If the
relative phases from both inputs to a given output pair of a hybrid were close to 0 and 180
degrees apart, the operation of that piece of the adding circuit could be verified. On the
other hand, the absolute phase delays from one input to the hybrid outputs was not a critical measurement of the circuit's performance. This is because each sum and difference
output is amplified and detected separately, and does not need to be in phase with any
other. The phase delays to the "straight through" outputs (the 0nes that do not pass through
a hybrid) really do not matter, but were measured to check tl eir adherence to the simula-
tion results. In addition, the reflection coefficients from the inputs to each output were
measured to determine how well the coax-to-microstrip lautnchers were performing.
The first results from the calibrated analyzer measurements showed that the reflection
coefficients at the inputs and outputs were larger than expected. Apparently, the reflections
at the edges of the circuit were amplified by the air gaps and the tab connectors being used
improperly. These reflections limited the amount of power 'being transferred through the
circuit and thus increased its loss. This would need to be remedied, for large losses which
are unequal in the two linearly polarized input signals will cause unequal phase delays
through the hybrids.
The measurements of the phase delays were favorable. The relative phase difference
for an input pair to a sum and difference output were 180 (+/- 14) degrees apart over a
500MHz bandwidth, as predicted by the simulation. However, these delays were not absolutely 0 and 180 degrees. Instead, the phase difference at each output from both input ports
seemed to be skewed an equal amount away from 180 degrees. In the case of the circular
polarization hybrid, both were skewed by 60 degrees and in the linear polarization hybrid,
they were delayed by 20 degrees. The absolute phase delays from the pair of inputs to both
outputs seemed to behave more like the simulation. In their case, the phase differences
were 0 (+/- 10) degrees from one input, and plus-or-minus 180 (+/- 15) degrees.
At this point in time during the internship at JPL, the deadline was approaching for
returning to classes. Therefore, all the microstrip circuits were redesigned at 18 GHz and
re-simulated. This enabled many of the problems encountered in the first prototype design
to be corrected.
Chapter 5
Conclusions
5.1 Test Results
The results from the tests outlined in the last chapter demonstrate the functionality of the
bandpass filter design used here. The parallel coupled line design is one that has a rich history of use and testing, and is reliably simulated. It was noted that the shifted center frequency in the first tests of the 13.8 GHz filters was due to the fringe capacitance off the
ends of the lines. To verify that this was the case, the filter design was re-simulated without trimming the lines. The results of this new simulation proved that without shortening
the lengths of the lines, the center frequency would indeed be shifted almost 1 GHz, as in
the tests of the constructed circuit.
As mentioned at the end of Chapter 4, the filters were re-designed to support an 18
GHz center frequency. In the AutoCAD layout of this circuit, the lines were doublechecked to make certain that the correct amount was trimmed. According to the MDS simulation of this layout, as long as the coaxial-to-microstrip connectors were performing correctly, the filters will work as planned.
The results of the adding circuit tests were slightly harder to interpret. As mentioned
above, the relative phase delays were skewed by 60 and 20 degrees. The most likely
source of this disturbance was determined to be the extra loss caused by the mismatches in
the coaxial connectors. These mismatches caused different losses in each input channel
and thus forced the hybrid couplers to output the sum and difference of unbalanced signals. This, in turn, caused the incorrect phase response between the input pairs and the two
outputs of each hybrid. Other than the phase problems, the adding circuit VSWR's were
larger than expected. Again, this is because the simulation assumed a perfect match
between the coaxial inputs and the microstrip inputs.
5.2 Trials And Tribulations
In designing an enclosure for the 13.8 GHz adding circuit, several problems were encountered. First, a box supplied by JPL stores was found to fit the circuit perfectly, except that
after building the box interface, the tabs of the connectors did not quite reach the surface
of the microstrip. This was because the thickness of the box was just enough to cause the
tabs to be too short. In order to fix this under the time pressure of the internship assignment, small copper extensions were cut and soldered onto the ends of the tabs to reach the
microstrip. This may have been another factor degrading the performance of the coaxialmicrostrip interface. Of course, in the mechanical design for the 18 GHz circuit, a custom
box was designed at a thickness given to safely support the custom thin-microstrip SMA
connectors ordered. These custom connectors were designed with a transition interior to
the hardware that would taper the dielectric radius of a coaxial cable to a thickness on the
order of 20 mils (which was the thickness of the re-designed 18 GHz adding circuit)
around the microstrip launching pin.
In the re-design of the actual microstrip adding circuit at 18 GHz, the substrate thickness was increased so that even with the new connectors, precise alignment would not be a
problem. Therefore, the mismatches between the connectors and the microstrip would be
minimized. However, this posed several difficulties. The most important of these was the
fact that as the dielectric becomes thicker, the width of the transmission lines must
increase to maintain the same impedance. Thus, because the concern over length errors
causing smaller phase errors was global, the line impedance was changed rather than the
dielectric constant. It was a straightforward matter to scale up the impedance of all the
lines in the microstrip circuit. The only addition to be made was a transition from the 50
ohm SMA connectors to the increased impedance of the main microstrip circuit (150
ohms). To do this, a simple quarter-wavelength step transformer was used, designed from
the equations in [13].
5.3 Further Study
The basic design of a spaceborne radiometer, including its block diagrams and the beginnings of detailed subsystem design have been presented here. Many opportunities exist for
the continuation and extension of this work. For instance, the design of the microstrip circuits (which were used as a smaller, lighter, and more parallel option to the operation of
the airborne radiometers from JPL) at 18 GHz need to be verified in ground tests before
they can be integrated into the existing airborne radiometers. The stability over wide temperature ranges of the microstrip circuits and of the spaceborne radiometer as a whole
need to be tested. This would include monitoring the gains of the various amplifiers as
well as the mechanical interface between different components, such as the microstrip circuits and the rest of the radiometer.
The design of the radiometer should be updated for use on a particular spaceborne
platform. Here, a generic orbiting satellite was used in the footprint and integration time
calculations, yet a more specific set of specifications should be used in the design for a
true spaceborne test. The details involved in this, with respect to how the radiometer is
mounted, could easily constitute another design study. Similarly, based on the actual spaceborne platform, a more detailed calibration scheme needs to be determined. This might
include what would be used for the cold and ambient loads, how the noise diode would be
calibrated and how often, etc.
Appendix A
Spaceborne Polarimetric Radiometer Types At 18 GHz
Gyrator
Correlator
Parameter
Bandwidth
100 MHz
180 MHz
Tsys + Scene
450 K
800 K
Integration Time / Pixel
55 ms
18 ms
RMS Noise, AT
0.28 K
0.4 K
Polarization Isolation
-25 dB
-20 dB
Gain/Phase Stability
Medium
Good
Calibration Stability
< 0.5 K
< 0.5 K
Absolute Calibration
< 1K
<2 K
Power
10 Watts
4 Watts
Size
Digital circuitry reduces
size
Smallest option (nonparallel operation)
Table A.1: Spaceborne Radiometer Comparisons
Parameter
6-Channel Polarimetric
Bandwidth
500 MHz
Tsys + Scene
450 K
Integration Time / Pixel
55 ms
RMS Noise, AT
0.17 K
Polarization Isolation
-25 dB
Gain/Phase Stability
Good
Calibration Stability
< 0.5 K
Absolute Calibration
< 1K
Power
5 Watts
Table A.2: Spaceborne Radiometer Comparisons (continued)
Parameter
Size
6-Channel Polarimetric
Moderate due to 6 channels
of microstrip and analog
hardware
Table A.2: Spaceborne Radiometer Comparisons (continued)
Appendix B
Footprint / Swath Coverage Calculations
At 18 GHz, the wavelength is X = 0.01667 m. The spacecraft height is h = 800 km, and
the incidence angle is Oi = 650. The pointing angle of the antenna, with respect to nadir,
is given as:
) = asin
63 78
sin .
(B.1)
Substituting the values above, we find the pointing angle to be: 9p
= 53.60. Continu-
ing, the footprint calculations are as follows:
Swath width, W = 4 x 6378 x sin (O - 9Op) /2) = 2533 km.
Range, R = 6378 x
sin (i i
sin (Op)
p)
1566 km.
We want the footprint width to be 50 km, so solving for the bea mwidth in the next
equation gives us:
Footprint Width, Fw = R - tan (OB) = 50 km, so:
Beamwidth, eB = 0.0318 rad, and since ®B = 1.22 x
(wheire D is the antenna
diameter), we have D = 0.61 m.
Footprint Scanned Width, Fws = 1.4 x Fw= 70 km.
Footprint Length, L = Fw/ (cos (Op)) = 84.2 km.
Circumf nXx W
= 114.
Number of pixels in circular scan, P = Circumf
Fws
Fws
Seconds per revolution of scanning assembly, Srev = ( (0.5 x FL) /v) = 6.3 s (where
v is the velocity of the spacecraft, given as 6.7 km/sec).
Thus, the maximum integration time per pixel,
t
=
rv= 55 ms.
P
Appendix C
Calibration Measurement Procedure
C.1 Overview
The calibration of the radiometer occurs in two distinct stages. First, the noise diode is
calibrated, so that an accurate number can be used for its temperature. Then, during the
radiometer scanning, the equivalent noise temperature for the system is calculated to
establish a calibration line.
T_
T
Vout
Figure C.I: Simplified Radiometer Block Diagram
In the above figure, each element has a noise temperature associated with it. In addition, the path from the antenna to the noise diode directional coupler has a loss; similarly,
the other potentially lossy element (with the exception of the directional coupler loss
which is negligible) is the microwave circulator, which has a loss term associated with it
as well.
C.2 Noise Diode Calibration
In this process, the objective is to characterize the noise temperature of the diode used for
the calibration of the radiometer system temperature. Because its stable noise temperature
is a crucial element in the system temperature calibration, an accurate representation is
crucial here. As in previous spaceborne configurations, the hot and cold load calibration
measurements would probably be done with a reflector to cold space and some type of hot
reference load. In the tests at JPL, this calibration was done on the ground prior to a flight
on the DC-8 with a cold load (bath of liquid nitrogen) and an ambient load.
The first measurement is with the Dicke reference load:
(C.1)
Vout = Vref = (Tref + Trad) G +Z.
Next are the two load plus noise measurements. First, the ambient load plus noise
diode measurement is taken. For the spaceborne application, this would be the "hot" load.
Vout = Vamb =
+T L 1- L
+s
+Tnd
(1 -L
+
ad (G) +Z
(C.2)
Next, the cold load plus noise diode measurement is taken:
+ TL
Vout = Vcold
-
+ Tnd
+ Ts 1 L
Ls)j+
+TLP
T
p
+ Trad (G) + Z (C.3)
We then combine equations C. 1 through C.3 in the following manner:
(C2)-
(Cl)
+ TL
+ T ndL
-
(Tcold
+ Ts1 -
1
- TrefC
(C.4)
Tamb ) L L s
Then, cancelling the denominator, and subtracting all the numerator elements except
the noise diode term,
Tnd
L-
(C4)
Tcold Tamb
(C4)
s-LpL
-
bLs(
1(C.5)
)+Ts( 1
T
".
(C.5)
And thus:
Tnd = (C5) L s .
(C.6)
C.3 Radiometer System Temperature Calibration
The equivalent noise temperature of the radiometer is calibrated once each circular scan of
the spaceborne antenna assembly. It is done with four measurements: one with the Dicke
switch to the reference load, two of an ambient load (with and without the noise diode
coupled in), and one with the radiometer off (to calculate the zero offset).
First take the reference load measurement,
Vout
=
(C.1)
Trad) G + Z.
Vref = (Tr+
Next are the two ambient load measurements. On the spaceborne platform, these can
be implemented by a reflector to cold space.
+
Vout = Vamb +n
Vout =
Vamb
+ TndL + T, 1
L
+ TL
n=
L -
4
+ Ts 1 -
+ Trad ) (G) + Z (C.2)
+ Tad (G) +Z.
(C.3)
Finally, the zero offset measurement is taken:
Vout = Vo
(C.4)
= Z.
the four measurements, then cancelling terms to solve for the unknown radiometer system
temperature.
(Cl) - (C4) _ Tef+ Trad
(C3) - (C2)
Týd
=(C1)
(C.5)
Tnd
Tnd
(C3) -- (C4)
(d
(C2) LS -
ref
(C.6)
Appendix D
6-Channel Radiometer Block Diagram
>
I
V
.+
v
OC
I.
Appendix E
13.8 GHz Band-Pass Filter Simulation
E.1 HP Microwave Design Simulator Layout
0>
N-If -,X
(1 •
z
252
V:C
r
="
u
.n Ij
W
2
Uj
KxI
Q
I,~
Ct
I
In W W
EE
nr
-de
I
LO
00
XU
II
E.2 Simulation Frequency Response
t.P
JDa dase
T I
Iter
LJA
Qual ifier=~
-IL
N
Clu
-c
1
o
I
11.0
GHz
freq
I
17.0
GHzA
Appendix F
13.8 GHz Adding Circuit Simulation
F.1 Adding Circuit Block Diagram
T..
Tn
TL
TR
Tp
TM
T, •
'V
F.2 MDS Layout
IS
BV
. IS .
oi~
19'
*U
m
Iam
.1
F.3 MDS Simulation Results
CM
NN
m m
Lo -
L
L
NN
mm
m m
II
@3
4-
E'021
s )
B2S/BE
seP4d
L2S/LE s ) 2sv4d
0'0920"2
0
0
Im
NN
L Mr,
ar
G3 w
L- L.
NN
=DU
mm
-a
(.•or
-a
3
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80' I
( 3BOS,8SS)aSpyd
ýjS/ýGS)2sUtd
(,O
O*OG2
0*9-
L
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NN
cu
o
NSIN
mm)
m
m m
• 0201
80CH-
(LES/82S)G•s24d
(LES/BE s )as24d
0 '28
04911-
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