Journal of Electrical and Control Engineering
Feb. 2013, Vol. 3 Iss. 1, PP. 13-18
Analysis of Power Line Communication in Vehicle:
towards a Vehicle PLC Demonstrator
Fabienne Nouvel, Philippe Tanguy
IETR/INSA, 20 Avenue des Buttes de Coesmes, 35709 Rennes, France
fabienne.nouvel @insa-rennes.fr; philippe.tanguy @insa-rennes.fr
Abstract- This paper deals with in-vehicle power lines and the
use of emerging power line technology (PLC). It appears that
with the increase of electronic devices unit (ECU) there is a
wire harness bottleneck. So, PLC seems to be a promising invehicle network for high data rates applications. In-vehicle
PLC channels and noise measurements are presented. We
discuss the PHY layer PLC system parameters based on these
measurements. Finally, an embedded demonstrator is
proposed in order to improve them in a real in-vehicle channel
environment.
Taking into account those in-vehicle measurements, the
channel characteristics will allow us to define the best
parameters for the signal processing and communication
system, like the bandwidth, the modulation, the equalization.
This study is discussed in Section 3.
After simulation using the measured channels, it seems
to be interesting to test several algorithms under real
channel while keeping flexibility. That’s why we propose to
study a fast prototyping solution to perform this task using
software defined radio principle (SDR). In this case, the
PHY layer is developed in software thanks to Maltab. We
used the SDR platform from ETTUS. This demonstrator is
presented in Section 4. Finally, we will conclude the paper
in Section 5.
Keywords- Power Line Communication; OFDM; Impulsive
Noises; Vehicle Environment
I.
INTRODUCTION
The number of electronic devices (ECU) in vehicles has
experienced an exponential growth in the last decades [1] [2].
It has leaded the automotive industry to introduce dedicated
buses as CAN or Flexray. Although they reduce the amount
of wires, they use specific wires. However, the use of the
power distribution channels inside vehicles both for power
and communication purposes is a promising alternative. It
would answer the vehicle requirements namely cost,
decrease of the amount of wires and weight, and offer more
flexibility and bandwidth.
II.
MEASUREMENT METHODOLOGY
In the first part of this section, we focus on PLC
channels measurements. We will analyse them both in the
frequency and time domains. Additionally, tests with indoor
PLC modems prove the feasibility of such solutions under
in-vehicle PLC wires.
A. Setup Measurement
Taking into account both the In-Vehicle Infotainment
(IVI) and X-by-wire (X means any mechanical system)
requirements, we can observe the necessity to find a limited
set of networks which answer to the growing of the multiple
applications. These new networks may be able to
dynamically optimize their parameters according to the
loads on the network (active ECUs), the data rate needs
(bandwidth sharing), the state of the vehicle (motor
ON/OFF, vehicle in motion, speed),... Among alternatives to
existing on-board networks, power line communications
(PLC) seem attractive. Many studies are carried out on PLC
and focus both on channels and noise in order to optimize
the PHY layer and the MAC layer [3].
Characterization of PLC channels has been reported
according to two major scenarios on four vehicles: front to
rear, front to front, presented in detail in [6] [8]. Other
similar studies have been carried out in [9] [10] and focus
both on PLC channels, impulsive noises, and PHY layer.
The main objectives here are to obtain a data base of
measurements on several different vehicles (gasoline, diesel,
and next electric, recent and old vehicles). The test bed is
represented in Fig. 1. The points [A..I] represent both the
communication and measurement nodes; a link between
point X to Y point is called XY path. The frequency ranges
go from DC up to 50 MHz. The vehicles are always in
motion and equipments operate randomly. The vehicles we
have considered are two 407 SW (recent car, diesel and
gasoline), a Renault Laguna II Estate (gasoline), a Renault
Espace (diesel) and a Citroën C3 (gasoline).
In indoor and during the last decade, the main thrust of
the research into PLC has been focused on low voltage (LV)
electrical power distribution networks, which have also
geographically the widest spread. Today, PLC focuses both
on high and medium voltage (HV and MV). More recently,
many studies are carried out on DC (12/42 V) voltage for
embedded application in vehicles (cars, aircraft) [4] [5] [6] [7]. It
is necessary to study in-vehicle DC channels to extract the
main parameters for the PHY and MAC layers. Section 2
provides a description of the experimentations under invehicle DC electrical wires and wireless environment. We
will focus on both transfer functions and noises under those
DC lines.
In order to analyse the channel transfer function, the Sparameters are recorded using a full 4 ports Vector Network
Analyzer (VNA) with 50 Ω reference impedance and a PC
interfaced to remote the device. The S-parameters are
recorded about every 10 seconds for 3 different paths: GF,
GH and HD. We record the S-parameters during about 10
minutes while the car is moving with the equipments like
wipers, lights, brakes. These paths have been chosen in
© American V-King Scientific Publishing
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Journal of Electrical and Control Engineering
Feb. 2013, Vol. 3 Iss. 1, PP. 13-18
order to analyse the differences between front to front and
rear to front paths.
36 dB in the considered band. The maximal attenuation
occurs with the longest past GF. The coherence bandwidth
BC0.9 (90% of the maximum sub-carriers autocorrelation)
can be calculated from the autocorrelation of the channel
frequency obtained with different paths [11] [12]. The channel
frequency response H(f) depends of both the sources and
loads and is correlated with the transfer function S21 by the
approximated relation:
Regarding the noise, two different noise studies have
been carried out on the same paths. The first consists of the
measurement of the power spectrum in the frequency
domain at each point during 10 minutes every 10 seconds
while the vehicle is in motion. The second is a measurement
in the time domain. To do so, a digital storage oscilloscope
(DSO) has been used to record at each point the signal over
the DC line. With this testbed we are able to record two
signals at two different points at the same time. Thus we can
observe the level of noise at the two different points
simultaneously.
H ( f ) ≈ S 21
(1)
As presented in [12], the mean BC0.9 is greater than 500
KHz and can reach 2 MHz. Furthermore, we have observed
its value is not correlated with the path’s length. For
example, BC0.9 values are respectively equal to 533 KHz
and 4.7 MHz for paths GF (≅ the longest path) on 407SW
and Laguna vehicles and equal to 2 MHz and 744 KHz for
path HD (≅ the shortest path) on the 407 SW and Laguna
vehicles.
As for indoor measurements, three main parameters will
be derived from the results: the insertion loss (ATT); the
coherence bandwidth (BC0.9) which is a direct measure of
the channel selectivity over a value of 0.9; the RMS delay
spread ( τ RMS ) that estimates the channel time dispersion.
Additionally, the RMS delay spread is calculated as
defined in [3] [11]. In order to define the maximum delay
spread a threshold of -30dB has been chosen. If we compute
the cumulative density function of the RMS delay, we can
observe that 90 % of the channel measurements have a
τ RMS delay spread lower than 210 ns with a smallest value
of about 50 ns.
If we compare our results with the results obtained in
indoor [11], BC0.9 in vehicles is as twice as large as those
obtained in indoor; the mean BC0.9 is of the order of 291.9
KHz in indoor. Keeping the same transmission bandwidth,
one can suggest the sub-channel spacing can be reduced. In
the case of Orthogonal Frequency Division Multiplex
(OFDM) technique used, the FFT size may be reduced and
the OFDM symbol duration will be shorter. With regard to
delay spread, the delay spread obtained in vehicles
(maximum value of 242 ns) is two times shorter than the
mean value in indoor (0.413 µs). This will allow us to
reduce the Cyclic Prefix (CP) length and therefore increase
the data rate. These initial results confirm that the PLC
communication parameters defined for indoor must be
optimized for in-vehicle PLC.
Fig. 1 In vehicle Test-bed for channel and noise measurements
B. Channel Measurements
Fig. 2 illustrates the channel transfer function obtained
on the vehicles in the [0-30] MHz band. Additional
measurements have been performed up to 50 MHz [8].
C. Noise Measurements
Fig. 3 illustrates the noise we observed on the DC
vehicle lines.
Fig. 2 Examples of transfer function for 3 different paths
Taking into account all the database measurements, we
have observed an insertion loss ATT of about -15 dB up to -
Fig. 3 Noise on the different paths at different points
© American V-King Scientific Publishing
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Journal of Electrical and Control Engineering
Feb. 2013, Vol. 3 Iss. 1, PP. 13-18
We can see the high level in the [0-5 ] MHz and in the
[12-15] MHz band, which overlap with the bands used by
indoor PLC modems. We can notice the low level in the
[30-50] MHz band. Like in [13] we have applied to the
noise recordings in-vehicle a time frequency analysis. We
have computed the spectrogram with short-time Fourier
transform and an Hamming window of length equal to the
length of HPAV OFDM symbol (40.96 µs) and an FFT size
of 3072 points [14]. We remark that some of them are always
present in any scenarios. In [15], the authors have presented
a detailed analysis of the periodic noise on in-vehicle power
lines in the frequency band up to 25 MHz The employed
methodology exploits the harmonically related peaks that
appear in the estimated PSD (Power Spectral Density) when
periodic noise components are present. It has been found
that the number of periodic components varies from one or
two (when the engine is OFF) up to 7. Two of them have
been found in all the vehicles. A common periodic noise
component with a repetition rate of approximately 64.75
kHz (with variations of up to hundreds of Hz between cars)
has been found in the noise registers captured in the cars in
all the engine states. Such behaviours have been observed in
our measurements with about the same rate. This value can
be used in future work to define the best channel coding.
For DC-line, we assume first a constant transmit power
signal density (PSD) of -60 dBm/Hz in the [4-30] MHz
range and 0dBm outside. The additive AWGN has a PSD of
-110dBm/Hz. We consider a sub-carrier spacing of 24.4
KHz and a sampling rate of 75 MHz. These parameters are
close to HPAV settings [14]. A bit loading algorithm with a
maximum of 10 bits per hertz is accomplished. For the
channel, we have considered the channel’s measurements
described in previous Section 2.
Taking into account those default values, two parameters
are studied and optimised: the guard interval CP and the
FFT size.
A. Guard Interval CP
As the CP length will impact the capacity, its duration is
first optimized in conjunction with the bit loading algorithm
presented in [21]. In fact, the maximum capacity for the
optimal CP is given by:
ROFDM
2
 N −1


Hm σ 2
N FFT


  (2)
=
∆f ∑ log 2 1 +
 Γ(σ n2 + N ISI + ICI )  
N FFT +CP  m =0



With σ the signal power, σn the noise power, H m the
channel coefficient at frequency m, Γ the margin gain for a
BER of 10-3, N the number of used sub-carriers (for
example 1148 tones with a FFT size of 3072). The
interferences N ISI + ICI are assumed first null. Considering
Taking into account those results, we can now focus on
the PHY layer parameters for the communication system.
III. IN-VEHICLE PLC PARAMETERS
In this sub-section, parameters that maximize
performances of the OFDM system are assessed. In our
study, OFDM technique has been adopted as for indoor PLC,
thanks to its high frequency diversity. OFDM modulation
can be realized through the IFFT/FFT processing block to
which the original stream is applied. Several complementary
operations are achieved to the information bits before they
are submitted to the IFFT processing. As presented in
Section 2, the in-vehicle channels are frequency selective,
noisy and multi paths will affect the transmission. For these
reasons, OFDM is a good candidate. This solution is applied
in other systems like in [16] [17]. Additional information on
this technique can be obtained in [18].
a SNR of 40dB and a BER of 10-3, we have noticed the
capacity achieves a maximum of 500 Mbps/s with a CP of
200 ns (15 samples), 186 (14 samples) and 133 ns (10
samples) respectively for paths GF, GH and HD. The higher
the channel attenuation is, the higher the value for CP is that
maximizes capacity as for high SNR the performances are
most driven by interference mitigated by long CP.
The transmitted OFDM waveform can be expressed as:
s (t ) =
1
Np
N p −1
∑ R {c
m =0
eal
m
∏(t )e 2 jπFmt
}
(1)
With NFFT the FFT size , Np= NFFT /2 , cm the complex
symbol on sub-carrier Fm. The sub-carrier spacing ∆f is
defined as 1/TOFDM with TOFDM the OFDM symbol duration.
In order to generate a real OFDM signal, the Hermitian
symmetry is first applied before the IFFT of size NFFT.
Fig. 4 Throughput according to engine state, CP optimization and BER
constraint
The OFDM parameters are based on home network ones.
For PLC, several standards with different kinds of multicarrier modulations like HPAV and HD-PLC [19] are already
proposed and investigated. Those standards have been
applied in car to measure data throughput on DC channels
[20]
.
Then it is possible to combine the CP length adaptation
with bit loading algorithm [10] while keeping 1148 used tones.
Three different engine states are considered as previously
discussed. Fig. 4 shows the theoretical data rate achieved
with this algorithm. We can observe the data rate decreases
when the engine is turn on, except for paths HD and FD.
© American V-King Scientific Publishing
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Journal of Electrical and Control Engineering
Feb. 2013, Vol. 3 Iss. 1, PP. 13-18
These two paths are located in the front area and closed
together. These simulations are closed to the measurements
carried out in [6]. These comparisons between
measurements and simulations demonstrate the accuracy of
our functional model.
A. Demonstrator USRP2 Board
The VPLC platform is based on USRP2 boards
combined with daughter boards [25]. Using these daughter
boards, the possible operation frequency range is very
modular (from DC up to 5.9 GHz). The mother board
includes ADC, DAC, a FPGA (Xilinx Spartan XC3S2000)
and a Gigabit-Ethernet interface with the PC. The two slots
of the board are for the front-end daughter boards. The two
14-bit ADC and the two 16-bit DAC of the daughter board
are independent and can work up to 100 Msps. We can also
combine them if we apply the complex baseband OFDM
symbol (I and Q) at the DAC entries rather than a real
baseband DMT symbol (Q=0).
B. FFT Size Optimisation
We have considered the FFT size for the same scenarios.
Considering the BC0.9 we have obtained, the FFT size may
be reduced. Four FFT sizes, from 3072/1536 used (∆f =
24,4 KHz< BC0.9) down to 128/64 used (∆f = 585 KHz <
BC0.9) tones are rather interesting. Fig 5 gives the data rate
we can achieve using those different FFT sizes. The optimal
CP is respectively equal to 14 and 50 samples for FFT= 128
and FFT=256, 1024, 3072.
To interface with the DC lines, we use the LFTX and
LFRX cards because they allow transmitting and receiving
signal from DC up to 30 MHz. These daughter boards
include differential amplifiers and low pass filters for
antialiasing. Fig. 6 illustrates the demonstrator. The outputs
are linked to the DC lines through a lower band filter and
transformer.
We can observe that when the FFT size increases (FFT >
1024), the ratio (NFFT/(NFFT+CP)) is closed to 1, the data
rate tends toward a fixed value. We can conclude it is not
necessary to increase the FFT size to increase the data rate.
If we consider a BER of 10-3, a SNR of 35 dB is necessary
for all the FFT sizes while keeping the sub-carrier spacing
lower than the BC0.9.
Fig. 6 VPLC platform with USRP2 Boards
B. PHY layer
The OFDM symbols are generated and organized in
frames. It is necessary to generate a synchronization symbol
in order to detect the beginning of the useful signal. The
transmitted frame is given in Fig. 7.The number of
OFDM/DMT symbols can be configured by the user
according to the experiments.
Fig. 5 Data rate according to the FFT size
Following this system communication analysis, it is now
necessary to integrate these communication solutions in an
embedded network. A demonstrator using the parameters
will be presented in next section.
IV. IN-VEHICLE DEMONSTRATOR
Software-defined radio (SDR) [22] has been defined to
help the wireless operators to maximize their investments in
multiple mobile standards and base stations In our vehicular
system, SDR can be considered for a “wired”
communication system, where some of its functional
components, such as modulations, equalization, etc., will be
implemented in software. This makes it possible to
configure the signal according to the requirements of the
application and the characteristics of the communication
channel (wired or wireless). The software generated signal,
thanks to Matlab and the GNU-Radio user interface and
according to the parameters defined previously, is then
applied to an USRP platform connected to a host computer
through Ethernet connections. Next part will describe our
VPLC (Vehicle PLC) demonstrator using this approach.
Details of the implementation are given in [23] and [24].
Fig. 7 PLC DMT frame
The number of DMT symbols is configured according to
the performances we want to test, with a default value of 20.
The time synchronization symbol includes a pseudo noise
sequence with good autocorrelation property of length equal
to (L=NFFT/4) and the CP. The synchronization symbol
includes consequently four sequences. Then all the DMT
(real OFDM generation) symbols of the file are transmitted
through the board. The synchronization symbol is organized
as:
[ P P –P –P]
© American V-King Scientific Publishing
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Journal of Electrical and Control Engineering
Feb. 2013, Vol. 3 Iss. 1, PP. 13-18
to front, through the battery with 3 CP lengths (139/60/20
samples). First, the SNR estimation is performed over the
bandwidth using QPSK symbols on each sub-carrier. We
use the VPLC demonstrator and the DMT signal is software
generated. As the channel can be considered as timeinvariant, the bit loading algorithm will allocate the bits on
the subcarriers at the transmitter side. Taking into account a
referred bit error rate of 10-3 and a transmitted power of 80dBm/Hz, we can achieve a mean data rate of about 30
Mbps when the motor is OFF and 7 Mbps when it is ON.
This lower result is caused by the lowest SNR when the
motor is ON (lower than 5 dB). When we modify the CP,
we obtain a little variation according to its length. We
respectively obtain 33 Mbps and 30 Mbps with CP=60 and
CP=139.
With P the IFFT of the pseudo random sequence of size
L.
At the receiver, the synchronization is performed on the
first symbol using the Minn & Bhargava algorithm [26] in the
time domain. The four sequences of the synchronization
symbol are correlated and generate a peak when the sliding
window of size M includes these four sequences and zero
everywhere else.
The frequency synchronization is performed by using an
external frequency reference applied as a board input. This
point will be studied in a future work.
C. Experiments
The two USRP2 boards are arranged in the vehicle
according to Fig. 1 at points G, F, H and D. We will focus
on link GF. In Fig. 8, we can observe the spectrum of the
LFTX daughterboard output with a DSP of -80dBm/Hz. In
this figure, the bandwidth [2-12.5] MHz is used. However it
is possible to transmit up to 25 MHz. We can observe there
are no notches compared to HPAV standard as we do not
have to consider the same electromagnetic constraints.
Then using the resulting bit loading, we have transmitted
a file of 107 samples, equivalent to about 120 frames of
DMT symbols including synchronization, equalization and
data DMT symbols. We obtain no error when the motor is
OFF. When the motor is ON, a lot of the frames are
undetected (nearly 40 %). This result is due to the fact that
the noise power is closed to the signal power and the
attenuation is about 40 dB on path GF. An additional
amplifier will be used to increase the signal to noise ratio.
If we analyse the best HD path, this last one achieves a
mean data rate of 70 Mbps and no errors are detected during
the two scenarios.
V.
CONCLUSIONS
We have presented a new in-vehicle embedded network
architecture based on PLC communications. The
measurements show that the proposed technologies are
suitable for the bit rates required by the multi-media
applications. PLC can be applied to reduce the number of
wires or to redundancy. We have both studied the channels
and the noise. We have deduced the main parameters for an
OFDM modulation, which is a god candidate for such
selective channels. In order to optimize the network layers,
the proposed demonstrator allows us to explore various
configurations by adjusting parameters to fully meet
expected system requirements.
Fig. 8 TX spectrum, LFTX output
In Fig. 9, the transmitted signal at the point F when the
OFDM signal is transmitted from the point G is given. We
can see the different OFDM frames and periodic noises
which will corrupt the signal. In the [5.2106 -5.4106] time
window, the signal is completely masked by impulsive
noises. It may result in none synchronization detection.
ACKNOWLEDGMENT
This work has been carried out by the CIFAER project,
initiated and supported by the ANR and by the French
Premium Cars Competitiveness Cluster ID4car.
Impulsive noises
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