IEEE TRANSACTIONS ON TERAHERTZ SCIENCE AND TECHNOLOGY, VOL. 12, NO. 5, SEPTEMBER 2022
527
210–365 GHz 90° Differential Phase Shifter
for Wideband Circular Polarizer
Sho Masui , Yutaka Hasegawa , Hideo Ogawa, Takafumi Kojima , Member, IEEE, and Toshikazu Onishi
Abstract—In this article, a wideband 90° differential phase
shifter in the 210–365 GHz band with a fractional bandwidth of
54% was designed, fabricated, and measured to develop a wideband
circular polarizer for millimeter/submillimeter wavelengths. To
achieve a wide bandwidth, we designed a 90° differential phase
shifter with a flat phase-difference characteristic by combining
three different types of 90° differential phase shifters to cancel
out the frequency dependence of the phase difference between
the two polarizations. The developed phase shifter has a phase
difference of 90 ± 6° and a return loss better than 17 dB for both
polarizations over the frequencies 210–365 GHz, in addition to
waveguide transitions necessary for S-parameter measurements.
The developed phase shifter can be used as an alternative for
achieving simultaneous observations of dual-polarization signals at
230 and 345 GHz, as proposed by the next-generation event horizon
telescope, which aims to observe black holes.
Index Terms—Phase shifters, polarization splitter, radio
astronomy, submillimeter-wave waveguide circuits.
I. INTRODUCTION
N SATELLITE communication and radio astronomy, a circular polarizer is essential for separately observing dualpolarized radio waves, which is capable of separating two
circular polarizations with left and right rotations. In radio
astronomy, in particular, to decrease the effects of the differences
in parallactic angles, circular polarizers are generally used in
very long baseline interferometry (VLBI) observations with
distant telescopes to achieve a high angular resolution [1]–[5].
Recently, the event horizon telescope (EHT) has successfully
observed a black hole by correlating signals from telescopes
around the world, taking advantage of the high resolution of
VLBI observations [6], [7]. Each telescope participating in the
I
Manuscript received 23 March 2022; revised 27 May 2022; accepted 28 June
2022. Date of publication 18 July 2022; date of current version 3 September
2022. This work was supported in part by JSPS KAKENHI under Grant
JP18H05440, Grant JP20J23670, and Grant JP21K03629, in part by the Grant
of Joint Development Research supported by the Research Coordination Committee, National Astronomical Observatory of Japan, and National Institutes of
Natural Sciences. (Corresponding author: Sho Masui.)
Sho Masui is with the Osaka Metropolitan University, Sakai 599-8531,
Japan, and also with the National Astronomical Observatory of Japan, Mitaka
181-8588, Japan (e-mail: s_s.masui@omu.ac.jp).
Yutaka Hasegawa, Hideo Ogawa, and Toshikazu Onishi are with Osaka Metropolitan University, Sakai 599-8531, Japan (e-mail: y.hasegawa@
omu.ac.jp; ogawah@omu.ac.jp; tonishi@omu.ac.jp).
Takafumi Kojima is with the National Astronomical Observatory of Japan and
the Graduate University for Advanced Studies, Mitaka 181-8588, Japan (e-mail:
t.kojima@nao.ac.jp).
Color versions of one or more figures in this article are available at
https://doi.org/10.1109/TTHZ.2022.3191851.
Digital Object Identifier 10.1109/TTHZ.2022.3191851
EHT project has cryogenic receivers installed to detect circularly
polarized signals with a high sensitivity. For instance, the Greenland telescope (GLT) [5], [8] has a dual circular polarization
receiver installed, which uses a simple and compact waveguide
stepped septum-type circular polarizer (SST-CP) with a fractional bandwidth of approximately 20%.
Next-generation EHT (ngEHT) is a new EHT project for
achieving simultaneous observations with dual polarization and
dual frequency in a 345 GHz band, in addition to the previously
successful 230 GHz band. The detailed scientific goals and
instrument requirements of the ngEHT are described elsewhere
[9]. One possible receiver system for these observations is to
implement a quasi-optical filter [10], [11] to separate the 230
and 345 GHz bands and dual-polarization receivers in each
frequency band. However, there are possible disadvantages in
terms of compactness and pointing errors between bands.
In recent years, wideband waveguide circuits [12]–[14] and
superconductor—insulator—superconductor mixers [15], [16]
with fractional bandwidths exceeding 50% have been developed for next-generation receivers, such as the Atacama large
millimeter/submillimeter array (ALMA). In [17] and [18], a
dual-band receiver with a single horn and single polarization
was developed for the simultaneous observations of 12 CO, 13 CO,
and C18 O (J = 2–1 and J = 3–2) in the 230 and 345 GHz
bands. This wideband receiver allows simultaneous observations
in a compact system without pointing errors between bands.
This dual-band receiver technology can be applied to further
observational studies on black holes. However, to perform observations with dual-band and dual circular polarization, a circular
polarizer that meets the requirement in terms of bandwidth
has not yet been developed in millimeter/submillimeter wave
bands.
Our final goal is to develop a low-loss wideband circular
polarizer in the 210–365 GHz band. Two types of circular polarizers have been reported in millimeter/submillimeter wave bands
based on stepped septum structures [5], [19] and a combination
of a 90° differential phase shifter and orthomode transducer
(OMT) [4]. The 90° differential phase shifter is a waveguide
device that produces 90° phase difference between two orthogonal polarizations at the output port, below simply indicated as a
phase shifter. The OMT is a waveguide device that separates two
orthogonal polarizations. In our previous study, the bandwidth of
the SST-CP used in the GLT was limited owing to the occurrence
of high-order modes [5], [19]. In this study, we applied the
concept of a wideband circular polarizer combining a waveguide
phase shifter, a 45° waveguide transition, and an OMT [4], as
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Fig. 1. Schematic diagram of a circular polarizer combining a 90° differential
phase shifter, a 45◦ waveguide transition, and OMT. The schematics (a)–(c)
represent the flow of circular polarizations being converted to orthogonal polarizations. The arrows in (a) and (b) depict orthogonal electric vectors in the square
waveguide only for Pol-right. At the input of the phase shifter, as shown in (a),
there is a 90° difference between vertical (V-pol) and horizontal polarizations
(H-pol), while they are in phase at the output in (b).
shown in Fig. 1. The phase shifter converts an input circular
polarization to a linear polarization, and then the OMT separates
the respective linear polarizations [20]. Note that the square
waveguide of the phase shifter physically must be connected
with a 45° rotation relative to the one for the OMT. Hence, they
are connected via a 45° waveguide transition that converts V-pol
and H-pol into one linear polarization by the composition of their
electric vectors, as shown in Fig. 1. High-performance wideband
OMTs in the millimeter/submillimeter wavelength range have
already been demonstrated and described in [12] and [13]. We
plan to develop a high-performance wideband OMT at 210–
365 GHz based on the design concept. Therefore, this article focuses on the development of a phase shifter in the 210–365 GHz
band with a fractional bandwidth of 54%, which significantly improves the wideband nature of circular polarizers. However, it is
extremely difficult for conventional circuits that use only a single
type of phase shifter to obtain a constant 90° phase difference
because the frequency dependence of the phase difference is very
large. To compensate for the frequency dependence [21]–[25],
we employed a waveguide phase shifter that combined three
different types of waveguide circuit elements. These circuit
elements can cancel out each other’s frequency dependence, and
the entire phase difference can be flattened to approximately
90°. Furthermore, we devised a structural design suitable for
fabrication by direct machining with high processing accuracy,
for example, a square instead of a circle waveguide was adopted.
The newly developed phase shifter has the widest bandwidth
compared to previous shifters in the millimeter/submillimeter
wave bands. In addition, when combined with an OMT,
high-performance circular polarization characteristics can be
expected.
The rest of this article is organized as follows. Section II
describes the design of the wideband phase shifter. In Section III,
we present the fabrication of the phase shifter and measurement
results of the S-parameters. In Section IV, we discuss the discrepancy between the measurement and simulation and verify the
result using an additional machining process. Finally, Section V
concludes this article.
Fig. 2. (a) Simulation model of the designed phase shifter and definition of
the planes, polarizations. (b) Cross sections of the phase shifter and definition
of the parameters. Optimized parameters of the phase shifter are summarized in
Table I.
II. DESIGN OF WIDEBAND PHASE SHIFTER
In this section, we describe a design of a phase shifter in
the 210–365 GHz band. One of the key characteristics of the
phase shifter is the phase difference between both polarizations
at the output port. The closer the phase difference of the phase
shifter is to 90°, the lower the cross-polarization loss. The crosspolarization loss in dB is given by [1]
tan (Δ∅ − 90◦ ) [dB]
(1)
Xpol = −20 log 2
where Δ∅ is the phase difference between both polarizations.
Δ∅ is defined by the following equation:
Δ∅ = ∅V − ∅H = βV lV − βH lH
(2)
where ∅V and ∅H are the phase angle of the vertical (V-pol) and
horizontal polarizations (H-pol), respectively, βV and βH are the
propagation constants of V-pol and H-pol, respectively, and lV
and lH are the effective lengths of the phase shifter in V-pol and
H-pol, respectively. The orientations of these polarizations are
shown in Fig. 2(a). Our target is to develop a receiver system with
a cross-polarization loss better than 20 dB. Since it is affected by
RF components, such as optics, corrugated horn, and waveguide
components. Therefore, the cross-polarization loss of waveguide
components should be minimized to be 25 dB or less. Equation
(1) indicates that the phase difference of the phase shifter must be
within 90 ± 6.4° to achieve the cross-polarization loss of 25 dB.
A conventional phase shifter has a single waveguide circuit
element, such as a corrugation or ridge, to have different phase
constants (βV , βH ) between polarizations [1], [2], [26], [27].
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MASUI et al.: 210–365 GHZ 90° DIFFERENTIAL PHASE SHIFTER FOR WIDEBAND CIRCULAR POLARIZER
With such a phase shifter, it is difficult to achieve a 90° phase difference over a wide fractional bandwidth exceeding 30%. This
is because the phase-difference characteristics of conventional
phase shifters exhibit quadratic behavior and are designed to be
closest to 90° only in the desired bandwidth. In our design, to
develop a wideband phase-difference characteristics, we aimed
to obtain a cubic behavior, which has one more extremum than
a quadratic function. To achieve the above characteristics, we
decided to combine the three types of phase shifters whose
phase-difference characteristics cancel each other [21]–[25]. In
the 210–365 GHz millimeter/submillimeter band, fabrication
errors have a significant impact on the phase or magnitude characteristics. Therefore, we devised a structural design suitable for
fabrication by direct machining with high processing accuracy.
For example, we avoided the use of H-plane ridge waveguides,
which are difficult to fabricate using machining technologies.
In our simulations, we canceled the frequency characteristics
by utilizing the double-ridge, E-plane corrugated, and H-plane
corrugated phase shifters, as shown in Fig. 2. As the waveguide
size of the phase shifter decreased, the frequency dependence of
the propagation constant on the low-frequency side increased. In
addition, if the waveguide size is large, there is a risk of higher
order modes being generated. In the simulation, we were careful
not to generate higher order modes within the 210–365 GHz and,
finally, set the waveguide size to 0.840 mm × 0.840 mm.
Because a double-ridge waveguide has a lower cutoff frequency than a rectangular waveguide, the propagation constant
of the ridge waveguide at a certain frequency is known to be
larger [28]. Therefore, the propagation constant of the V-pol, the
electric field in the same direction as the ridge, becomes large,
while it does not have a significant effect on the H-pol. Consequently, this creates a phase difference between the polarizations. For the simulation, we used high frequency structure simulator (HFSS), a three-dimensional (3-D) finite-element analysis
software by ANSYS [29]. The phase angle of the double-ridge
phase shifter for both polarizations and the phase difference
between the polarizations are shown in Fig. 3(a). The phase
difference between polarizations in a double-ridge waveguide
is discussed in detail in [26], where it is argued that the phase
difference increases with the height of the ridge. Therefore, in
the simulation, we first set a large phase difference with rh , one
of the parameters of the ridge in Fig. 2(b), and then connected the
double-ridge step. These heights and lengths were determined to
optimize the overall phase difference and impedance matching
in order to achieve a phase difference of 90° at around 275 GHz.
As a result, we obtained the phase-difference characteristic, as
shown in Fig. 3(a).
Next, we considered qualitatively the frequency dependence
of the phase difference of the E-plane corrugated phase shifter.
The propagation constant of the TEmn mode in a rectangular
waveguide is given by
mπ 2 nπ 2
ω 2 με −
−
(3)
β =
a
b
where μ is the permeability of the vacuum, ε is the dielectric
constant of the vacuum, and a and b are the waveguide heights
[see Fig. 2(b)]. We assumed that the V-pol and H-pol signals
529
Fig. 3. Simulation results of the phase shifters. (a) Phase difference of the
double-ridge phase shifter between polarizations and phase angle of V-pol and
H-pol. (b) Phase difference of the E-plane corrugated phase shifter and phase
angle of V-pol and H-pol. In the optimized design, Ncorr1 = 20. (c) Phase
difference of the H-plane corrugated phase shifter. In the optimized design, ch
= 0.4.
correspond to the TE10 (m = 1 and n = 0) and TE01 (m = 0 and
n = 1) modes, respectively. From (3), the propagation constants
β V and β H of V-pol and H-pol depend on a and b, respectively. In
V-pol, the propagation constant β V of the corrugated waveguide
monotonically increases with frequency, and it is the same as
that of a square waveguide (0.840 mm × 0.840 mm) without
the corrugation because height a is kept constant. The electric
field penetrates the corrugation, which behaves as an inductive
stub, therefore increasing the effective path length lV defined in
(2). This effect is greater at higher frequencies. Thus, the phase
angle ∅V shifts relatively large at higher frequencies compared
to the square waveguide without the corrugation. On the other
hand, for H-pol, height b at the corrugation becomes larger;
thus, the effective propagation constant β H of the corrugated
waveguide increases on the low-frequency side in accordance
with (3), whereas it weakly varies on the high-frequency side. As
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TABLE I
SUMMARY OF THE OPTIMIZED MAJOR PARAMETERS
a result, the corrugated waveguide creates large shift of the phase
angle ∅H on low-frequency side compared to that of the square
waveguide without the corrugation. Thus, as shown in Fig. 3(b),
the phase angles ∅H and ∅V become larger on the low- and
high-frequency side, respectively, which facilitates compensation of the frequency dependence in the double-ridge waveguide.
In this design, we adjusted h1 and w1 of the corrugations and
constructed a unit cell in which the phase difference of the corrugations was 0° at the frequency at which the phase-difference
characteristic of the double ridge was around 90°. We then optimized the number of corrugations Ncorr1 to cancel the frequency
characteristics of the double ridge [see Fig. 3(b)]. While the
working principles of the H-plane and E-plane corrugated phase
shifters are similar, the H-plane is designed to have a large phase
difference only at high frequencies by adjusting parameter ch ,
as shown in Fig. 3(c). Since the number of H-plane corrugations
Ncorr2 changes the magnitude of the phase difference in Fig. 3(c),
it was determined based on the obtained phase differences of
double-ridge and E-plane corrugated ones.
A simulation model of the designed phase shifter that combines the above double-ridge, E-plane corrugated, and H-plane
corrugated shifters is shown in Fig. 2. The disadvantage of this
phase shifter is that the length of the circuit is longer owing to the
combination of the three different phase shifters. To make the
length as short as possible, the H-plane corrugated phase shifter
was inserted between the double-ridge and E-plane corrugated
shifters. After combining them, the entire phase shifter was
optimized. The overall length of the optimized phase shifter was
approximately 7.7 mm, which corresponds to approximately
5.8 times the guide wavelength at center frequency. The final
simulation results are shown in Fig. 4, and optimized parameters
of the phase shifter are summarized in Table I. In the simulation,
the insertion loss was analyzed by considering the conductivity
of aluminum and roughness of the waveguide surface (0.075 μm)
for our machining process, which was obtained in previous
studies [17], [30]. The simulation results show a phase delay
of 90 ± 6° and a return loss better than 20 dB, which meets our
design requirements.
III. FABRICATION AND MEASUREMENT OF S-PARAMETER
The phase shifter is fabricated as two E-plane split blocks
because it is designed for direct machining, which is a highprecision manufacturing method, as mentioned in Section II.
The developed phase shifter is connected to an OMT; therefore,
the waveguide flange of the phase shifter must correspond to
that of the OMT. Waveguide circuits above the 325 GHz band
are relatively smaller than the standard waveguide interface
(UG-387). Hence, adopting UG-387 creates unnecessary path
Fig. 4. Simulation results of S-parameter and phase difference for the
designed.
Fig. 5. (Left) Photograph of the fabricated phase shifter with a Japanese
transit coin. (Right) A photograph of the fabricated phase shifter combined
the waveguide transitions to connect to the measurement system.
lengths or can cause unnecessary modes. To avoid this problem,
for the fabrication of the OMT and phase shifter, we adopted
the miniature interface proposed by [31] as a waveguide flange
above the 325 GHz band. The fabricated phase shifter is illustrated in Fig. 5. Considering that the square waveguide of the
phase shifter is connected with 45° rotation relative to the one
of the OMT, the flange of the phase shifter was fabricated in an
octagonal shape. The length in the direction of the radio wave
propagation is approximately 7.7 mm. As mentioned earlier,
the size of the input/output square waveguide was 0.840 mm ×
0.840 mm.
A PNA-X vector network analyzer (VNA) from the National Institute of Information and Communications Technology
(NICT) and the National Astronomical Observatory of Japan
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MASUI et al.: 210–365 GHZ 90° DIFFERENTIAL PHASE SHIFTER FOR WIDEBAND CIRCULAR POLARIZER
Fig. 6.
531
Schematic diagram of the measurement system in NICT and NAOJ.
(NAOJ) was used for the phase-shifter measurements. Because
of the wide measurement range, we carried out the S-parameter
measurement of the phase shifter using two types of extenders,
the WR-3.4 (0.864 mm × 0.432 mm) extender of NICT for
220–330 GHz and the WR-2.2 (0.559 mm × 0.279 mm) extender of NAOJ for 325–500 GHz. To minimize the mismatch
with the extenders at the flange interface, waveguide transitions
with waveguide sizes from square (0.840 mm × 0.840 mm) to
WR-3.4 and from square to WR-2.2, with a length of 15 mm,
were prepared. In addition, because the waveguide flange of the
extenders is UG-387, which is the waveguide interface standard,
the waveguide transition converts UG-387 to our miniflange, as
shown in Fig. 5 (right). The transition was fabricated with a
length such that the expected return loss was higher than 30 dB
when back-to-back connection so that the measurement of return
loss for the phase shifter would not be affected. However, in
the back-to-back configuration, the return losses were measured
approximately to be 25 dB. Thus, these return losses for the
transitions may affect the measurement of the return loss for the
phase shifter. The schematic diagram of the measurement setup
for the phase shifter is shown in Fig. 6. Short-open-load-through
calibration was performed at the waveguide port of the extenders.
Note that only one polarization is available in this measurement
setup, and thus, the phase shifter must be rotated to measure both
polarizations. The measurement results of the S-parameter for
the phase shifter are shown in Fig. 7. The upper graph in Fig. 7
shows the insertion and return losses for both polarizations. In
these results, the insertion loss of the waveguide transitions was
removed from the measured S11 and S21 , whereas no correction
was made in the return loss of the waveguide transitions. The
averaged insertion losses of V-pol and H-pol in 210–365 GHz
are approximately 0.29 dB and 0.14 dB, respectively. However,
the return loss of 17 dB was 3 dB lower than the simulation
results. One of the reasons could be the effect of the transitions.
Although there is an uncertainty in the measurement owing to
the waveguide transition, the results show that the overall return
loss shows acceptable performance for the receiver operation.
The bottom graph of Fig. 7 shows the phase difference. This
performance is approximately 90 ± 10° in the 210–365 GHz
band, which corresponds to a cross-polarization loss of approximately 20 dB from (1). Comparing the simulation and
Fig. 7. (a) Measurement results of the S-parameter for phase shifter.
(b) Comparison of designed and measured phase difference of phase shifter.
measurement results, we found that the overall phase difference
in the measurement was approximately 5° higher than that in
the simulation. Through 3-D inspections of this component,
we found that the discrepancy in this result was caused by a
slight fabrication error. A detailed investigation is presented
Section IV.
IV. DISCUSSION AND ADDITIONAL PROCESS
The VK-X3000 3-D surface profiler produced by Keyence
Corp. was used to measure the structural dimensions of the
phase shifter. The measurement results are shown in Fig. 8.
Fig. 8(a) and (b) shows the results for the double-ridge and
corrugated parts, respectively. Owing to time constraints, we had
to select a limited number of places where we could measure the
dimensions and check for deviations from the design values. It
was confirmed that the worst error was approximately 4 μm, and
most of the parts were precisely fabricated within approximately
3 μm. Fig. 8(c) shows an enlarged E-plane corrugated part.
From these results, the dimensions of the corrugations were in
good agreement with the design, whereas we found that burrs,
which were assumed to be generated during the milling process,
existed at the convex edges. Throughout the detailed inspection
of the corrugations, burrs were found in most corrugations of
the E-plane, and their sizes were approximately 3–5 μm. To
verify the effect of burrs on the phase shifter, a model with
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TABLE II
COMPARISON OF PHASE-SHIFTER’S CHARACTERISTICS IN MILLIMETER/SUBMILLIMETER WAVE BANDS
Fractional bandwidth is defined as the frequency bandwidth divided by the center frequency multiplied by 100. The frequency bandwidth is defined as the difference between
maximum and minimum frequencies and the center frequency is defined as half of the sum of the maximum and minimum frequencies. a These are calculated values from (1) and
the phase difference.
Fig. 8. Measurement results of 3-D parameter for the phase shifter. (a) Enlarged view of the ridge section. (b) Enlarged view of the corrugation section.
(c) Enlarged view of the burr found in the corrugation.
Fig. 9. (a) Model of phase shifter considering burrs. (b) Simulation results
when the height of the burrs is varied (1, 3, and 5 µm).
burrs was created, as shown in Fig. 9(a), and simulations were
performed with burr heights of 1, 3, and 5 μm. The results show
that these small burrs have a significant effect on the phase
difference but do not affect the return loss. The simulation results
for the phase difference are shown in Fig. 9(b). Because the
actual phase shifter may have some areas without burrs, we
could not completely reproduce the measurement results with
the model that considered burrs. However, because the effect of
burrs on the simulated phase difference shows the same direction
as the deviation of the measurement results, this suggests that
the burrs mainly cause this degradation. Therefore, additional
processing of the phase shifter was performed to remove burrs.
After additional machining, the corrugated area was inspected
again using a 3-D measuring instrument, and the burrs were
removed. We then repeated the measurements with the VNA and
extenders and obtained the results, as shown in Fig. 10. As shown
in the simulation, there was no significant effect on the return
loss, as shown in Fig. 10(a). The measured phase difference
was 90 ± 6°, which corresponds to a cross-polarization loss of
approximately 25 dB from (1) in the 210–365 GHz band. The
phase difference shifted by approximately 5° after the additional
process, and the measurement results closely resembled the
simulation results. The performance of the developed wideband
phase shifter is comparable to other millimeter/submillimeter
circular polarizers and phase shifters above about 100 GHz (see
Table II).
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MASUI et al.: 210–365 GHZ 90° DIFFERENTIAL PHASE SHIFTER FOR WIDEBAND CIRCULAR POLARIZER
533
and measure its structural parameters. The authors like to thank
Editage (www.editage.com) for English language editing.
REFERENCES
Fig. 10. (a) Comparison of S11 before and after additional processing of the
phase shifter. (b) Comparison of the designed and measured phase difference of
the phase shifter after additional processing.
V. CONCLUSION
In this article, a wideband phase shifter in the 210–365 GHz
band with a fractional bandwidth of 54% was designed, fabricated, and measured to develop a wideband circular polarizer with millimeter/submillimeter wavelengths. To achieve the
above wideband phase shifter, we designed a phase shifter that
combines a double-ridge-type phase shifter, an E-plane phase
shifter, and an H-plane phase shifter to cancel out the frequency
dependence of the phase difference between the two polarizations. It was fabricated using miniature waveguide interfaces
suitable for circuit sizes in millimeter/submillimeter wave bands.
Although the performance varied owing to the presence of burrs
caused by manufacturing errors, after additional machining, the
final results were close to those of the simulation. According
to the final measurement results, the return loss is better than
17 dB, and the phase difference is 90 ± 6°, corresponding to a
cross-polarization loss of 25 dB. In the future, it is expected that
a wideband circular polarizer with good performance will be
achieved when combined with OMT. We believe that this development will contribute to further VLBI observational studies.
ACKNOWLEDGMENT
The authors would like to thank I. Watanabe and S. Ochiai
of NICT. They helped us measure the S-parameter of the phase
shifter. M. Ishino of KMCO helped us fabricate the phase shifter
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Sho Masui received the B.S. and M.S. degrees in
physics from Osaka Prefecture University, Osaka,
Japan, in 2018 and 2020, respectively. He is currently
working on the development of wideband heterodyne
receiver system for next-generation radio telescope
toward the Ph.D. degree with Osaka Metropolitan
University, Osaka, Japan.
From 2017 to 2021, he developed a wideband
heterodyne receiver system for an Osaka 1.85 m
millimeter–submillimeter telescope. From 2021 to
2022, he joined the National Astronomical Observatory of Japan as a graduate student and has worked on the development of superconducting devices. His research interests include the design and measurement
of wideband waveguide circuits and wideband and compact superconducting
circuits.
Yutaka Hasegawa received the B.S., M.S., and D.S.
degrees in physics from the Graduate School of Science, Osaka Prefecture University, Osaka, Japan, in
2012, 2014, and 2017, respectively.
From 2017 to 2020, he worked as an Aerospace
Project Research Associate with ISAS/JAXA, where
he was engaged in development project management
of a new JAXA 54-m deep-space radio telecommunication antenna (GREAT/MDSS), especially on the
low-noise receiver systems. He is currently a Postdoctoral Researcher with Osaka Metropolitan University,
Osaka, Japan, and engage in the development of various radio receiver system,
especially for the radio astronomical molecules lines observation.
Hideo Ogawa received the B.S., M.S., and D.S.
degrees in physics from Nagoya University, Nagoya,
Japan, in 1965, 1967, and 1992, respectively.
From 1970 to 1999, he belonged to Astrophysics
Laboratory, Nagoya University, where he was involved in developing astronomical instrumentation
for millimeter-wave astronomy. In 1999, he joined the
Astrophysics Laboratory, Osaka Prefecture University, Osaka, Japan. He is currently a Guest Professor
with Osaka Metropolitan University, Osaka, Japan.
His current research interests include the development of astronomical telescopes for millimeter and submillimeter waves.
Takafumi Kojima (Member, IEEE) received the
B.S., M.S., and D.S. degrees in physics from Osaka
Prefecture University, Osaka, Japan, in 2005, 2007,
and 2010, respectively.
From 2007 to 2010, he joined the National Astronomical Observatory of Japan (NAOJ) as a graduate student and worked on ALMA band 10 SIS
mixer development. From 2010 to 2012, he was
with Microsystem Integration Laboratories, Nippon
Telegraph and Telephone Corporation, Atsugi, Japan,
where he was engaged in research on signal processing for a millimeter-wave near-field imaging system. He is currently an Associate
Professor with Advanced Technology Center, NAOJ, Mitaka, Japan. His research
interests include wideband superconducting circuits for radio astronomy from
microwave to submillimeter waves.
Dr. Kojima was a recipient of the 2018 IEEE MTT-S Japan Young Engineer Award, Young Scientists’ Award, and 2021 Commendation for Science
and Technology by the Minister of Education, Culture, Sports, Science, and
Technology, Japan, in 2021.
Toshikazu Onishi received the B.S. and M.S. degrees
in physics and the Ph.D. degree in science from
Nagoya University, Nagoya, Japan, in 1991, 1993,
and 1996, respectively.
With Nagoya University, he was a Postdoctoral Fellow from 1996 to 1999, an Assistant Professor from
1999 to 2003, and an Associate Professor from 2003
to 2009. His work includes the development of radio
telescope systems, management of the NANTEN and
NANTEN2 telescopes, and research on star formation
using radio telescopes. From 2009 to 2022, he was a
Professor with Osaka Prefecture University, Sakai, Japan. He is currently a
Professor with Osaka Metropolitan University, Osaka, Japan. He is working on
the development of radio telescope systems and star formation studies using
various telescopes, including the ALMA and 1.85 m telescopes with Osaka
Prefecture University. He contributed to the ALMA as a member of the ALMA
Board.
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