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as shown in Figure 3.6(c). A current can then be generated between the source and
drain terminals. Since a voltage must be applied to the gate to create the inversion
charge, this transistor is called an enhancement-mode MOSFET. Also, since the
carriers in the inversion layer are electrons, this device is also called an n-channel
MOSFET (NMOS).
The source terminal supplies carriers that flow through the channel, and the
drain terminal allows the carriers to drain from the channel. For the n-channel
MOSFET, electrons flow from the source to the drain with an applied drain-to-source
voltage, which means the conventional current enters the drain and leaves the source.
The magnitude of the current is a function of the amount of charge in the inversion
layer, which in turn is a function of the applied gate voltage. Since the gate terminal
is separated from the channel by an oxide or insulator, there is no gate current.
Similarly, since the channel and substrate are separated by a space-charge region,
there is essentially no current through the substrate.
3.1.3
Ideal MOSFET Current–Voltage
Characteristics—NMOS Device
The threshold voltage of the n-channel MOSFET, denoted as VT N , is defined2 as the
applied gate voltage needed to create an inversion charge in which the density is
equal to the concentration of majority carriers in the semiconductor substrate. In
simple terms, we can think of the threshold voltage as the gate voltage required to
“turn on” the transistor.
For the n-channel enhancement-mode MOSFET, the threshold voltage is positive because a positive gate voltage is required to create the inversion charge. If the
gate voltage is less than the threshold voltage, the current in the device is essentially
zero. If the gate voltage is greater than the threshold voltage, a drain-to-source
current is generated as the drain-to-source voltage is applied. The gate and drain
voltages are measured with respect to the source.
Figure 3.7(a) shows an n-channel enhancement-mode MOSFET with the source
and substrate terminals connected to ground. The gate-to-source voltage is less than
the threshold voltage, and there is a small drain-to-source voltage. With this bias
configuration, there is no electron inversion layer, the drain-to-substrate pn junction
is reverse biased, and the drain current is zero (neglecting pn junction leakage
currents).
Figure 3.7(b) shows the same MOSFET with an applied gate voltage greater
than the threshold voltage. In this situation, an electron inversion layer is created and,
when a small drain voltage is applied, electrons in the inversion layer flow from the
source to the positive drain terminal. The conventional current enters the drain
terminal and leaves the source terminal. Note that a positive drain voltage creates a
reverse-biased drain-to-substrate pn junction, so current flows through the channel
region and not through a pn junction.
The iD versus v DS characteristics3 for small values of v DS are shown in Figure 3.8. When vG S < VT N , the drain current is zero. When vG S is greater than VT N ,
2
The usual notation for threshold voltage is VT . However, since we have defined the thermal voltage as
VT = kT /q , we will use VT N for the threshold voltage of the n-channel device.
The voltage notation v DS and vG S , with the dual subscript, denotes the voltage between the drain (D) and
source (S) and between the gate (G) and source (S), respectively. Implicit in the notation is that the first
subscript is positive with respect to the second subscript.
3
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vGS1 > VTN
Oxide
S
vGS1
Channel
inversion
charge
iD
vDS
vDS
Depletion
region
p-type
iD
Oxide
S
Channel
inversion
charge
p-type
L
iD
iD
vDS
vDS
(a)
(b)
vGS1
vGS1
iD
Oxide
S
vDS (sat)
S
Oxide
vDS (sat)
vDS > vDS (sat)
E-field depletion region
Channel
inversion
charge
p-type
iD
iD
Channel
inversion
charge
p-type
iD
Saturation
region
vDS (sat)
vDS
(c)
vDS (sat)
vDS
(d)
Figure 3.9 Cross section and i D versus v DS curve for an n-channel enhancement-mode
MOSFET when vG S > VT N for (a) a small v DS value, (b) a larger v DS value but for
v DS < v DS (sat) , (c) v DS = v DS (sat), and (d) v DS > v DS (sat)
generate the family of curves for this n-channel enhancement mode MOSFET as
shown in Figure 3.10.
Although the derivation of the current–voltage characteristics of the MOSFET
is beyond the scope of this text, we can define the relationships. The region for
which v DS < v DS (sat) is known as the nonsaturation or triode region. The ideal
current–voltage characteristics in this region are described by the equation
i D = K n 2(vG S − VT N )v DS − v 2DS
(3.2(a))
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iD
vDS (sat) = vGS – VTN
Nonsaturation
region,
vDS < vDS (sat)
vGS5 > vGS4
Saturation region
vDS > vDS (sat)
vGS4 > vGS3
vGS3 > vGS2
vGS2 > vGS1
vGS1 > VTN > 0
vDS
Figure 3.10 Family of iD versus v DS curves for an n-channel enhancement-mode MOSFET.
Note that the v DS (sat) voltage is a single point on each of the curves. This point denotes the
transition between the nonsaturation region and the saturation region
In the saturation region, the ideal current–voltage characteristics for vG S > VT N are
described by the equation
i D = K n (vG S − VT N )2
(3.2(b))
In the saturation region, since the ideal drain current is independent of the drain-tosource voltage, the incremental or small-signal resistance is infinite. We see that
r0 = v DS /i D |vG S =const. = ∞
The parameter Kn is sometimes called the transconduction parameter for the
n-channel device. However, this term is not to be confused with the small-signal
transconductance parameter introduced in the next chapter. For simplicity, we will
refer to this parameter as the conduction parameter, which for an n-channel device
is given by
W μn Cox
Kn =
(3.3(a))
2L
where Cox is the oxide capacitance per unit area. The capacitance is given by
Cox = ox /tox
where tox is the oxide thickness and ox is the oxide permittivity. For silicon devices,
ox = (3.9)(8.85 × 10−14 ) F/cm. The parameter μn is the mobility of the electrons
in the inversion layer. The channel width W and channel length L were shown in
Figure 3.5(a).
As Equation (3.3(a)) indicates, the conduction parameter is a function of both
electrical and geometric parameters. The oxide capacitance and carrier mobility are
essentially constants for a given fabrication technology. However, the geometry, or
width-to-length ratio W/L, is a variable in the design of MOSFETs that is used to
produce specific current–voltage characteristics in MOSFET circuits.
We can rewrite the conduction parameter in the form
Kn =
kn W
·
2 L
(3.3(b))
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In order to fabricate n-channel and p-channel devices that are electrically equivalent, the magnitude of the threshold voltages must be equal, and the n-channel and
p-channel conduction parameters must be equal. Since, in general, μn , and μ p are not
equal, the design of equivalent transistors involves adjusting the width-to-length
ratios of the transistors.
3.1.8
Summary of Transistor Operation
We have presented a first-order model of the operation of the MOS transistor. For an
n-channel enhancement-mode MOSFET, a positive gate-to-source voltage, greater
than the threshold voltage VT N , must be applied to induce an electron inversion layer.
For vG S > VT N , the device is turned on. For an n-channel depletion-mode device, a
channel between the source and drain exists even for vG S = 0. The threshold voltage
is negative, so that a negative value of v GS is required to turn the device off.
For a p-channel device, all voltage polarities and current directions are reversed
compared to the NMOS device. For the p-channel enhancement-mode transistor,
VT P < 0 and for the depletion-mode PMOS transistor, VT P > 0.
Table 3.1 lists the first-order equations that describe the i–v relationships in
MOS devices. We note that K n and K p are positive values and that the drain current
i D is positive into the drain for the NMOS device and positive out of the drain for the
PMOS device.
Table 3.1
Summary of the MOSFET current–voltage relationships
NMOS
PMOS
Nonsaturation region (v DS < v DS (sat))
i D = K n [2(vG S − VT N )v DS − v 2DS ]
Saturation region (v DS > v DS (sat))
i D = K n (vG S − VT N )2
Transition point
v DS (sat) = vG S − VT N
Enhancement mode
VT N > 0
Depletion mode
VT N < 0
Nonsaturation region (v S D < v S D (sat))
i D = K p [2(v SG + VT P )v S D − v S2 D ]
Saturation region (v S D > v S D (sat))
i D = K p (v SG + VT P )2
Transition point
v S D (sat) = v SG + VT P
Enhancement mode
VT P < 0
Depletion mode
VT P > 0
3.1.9
Short-Channel Effects
The current–voltage relations given by Equations (3.2(a)) and (3.2(b)) for the n-channel
device and Equations (3.4(a)) and (3.4(b)) for the p-channel device are the ideal
relations for long-channel devices. A long-channel device is generally one whose
channel length is greater than 2 μm. In this device, the horizontal electric field in the
channel induced by the drain voltage and the vertical electric field induced by the
gate voltage can be treated independently. However, the channel length of presentday devices is on the order of 0.2 μm or less.
There are several effects in these short-channel devices that influence and
change the long-channel current–voltage characteristics. One such effect is a variation
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in threshold voltage. The value of threshold voltage is a function of the channel
length. This variation must be considered in the design and fabrication of these
devices. The threshold voltage also becomes a function of the drain voltage. As the
drain voltage increases, the effective threshold voltage decreases. This effect also
influences the current–voltage characteristics.
The process conduction parameters, kn and k p , are directly related to the carrier
mobility. We have assumed that the carrier mobilities and corresponding process
conduction parameters are constant. However, the carrier mobility values are functions of the vertical electric field in the inversion layer. As the gate voltage and vertical electric field increase, the carrier mobility decreases. This result, again, directly
influences the current–voltage characteristics of the device.
Another effect that occurs in short-channel devices is velocity saturation. As the
horizontal electric field increases, the velocity of the carriers reaches a constant value
and will no longer increase with an increase in drain voltage. Velocity saturation will
lower the VDS (sat) voltage value. The drain current will reach its saturation value at
a smaller VDS voltage. The drain current also becomes approximately a linear function of the gate voltage in the saturation region rather than the quadratic function of
gate voltage in the long-channel characteristics.
Although the analysis of modern MOSFET circuits must take into account these
short-channel effects, we will use the long-channel current–voltage relations in this
introductory text. We will still be able to obtain a good basic understanding of the operation and characteristics of these devices, and we can still obtain a good basic understanding of the operation and characteristics of MOSFET circuits using the ideal
long-channel current–voltage relations.
3.1.10
Additional Nonideal
Current–Voltage Characteristics
The five nonideal effects in the current–voltage characteristics of MOS transistors
are: the finite output resistance in the saturation region, the body effect, subthreshold
conduction, breakdown effects, and temperature effects. This section will examine
each of these effects.
Finite Output Resistance
In the ideal case, when a MOSFET is biased in the saturation region, the drain
current iD is independent of drain-to-source voltage v DS. However, in actual
MOSFET iD versus v DS characteristics, a nonzero slope does exist beyond the saturation point. For v DS > v DS (sat), the actual point in the channel at which the inversion charge goes to zero moves away from the drain terminal (see Figure
3.9(d)). The effective channel length decreases, producing the phenomenon called
channel length modulation.
An exaggerated view of the current–voltage characteristics is shown in Figure 3.20.
The curves can be extrapolated so that they intercept the voltage axis at a point
v DS = −V A . The voltage V A is usually defined as a positive quantity. The slope of the
curve in the saturation region can be described by expressing the i D versus v DS characteristic in the form, for an n-channel device,
i D = K n [(vG S − VT N )2 (1 + λv DS )]
where λ is a positive quantity called the channel-length modulation parameter.
(3.7)
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iD
1
Slope = r
o
–VA = – 1
λ
vDS
0
Figure 3.20 Family of i D versus v DS curves showing the effect of channel length
modulation producing a finite output resistance
The parameters λ and V A are related. From Equation (3.7), we have
(1 + λ v DS ) = 0 at the extrapolated point where i D = 0. At this point, v DS = −V A ,
which means that V A = 1/λ.
The output resistance due to the channel length modulation is defined as
∂i D −1 ro =
(3.8)
∂v DS
vG S =const.
From Equation (3.7), the output resistance, evaluated at the Q-point, is
ro = [λK n (VG S Q − VT N )2 ]−1
(3.9(a))
or
ro ∼
= [λ I D Q ]−1 =
1
VA
=
λ IDQ
IDQ
(3.9(b))
The output resistance ro is also a factor in the small-signal equivalent circuit of the
MOSFET, which is discussed in the next chapter.
Body Effect
Up to this point, we have assumed that the substrate, or body, is connected to the
source. For this bias condition, the threshold voltage is a constant.
In integrated circuits, however, the substrates of all n-channel MOSFETs are
usually common and are tied to the most negative potential in the circuit. An example of two n-channel MOSFETs in series is shown in Figure 3.21. The p-type
S1
VI
VO
D1
n+
VDD
S2
D2
n+
M1
n+
M2
p-substrate
Figure 3.21 Two n-channel MOSFETs fabricated in series in the same substrate. The source
terminal, S2 , of the transistor M2 is more than likely not at ground potential.
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D
G
08:37 PM
–
+
vGS
B
vSB
–
+
S
Figure 3.22 An n-channel
enhancement-mode
MOSFET with a substrate
voltage
substrate is common to the two transistors, and the drain of M1 is common to
the source of M2 . When the two transistors are conducting, there is a nonzero drainto-source voltage on M1 , which means that the source of M2 is not at the same
potential as the substrate. These bias conditions mean that a zero or reverse-bias
voltage exists across the source–substrate pn junction, and a change in the source–
substrate junction voltage changes the threshold voltage. This is called the body
effect. The same situation exists in p-channel devices.
For example, consider the n-channel device shown in Figure 3.22. To maintain a
zero- or reverse-biased source–substrate pn junction, we must have v S B ≥ 0. The
threshold voltage for this condition is given by
VT N = VT N O + γ 2φ f + v S B − 2φ f
(3.10)
where VT N O is the threshold voltage for v S B = 0; γ , called the bulk threshold or
body-effect parameter, is related to device properties, and is typically on the order
of 0.5 V1/2; and φ f is a semiconductor parameter, typically on the order of 0.35 V,
and is a function of the semiconductor doping. We see from Equation (3.10) that the
threshold voltage in n-channel devices increases due to this body effect.
The body effect can cause a degradation in circuit performance because of the
changing threshold voltage. However, we will generally neglect the body effect in
our circuit analyses, for simplicity.
Subthreshold Conduction
If we consider the ideal current-voltage relationship for the n-channel MOSFET
biased in the saturation region, we have, from Equation (3.2(b)),
i D = K n (vG S − VT N )2
iD
Experimental
Ideal
VTN
vGS
√
Figure 3.23 Plot of i D
versus vG S for a MOSFET
biased in the saturation
region showing subthreshold
conduction. Experimentally, a
subthreshold current exists
even for vG S < VT N .
Taking the square root of both sides of the equation, we obtain
i D = K n (vG S − VT N )
(3.11)
√
From Equation (3.11), we see that i d is a linear function of vG S . Figure 3.23 shows
a plot of this ideal relationship.
Also plotted in Figure 3.23 are experimental results, which show that when vG S
is slightly less than VT N , the drain current is not zero, as previously assumed. This
current is called the subthreshold current. The effect may not be significant for a
single device, but if thousands or millions of devices on an integrated circuit are
biased just slightly below the threshold voltage, the power supply current will not
be zero but may contribute to significant power dissipation in the integrated circuit.
One example of this is a dynamic random access memory (DRAM), as we will see in
Chapter 16.
In this text, for simplicity we will not specifically consider the subthreshold
current. However, when a MOSFET in a circuit is to be turned off, the “proper”
design of the circuit must involve biasing the device at least a few tenths of a volt
below the threshold voltage to achieve “true” cutoff.
Breakdown Effects
Several possible breakdown effects may occur in a MOSFET. The drain-to-substrate
pn junction may break down if the applied drain voltage is too high and avalanche
multiplication occurs. This breakdown is the same reverse-biased pn junction breakdown discussed in Chapter 1 in Section 1.2.5.
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VDD
VDD
iD
RD
R1
CC
vG
+
vi
+
–
R2
vGS
ID
R1
vO
+
vDS
–
VDD = 5 V
RD
+
VDS
–
0
VG
+
R2
VGS
–
(a)
–
R1 = 30 kΩ
( )
20
(5)
50
=2V
R2 = 20 kΩ
VG =
RD = 20 kΩ
+
ID = 0.1 mA
+
VDS = 5 – (0.1)(20)
– =3V
VGS = 2 V
–
(b)
Figure 3.24 An NMOS common-
Figure 3.25 (a) The dc equivalent circuit of the NMOS common-source
source circuit
circuit and (b) the NMOS circuit for Example 3.3, showing current and
voltage values
enhancement-mode MOSFET. The source terminal is at ground potential and is
common to both the input and output portions of the circuit. The coupling capacitor
CC acts as an open circuit to dc but it allows the signal voltage to be coupled to the
gate of the MOSFET.
The dc equivalent circuit is shown in Figure 3.25(a). In the following dc analyses, we again use the notation for dc currents and voltages. Since the gate current
into the transistor is zero, the voltage at the gate is given by a voltage divider, which
can be written as
R2
VG = VG S =
VD D
(3.12)
R1 + R2
Assuming that the gate-to-source voltage given by Equation (3.12) is greater
than VT N , and that the transistor is biased in the saturation region, the drain current is
I D = K n (VG S − VT N )2
(3.13)
The drain-to-source voltage is
VDS = VD D − I D R D
(3.14)
If VDS > VDS (sat) = VG S − VT N , then the transistor is biased in the saturation
region, as we initially assumed, and our analysis is correct. If VDS < VDS (sat), then
the transistor is biased in the nonsaturation region, and the drain current is given by
the more complicated characteristic Equation (3.2(a)).
The power dissipated in the transistor, since there is no gate current, is simply
given by
PT = I D VDS
(3.15)
EXAMPLE 3.3
Objective: Calculate the drain current and drain-to-source voltage of a common-
source circuit with an n-channel enhancement-mode MOSFET. Find the power
dissipated in the transistor.
For the circuit shown in Figure 3.25(a), assume that R1 = 30 k, R2 = 20 k,
R D = 20 k, VD D = 5 V, VT N = 1 V, and K n = 0.1 mA/V2 .
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157
Since VDS = 4.74 V > VDS (sat) = VG S − VT N = 2.24 − 0.8 = 1.44 V , the transistor is biased in the saturation region, as initially assumed.
Trade-offs: Note that even if kn changes, the drain current remains constant. For
76 ≤ kn ≤ 84 μA/V2 , the variation in VG S Q is 2.209 ≤ VG S Q ≤ 2.281 V and the
variation in VDS Q is 4.709 ≤ VDS Q ≤ 4.781 V. The variation in VDS Q is ±0.87 percent even with a ±5 percent variation in kn . This stability effect is one advantage of
using constant current biasing.
+3 V
Comment: MOSFET circuits can be biased by using constant-current sources,
RS
which in turn are designed by using other MOS transistors, as we will see. Biasing
with current sources tends to stabilize circuits against variations in device or circuit
parameters.
EXERCISE PROBLEM
RD
Ex 3.8: (a) Consider the circuit shown in Figure 3.33. The transistor parameters
are VT P = −0.40 V and K p = 30 μA/V2 . Design the circuit such that
I D Q = 60 μA and VS D Q = 2.5 V. (b) Determine the variation in Q-point values if
the VT P and K p parameters vary by ±5 percent. (Ans. (a) R S = 19.77 k,
R D = 38.57 k; (b) 58.2 ≤ I D Q ≤ 61.08 μA, 2.437 ≤ VS D Q ≤ 2.605 V)
n-Channel Enhancement-Load Device
An enhancement-mode MOSFET connected in a configuration such as that shown in
Figure 3.34 can be used as a nonlinear resistor. A transistor with this connection is
called an enhancement-load device. Since the transistor is an enhancement mode
device, VT N > 0. Also, for this circuit, v DS = vG S > v DS (sat) = vG S − VT N , which
means that the transistor is always biased in the saturation region. The general iD versus v DS characteristics can then be written as
i D = K n (vG S − VT N )2 = K n (v DS − VT N )2
(3.21)
Figure 3.35 shows a plot of Equation (3.21) for the case when K n = 1 mA/V2 and
VT N = 1 V.
9
vDS (sat) = vGS – VTN
8
7
Transistor
characteristics
6
5
4
3
2
1
0
Figure 3.33 Circuit for
Exercise Ex 3.8
VDD
iD
+
vGS
+
vDS
–
–
Figure 3.34 Enhancement-
mode NMOS device with the
gate connected to the drain
iD (mA)
10
–3 V
1
2
3
4
5
vDS (V)
Figure 3.35 Current–voltage characteristic of an enhancement load device
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Chapter 3 The Field-Effect Transistor
vGS = 0
vGS = VTN
vGS > VTN
(b)
(c)
n-channel
Semi-insulating substrate
(a)
Figure 3.58 Channel space-charge region of an enhancement-mode MESFET for:
(a) vG S = 0, (b) vG S = VT N , and (c) vG S > VT N
these devices include: higher electron mobility in GaAs, hence smaller transit time
and faster response; and decreased parasitic capacitance and a simplified fabrication
process, resulting from the semi-insulating GaAs substrate.
In the MESFET in Figure 3.57, a reverse-bias gate-to-source voltage induces a
space-charge region under the metal gate, which modulates the channel conductance,
as in the case of the pn JFET. If a negative applied gate voltage is sufficiently large,
the space-charge region will eventually reach the substrate. Again, pinchoff will
occur. Also, the device shown in the figure is a depletion mode device, since a gate
voltage must be applied to pinch off the channel, that is, to turn the device off.
In another type of MESFET, the channel is pinched off even at vG S = 0, as
shown in Figure 3.58(a). For this MESFET, the channel thickness is smaller than the
zero-biased space-charge width. To open a channel, the depletion region must be reduced; that is, a forward-biased voltage must be applied to the gate–semiconductor
junction. When a slightly forward-bias voltage is applied, the depletion region extends just to the width of the channel as shown in Figure 3.58(b). The threshold
voltage is the gate-to-source voltage required to create the pinchoff condition.
The threshold voltage for this n-channel MESFET is positive, in contrast to the
negative threshold voltage of the n-channel depletion-mode device. If a larger
forward-bias voltage is applied, the channel region opens, as shown in Figure
3.58(c). The applied forward-bias gate voltage is limited to a few tenths of a volt before a significant gate current occurs.
This device is an n-channel enhancement-mode MESFET. Enhancementmode p-channel MESFETs and enhancement-mode pn JFETs have also been fabricated. The advantage of enhancement-mode MESFETs is that circuits can be
designed in which the voltage polarities on the gate and drain are the same. However,
the output voltage swing of these devices is quite small.
D
iD
G
+
+
vDS
–
vGS
S
(a)
S
vGS
3.6.2
Current–Voltage Characteristics
The circuit symbols for the n-channel and p-channel JFETs are shown in Figure 3.59,
along with the gate-to-source voltages and current directions. The ideal current–
voltage characteristics, when the transistor is biased in the saturation region, are
described by
vG S 2
i D = I DSS 1 −
(3.32)
VP
where I DSS is the saturation current when vG S = 0, and V P is the pinchoff voltage.
–
G
–
+
+
vSD
–
iD
D
(b)
Figure 3.59 Circuit symbols
for: (a) n-channel JFET and
(b) p-channel JFET
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iD
iD
IDSS
Saturation region
vGS ≤ VP
vGS = 0
IDSS
Saturation region
vGS = 0
–1
+1
–2
+2
–3
+3
vDS
(a)
vGS ≥ VP
vSD
(b)
Figure 3.60 Current–voltage characteristics for: (a) n-channel JFET and (b) p-channel JFET
The current–voltage characteristics for n-channel and p-channel JFETs are
shown in Figures 3.60(a) and 3.60(b), respectively. Note that the pinchoff voltage V P
for the n-channel JFET is negative and the gate-to-source voltage vG S is usually negative; therefore, the ratio vG S /V P is positive. Similarly, the pinchoff voltage V P for
the p-channel JFET is positive and the gate-to-source voltage vG S must be positive,
and therefore the ratio vG S /V P is positive.
For the n-channel device, the saturation region occurs when v DS ≥ v DS (sat)
where
v DS (sat) = vG S − V P
(3.33)
For the p-channel device, the saturation region occurs when v S D ≥ v S D (sat) where
v S D (sat) = V P − vG S
(3.34)
EXAMPLE 3.19
Objective: Calculate i D and v DS (sat) in an n-channel pn JFET.
Assume the saturation current is I DSS = 2 mA and the pinchoff voltage is
V P = −3.5 V. Calculate i D and v DS (sat) for vG S = 0, V P /4, and V P /2.
Solution: From Equation (3.32), we have
2
vG S 2
vG S
i D = I DSS 1 −
= (2) 1 −
VP
(−3.5)
Therefore, for vG S = 0, V P /4, and V P /2, we obtain
i D = 2, 1.13, and 0.5 mA
From Equation (3.33), we have
v DS (sat) = vG S − V P = vG S − (−3.5)
Therefore, for vG S = 0, V P /4, and V P /2, we obtain
v DS (sat) = 3.5, 2.63, and 1.75 V
Comment: The current capability of a JFET can be increased by increasing the
value of I DSS , which is a function of the transistor width.
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EXERCISE PROBLEM
3.19: The parameters of an n-channel JFET are I DSS = 12 mA,
V P = −4.5 V, and λ = 0. Determine VDS (sat) for VG S = −1.2 V, and calculate
I D for VDS > VDS (sat). (Ans. VDS (sat) = 3.3 V, I D = 6.45 mA)
Ex
As in the case of the MOSFET, the i D versus v DS characteristic for the JFET
may have a nonzero slope beyond the saturation point. This nonzero slope can be described through the following equation:
vG S 2
i D = I DSS 1 −
(1 + λ v DS )
(3.35)
VP
The output resistance ro is defined as
∂i D −1 ro =
∂v DS
(3.36)
vG S =const.
Using Equation (3.35), we find that
−1
VG S Q 2
ro = λI DSS 1 −
VP
(3.37(a))
or
ro ∼
= [λI D Q ]−1 =
1
λI D Q
(3.37(b))
The output resistance will be considered again when we discuss the small-signal
equivalent circuit of the JFET in the next chapter.
Enhancement-mode GaAs MESFETs can be fabricated with current–voltage
characteristics much like those of the enhancement-mode MOSFET. Therefore, for
the ideal enhancement-mode MESFET biased in the saturation region, we can write
i D = K n (vG S − VT N )2
(3.38(a))
For the ideal enhancement-mode MESFET biased in the nonsaturation region,
i D = K n 2(vG S − VT N )v DS − v 2DS
(3.38(b))
where K n is the conduction parameter and VT N is the threshold voltage, which in this
case is equivalent to the pinchoff voltage. For an n-channel enhancement-mode
MESFET, the threshold voltage is positive.
3.6.3
Common JFET Configurations: dc Analysis
There are several common JFET circuit configurations. We will look at a few of
these, using examples, and will illustrate the dc analysis and design of such circuits.
DESIGN EXAMPLE 3.20
Objective: Design a JFET circuit with a voltage divider biasing circuit.
Consider the circuit shown in Figure 3.61(a) with transistor parameters
I DSS = 12 mA, V P = −3.5 V, and λ = 0. Let R1 + R2 = 100 k. Design the circuit such that the dc drain current is I D = 5 mA and the dc drain-to-source voltage is
VDS = 5 V.
185
206
Part 1 Semiconductor Devices and Basic Applications
4.1
THE MOSFET AMPLIFIER
Objective: • Investigate the process by which a single-transistor
circuit can amplify a small, time-varying input signal and develop the
small-signal models of the transistor that are used in the analysis of
linear amplifiers.
In this chapter, we will be considering signals, analog circuits, and amplifiers. A
signal contains some type of information. For example, sound waves produced by a
speaking human contain the information the person is conveying to another person.
A sound wave is an analog signal. The magnitude of an analog signal can take on
any value, within limits, and may vary continuously with time. Electronic circuits
that process analog signals are called analog circuits. One example of an analog circuit is a linear amplifier. A linear amplifier magnifies an input signal and produces
an output signal whose magnitude is larger and directly proportional to the input
signal.
In this chapter, we analyze and design linear amplifiers that use field-effect transistors as the amplifying device. The term small signal means that we can linearize
the ac equivalent circuit. We will define what is meant by small signal in the case of
MOSFET circuits. The term linear amplifiers means that we can use superposition so
that the dc analysis and ac analysis of the circuits can be performed separately and
the total response is the sum of the two individual responses.
The mechanism with which MOSFET circuits amplify small time-varying
signals was introduced in the last chapter. In this section, we will expand that
discussion using the graphical technique, dc load line, and ac load line. In the
process, we will develop the various small-signal parameters of linear circuits and
the corresponding equivalent circuits.
4.1.1
VDD
iD
RD
vO
+
vi
+
–
vGS –
+
vDS
–
+
VGSQ
–
Figure 4.1 NMOS commonsource circuit with timevarying signal source in
series with gate dc source
Graphical Analysis, Load Lines,
and Small-Signal Parameters
Figure 4.1 shows an NMOS common-source circuit with a time-varying voltage
source in series with the dc source. We assume the time-varying input signal is sinusoidal. Figure 4.2 shows the transistor characteristics, dc load line, and Q-point,
where the dc load line and Q-point are functions of vG S , VD D , RD, and the transistor
parameters. For the output voltage to be a linear function of the input voltage, the
transistor must be biased in the saturation region. (Note that, although we primarily
use n-channel, enhancement-mode MOSFETs in our discussions, the same results
apply to the other MOSFETs.)
Also shown in Figure 4.2 are the sinusoidal variations in the gate-to-source voltage, drain current, and drain-to-source voltage, as a result of the sinusoidal source vi.
The total gate-to-source voltage is the sum of VG S Q and vi . As vi increases, the
instantaneous value of vG S increases, and the bias point moves up the load line. A
larger value of vG S means a larger drain current and a smaller value of v DS . For a
negative vi (the negative portion of the sine wave), the instantaneous value of vG S
decreases below the quiescent value, and the bias point moves down the load line. A
smaller vG S value means a smaller drain current and increased value of v DS . Once the
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Chapter 4 Basic FET Amplifiers
iD
vDS (sat)
Q-point
Time
vi
IDQ
Time
VGSQ
VDSQ
VDD
vDS
Time
Figure 4.2 Common-source transistor characteristics, dc load line, and sinusoidal variation
in gate-to-source voltage, drain current, and drain-to-source voltage
Q-point is established, we can develop a mathematical model for the sinusoidal, or
small-signal, variations in gate-to-source voltage, drain-to-source voltage, and drain
current.
The time-varying signal source vi in Figure 4.1 generates a time-varying
component of the gate-to-source voltage. In this case, vgs = vi , where vgs is the timevarying component of the gate-to-source voltage. For the FET to operate as a linear
amplifier, the transistor must be biased in the saturation region, and the instantaneous
drain current and drain-to-source voltage must also be confined to the saturation
region.
When symmetrical sinusoidal signals are applied to the input of an amplifier,
symmetrical sinusoidal signals are generated at the output, as long as the amplifier
operation remains linear. We can use the load line to determine the maximum output
symmetrical swing. If the output exceeds this limit, a portion of the output signal will
be clipped and signal distortion will occur.
In the case of FET amplifiers, the output signal must avoid cutoff (i D = 0) and
must stay in the saturation region (v DS > v DS (sat)). This maximum range of output
signal can be determined from the load line in Figure 4.2.
Transistor Parameters
We will be dealing with time-varying as well as dc currents and voltages in this
chapter. Table 4.1 gives a summary of notation that will be used. This notation was
Table 4.1
Summary of notation
Variable
Meaning
iD, vGS
ID, VGS
id, vgs
Id, Vgs
Total instantaneous values
DC values
Instantaneous ac values
Phasor values
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Part 1 Semiconductor Devices and Basic Applications
discussed in the Prologue, but is repeated here for convenience. A lowercase letter
with an upper case subscript, such as i D or vG S , indicates a total instantaneous value.
An uppercase letter with an uppercase subscript, such as ID or VG S , indicates a dc
quantity. A lowercase letter with a lowercase subscript, such as i d and vgs , indicates
an instantaneous value of an ac signal. Finally, an uppercase letter with a lowercase
subscript, such as Id or Vgs , indicates a phasor quantity. The phasor notation, which
is also reviewed in the Prologue, becomes especially important in Chapter 7 during
the discussion of frequency response. However, the phasor notation will generally be
used in this chapter in order to be consistent with the overall ac analysis.
From Figure 4.1, we see that the instantaneous gate-to-source voltage is
vG S = VG S Q + vi = VG S Q + vgs
(4.1)
where VGSQ is the dc component and vgs is the ac component. The instantaneous
drain current is
i D = K n (vG S − VT N )2
(4.2)
Substituting Equation (4.1) into (4.2) produces
i D = K n [VG S Q + vgs − VT N ]2 = K n [(VG S Q − VT N ) + vgs ]2
(4.3(a))
2
i D = K n (VG S Q − VT N )2 + 2K n (VG S Q − VT N )vgs + K n vgs
(4.3(b))
or
The first term in Equation (4.3(b)) is the dc or quiescent drain current IDQ, the
second term is the time-varying drain current component that is linearly related to the
signal vgs, and the third term is proportional to the square of the signal voltage. For
a sinusoidal input signal, the squared term produces undesirable harmonics, or nonlinear distortion, in the output voltage. To minimize these harmonics, we require
vgs 2(VG S Q − VT N )
(4.4)
which means that the third term in Equation (4.3(b)) will be much smaller than the
second term. Equation (4.4) represents the small-signal condition that must be satisfied for linear amplifiers.
2
Neglecting the vgs
term, we can write Equation (4.3(b)) as
i D = I D Q + id
(4.5)
Again, small-signal implies linearity so that the total current can be separated into
a dc component and an ac component. The ac component of the drain current is
given by
i d = 2K n (VG S Q − VT N )vgs
(4.6)
The small-signal drain current is related to the small-signal gate-to-source voltage by the transconductance gm. The relationship is
id
gm =
= 2K n (VG S Q − VT N )
(4.7)
vgs
The transconductance is a transfer coefficient relating output current to input voltage
and can be thought of as representing the gain of the transistor.
The transconductance can also be obtained from the derivative
∂i D gm =
= 2K n (VG S Q − VT N )
(4.8(a))
∂vG S vG S =VG S Q =const.
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Chapter 4 Basic FET Amplifiers
iD
Time
Slope = gm
IDQ
Time
VTN
VGS
vGS
Figure 4.3 Drain current versus gate-to-source voltage characteristics, with superimposed
sinusoidal signals
which can be written as
gm = 2 K n I D Q
(4.8(b))
The drain current versus gate-to-source voltage for the transistor biased in the
saturation region is given in Equation (4.2) and is shown in Figure 4.3. The transconductance gm is the slope of the curve. If the time-varying signal vgs is sufficiently
small, the transconductance gm is a constant. With the Q-point in the saturation
region, the transistor operates as a current source that is linearly controlled by vgs. If
the Q-point moves into the nonsaturation region, the transistor no longer operates as
a linearly controlled current source.
As shown in Equation (4.8(a)), the transconductance is directly proportional to
the conduction parameter Kn, which in turn is a function of the width-to-length ratio.
Therefore, increasing the width of the transistor increases the transconductance, or
gain, of the transistor.
EXAMPLE 4.1
Objective: Calculate the transconductance of an n-channel MOSFET.
Consider an n-channel MOSFET with parameters VT N = 0.4 V, kn = 100 μA/V2 ,
and W/L = 25. Assume the drain current is I D = 0.40 mA.
Solution: The conduction parameter is
Kn =
kn W
0.1
·
=(
)(25) = 1.25 mA/V2
2 L
2
Assuming the transistor is biased in the saturation region, the transconductance is
determined from Equation (4.8(b)) as
√
gm = 2 K n I D Q = 2 (1.25)(0.4) = 1.41 mA/V
Comment: The value of the transconductance can be increased by increasing the
transistor W/L ratio and also by increasing the quiescent drain current.
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EXERCISE PROBLEM
Ex 4.1: For an n-channel MOSFET biased in the saturation region, the parameters
are kn = 100 μA/V2 , VT N = 0.6 V, and I D Q = 0.8 mA. Determine the transistor
width-to-length ratio such that the transconductance is gm = 1.8 mA/V. (Ans. 20.25)
AC Equivalent Circuit
From Figure 4.l, we see that the output voltage is
v DS = v O = VD D − i D R D
(4.9)
Using Equation (4.5), we obtain
v O = VD D − (I D Q + i d )R D = (VD D − I D Q R D ) − i d R D
(4.10)
The output voltage is also a combination of dc and ac values. The time-varying
output signal is the time-varying drain-to-source voltage, or
vo = vds = −i d R D
(4.11)
Also, from Equations (4.6) and (4.7), we have
i d = gm vgs
(4.12)
In summary, the following relationships exist between the time-varying signals
for the circuit in Figure 4.1. The equations are given in terms of the instantaneous
ac values, as well as the phasors. We have
vgs = vi
or
Vgs = Vi
(4.13)
i d = gm vgs
or
Id = gm Vgs
(4.14)
vds = −i d R D
or
Vds = −Id R D
(4.15)
and
vo
id
vi
+
–
+
vgs
RD
–
Figure 4.4 AC equivalent
circuit of common-source
amplifier with NMOS
transistor
and
The ac equivalent circuit in Figure 4.4 is developed by setting the dc sources in
Figure 4.l equal to zero. The small-signal relationships are given in Equations (4.13),
(4.14), and (4.15). As shown in Figure 4.l, the drain current, which is composed of ac
signals superimposed on the quiescent value, flows through the voltage source VD D .
Since the voltage across this source is assumed to be constant, the sinusoidal current
produces no sinusoidal voltage component across this element. The equivalent ac
impedance is therefore zero, or a short circuit. Consequently, in the ac equivalent
circuit, the dc voltage sources are equal to zero. We say that the node connecting RD
and VD D is at signal ground.
4.1.2
Small-Signal Equivalent Circuit
Now that we have the ac equivalent circuit for the NMOS amplifier circuit, (Figure
4.4), we must develop a small-signal equivalent circuit for the transistor.
Initially, we assume that the signal frequency is sufficiently low so that any
capacitance at the gate terminal can be neglected. The input to the gate thus appears
as an open circuit, or an infinite resistance. Equation (4.14) relates the small-signal
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Chapter 4 Basic FET Amplifiers
D
id (Id)
G
+
G
+
vds (Vds)
+
D
vgs (Vgs)
vgs (Vgs)
–
–
gmvgs id (Id)
(gmVgs)
+
vds (Vds)
–
–
S
S
(a)
(b)
Figure 4.5 (a) Common-source NMOS transistor with small-signal parameters and
(b) simplified small-signal equivalent circuit for NMOS transistor
drain current to the small-signal input voltage, and Equation (4.7) shows that the
transconductance gm is a function of the Q-point. The resulting simplified smallsignal equivalent circuit for the NMOS device is shown in Figure 4.5. (The phasor
components are in parentheses.)
This small-signal equivalent circuit can also be expanded to take into account the
finite output resistance of a MOSFET biased in the saturation region. This effect, discussed in the last chapter, is a result of the nonzero slope in the iD versus v DS curve.
We know that
i D = K n [(vG S − VT N )2 (1 + λv DS )]
(4.16)
where λ is the channel-length modulation parameter and is a positive quantity. The
small-signal output resistance, as previously defined, is
∂i D −1 ro =
(4.17)
∂v
DS
vG S =VG S Q =const.
ro = [λK n (VG S Q − VT N )2 ]−1 ∼
= [λI D Q ]−1
+
–
Vgs
+
–
gmVgs
ro
+
+
Id
gmVgs
Vgs
ro
Vds
–
S
(4.18)
This small-signal output resistance is also a function of the Q-point parameters.
The expanded small-signal equivalent circuit of the n-channel MOSFET is
shown in Figure 4.6 in phasor notation. Note that this equivalent circuit is a transconductance amplifier in that the input signal is a voltage and the output signal is a
current. This equivalent circuit can now be inserted into the amplifier ac equivalent
circuit in Figure 4.4 to produce the circuit in Figure 4.7.
Vi
D
–
or
+
G
Vo
RD Vds
–
Figure 4.7 Small-signal equivalent circuit of common-source circuit with NMOS transistor
model
EXAMPLE 4.2
Objective: Determine the small-signal voltage gain of a MOSFET circuit.
For the circuit in Figure 4.l, assume parameters are: VG S Q = 2.12 V, VD D = 5 V,
and R D = 2.5 k. Assume transistor parameters are: VT N = 1 V. K n = 0.80
mA/V2, and λ = 0.02 V−1 . Assume the transistor is biased in the saturation region.
Figure 4.6 Expanded smallsignal equivalent circuit,
including output resistance,
for NMOS transistor
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Solution: The quiescent values are
IDQ ∼
= K n (VG S Q − VT N )2 = (0.8)(2.12 − 1)2 = 1.0 mA
and
VDS Q = VD D − I D Q R D = 5 − (1)(2.5) = 2.5 V
Therefore,
VDS Q = 2.5 V > VDS (sat) = VG S − VT N = 1.82 − 1 = 0.82 V
which means that the transistor is biased in the saturation region, as initially
assumed, and as required for a linear amplifier. The transconductance is
gm = 2K n (VG S Q − VT N ) = 2(0.8)(2.12 − 1) = 1.79 mA/V
and the output resistance is
ro = [λI D Q ]−1 = [(0.02)(1)]−1 = 50 k
From Figure 4.7, the output voltage is
Vo = −gm Vgs (ro R D )
Since Vgs = Vi , the small-signal voltage gain is
Av =
Vo
= −gm (ro Rd ) = −(1.79)(502.5) = −4.26
Vi
Comment: The magnitude of the ac output voltage is 4.26 times larger than the mag-
nitude of the input voltage. Hence, we have an amplifier. Note that the small-signal
voltage gain contains a minus sign, which means that the sinusoidal output voltage is
180 degrees out of phase with respect to the input sinusoidal signal.
EXERCISE PROBLEM
Ex 4.2: For the circuit shown in Figure 4.1, VD D = 3.3 V and R D = 10 k. The
transistor parameters are VT N = 0.4 V, kn = 100 μA/V2 , W/L = 50, and
λ = 0.025 V−1 . Assume the transistor is biased such that I D Q = 0.25 mA. (a) Verify
that the transistor is biased in the saturation region. (b) Determine the smallsignal parameters gm and ro . (c) Determine the small-signal voltage gain.
(Ans. (a) VG S Q = 0.716 V and VDS Q = 0.8 V so that VDS > VDS (sat);
(b) gm = 1.58 mA/V, ro = 160 k; (c) −14.9)
Problem-Solving Technique: MOSFET AC Analysis
Since we are dealing with linear amplifiers, superposition applies, which means
that we can perform the dc and ac analyses separately. The analysis of the MOSFET
amplifier proceeds as follows:
1.
2.
3.
Analyze the circuit with only the dc sources present. This solution is the dc or
quiescent solution. The transistor must be biased in the saturation region in
order to produce a linear amplifier.
Replace each element in the circuit with its small-signal model, which means
replacing the transistor by its small-signal equivalent circuit.
Analyze the small-signal equivalent circuit, setting the dc source components
equal to zero, to produce the response of the circuit to the time-varying input
signals only.
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Id
G
–
gmVsg
Vsg
ro
+
D
G
–
+
Vsd
Vgs
+
–
S
Id
D
+
gmVgs
ro
Vds
–
S
(a)
(b)
Figure 4.11 Small signal equivalent circuit of a p-channel MOSFET showing (a) the
conventional voltage polarities and current directions and (b) the case when the voltage
polarities and current directions are reversed.
This expression for the small-signal voltage gain of the p-channel MOSFET
amplifier is exactly the same as that for the n-channel MOSFET amplifier. The
negative sign indicates that a 180-degree phase reversal exists between the ouput and
input signals, for both the PMOS and the NMOS circuit.
We may note that if the polarity of the small-signal gate-to-source voltage is
reversed, then the small-signal drain current direction is reversed. This change of polarity is shown in Figure 4.11. Figure 4.11(a) shows the conventional voltage polarity and current directions in a PMOS transistor. If the control voltage polarity is reversed as shown in Figure 4.11(b), then the dependent current direction is also
reversed. The equivalent circuit shown in Figure 4.11(b) is then the same as that of
the NMOS transistor. However, the author prefers to use the small-signal equivalent
circuit in Figure 4.9 to be consistent with the voltage polarities and current directions
of the PMOS transistor.
Modeling the Body Effect
4.1.3
As mentioned in Section 3.1.9, Chapter 3, the body effect occurs in a MOSFET in
which the substrate, or body, is not directly connected to the source. For an NMOS
device, the body is connected to the most negative potential in the circuit and will be
at signal ground. Figure 4.12(a) shows the four-terminal MOSFET with dc voltages
and Figure 4.12(b) shows the device with ac voltages. Keep in mind that vSB must be
greater than or equal to zero. The simplified current-voltage relation is
i D = K n (vG S − VT N )2
(4.22)
and the threshold voltage is given by
VT N = VT N O + γ 2φ f + v S B − 2φ f
D
D
id
ID
G
–
VSB
+
VGS – +
S
(a)
B
G
+
+
vbs
B
vgs – –
S
(b)
Figure 4.12 The four-terminal NMOS device with (a) dc voltages and (b) ac voltages
(4.23)
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If an ac component exists in the source-to-body voltage, vSB, there will be an ac
component induced in the threshold voltage, which causes an ac component in the
drain current. Thus, a back-gate transconductance can be defined as
∂ VT N ∂i D −∂i D ∂i D
gmb =
=
=
−
·
(4.24)
∂v B S Q - pt
∂v S B Q - pt
∂ VT N
∂v S B Q - pt
Using Equation (4.22), we find
∂i D
= −2K n (vG S − VT N ) = −gm
∂ VT N
and using Equation (4.23), we find
∂ VT N
γ
= ≡η
∂v S B
2 2φ f + v S B
The back-gate transconductance is then
(4.25(a))
(4.25(b))
gmb = −(−gm ) · (η) = gm η
(4.26)
Including the body effect, the small-signal equivalent circuit of the MOSFET is
shown in Figure 4.13. We note the direction of the current and the polarity of the
small-signal source-to-body voltage. If vbs > 0, then v S B decreases, VT N decreases,
and iD increases. The current direction and voltage polarity are thus consistent.
For φ f = 0.35 V and γ = 0.35 V1/2 , the value of η from Equation (4.25(b)) is
η∼
= 0.23. Therefore, η will be in the range 0 ≤ η ≤ 0.23. The value of vbs will
depend on the particular circuit.
In general, we will neglect gmb in our hand analyses and designs, but will investigate the body effect in PSpice analyses.
G
D
+
vgs
gmsvgs
gmbvbs
vds
–
–
S
+
ro
–
vbs
+
B
Figure 4.13 Small-signal equivalent circuit of NMOS device including body effect
Test Your Understanding
TYU 4.1 The parameters of an n-channel MOSFET are: VT N = 0.6 V, kn =
100 μA/V2 , and λ = 0.015 V−1 . The transistor is biased in the saturation region with
I D Q = 1.2 mA. (a) Design the width-to-length ratio such that the transconductance is
gm = 2.5 mA/V. (b) Determine the small-signal output resistance ro . (Ans. (a) 26.0,
(b) 55.6 k).
TYU 4.2 For the circuit shown in Figure 4.1, VD D = 3.3 V and R D = 8 k.
The transistor parameters are VT N = 0.4 V, K n = 0.5 mA/V2 , and λ = 0.02 V−1 .
(a) Determine VG S Q and VDS Q for I D Q = 0.15 mA. (b) Calculate gm , ro , and the smallsignal voltage gain. (Ans. (a) VG S Q = 0.948 V, VDS Q = 2.1 V; (b) gm = 0.548 mA/V,
ro = 333 k, Av = −4.28).
215
216
Part 1 Semiconductor Devices and Basic Applications
TYU 4.3 For the circuit in Figure 4.1, the circuit and transistor parameters are given
in Exercise TYU 4.2. If vi = 25 sin ωt (mV), determine i D and υ DS . (Ans.
i D = 0.15 + 0.0137 sin ωt (mA), υ DS = 2.1 − 0.11 sin ωt (V)).
TYU 4.4 The parameters for the circuit in Figure 4.8 are VD D = 5 V and R D = 5 k.
The transistor parameters are VT P = −0.4 V, K p = 0.4 mA/V2 , and λ = 0. (a) Determine VSG Q and I D Q such that VS D Q = 3 V. (b) Calculate gm and the small-signal
voltage gain. (Ans. (a) I D Q = 0.4 mA, VSG Q = 1.4 V; (b) gm = 0.8 mA/V,
Av = −4).
TYU 4.5 A transistor has the same parameters as those given in Exercise Ex4.1. In
addition, the body effect coefficient is γ = 0.40 V1/2 and φ f = 0.35 V. Determine
the value of η and the back-gate transconductance gmb for (a) v S B = 1 V and
(b) v S B = 3 V. (Ans. (a) η = 0.153, (b) η = 0.104).
4.2
BASIC TRANSISTOR AMPLIFIER
CONFIGURATIONS
Objective: • Discuss the three basic transistor amplifier configurations.
As we have seen, the MOSFET is a three-terminal device. Three basic singletransistor amplifier configurations can be formed, depending on which of the three
transistor terminals is used as signal ground. These three basic configurations are
appropriately called common source, common drain (source follower), and
common gate.
The input and output resistance characteristics of amplifiers are important in
determining loading effects. These parameters, as well as voltage gain, for the three
basic MOSFET circuit configurations will be determined in the following sections.
The characteristics of the three types of amplifiers will then allow us to understand
under what condition each amplifier is most useful.
Initially, we will consider MOSFET amplifier circuits that emphasize discrete
designs, in that resistor biasing will be used. The purpose is to become familiar
with basic MOSFET amplifier designs and their characteristics. In Section 4.7, we
will begin to consider integrated circuit MOSFET designs that involve all-transistor circuits and current source biasing. These initial designs provide an introduction to more advanced MOS amplifier designs that will be considered in Part 2 of
the text.
4.3
THE COMMON-SOURCE AMPLIFIER
Objective: • Analyze the common-source amplifier and become
familiar with the general characteristics of this circuit.
In this section, we consider the first of the three basic circuits—the common-source
amplifier. We will analyze several basic common-source circuits, and will determine
small-signal voltage gain and input and output impedances.
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A Basic Common-Source Configuration
4.3.1
Figure 4.14 shows the basic common-source circuit with voltage-divider biasing. We
see that the source is at ground potential—hence the name common source. The signal from the signal source is coupled into the gate of the transistor through the coupling capacitor CC, which provides dc isolation between the amplifier and the signal
source. The dc transistor biasing is established by R1 and R2, and is not disturbed
when the signal source is capacitively coupled to the amplifier.
VDD
RD
R1
vi
vO
CC1
RSi
+
–
R2
Figure 4.14 Common-source circuit with voltage divider biasing and coupling capacitor
If the signal source is a sinusoidal voltage at frequency f, then the magnitude
of the capacitor impedance is |Z C | = [1/(2π f CC )]. For example, assume that
CC = 10 μF and f = 2 kHz. The magnitude of the capacitor impedance is then
|Z C | =
1
1
∼
=
= 8
3
2π f CC
2π(2 × 10 )(10 × 10−6 )
The magnitude of this impedance is generally much less than the Thevenin resistance
at the capacitor terminals. We can therefore assume that the capacitor is essentially a
short circuit to signals with frequencies greater than 2 kHz. We will also neglect, in
this chapter, any capacitance effects within the transistor.
For the circuit shown in Figure 4.14, assume that the transistor is biased in the saturation region by resistors R1 and R2, and that the signal frequency is sufficiently large
for the coupling capacitor to act essentially as a short circuit. The signal source is represented by a Thevenin equivalent circuit, in which the signal voltage source vi is in
series with an equivalent source resistance RSi. As we will see, RSi should be much less
than the amplifier input resistance, Ri = R1 R2 , in order to minimize loading effects.
Figure 4.15 shows the resulting small-signal equivalent circuit. The smallsignal variables, such as the input signal voltage Vi, are given in phasor form.
RSi
Ri
Ro
G
D
+
Vi
+
–
Id
R1 ⎜⎜R2 Vgs
gmVgs
ro
+
Vo
RD Vds
–
–
S
Figure 4.15 Small-signal equivalent circuit, assuming coupling capacitor acts as a short circuit
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iD
ID (max) =
VDD
RD
VDS (sat) = VGS – VTN
Transition point
Q-point
dc load line, slope = –
IDQ
VDSQ
VDD
1
RD
vDS
Figure 4.16 DC load line and transition point separating saturation and nonsaturation regions
Since the source is at ground potential, there is no body effect. The output voltage
is
Vo = −gm Vgs (ro R D )
The input gate-to-source voltage is
Ri
Vgs =
· Vi
Ri + R Si
so the small-signal voltage gain is
Vo
Ri
Av =
= −gm (ro R D ) ·
Vi
Ri + R Si
(4.27)
(4.28)
(4.29)
We can also relate the ac drain current to the ac drain-to-source voltage, as
Vds = −Id (R D ).
Figure 4.16 shows the dc load line, the transition point (that separates the saturation bias region and nonsaturation bias region), and the Q-point, which is in the
saturation region. In order to provide the maximum symmetrical output voltage swing
and keep the transistor biased in the saturation region, the Q-point must be near the
middle of the saturation region. At the same time, the input signal must be small
enough for the amplifier to remain linear.
The input and output resistances of the amplifier can be determined from Figure 4.15. The input resistance to the amplifier is Ri = R1 R2 . Since the low-frequency
input resistance looking into the gate of the MOSFET is essentially infinite, the input
resistance is only a function of the bias resistors. The output resistance looking back
into the output terminals is found by setting the independent input source Vi equal to
zero, which means that Vgs = 0. The output resistance is therefore Ro = R D ro .
EXAMPLE 4.3
Objective: Determine the small-signal voltage gain and input and output resistances
of a common-source amplifier.
For the circuit shown in Figure 4.14, the parameters are: VD D = 3.3 V,
R D = 10 k, R1 = 140 k, R2 = 60 k, and RSi = 4 k. The transistor parameters
are: VT N = 0.4 V, K n = 0.5 mA/V2 , and λ = 0.02 V−1 .
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Solution (dc calculations): The dc or quiescent gate-to-source voltage is
VG S Q =
R2
60
(VD D ) =
(3.3) = 0.99 V
R1 + R2
140 + 60
The quiescent drain current is
I D Q = K n (VG S Q − VT N )2 = (0.5)(0.99 − 0.4)2 = 0.174 mA
and the quiescent drain-to-source voltage is
VDS Q = VD D − I D Q R D = 3.3 − (0.174)(10) = 1.56 V
Since VDS Q > VG S Q − VT N , the transistor is biased in the saturation region.
Small-signal Voltage Gain: The small-signal transconductance gm is then
√
gm = 2 K n I D Q = 2 (0.5)(0.174) = 0.590 mA/V
and the small-signal output resistance is
ro =
1
1
=
= 287 k
λI Q
(0.02)(0.174)
The input resistance to the amplifier is
Ri = R1 R2 = 14060 = 42 k
From Figure 4.15 and Equation (4.29), the small-signal voltage gain is
Ri
42
= −(0.59)(28710)
Av = −gm (ro R D )
Ri + R Si
42 + 4
or
Av = −5.21
Input and Output Resistances: As already calculated, the amplifier input resis-
tance is
Ri = R1 R2 = 14060 = 42 k
and the amplifier output resistance is
Ro = R D ro = 10287 = 9.66 k
Comment: The resulting Q-point is not in the center of the saturation region. There-
fore, this circuit does not achieve the maximum symmetrical output voltage swing in
this case.
Discussion: The small-signal input gate-to-source voltage is
Vgs =
Ri
Ri + R Si
· Vi =
42
42 + 4
· Vi = (0.913) · Vi
Since RSi is not zero, the amplifier input signal Vgs is approximately 91 percent of
the signal voltage. This is called a loading effect. Even though the input resistance to
the gate of the transistor is essentially infinite, the bias resistors greatly influence the
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amplifier input resistance and loading effect. This loading effect can be eliminated
or minimized when current source biasing is considered.
EXERCISE PROBLEM
Ex 4.3: The parameters of the circuit shown in Figure 4.14 are VD D = 5 V,
R1 = 520 k, R2 = 320 k, R D = 10 k, and R Si = 0. Assume transistor parameters of VT N = 0.8 V, K n = 0.20 mA/V2, and λ = 0. (a) Determine the smallsignal transistor parameters gm and ro. (b) Find the small-signal voltage gain.
(c) Calculate the input and output resistances Ri and Ro (see Figure 4.15). (Ans.
(a) gm = 0.442 mA/V, ro = ∞; (b) Av = −4.42; (c) Ri = 198 k, Ro = R D =
10 k)
DESIGN EXAMPLE 4.4
Objective: Design the bias of a MOSFET circuit such that the Q-point is in the
middle of the saturation region. Determine the resulting small-signal voltage gain.
Specifications: The circuit to be designed has the configuration shown in
Figure 4.17. Let R1 R2 = 100 k. Design the circuit such that the Q-point is
I D Q = 2 mA and the Q-point is in the middle of the saturation region.
VDD = 12 V
R1
CC
vi
+
–
RD
ID = 2 mA
vO
R2
Figure 4.17 Common-source NMOS transistor circuit
Choices: A transistor with nominal parameters VT N = 1 V, kn = 80 μA/V2 ,
W/L = 25, and λ = 0.015 V−1 is available.
Solution (dc design): The load line and the desired Q-point are given in Figure 4.18.
If the Q-point is to be in the middle of the saturation region, the current at the transition point must be 4 mA.
The conductivity parameter is
k W
0.080
Kn = n ·
=
(25) = 1 mA/V2
2 L
2
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Chapter 4 Basic FET Amplifiers
iD
Load line, slope = – 1
RD
VDS (sat) = VGS – VTN
4 mA
Q-point
VGSQ = 2.41 V
2 mA
VDSt (sat)
VDSQ = 7 V
VDD = 12 V vDS
Figure 4.18 DC load line and transition point for NMOS circuit shown in Figure 4.17
We can now calculate VDS (sat) at the transition point. The subscript t indicates
transition point values. To determine VG St , we use
I Dt = 4 = K n (VG St − VT N )2 = 1(VG St − 1)2
which yields
VG St = 3 V
Therefore
VDSt = VG St − VT N = 3 − 1 = 2 V
If the Q-point is in the middle of the saturation region, then VDS Q = 7 V, which
would yield a 10 V peak-to-peak symmetrical output voltage. From Figure 4.17, we
can write
VDS Q = VD D − I D Q R D
or
RD =
VD D − VDS Q
12 − 7
= 2.5 k
=
IDQ
2
We can determine the required quiescent gate-to-source voltage from the current
equation, as follows:
I D Q = 2 = K n (VG S Q − VT N )2 = (1)(VG S Q − 1)2
or
VG S Q = 2.41 V
Then
R2
1
R1 R2
VG S Q = 2.41 =
(VD D ) =
(VD D )
R1 + R2
R1
R1 + R2
(100)(12)
Ri
=
· VD D =
R1
R1
which yields
R1 = 498 k and
R2 = 125 k
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Solution (ac analysis): The small-signal transistor parameters are
gm = 2 K n I D Q = 2 (1)(2) = 2.83 mA/V
and
ro =
1
1
=
= 33.3 k
λI D Q
(0.015)(2)
The small-signal equivalent circuit is the same as shown in Figure 4.7. The smallsignal voltage gain is
Av =
Vo
= −gm (ro R D ) = −(2.83)(33.32.5)
Vi
or
Av = −6.58
Comment: Establishing the Q-point in the middle of the saturation region allows the
maximum symmetrical swing in the output voltage, while keeping the transistor
biased in the saturation region.
EXERCISE PROBLEM
Ex 4.4: Consider the circuit shown in Figure 4.14. Assume transistor parameters
of VT N = 0.8 V, K n = 0.20 mA/V2, and λ = 0. Let VD D = 5 V, Ri = R1 R2 =
200 k, and R Si = 0. Design the circuit such that I D Q = 0.5 mA and the Q-point
is in the center of the saturation region. Find the small-signal voltage gain.
(Ans. R D = 2.76 k, R1 = 420 k, R2 = 382 k, Av = −1.75)
Common-Source Amplifier with Source Resistor
4.3.2
A source resistor RS tends to stabilize the Q-point against variations in transistor
parameters (Figure 4.19). If, for example, the value of the conduction parameter
varies from one transistor to another, the Q-point will not vary as much if a source
+5 V
R1 = 165 kΩ
CC
vi
+
–
R2 = 35 kΩ
RD = 7 kΩ
vO
RS = 0.5 kΩ
–5 V
Figure 4.19 Common-source circuit with source resistor and positive and negative supply
voltages
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Chapter 4 Basic FET Amplifiers
resistor is included in the circuit. However, as shown in the following example, a
source resistor also reduces the signal gain.
The circuit in Figure 4.19 is an example of a situation in which the body effect
should be taken into account. The substrate (not shown) would normally be
connected to the −5 V supply, so that the body and substrate terminals are not at the
same potential. However, in the following example, we will neglect this effect.
The circuit shown in Figure 4.20(a) is a PMOS version of the common-source
amplifier with a source resistor R S included.
EXAMPLE 4.5
Objective: Determine the small-signal voltage gain of a PMOS transistor circuit.
Consider the circuit shown in Figure 4.20(a). The transistor parameters are
K p = 0.80 mA/V2 , VT P = −0.5 V, and λ = 0. The quiescent drain current is found
to be I D Q = 0.297 mA.
The small-signal equivalent circuit is shown in Figure 4.20(b). To sketch the
small-signal equivalent circuit, start with the three terminals of the transistor, draw in
the transistor equivalent circuit between these three terminals, and then sketch in the
other circuit elements around the transistor.
Solution: The small-signal output voltage is
Vo = +gm Vsg R D
Writing a KVL equation from the input around the gate–source loop, we find
Vi = −Vsg − gm Vsg R S
or
Vsg =
−Vi
1 + gm R S
Substituting this expression for Vsg into the output voltage equation, we find the
small-signal voltage gain as
Av =
Vo
−gm R D
=
Vi
1 + gm R S
+3 V
G
R1 = 100 kΩ
RS = 3 kΩ
Vsg
Cc
vi
+
–
D
vo
R2 = 200 kΩ
Vi
+
–
gmVsg
+
R1 ⎜⎜ R2
S
RD = 10 kΩ
RS
–3 V
(a)
Vo
–
(b)
Figure 4.20 (a) PMOS circuit for Example 4.5, and (b) small-signal equivalent circuit
RD
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The small-signal transconductance is
gm = 2 K p I D Q = 2 (0.80)(0.297) = 0.975 mA/V
We then find the small-signal voltage gain as
Av =
−(0.975)(10)
1 + (0.975)(3)
or
Av = −2.48
Comment: The analysis of a PMOS transistor circuit is essentially the same as that
of an NMOS transistor circuit. The voltage gain of a MOS transistor circuit that contains a source resistor is degraded compared to a circuit without a source resistor.
However, the Q-point tends to be stabilized.
Discussion: We mentioned that including a source resistor tends to stabilize the
circuit characteristics against any changes in transistor parameters. If, for example, the conduction parameter K p varies by ±10 percent, we find the following
results.
Kp(mA/V2)
gm(mA/V)
Av(V/V)
0.72
0.80
0.88
0.9121
0.9749
1.035
−2.441
−2.484
−2.521
With a ±10 percent variation in K p , there is less than a ±1.8 percent variation in the
voltage gain.
EXERCISE PROBLEM
Ex 4.5: For the circuit shown in Figure 4.19, the transistor parameters are
VT N = 0.8 V, K n = 1 mA/V2 , and λ = 0. (a) From the dc analysis, find I D Q and
VDS Q . (b) Determine the small-signal voltage gain. (Ans. (a) I D Q = 0.494 mA,
VDS Q = 6.30 V; (b) Av = −5.78).
4.3.3
Common-Source Circuit
with Source Bypass Capacitor
A source bypass capacitor added to the common-source circuit with a source resistor
will minimize the loss in the small-signal voltage gain, while maintaining the Qpoint stability. The Q-point stability can be further increased by replacing the source
resistor with a constant-current source. The resulting circuit is shown in Figure 4.21,
assuming an ideal signal source. If the signal frequency is sufficiently large so that
the bypass capacitor acts essentially as an ac short-circuit, the source will be held at
signal ground.
EXAMPLE 4.6
Objective: Determine the small-signal voltage gain of a circuit biased with a constant-
current source and incorporating a source bypass capacitor.
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Chapter 4 Basic FET Amplifiers
+5 V
RD = 7 kΩ
vO
vi
+
–
RG = 200 kΩ
CS
IQ = 0.5 mA
–5 V
Figure 4.21 NMOS common-source circuit with source bypass capacitor
For the circuit shown in Figure 4.21, the transistor parameters are: VT N = 0.8 V,
K n = 1 mA/V2, and λ = 0.
Solution: Since the dc gate current is zero, the dc voltage at the source terminal is
VS = −VG S Q , and the gate-to-source voltage is determined from
I D Q = I Q = K n (VG S Q − VT N )2
or
0.5 = (1)(VG S Q − 0.8)2
which yields
VG S Q = −VS = 1.51 V
The quiescent drain-to-source voltage is
VDS Q = VD D − I D Q R D − VS = 5 − (0.5)(7) − (−1.51) = 3.01 V
The transistor is therefore biased in the saturation region.
The small-signal equivalent circuit is shown in Figure 4.22. The output voltage is
Vo = −gm Vgs R D
Vo
+
Vi
+
–
RG Vgs
gmVgs
RD = 7 kΩ
–
Flgure 4.22 Small-signal equivalent circuit, assuming the source bypass capacitor acts as a
short circuit
Since Vgs = Vi , the small-signal voltage gain is
Vo
Av =
= −gm R D = −(1.414)(7) = −9.9
Vi
Comment: Comparing the small-signal voltage gain of 9.9 in this example to the
2.48 calculated in Example 4.5, we see that the magnitude of the gain increases when
a source bypass capacitor is included.
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EXERCISE PROBLEM
Ex 4.6: The common-source amplifier in Figure 4.23 has transistor parameters
k p = 40 μA/V2 , W/L = 40, VT P = −0.4 V, and λ = 0.02 V−1 . (a) Determine
I D Q and VS D Q . (b) Find the small-signal voltage gain. (Ans. (a) I D Q = 1.16 mA,
VS D Q = 2.29 V; (b) Av = −3.68)
+3 V
RS = 1.2 kΩ
CC1
+
–
vi
CS
vO
RG = 100 kΩ
RD = 2 kΩ
–3 V
Figure 4.23 Figure for Exercise Ex 4.6
Test Your Understanding
TYU 4.6 Consider the common-source amplifier in Figure 4.24 with transistor
parameters VT N = 1.8 V, K n = 0.15 mA/V2, and λ = 0. (a) Calculate I D Q and
VDS Q . (b) Determine the small-signal voltage gain. (c) Discuss the purpose of RG and
its effect on the small-signal operation of the amplifier. (Ans. (a) I D Q = 1.05 mA,
VDS Q = 4.45 V; (b) Av = −2.65)
+5 V
VDD = 15 V
RS
RD = 10 kΩ
RG = 5 MΩ
CC1
CS
CC2
RL = 5 kΩ
vi
CC
vo
+
–
vi
+
–
vO
RG = 1 MΩ
RD
–5 V
Figure 4.24 Figure for Exercise TYU 4.6
Figure 4.25 Figure for Exercise TYU 4.7
TYU 4.7 The parameters of the transistor shown in Figure 4.25 are: VT P = +0.8 V,
K p = 0.5 mA/V2, and λ = 0.02 V−1 . (a) Determine RS and RD such that I D Q =
0.8 mA and VS D Q = 3 V. (b) Find the small-signal voltage gain. (Ans. (a) R S =
5.67 k, R D = 3.08 k; (b) Av = −3.71)
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Chapter 4 Basic FET Amplifiers
4.4
THE COMMON-DRAIN
(SOURCE-FOLLOWER) AMPLIFIER
Objective: • Analyze the common-drain (source-follower) amplifier
and become familiar with the general characteristics of this circuit.
The second type of MOSFET amplifier to be considered is the common-drain
circuit. An example of this circuit configuration is shown in Figure 4.26. As seen in
the figure, the output signal is taken off the source with respect to ground and the drain
is connected directly to VD D . Since VD D becomes signal ground in the ac equivalent
circuit, we have the name common drain. The more common name is source follower.
The reason for this name will become apparent as we proceed through the analysis.
VDD
R1
RSi
CC
vO
vi
+
–
R2
RS
Flgure 4.26 NMOS source-follower or common-drain amplifier
Small-Signal Voltage Gain
4.4.1
The dc analysis of the circuit is exactly the same as we have already seen, so we will
concentrate on the small-signal analysis. The small-signal equivalent circuit, assuming the coupling capacitor acts as a short circuit, is shown in Figure 4.27(a). The
drain is at signal ground, and the small-signal resistance ro of the transistor is in
RSi
Vin
D
G
Ri
+
RS i
gmVgs
Vgs
Vi
+
–
R1⎪⎪ R2
–
S
ro
Vo
Vi
+
–
Vin
+ Vgs –
R1⎪⎪R2
gmVgs
RS
(a)
Vo
(b)
Figure 4.27 (a) Small-signal equivalent circuit of NMOS source follower and (b) small-signal
equivalent circuit of NMOS source follower with all signal grounds at a common point
RS
ro
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parallel with the dependent current source. Figure 4.27(b) is the same equivalent
circuit, but with all signal grounds at a common point. We are again neglecting the
body effect. The output voltage is
Vo = (gm Vgs )(R S ro )
(4.30)
Writing a KVL equation from input to output results in the following:
Vin = Vgs + Vo = Vgs + gm Vgs (R S ro )
Therefore, the gate-to-source voltage is
1
Vin
gm
Vgs =
= 1
· Vin
1 + gm (R S ro )
+ (R S ro )
gm
(4.31(a))
(4.31(b))
Equation (4.31(b)) is written in the form of a voltage-divider equation, in which
the gate-to-source of the NMOS device looks like a resistance with a value of 1/gm .
More accurately, the effective resistance looking into the source terminal (ignoring
ro) is 1/gm. The voltage Vin is related to the source input voltage Vi by
Ri
· Vi
Vin =
(4.32)
Ri + R Si
where Ri = R1 R2 is the input resistance to the amplifier.
Substituting Equations (4.31(b)) and (4.32) into (4.30), we have the small-signal
voltage gain:
Vo
gm (R S ro )
Ri
·
Av =
=
(4.33(a))
Vi
1 + gm (R S ro )
Ri + R Si
or
Av =
R S ro
·
1
+ R S ro
gm
Ri
Ri + R Si
(4.33(b))
which again is written in the form of a voltage-divider equation. An inspection of Equation 4.33(b) shows that the magnitude of the voltage gain is always less than unity.
EXAMPLE 4.7
Objective: Calculate the small-signal voltage gain of the source-follower circuit in
Figure 4.26.
Assume the circuit parameters are VD D = 12 V, R1 = 162 k, R2 = 463 k,
and R S = 0.75 k, and the transistor parameters are VT N = 1.5 V, K n = 4 mA/V2,
and λ = 0.01 V−1 . Also assume R Si = 4 k.
Solution: The dc analysis results are I D Q = 7.97 mA and VG S Q = 2.91 V. The
small-signal transconductance is therefore
gm = 2K n (VG S Q − VT N ) = 2(4)(2.91 − 1.5) = 11.3 mA/V
and the small-signal transistor resistance is
ro ∼
= [λI D Q ]−1 = [(0.01)(7.97)]−1 = 12.5 k
The amplifier input resistance is
Ri = R1 R2 = 162 463 = 120 k