Design and analysis of quasi-lumped notch filter
with extended upper pass band
Mohit, Satish Jhariya, Shailendra Singh,✉
and Raghvendra Kumar Rai
Product Development and Innovation Center, Bharat Electronics Limited, Bangalore, India
✉ E-mail: shailendra@bel.co.in
This paper presents a hybrid technique-based compact narrow band stop
filter design for broadband applications. It provides the rejection of 60
dB at the centre frequency of 2.25 GHz without any stop band harmonics over the required frequency band of operation, i.e. 1–10 GHz.
This filter is highly appropriated for the rejection of undesired bands for
RADAR, electronic warfare, and military communications.
Introduction: Band stop filters (BSFs) are used in wide range of microwave applications to reject undesired frequencies and pass other frequencies, which are generated from active and passive devices of the
circuit. BSF realized by lumped elements will not repeat its response
over the frequency, but it is very difficult to get the high-frequency
lumped inductor required for the realization of BSF. BSF realized by
distributed circuit has no restriction in terms of availability of components but will repeat its response on all harmonic (λ/2 resonator)
or odd harmonic (λ/4 resonator) frequencies. This behaviour is not
suitable for broadband applications because it is required to have no
such harmonic resonances over the band other than the undesired frequency band. These drawbacks of conventional topologies as mentioned
in the literature makes these kind of BSFs unsuitable for broadband
applications.
Multiple designs and methods are proposed in [1] to design BSF but
they will repeat the response on harmonic frequencies. Different methods are proposed in the literature to enhance the stopband bandwidth.
Double spur line and double stub loaded BSF [2], increases the stopband bandwidth by using spurs. Multiple shunt connected open stubs of
different lengths [3], and anti-coupled resonator with quarter wave transformers are used to achieve wider and deeper rejection [4]. In [5] open
stub technique is proposed to suppress second harmonics of band pass
and band stop filters. A compact, narrow band, and highly selective band
stop filter is realized by using half wavelength stepped impedance hairpin resonators [6]. Stepped impedance resonators will shift the second
stop band to the higher side but higher-order harmonics are not suppressed.
In literature many harmonic rejection methods have been proposed
for bandpass filters, i.e. by using EBG [7], split ring resonator [8], and
integration of LPF [9] but very few items are available corresponding to band stop filter. Above techniques mentioned in the literature
for the suppression of bandpass filter harmonics are not sufficient for
present requirements, because they provide a large bandwidth of rejection. As per current requirement, BSF should reject unwanted 860
MHz (2.25 GHz ± 430 MHz) 3 dB frequency in the band of 1–10
GHz [10]. A distributed element-based “synthesis by optimization technique” is proposed to achieve band stop filters characteristics with extended upper passbands [11]. It is suitable to meet the electrical requirements but design complexity and size are not suitable for the required
application.
The main objective of this paper is to propose a technique for designing a compact narrow-band stop filter having at least 4th order harmonic
suppression with wideband impedance matching.
“Design and circuit behavioural modelling” section of this paper discusses the filter design & circuit behavioural modelling. “Simulation and
result analysis” section discusses the simulation of the proposed filter.
The simulation method and the results are discussed with important realization constraints for achieving the measured results in good agreement
with the simulated results. “Fabrication and measurement” section discusses practical result achievement and analysis. Conclusions and results
are summarized in “Conclusion” section.
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Fig. 1 Basic BSF topology
Design and circuit behavioural modelling: Figure 1 shows the basic
topology of BSF. The calculated values of reactive elements, i.e. L &
C, for the realization of 3rd order Chebyshev’s [1] BSF having 38.22
%, 3 dB fractional bandwidth (FBW) at 2.25 GHz are C1 = 1.4 pF, L1
= 3.574nH, C2 = 0.5 pF, and L2 = 10 nH. These values of inductors
and capacitors can be realized either by lumped components, distributed
components or a combination of both. Since wideband lumped ceramics
capacitors (e.g. ATC600L) are, readily available but wideband lumped
inductors are not available as per the design requirement, we cannot realize this filter using lumped elements. This filter can be realized by distributed elements but its response will repeat at harmonic frequencies.
So, to solve this problem, a quasi-lumped hybrid realization topology is
proposed. In this topology, used capacitors are lumped, while inductors
are realized by two different techniques, i.e. planar (microstrip line) and
non-planar (coil based).
Selection of the planar and non-planar realization of the inductors
is done based on the value of inductance. For high value of inductance,
coil inductor provides stable realization and tuning flexibility. Low value
coil-based realization would be highly sensitive to the wire length thus
it is realized using planar technology.
In the proposed design, a high impedance (110 ) microstrip line is
considered for realizing series connected parallel resonant circuit inductance L1. The line impedance is limited to 110 due to fabrication limitations. The length required for the realization of required inductance
[12] at resonance frequency ( f0 ) is calculated by Equation (1)
l=
l=
λH × sin−1
102.26 × sin−1
2π
ω×L
ZH
2π×2.25×109 ×3.574×10−9
110
(1)
2π
l = 7.77mm,
where L is required value of inductance, ω is angular frequency, λH and
ZH are the wavelength and characteristics impedance associated with the
high impedance micro strip line respectively. Inductor L2 (10 nH) cannot
be realized using 110 microstrip line. It is realized by wrapping 6
turns (N) of 70 µm diameter wire on foam having diameter (D) of 0.29
mm. Number of turns are required for realizing the required amount of
inductance is calculated by using ADS2022.
Circuit behavioural modelling: This band stop filter circuit shown in
Figure 1 can be divided in two types of resonant circuits, i.e. serie s and
parallel. Behaviour of these resonant circuits around its fundamental resonance frequency will give clarity on its functionality. If the observation
frequency ( f ) is less than the resonance frequency ( f0 ), series connected
parallel LC resonance circuit will act as an inductor and shunt connected
series LC resonance circuit will act as a capacitor. Thus, it will behave
as a low pass filter as shown in Figure 2a. In contrast to this, if the observation frequency ( f ) is greater than the resonance frequency ( f0 ),
series connected parallel LC resonance circuit will act as a capacitor
and shunt connected series LC resonance circuit will act as an inductor.
Thus, BSF circuit of Figure 1 will behave as a high pass filter as shown in
Figure 2b.
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Fig. 3 1st order BSF simulated and fabricated layout. (Dimensions (mm) in
layout are: W1 = 0.15, W2 = W3 = 0.76, L1 = 3.71, L2 = 2, L3 = 0.4, L4
= 1.05)
Simulation and result analysis: The proposed 3-pole filter is designed
and simulated in three steps-1) single pole series connected parallel resonant circuit, 2) single pole shunt connected series resonant circuit, and
3) three pole proposed design.
In the first stage, initially single section of parallel LC resonance circuit is simulated separately. The single section parallel resonant circuit
was realized by the inductor on one side and the capacitor in line with
the 50 ohm line. This realization provides matching below the resonant
frequency but large mismatch above it. To counter this problem required
inductor length is split into two parts, i.e. upper and lower. Length of the
lower part of inductive line is approximately half of the required inductive length for distributed inductor and upper line is provided with proper
soldering pads for adding the capacitor. Extra gap capacitance will also
be added in parallel with lumped capacitor resulting in an increased Qfactor. This increase in Q-factor will decrease the BW of the resonator.
Q-factor of this resonator can be decreased by reducing the lumped capacitor value and increasing the inductive line length for maintaining
the constant resonance frequency. Ratio of upper (2 × L4) to lower line
lengths (L3) which are used in realizing the inductor will also affect the
Q-factor of the resonator. Layout of the structure is shown in Figure 3b.
Simulated circuit and EM response of the first stage single section series connected parallel resonance circuit are shown in Figure 4a. EM
simulated 3, 10, and 15 dB rejection BWs are mentioned in Table 1.
Rejection of 30.61 dB is achieved at 2.25 GHz.
Similarly, in second stage, single section of shunt connected series
LC resonance circuit is simulated separately and response is shown in
Figure 4b. Here, used capacitor C2 is lumped while inductor L2 is wire
wound air core inductor. EM simulated 3, 10, and 15 dB rejection BWs
are mentioned in Table 1. Rejection of 21.65 dB is achieved at the resonant frequency of 2.24 GHz.
In final stage, three (N = 3) pole band stop filter is realized by cascading the series connected parallel resonant circuit and shunt connected
series resonant circuit. In shunt connected series resonant circuit, L2 is
tuneable. The required amount of tunability in terms of design frequency
and bandwidth can be achieved by using L2.
This 3rd order BSF schematic and layout are shown in Figures 5a
and 5b respectively. Simulated response of proposed 3rd order filter is
shown in Figure 6. EM simulated 3, 10, and 15 dB rejection BWs are
mentioned in Table 1. Rejection of 73.20 dB is achieved at 2.27 GHz. It
2
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Fig. 4 ADS simulated EM and circuit response of single pole circuits (a) series connected parallel resonant circuit, (b) shunt connected series resonant
circuit
Table 1. Rejection BW of 1st order and 3rd order BRF’s
a
b
c
BW
Sim. (%FBW Meas. (%FBW Sim. (%FBW Sim. (%FBW Meas. (%FBW
@ f0 )
@ f0 )
@ f0 )
@ f0 )
(MHz)
@ f0 )
3 dB
10 dB
370
407
425
620
610
(16.44)
(17.54)
(18.97)
(27.31)
(27.31)
124
120
138
440
420
(5.55)
(5.17)
(6.16)
(19.35)
(18.42)
15 dB
68
60
69
360
350
(3.02)
(2.59)
(3.08)
(15.86)
(15.35)
* a - 1st order series connected parallel resonator; b - 1st order shunt connected series
resonator; c – 3rd order proposed filter.
is observed that 3rd order filter BW is more as compared to combined
BW of individual resonators. This increase in BW is due to cascading
of multiple band reject resonant circuits. Cascading will increase the
rejection bandwidth of band reject filter where as it will decrease the
pass band bandwidth in case of band pass filters.
Fabrication and measurement: The proposed filter is designed on a 10
mil (0.252 mm) thick Rogers soft substrate (RT Duriod- 5880) with a
design dielectric constant(Dk) of 2.2, electric loss tangent of 0.0009 and
copper cladding of 1 ounce. This particular material is selected for the
design due to its strict tolerance (<±0.02) in terms of Dk and electric
loss tangent (<0.0005) over a broad frequency range. The minimized
variation in material ensures that filter response from unit to unit would
be same for a large production quantity.
Thickness of substrate used for the fabrication is very less and it needs
mechanical support. This PCB can be attached to metallic housing by
direct substrate attachment process [13] or by using a separate carrier
plate, which will integrate with metallic housing. If the attachment of
PCB and carrier plate is not proper, resulting air voids will form be-
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Fig. 2 Behaviour of BSF circuit for (a) f < f0 , and (b) f > f0
Table 2. Performance comparison of proposed filter with other
works
Fig. 5 3rd order BSF (a) ADS schematic, (b) ADS layout, and (c) prototype.
Schematic dimensions (mm) are: Wpd = 0.15, r = 0.5, ll = 1.8, Lc = 0.4,
Wc = 0.76, C1 = 1.4 pF, Wl = 0.15, C2 = 0.5 pF
Reference
Size (λ0 × λ0 ) HS Order UPB/ f0 TAF OF
LR @ f0
[2]
0.224 x 0.194
N.A.
N.A.
No
4
50 dB @ 3.50 GHz
[3]
0.384 x 0.614
2nd
∼ 3.0
No
5
20 dB @ 11.50 GHz
[5]
0.136 x 0.179
3rd
N.A.
No
3
50 dB @ 3 GHz
th
∼ 5.5
No
5
58 dB @ 1 GHz
th
∼ 4.4
Yes
3
58.46 dB @ 2.28 GHz
[11]
N.A.
Proposed Work 0.076 x 0.030
5
4
* f0 - band reject filter design frequency; HS - harmonic suppression; UPB – upper
passband BW, i.e. passband after band stop characteristics; λ0 – free space wavelength
in mm at f0 ; TAF - tunability after fabrication; OF - order of the filter; LR - level
of rejection
Fig. 6 Simulated response of filter structure (N = 3) in ADS
tween them. A metallic base plate or an electroless nickel immersion
gold (ENIG) plated Kovar material of 0.2 mm thickness can be used as
a carrier plate. This carrier plate attached filter PCB with the mounted
capacitors and inductors is placed in a metallic housing. The specific
material and plating is required for the proper attachment of PCB to
avoid air voids, which may result in discontinuity of ground, leading to
deviation from predicted response.
The recommended air gap between the PCB and the top cover of the
housing is at least 10 times the thickness of the substrate, to avoid the
effect of top metal loading on micro strip line. This enclosing can also
act as a cavity. Therefore, it should also be ensured that the cavity should
not have the oscillations within the band of interest. If any oscillations
are present, then it can be shifted out of band by protruding the cavity.
Fabricated models of 1st order and 3rd order BSF filter are shown in
Figures 3b and 5c respectively. De-embedded dimension of the structure
is 10 × 4 mm2 .
Measured results of the first-order series connected parallel resonant
filter and third-order proposed filter are shown in Figure 7. Measured
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3, 10, and 15 dB rejection bandwidth of 1st order and 3rd order proposed structure is given in Table 1. Rejection of 20.88 and 58.36 dB is
achieved at 2.32 and 2.28 GHz respectively. Return loss over the specified band 1–10 GHz (except near rejection band) is better than 10 dB,
and the insertion loss for the passband is ≤1 dB. The measured results
are matching with the EM results of the design. Comparative study of
the proposed filter with the prior arts is as shown in the Table 2. It can
be observed from the table that the proposed structure is compact and
has a similar or superior rejection in a lower order design than the previous one.
Conclusion: A novel quasi lumped hybrid technique is proposed to
achieve the objective of a compact narrow-band stop filter having at
least 4th order harmonic suppression with wideband impedance matching. Proposed technique is very much suitable for the industrial mass
production requirement. Realization method for inductors provides the
flexibility of tuning which is helpful in the frequency adjustment. It suppresses the in band harmonic responses at least up to 4th order with <1
dB insertion loss without deteriorating the performance of the system.
The EM simulated 3 dB %FBW of the filter is in good agreement with
measured result which is 27.31% while computed FBW from classical
equations is 38.22%. In case of shunt connected series resonant circuit
very small reduction in Q-factor is observed. For achieving the desired
rejection bandwidth shunt-connected series resonance circuit inductor is
made tunable.
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Fig. 7 Measured response of (a) single section (N = 1), and (b) proposed
BSF
Author contributions: Mohit: Investigation; validation. Satish Jhariya:
Conceptualization; writing—original draft. Shailendra Singh: Conceptualization; methodology; validation; writing—original draft. Raghvendra Rai: Project administration; validation.
Conflict of interest statement: The authors declare no conflicts of interest.
Data availability statement: The data that support the findings of
this study are openly available in [repository name e.g. “figshare”] at
http://doi.org/[doi], reference number [reference number].
© 2024 Bharat Electronics Limited. Electronics Letters published by
John Wiley & Sons Ltd on behalf of The Institution of Engineering and
Technology.
This is an open access article under the terms of the Creative Commons
Attribution-NonCommercial-NoDerivs License, which permits use and
distribution in any medium, provided the original work is properly cited,
the use is non-commercial and no modifications or adaptations are made.
Received: 11 September 2023 Accepted: 21 December 2023
doi: 10.1049/ell2.13075
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By changing the resonance frequency of shunt connected series resonance circuit, stopband bandwidth and rejection of the filter can be controlled. By fixing the series, connected parallel resonant circuit response
and varying the shunt connect series inductor and capacitor components
approximate 5% shift in resonance frequency can be achieved. This shift
will also change the rejection BW and sharpness of the rejection. The
same technique can be used for the realization of narrow band pass filters with extended band reject requirements.