IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 3, MAY 1998
501
A Family of Continuous-Conduction-Mode
Power-Factor-Correction Controllers
Based on the General Pulse-Width Modulator
Zheren Lai, Student Member, IEEE, and Keyue Ma Smedley, Senior Member, IEEE
Abstract— This paper presents a family of constant-switching-frequency pulse-width-modulated controllers for singlephase power-factor-correction (PFC) circuits that operate at
continuous-conduction mode (CCM). Both trailing- and leadingedge pulse-width modulation (PWM) are used. These controllers
do not require the multiplier and rectified-line-voltage sensor,
which are needed by traditional control methods, and they
can be implemented with a unified control circuit. Controller
examples are analyzed and verified experimentally.
Index Terms— PFC, power factor correction, PWM, rectifier,
switching converter.
I. INTRODUCTION
A
SINGLE-phase diode bridge followed by a dc–dc converter with proper control forms a rectifier with active
power-factor correction (PFC). If the controller forces the
input current
to have the same shape as the input voltage
so that the input impedance appears to be resistive, that
rectifier is called a resistor emulator. The resistor emulator
not only requires a near-unity power factor, but also low
harmonic contents in the line current. There are two traditional
approaches to control a resistor emulator, namely, the voltagefollower approach and the multiplier approach [1].
The voltage-follower approach realizes a resistor emulator
with the constant-duty-ratio or the constant-ON-time control,
such as a flyback [2] or a Cuk [3] converter operating at
discontinuous-conduction mode (DCM), or a boost converter
at the boundary of DCM and continuous-conduction mode
(CCM) [4]. The control circuit is simply a voltage-mode pulsewidth-modulation (PWM) chip which does not even require a
current sensor. However, the DCM or the boundary operation
causes a large current stress on semiconductors and demands
more effort to attenuate the current ripple so as to have a
satisfactory low electromagnetic interference (EMI) to the line.
The input-current ripple of the DCM Cuk converter introduced
in [3] can be eliminated by inductor coupling, however,
the current stress on semiconductors remains the same. At
high-power applications, the current stress and current ripple
become too large for a single DCM converter to operate
efficiently. Therefore, the voltage-follower approach is not
suitable for high-power application.
Manuscript received December 16, 1996; revised August 18, 1997. Recommended by Associate Editor, T. Sloane.
The authors are with the Department of Electrical and Computer Engineering, University of California, Irvine, CA 92697 USA.
Publisher Item Identifier S 0885-8993(98)03426-7.
The multiplier approach requires relatively complicated
control circuitry. As shown in Fig. 1(a), this approach needs a
multiplier, current sensor, and sensor of the input voltage
The control method is based on the current mode control. The
current reference is the rectified line voltage with its amplitude
, the output of the
modulated by the modulation voltage
feedback compensator. In contrast to the voltage-follower approach, resistor emulators with the multiplier approach operate
in CCM so they are suitable for high-power applications.
Examples of the multiplier approach include the averagecurrent mode control [5] and the peak-current mode control
[6]. The peak-current mode control usually has larger current
distortion due to the current ripple, however, its current sensor
can be implemented with a current transformer (CT). The CT
has fewer losses, thus, it is more desirable in high-power
applications. Additional current distortion may result due to
the nonlinearity of the multiplier.
A number of papers have been dedicated to simplifying
the control of the CCM converters [7]–[11]. These papers
eliminated the multiplier and the input-voltage sensor in the
multiplier approach. Some methods fulfilled that purpose under
the penalty of higher current distortion [7]. The nonlinearcarrier (NLC) control, proposed in [8] and [9] for the boost
converter and other topologies, initiated a new approach for
the resistor emulator. This new approach utilizes the property
of the quasi-steady-state operation of the CCM converters to
simplify the control circuitry, hence, it may be called the quasisteady-state approach. The linear peak-current mode (LPCM)
control proposed in [10] is another example of this new
approach. It was found that the LPCM control was a subset
of the family of controllers proposed independently in [11],
shortly after [10] was published, as application examples of a
general PWM modulator [11]. The block diagram of this new
approach, shown in Fig. 1(b), demonstrates that these control
methods have the similar complexity as the traditional current
mode control for dc–dc converters. Advantages of the current
mode control, such as the cycle-by-cycle current limitation, are
preserved. This paper discusses the common property of the
quasi-steady-state approach and further analyzes and verifies
the family of controllers proposed in [11].
The common property of this quasi-steady-state approach is
discussed first in Section II. Then, the general PWM modulator
is reviewed in Section III. The generality of this modulator can
be demonstrated with the unified quasi-steady-state approach
for resistor emulators presented in Section IV. Section V takes
0885–8993/98$10.00  1998 IEEE
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 3, MAY 1998
(a)
(b)
Fig. 1. Block diagram of PFC circuits at CCM operation (a) with the multiplier-approach controller and (b) with the proposed controllers.
the simplest controller among this family as an example and
analyzes the stability of the control method and the linecurrent distortion when a small-ripple assumption is not valid.
The same treatment is applicable to other converters and
controllers. Section VI verifies this family of controllers experimentally with four control methods for resistor emulators
with various dc–dc converters. Finally, conclusions are drawn
in Section VII.
Notation conventions are as follows unless indicated explicitly. Capital letters are used for quantities associated with the
steady state of the dc–dc converter, lowercase letters represent
time-variant variables, and a quantity in a pair of angular
brackets is the local average of the quantity, i.e., the average
in each switching cycle.
conversion ratio of the dc–dc converter, and
is the modulation voltage, as shown in Fig. 1(b), for controlling the
amplitude of the line current. When the CCM dc–dc converter
is stable, the steady-state duty ratio
satisfies
(3)
is the output voltage and
’s for five topologies
where
are shown in Table I. Capital characters are used here to
indicate the switching-frequency steady state. Substituting (3)
into (2) yields the quasi-steady-state approximation
(4)
is a constant over a line cycle if the output capacitance is
large enough, therefore, if
is also a constant in that line
cycle,
is proportional to
and the emulated resistance
(5)
II. QUASI-STEADY-STATE APPROACH
A dc–dc converter will reach a steady state if its inputs and
load remain unchanged for a long time and the control law
governing the converter is stable. The line frequency in Fig. 1
is usually well below the switching frequency, hence, the input
voltage of the dc–dc converters can be approximated as a
constant in a few consecutive switching periods. The converter
tries to converge to its steady state if the control law is stable.
Even though the converter can never reach the true switchingfrequency steady state, it operates in the neighborhood, that is,
in the quasi-steady state. The important property of the quasisteady-state operation is that all quantities can be approximated
with their steady-state values.
The control objective of the resistor emulator is to force the
input current of the dc–dc converter to be proportional to the
input voltage so that the input impedance is resistive. In other
words, the local average of the input current
(1)
is the emulated resistance.
can be controlled by
where
modulating the duty ratio
This objective can be accomplished, in general, with the
control law
(2)
is the
due to the quasi-steady-state operation, where
equivalent current-sensing resistance,
is the voltage-
can regulate
so as to control the input current.
Voltage
Implementation of (2) relies on the PWM modulator. Various implementations have been shown in [8]–[10]. These
implementations as well as some new ones can be unified
with the general PWM modulator proposed in [11].
III. THE GENERAL PWM MODULATOR
The general PWM modulator is shown in Fig. 2(a). It
contains a constant-frequency clock generator, a flip flop (FF),
a comparator (CMP), and two stages of an integrator with
reset. The number in the circle represents a constant gain of
two.
and are control inputs to the modulator, and the
output can be taken either from or of FF. The integrator
performs normal integration unless the control input
is at
logical high state, which resets the integrator output to zero as
result. The time constant of the integrator is selected to equal
the switching period
The operation waveforms of the general modulator are
shown in Fig. 2(b). Each switching period is initiated by the
constant-frequency clock. The output of FF rises to logical
high state as a clock-pulse arrives and each integrator starts
integrating its input signal. When the voltage
at the
noninverting input of CMP reaches the voltage at the inverting
input, CMP outputs a logical high signal, resetting FF and
both integrators.
LAI AND SMEDLEY: FAMILY OF CONTROLLERS BASED ON PULSE-WIDTH MODULATOR
(a)
503
(b)
Fig. 2. The two-stage general PWM modulator and its operation waveforms.
Assuming that
leads to
and
are constant in a switching period
(6)
for time
from the beginning of a cycle to the moment when
Thus, the duty ratio of
satisfies
(7)
The modulation of ’s pulse width is called trailing-edge
modulation because the leading edge of
pulse, as shown
in Fig. 2(b), is synchronized by the clock, and the trailing
edge is modulated according to values of
and
On
the contrary, ’s pulse width is leading-edge modulated, and
its pulse duty ratio satisfies
(8)
In a resistor emulator,
and
can be approximated
as constants in a switching cycle, thus, (6)–(8) are valid. In
and
the next section, we will show how to derive
from (2) for various power topologies. The general modulator
provides the flexibility of choosing leading- or trailing-edge
modulation. For some applications, using leading-edge modulation may have more advantages over using trailing-edge
modulation.
IV. UNIFIED IMPLEMENTATION OF
QUASI-STEADY-STATE APPROACH
A family of resistor-emulator controllers described in the
format of (7) or (8) can be derived from (2) for various
topologies. In this section, the boost, buck–boost, and Cuk
topologies are used as examples to demonstrate the derivation
and the application of the general modulator.
A. The Boost Converter Topology
The power stage of the boost topology is given in Fig. 3,
where
represents the transfer function of the output
feedback compensator and
is the output of the current
sensor. In this topology
(9)
Substituting the above equation and
yields
in Table I to (2)
(10)
Fig. 3. The boost resistor emulator block diagram.
The trailing-edge modulation requires the control law to be
in the format of (7). Reorganizing (10) gives
(11)
and
Comparing to (7) yields
(11) is a control law of average-current mode control
because the sensed current is an averaged quantity, which is
the same as that in [5].
When the current ripple in the inductor is negligible, the
is approximately equal to the
average inductor current
instant inductor current
Thus, one can replace the averagecurrent sensor with a simpler instant-current sensor. In this
paper, we assume that the ripple is negligible so that the
average and instant-inductor-current controllers are equivalent,
called the inductor-current controller. The inductor-current
controller for the trailing-edge-modulated boost topology is
shown in Fig. 4(a). When this assumption is not valid, the
instant-inductor-current control will result in more line-current
distortion. The distortion due to the current ripple is analyzed
for one example shortly.
Inductor-current sensing is usually accomplished with a
resistor in series with the inductor, as shown in Fig. 4(a).
It is more efficient if a switch-current CT is used. For the
boost converter, it is a common practice to replace the inductor
current with the switch current in the control circuit because
they have the equal value when the switch is on. That results in
the instant-switch-current controller, shown in Fig. 4(b). The
instant-switch-current control is the same method as the LPCM
control [10]. The switch-current control has one disadvantage,
that is, the sensed current signal is subject to switching noise.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 3, MAY 1998
Equation (15) is the fundamental control law for the NLC
control. Fig. 4(c) shows the average-switch-current controlled
boost resistor emulator.
Up to this point, the three control circuits derived (four
if assume the average and instant inductor-current controls
are two different ones) are trailing-edge modulated. One can
also find the leading-edge modulation version of these three
controllers by converting each control law to the format of (8).
Notice (10) is already in this format. Comparing (10) to
(8) yields
and
equal to
and zero,
respectively. Assuming negligible current ripple, (10) is the
inductor-current control law. Fig. 4(d) illustrates the leadingedge-modulated inductor-current controller. The modulation
output is from
For the leading-edge modulation, the modulation decision is made during the off period of the active
switch, as shown in Fig. 8. During that period, the diode
current is equal to the inductor current, hence, one can use a
CT to sense the diode current instead of the inductor current.
The resulting instant-switch-current-controlled boost converter
is shown in Fig. 4(e). For noise-immunity reason, one can also
with
use the average-switch-current sensor. Replacing
TABLE I
INPUTS TO THE GENERAL PWM MODULATOR
FOR PROPOSED RESISTOR-EMULATOR CONTROLLERS
(16)
results in
(17)
Usually, one has to use a low-pass filter after the sensor
to avoid control error caused by the noise. Employing the
average-switch-current sensor, as used in the NLC control [8],
enhances the noise immunity.
The average-switch-current sensor, as shown in Fig. 4(c),
is equivalent to the switch-current sensor cascading with an
integrator with reset. Its output
(12)
where
is the sensing capacitance and is the turns ratio
of the CT (primary: secondary
). The equivalent
sensing resistance
(13)
The average-switch-current control law can be found by
integrating each term in (11) one more time. In fact, the
average switch current
relates the inductor current by
(14)
for the boost converter. Substituting
(11) leads to
from (14) into
in
(15)
The average-switch-current controller is shown in Fig. 4(f).
Another three controllers are derived for the boost converter that are leading-edge modulated. These controllers are
listed in Table I for application convenience. Inductor-current
controllers and switch-current controllers are listed separated
because they are not always equivalent, as shown later in the
Cuk converter. Notice that some controllers with
,
therefore, only one stage of integrator-with-reset is actually
necessary. Thus, the modulator can be further simplified for
these controllers.
The leading-edge modulation for the boost converter has
some advantages compared to the trailing-edge modulation.
1) The current-sensing circuitry for some boost-derived
topologies with multiple switches is simplified. Fig. 5
shows some of these topologies employed in literature
[14]–[16]. For trailing-edge modulation, at least two
switch-current CT’s are required due to multiple active
switches in the topology. For leading-edge modulation,
there is a common path for the two or more diode
currents, thus, they can be sensed with one CT, as shown
in each figure.
2) The switching ripple current in the output filtering
capacitor can be reduced. The boost converter is usually
used as a preregulator, and the postregulator is normally
trailing-edge modulated. This results in less switching
ripple current in the capacitor as indicated in [13].
3) For high-power applications where two boost converters have to operate in parallel, one may parallel a
leading-edge-modulated converter with a trailing-edgemodulated one. This may lead to inductor-current ripple
and line-current distortion cancellation.
LAI AND SMEDLEY: FAMILY OF CONTROLLERS BASED ON PULSE-WIDTH MODULATOR
505
(a)
(b)
(c)
(d)
(e)
(f)
Fig. 4. The boost resistor emulators with various control methods based on the general PWM modulator.
B. The Buck–Boost Topology
For the buck–boost topology, the derivation starts from the
general control law in (2) as well. As shown in Fig. 6
(18)
Replacing
in (2) with
and substituting
from
Table I directly leads to the average-switch-current control
law for trailing-edge modulation
(19)
For the buck–boost converter, (14) is still valid, hence, one
can find the inductor-current control law
(20)
When the switch is on,
, hence, replacing
yields the switch-current controller.
with
Leading-edge-modulated controllers can be derived with the
same procedures as for the boost converter. The derivation
results are listed in Table I.
C. The Cuk Topology
The Cuk as well as the Sepic and Zeta topologies are
slightly different from the previous two because they have
two inductors. We use the Cuk converter as an example to
demonstrate the derivation of the controllers.
As shown in Fig. 7, the input current is the same as the input
inductor current, thus, by substituting
, one can find that
(21)
which is the inductor-current control law. Attention needs to
be paid to the value of the switch current. Unlike the previous
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 3, MAY 1998
Fig. 7. The Cuk resistor emulator block diagram.
Fig. 8. Operation waveforms for the instant-switch-current-controlled boost
resistor emulator with leading-edge modulation.
Leading-edge-modulated controllers were also found and listed
in Table I.
The same procedures can be applied to other topologies
to find out their resistor-emulator controllers. For the Sepic
and Zeta converters, the derived results are the same as the
Cuk converter as given in Table I. Notice that at least one
controller for each topology in Table I can be implemented
with single-integrator modulators because
Fig. 5. Boost-derived PFC topologies employed in [14]–[16].
V. STABILITY
AND
DISTORTION ANALYSIS
In order for the resistor emulator to operate successfully, the
control law needs to be stable. Furthermore, the line current
will have distortion when instant-current control is employed
due to the switching-frequency inductor-current ripple. In this
section, we will take the switch-current-controlled boost topology with leading-edge modulation as an example to analyze
the stability and distortion. Analysis needs only to carry out for
dc steady state. Actual quantities can be approximated with the
steady-state results because of the quasi-steady-state operation.
Fig. 6. The buck–boost resistor emulator block diagram.
two topologies, the steady-state switch current
(22)
and the switch is on. Thus, one can find the switch-current
control law to be
(23)
The average-switch-current control law can be found by reThe control law happens to be the
placing
same as (19). These derivation results are listed in Table I.
A. Stability Analysis
Mapping theory [12] is used for this analysis. Assume the
the peak inductor current for
input voltage is a dc quantity
and the inductor current changes linearly
the th cycle is
with time, as shown in Fig. 8. The moment when FF is reset
and satisfies
(24)
LAI AND SMEDLEY: FAMILY OF CONTROLLERS BASED ON PULSE-WIDTH MODULATOR
Replacing
with
, where
507
, leads to
(25)
where
hand
is the switching frequency. On the other
(26)
hence
(27)
where
Fig. 9. Calculated line current at various power levels.
(28)
and
determines the stability of the current loop [12]. When
(29)
the circuit will converge to its steady state. Any perturbation
will disappear gradually. If the inductor current, as shown in
Fig. 8, is higher than the steady-state value, it takes longer
to reach
for a given value of
, thus, the
for
inductor has longer time to discharge so as to reduce its
reaches
value. If the inductor current is too small,
sooner, resulting a larger duty ratio, hence, the inductor current
increases. Subswitching-frequency harmonic oscillation occurs
when (29) is not satisfied.
The steady state can be found out by letting (26) equal
zero, therefore
Fig. 10. Theoretical THD versus power levels.
(30)
Combining (5), (28)–(30) gives another form of the stability
condition, that is,
(31)
In PFC applications, the stability is guaranteed when the line
0.5
The possible situation
voltage is at the region that
for the instability to occur is when the converter operates at the
region near the peak line voltage and simultaneously at light
load. The instability is localized and does not spread from one
line cycle to another. With certain minimal load condition, the
unstable situation can be excluded.
B. Line-Current Distortion
The line-current distortion is found by solving the steadystate expressions. Combining (25) with (30) yields the steadystate peak inductor current
(32)
The valley inductor current
is
For constant
and
the valley current is proportional
to the input voltage for the leading-edge-modulated boost
resistor emulator. In the trailing-edge-modulation version of
this method, the peak inductor current is proportional to , as
shown in [10], thus, paralleling two boost converters with these
two controllers, respectively, will result in inductor-current
ripple and line-current distortion cancellation. The average
inductor current is then found
(34)
Equation (34) can be used to find the actual average line
current. An example is given with the following circuit parameters. For
V
line voltage,
H
kHz, and
V
The normalized line current is
calculated and shown in Fig. 9 for
and
, respectively. Taking
as the full load, the line
current total harmonic distortion (THD) versus power level
is plotted in Fig. 10. The distortion improvement over the
peak-current control with multiplier approach [6] is obvious.
VI. EXPERIMENTAL VERIFICATIONS
(33)
Three converters with four different controllers were built
and tested.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 3, MAY 1998
(a)
(b)
(c)
Fig. 11. Experimental waveforms at different load power levels. Top: duty ratio (0.5/div). Middle: line voltage (110 Vrms ). Bottom: line current, 5 A/div,
2 A/div, and 0.5 A/div, respectively, for (a)–(c). (a) At full load, (b) 40% of full load, and (c) 10% of full load.
A. The Boost Resistor Emulator with Leading-Edge
Instant-Switch-Current Control
An experimental circuit has been built according to Fig. 4(e)
and tested. The line voltage is 110 V , and the output
voltage is 220 V
The switching frequency
is 100 kHz.
The inductance of the boost converter is 520 H An
EMI filter was inserted between the diode bridge and the
boost converter, similar to the one shown in Fig. 13, with
an inductance of 44 H and a capacitance of 0.68 F The
feedback compensator
has a proportional-integral (PI)
transfer function. Experimental waveforms for a full load of
350 W, 40% of full load, and 10% of the full load are
shown in Fig. 11(a)–(c), respectively. The waveforms in each
figure, from top to bottom, are the duty ratio measured with a
Tektronix time-to-voltage converter TVC501, line voltage, and
line current with its scale marked in the figure, respectively.
Notice that in Fig. 11(a) and (b) the current distortion is not
significant while in (c) the distortion is noticeable. The current
shape in Fig. 11(c) is similar to the calculated waveform shown
in Fig. 9. It is difficult to compare the measured THD with
the theoretical THD since the line voltage itself has some
distortion, however, one can still see the trend of distortion
as load decreases.
B. Flyback Resistor Emulator with Trailing-Edge
Average-Switch-Current Control
Fig. 12 shows the flyback PFC converter of trailing-edge
average-switch-current control (the feedback compensator is
not shown). As indicated in [10], this control method is
unconditionally stable.
An experimental circuit was built and tested. The major
63 kHz, the primary
circuit parameters are as follows:
Fig. 12. Flyback PFC converter with trailing-edge average-switch-current
control.
Fig. 13. Experimental waveforms of the flyback converter at 100-W power
output. Top: duty ratio (0.63/div). Middle: line voltage (110 Vrms ). Bottom:
line current (1 A/div).
inductance is 340 H, the flyback transformer turns ratio is
1:1,
H,
F, line voltage is 110 V ,
and the output voltage is regulated at 50 V The experimental
LAI AND SMEDLEY: FAMILY OF CONTROLLERS BASED ON PULSE-WIDTH MODULATOR
Fig. 14.
Cuk converter with instant-inductor-current control.
509
Fig. 16. Cuk converter with average-switch-current control.
Fig. 15. Experimental waveforms of the Cuk converter with
instant-inductor-current control at 150-W power output. Top: duty ratio
(0.5/div). Middle: line voltage (110 Vrms ). Bottom: line current (2 A/div).
Fig. 17. Experimental waveforms of the Cuk converter with average-switch-current control at 150-W power output. Top: duty ratio (0.5/div).
Middle: line voltage (110 Vrms ). Bottom: line current (2 A/div).
result is demonstrated in Fig. 13. The line-current distortion is
insignificant.
VII. CONCLUSIONS
C. Cuk Resistor Emulators
To verify the control methods for the Cuk, Sepic, and
Zeta converters, a trailing-edge instant-inductor-current and
an average-switch-current controlled Cuk PFC converter were
built and examined. The circuits are illustrated in Figs. 14 and
16, respectively, with their feedback compensators not shown.
The same power stage were used for both control methods. The
switching frequency is 100 kHz, and other major parameters
are shown in the figures. The line voltage was 110 V , and
both outputs were regulated at 50 V Figs. 15 and 17 are the
experimental waveforms at 150-W power output. From top to
bottom, the waveforms are the duty ratio, line voltage, and line
current, respectively. Notice that both line currents are closely
following the line voltage.
It was observed that for the average-switch-current control,
the line current had some ringing at around 2 kHz when the
rectified line voltage changed slope rate abruptly, i.e., when the
line voltage crosses zero or is at its peak for the distorted line
voltage. The ringing is still noticeable in Fig. 17 after resistive
damping is used to clean up the waveforms. With instantinductor-current control, the ringing is less severe for the same
power stage. In other words, different control methods may
provide different dynamic performance for the same power
stage. Further investigation of the dynamics is beyond the
scope of this paper.
A family of resistor emulator controllers for the quasisteady-state approach are presented in this paper based on the
general PWM modulator. The property and general control law
for the quasi-state-state approach are discussed. The derivation
procedures are given in detail with derivation results for five
commonly used converters listed in Table I. Both trailing- and
leading-edge modulation can be realized at constant switching
frequency. Leading-edge modulation can sometimes lead to
simpler control circuitry as demonstrated in the boost converter
example.
Both leading- and trailing-edge modulation have three basic
control circuits for one converter according to the way that
the current is measured. Physically, the average-switch-current
control is equivalent to the instant-switch-current control in the
sense that the former is the integration of the latter.
The PFC circuits are ideal resistor emulators when the
switching-frequency inductor-current ripple is zero. In practice, the line-current distortion due to the current ripple can be
analyzed for each specific control method.
For this family of controllers, the rectified-line-voltage
sensor, error amplifier in the current loop, and multiplier in
the voltage feedback loop that exist in a traditional CCM PFC
circuit are eliminated, hence, the control circuitry is simplified.
The performance of these PFC circuits is comparable to or
improved over those traditional CCM converters with the
multiplier-approach control. The most important advantage
of these controllers is that their implementation is unified.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 3, MAY 1998
One can see from the three converters and four control
methods experimentally verified in this paper that they share
an identical PWM modulator. Therefore, these controllers are
well suited for integrated-circuit implementation.
[14] P. N. Enjeti and R. Martinez, “A high performance single phase ac to dc
rectifier with input power factor correction,” in APEC’93, pp. 190–195.
[15] A. Pietiewicz and D. Tollik, “New high power single-phase power factor
corrector with soft-switching,” in Intelec’96, pp. 114–119.
[16] E. X. Yang, Y. Jiang, G. Hua, and F. C. Lee, “Isolated boost circuit for
power factor correction,” in APEC’93, pp. 196–203.
ACKNOWLEDGMENT
The authors would like to thank Dr. D. Maksimovic of
the University of Colorado at Boulder for correcting errors
in Table I in an earlier version of this paper.
REFERENCES
[1] J. Sebastian, M. Jaureguizar, and J. Uceda, “An overview of power factor
correction in single-phase off-line power supply systems,” in IECON’94,
pp. 1688–1693.
[2] R. Erickson, M. Madigan, and S. Singer, “Design of a simple highpower-factor rectifier based on the flyback converter,” in APEC’90, pp.
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Zheren Lai (S’95) received the B.S. and M.S.
degrees in electrical engineering from Zhejiang University, China, in 1989 and 1993, respectively. He
is currently working toward the Ph.D. degree at the
University of California (UCI), Irvine.
From 1989 to 1992, he worked on active power
filters. He was a Research Engineer at the Electrical
and Electronic Engineering Company, Hangzhou,
China, for a short period in 1993. Since joining
UCI in the fall of 1993, his research has been
concentrated in the control of switching converters, including switching audio power amplifiers and power-factor-corrected
rectifiers.
Keyue Ma Smedley (SM’97) received the B.S. and
M.S. degrees in electrical engineering from Zhejiang
University, China, in 1982 and 1985, respectively,
and the Master and Ph.D. degrees in electrical engineering from the California Institute of Technology,
Pasadena, in 1987 and 1991, respectively.
She was an Engineer at the Superconducting Super Collider from 1990 to 1992, where she designed
specific ac–dc conversion systems for all accelerator
rings. She joined the Faculty of Electrical and Computer Engineering at the University of California,
Irvine, in 1992, where she established a state-of-the-art power electronics
laboratory. Her research interests include modeling, control, topologies, and
integration of switching converters, inverters, class-D power amplifiers, softswitching techniques, power-factor-correction methods, laser current sources,
power conversion for alternative energy sources, etc. She currently holds the
U.S. patent for one-cycle control.
Dr. Smedley is the Chair of the Constitution and Bylaws Committee
of the IEEE Power Electronics Society, an Associate Editor of the IEEE
TRANSACTIONS ON POWER ELECTRONICS, an Associate Editor of the the Advisor
of the IEEE UCI Student Chapter, a Faculty Member of Eta Kappa Nu, and
a Member of the Power Sources Manufacturer’s Association.