IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 71, NO. 7, JULY 2023
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An Active Reconfigurable Intelligent Surface
Utilizing Phase-Reconfigurable
Reflection Amplifiers
Junhui Rao , Graduate Student Member, IEEE, Yujie Zhang , Member, IEEE,
Shiwen Tang , Graduate Student Member, IEEE, Zan Li , Student Member, IEEE,
Chi-Yuk Chiu , Senior Member, IEEE, and Ross Murch , Fellow, IEEE
Abstract— Active reconfigurable intelligent surfaces (RISs)
have recently been proposed to complement and generalize
passive RIS. In this work, the design for an active RIS based
on phase-reconfigurable reflection amplifiers is proposed. The
proposed active RIS elements’ design consists of a two-layer patch
antenna and a phase-reconfigurable reflection amplifier that is
realized by a cascade of a phase shifter and a reflection amplifier.
Theoretical and numerical analyses are used to quantify key
parameters that need to be met in the design of the reflection
amplifier and phase shifters. These show that a tradeoff between
the reflection amplifier’s gain and the phase shifter’s return and
insertion loss is required to be balanced to obtain a stable and
effective overall design. In particular, a reflection amplifier with
about 13-dB gain and a 2-bit phase shifter with only about 1.9dB insertion loss and −25-dB return loss are designed. The final
RIS element obtains 8.5-dB gain with four reconfigurable phases
ranging from 0◦ to 360◦ with small root mean square (rms)
gain and phase errors. An active RIS can be constructed by
concatenating these elements together, and an efficient analytical
method to calculate the scattered pattern by the active RIS is
also proposed. Experimental results for a 2 × 2 element active
RIS prototype are also provided. These show that the active RIS
provides received powers around 8.5 dB higher than that from
using an equivalent passive RIS. The results demonstrate the
feasibility of achieving gain in active RISs and help demonstrate
their promise as a technology for future high-capacity wireless
communication systems.
Index Terms— Active RIS, beam steering, beamforming, intelligent reflecting surface, reconfigurable intelligent surface (RIS),
sixth generation (6G).
Manuscript received 4 November 2022; revised 22 December 2022;
accepted 28 December 2022. Date of publication 24 January 2023; date of
current version 30 June 2023. This work was supported by the Hong Kong
Research Grants Council Collaborative Research Fund under Grant C601220G and in part by the General Research Fund under Grant 16207620.
(Corresponding authors: Ross Murch; Yujie Zhang.)
Junhui Rao, Yujie Zhang, Shiwen Tang, Zan Li, and Chi-Yuk Chiu are with
the Department of Electronic and Computer Engineering, The Hong Kong
University of Science and Technology, Hong Kong, China (e-mail:
jraoaa@connect.ust.hk; yzhangfy@connect.ust.hk; stangai@connect.ust.hk;
zligq@connect.ust.hk; eefrankie@ust.hk).
Ross Murch is with the Department of Electronic and Computer Engineering
and the Institute for Advanced Study (IAS), The Hong Kong University of
Science and Technology, Hong Kong, China (e-mail: eermurch@ust.hk).
Color versions of one or more figures in this article are available at
https://doi.org/10.1109/TMTT.2023.3237029.
Digital Object Identifier 10.1109/TMTT.2023.3237029
I. I NTRODUCTION
R
ECONFIGURABLE intelligent surfaces (RISs) have
been proposed and extensively developed in the past
few years [1], [2], [3], [4], [5], [6]. This is due to their
ability to configure the propagation environment smartly. RISs
are also seen as a promising technology for future sixthgeneration (6G) communication networks. Specifically, RIS
consists of a number of independent elements that can scatter
incident electromagnetic waves and impose specified phases.
By smartly configuring each element’s state, or scattering
phase, across the RIS, various scattering properties, such as
beamforming or interference nulling, can be achieved. Hence,
RIS can enhance the received signal-to-interference-plus-noise
ratio (SINR) [7] or provide physical-layer security [8], [9].
Furthermore, because of the straightforward structure of the
RIS, compared to traditional communication systems with RF
chains, RISs are promising for future energy and spectrumefficient wireless communication.
In conventional RIS, the elements are passive and configured with low-power RF switches. The noise introduced by
the passive elements can be ignored, and the power efficiency
is relatively high. By implementing RISs with large apertures
and high array gains, the performance of wireless communication systems has been shown to be enhanced significantly
in scenarios where the direct link between the transmitter
and receiver is blocked [10]. However, in another common
scenario where the direct link is present, the deployment of
RIS brings only minimal performance improvement resulting
from the “multiplicative fading” or “double fading” effect of
RIS. That is, the path loss of the transmission coefficient from
the transmitter through RIS to the receiver is the product of
the path losses due to the transmitter-RIS and RIS-receiver
links, and is thousands of times higher than that of the direct
link [11]. Therefore, passive RISs are ineffective in these
scenarios even though large-scale RISs with high array gain
are employed.
To compensate for the deficiencies of passive RIS and
also generalize the approach, the concept of active RIS has
recently been proposed [12], [13], [14], [15]. In active RISs,
the elements are not passive but are active and can scatter the
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incident signals with both amplification and tunable phases.
The feature of amplifying the incident waves strengthens the
signal scattered by active RISs and makes it comparable to
the signal from the direct path in scenarios where the “multiplicative fading” effect is a key factor. In addition, compared to
passive RISs, active RISs can have fewer elements to reach an
equivalent level of array gain or required signal-to-noise ratio
(SNR) at the receiver. This can help reduce the size of RISs
and the complexity of the channel estimation and controlling
system arising from a large number of independent elements.
Furthermore, another feature of active RIS is that, when its
amplifiers are turned off, it can become a passive RIS. Active
RIS can, therefore, be considered a generalization of passive
RISs.
In the elements of active RISs, the noise will also be
amplified due to the active devices. As a result, similar to
amplify-and-forward (AF) relays and massive multiple-input
multiple-output (MIMO), the received SNR, when active RIS
is present, scales by the order of O(N), instead of O(N 2 ) for
passive RIS, where N is the number of RIS elements. Due to
this limitation, in the situation where the direct link is blocked,
the improvements of system performance by active RIS and
passive RIS are similar [16]. However, even considering the
extra noise, in the scenario where the direct link is present,
communication aided by active RISs can achieve a substantial
improvement compared to passive RISs [12], [13].
Active RISs also share some characteristics with AF
relays [17], [18]. Both of them receive signals and then radiate
the amplified signals. However, the architectures are distinct.
Active RIS elements simultaneously receive and transmit
signals using a phase-reconfigurable reflection-type amplifier.
AF relays typically have the receiving antennas separate from
the transmitting antennas, and phase shifting of the transmit
signals is usually performed at the baseband so that a full
RF chain for each transmit antenna is required. Furthermore,
the need to reduce interference between the receiving and
transmitting antennas will also complicate the relay setup and
hardware in AF relays.
One possible method to implement active RISs is by using
an amplifying reflectarray [19], [20], [21], [22], [23], [24]
that can reflect the amplified scattered field with orthogonal
polarization to the incident waves. Two-port antennas with two
orthogonal polarizations can be utilized, and a conventional
two-port power amplifier (PA) can be connected between the
two ports. To maintain low mutual coupling between the
ports of the antenna, the polarization of the different ports is
required to be orthogonal. Therefore, one of the limitations
of this method is that it will alter the polarization of the
scattered waves. Besides, another limitation is that most of
these amplifying reflectarrays are focused on a fixed design,
lacking reconfigurable components, such as phase shifters.
Thus, they cannot achieve the controllable amplified scattering
required in active RIS.
In this article, we provide a detailed investigation into the
design methodology, fabrication, and measurement of active
RIS for use in wireless communication. It complements the
existing wireless system research on active RIS that has been
performed [12], [13]. A summary of our research contributions
and the unique features of the proposed design is given as
follows.
1) Architecture of Active Element: An active element architecture based on a phase-reconfigurable reflection amplifier is proposed. When concatenated into an RIS, the
resulting active RIS can scatter the incident waves
without altering their polarization. The architecture can
also function as a passive RIS when the amplifiers are
turned off.
2) Design Methodology of Phase-Reconfigurable Reflection
Amplifier: The phase-reconfigurable reflection amplifier
is achieved by a cascade of a phase shifter and a
reflection amplifier. However, due to the large reflection
coefficient of the reflection amplifier, the return loss and
the insertion loss of the phase shifter, and the gain of
the reflection amplifier need to be carefully designed.
Detailed theoretical and numerical analyses are provided
to specify the requirements.
3) Analytical Method to Calculate the Active RIS Scattered Pattern: Due to the limitation of electromagnetic
solvers in simulating active devices, an efficient analytical method for active RIS with a large number of
elements is proposed to find the scattered active RIS
pattern.
4) Prototype and Experimental Verification: A prototype
of the proposed active RIS is fabricated, and a testbed
was built to measure the scattered patterns to verify its
effectiveness.
This article is organized as follows. Section II describes the
active RIS architecture and design methodology of the phasereconfigurable reflection amplifier. Section III introduces the
design of the reflection amplifier, phase shifter, and scattering elements. Section IV provides analytical results for the
scattered pattern and its experimental verification by using a
prototype of the proposed active RIS. Section V compares the
proposed design with related work and provides discussions.
Finally, Section VI concludes this work.
II. D ESIGN M ETHODOLOGY
A. Active RIS Architecture
Fig. 1 shows the architecture of the proposed active RIS
formed by 4 × 4 identical and independently reconfigurable
elements. Each element consists of an antenna loaded by
a phase-reconfigurable reflection amplifier. When an external electromagnetic wave is incident on the surface of the
elements, they will receive and reradiate it back after its
magnitude is amplified and the required phase on each element
is imposed. The phases of the phase-reconfigurable reflection
amplifiers are adjusted by a controlling system, such as a
field-programmable gate array (FPGA) in our design. By configuring the elements with the desired phases, a beam with
amplification can be steered toward a variety of directions
(or any wavefront with an arbitrary phase across the RIS
element).
As shown in Fig. 2(a), the element consists of three key
components, which are the antenna element, the phase shifter,
and the reflection amplifier. A two-layer planar patch antenna
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Fig. 1.
Illustration of the proposed active RIS constructed from
4 × 4 elements. The coordinate system for the RIS is also shown and used
throughout this article. In the coordinate system shown on the active RIS,
we refer to azimuth as being the angle φ and elevation being the angle θ .
with relatively wide bandwidth and beamwidth is utilized to
collect the incident waves and transmit the amplified waves.
The phase shifter is based on a switched line structure to provide the reconfigurable phases, while the reflection amplifier
is based on a common-source topology leveraging a transistor
to amplify the incident waves. The concatenation of the phase
shifter and reflection amplifier forms a phase-reconfigurable
reflection amplifier that needs to be designed as a single unit
instead of two separate components, as shown in the system
diagram of the element in Fig. 2(b). This is because the effect
of the concatenation makes certain parameters in the design
sensitive to changes in the phase states, and more details are
provided in Section II-B.
B. Design Methodology of Phase-Reconfigurable
Reflection Amplifier
As shown in Fig. 2(b), the phase shifter is a two-port
component whose characteristics can be described by an
S-parameter matrix, while the reflection amplifier is the load
connected to port 2 of the phase shifter. Utilizing the two-port
network theory [25], the overall reflection coefficient of the
active element can be written as
S12 S21 amp
(1)
Γoverall = S11 +
1 − S22 amp
where S12 and S21 are the forward and reverse losses of the
phase shifter, and S11 and S22 are the reflection coefficients
of the phase shifter at the two ports. Γamp is the reflection
coefficient of the reflection amplifier that is connected to port
2. Because the return loss, S11 , in the phase shifter is usually
smaller than −10 dB and Γoverall should be much larger than
0 dB to reach a desirable gain, the first term in (1) can
Fig. 2. Proposed architecture of active RIS element. (a) Antenna, phase
shifter, and reflection amplifier. (b) Equivalent network of the phasereconfigurable reflection amplifier. Note that Ports 1 and 2 have also been
added as labels on the phase shifter.
be ignored. Γoverall can, therefore, be reduced to
Γoverall ≈
S12 S21 amp
.
1 − S22 amp
(2)
There are two main objectives in the design of the phasereconfigurable reflection amplifier. The first is to realize
a stable and relatively large reflection gain (magnitude of
Γoverall ) at port 1 of the phase shifter, as shown in Fig. 2(b).
Another objective is to evenly tune the reflected phases (phase
of Γoverall ) through 360◦ .
To meet the first objective, the magnitudes of the denominator and numerator in (2) should remain as constant as possible
across all phase states. The parameter S22 is sensitive to the
phase state of the phase shifter, and when combined with the
gain Γamp , the product S22 amp can be significant. Therefore,
to make certain the denominator remains constant across phase
states, it is necessary to make S22 amp close to zero so that the
denominator is nearly constant and close to unity. This needs
to be strictly set and is a key design parameter specific to the
overall design of the phase-reconfigurable reflection amplifier.
For the numerator, we can maintain gain stability by ensuring that the insertion loss is constant between states to maintain
the stability of the numerator. For convenience, we, therefore,
define
S12 = S21 = L PS eiθPS
(3)
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Fig. 3. Performance of the system when the gain of the reflection amplifier is fixed to 13 dB, and the insertion loss and return loss S22 of the phase shifter
vary: (a) rms phase error, (b) rms gain error, and (c) mean gain. The desired performance range is highlighted by the red rectangular outline.
where L PS and θPS are the insertion loss and the phase of
the phase shifter, respectively. For all states, we expect the
insertion loss L PS to be approximately constant.
Unlike conventional amplifiers with two ports, the reflection
amplifier only has one port (acts as input and output) and
amplifies the reflected signal. The reflection coefficient, Γamp ,
of the reflection amplifier can be written as
Γamp =
Z amp − Z 0
= gamp eiθamp
Z amp + Z 0
(4)
where Z 0 and Z amp are the characteristic impedance, usually
50 , and the input impedance of the reflection amplifier,
respectively, as shown in Fig. 2(a). To obtain a desirable
reflection gain, the absolute value of {Z amp + Z 0 } should be
as small as possible across the frequency band of interest.
Thus, the real part of input impedance Re{Z amp } has to be
a negative value, which can be realized by a tunnel diode
or transistor [26], [27], [28]. Γamp can also be expressed as
gamp eiθamp , where gamp and θamp are the magnitude and phase
of reflection coefficient, respectively.
When our design criteria are met, S22 amp is close to zero
so that further simplification allows Γoverall to be reduced to
Γoverall ≈ L PS gamp ei (θamp +2θps ) .
2
(5)
That is, the overall gain of the system is the gain of the
reflection amplifier minus double the insertion loss, a “double
loss” effect, and the final phase is also double that of the
phase shifter’s phase. Due to the small variation of insertion
loss over different states of the phase shifter, the final gain is
stable in this configuration. The phases can also be tuned to
desired values by appropriately designing the phase values of
2
the phase shifter. In other words, by ensuring that L PS gamp is
large enough, our two design objectives are met, and an active
RIS with good performance is obtained.
To better appreciate the element design, a numerical example of a reflection amplifier with 13-dB gain connected to
different phase shifters with various insertion loss S12 and
return loss S22 is shown in Fig. 3(a)–(c). The desired performance range of the system is highlighted by the red rectangular
outline. The root mean square (rms) phase error evaluates how
much the actual phases diverge from the expected values, and
the rms gain error quantifies the uniformity of the final gains
of different states. It can be seen that, for a reflection amplifier
with this level of gain, a phase shifter with a return loss S22
smaller than −25 dB and insertion loss smaller than 2 dB is
needed to realize about 10-dB final gain with acceptable rms
phase and gain error.
III. D ESIGN OF E LEMENT C OMPONENTS
In this section, we describe the specific designs, prototypes,
and measurements of the reflection amplifier, phase shifter, and
antenna, meeting the specifications that were found desirable
in Section II.
A. Reflection Amplifier
To realize reflection amplifier gain, Re{Z amp } is required to
be negative. To achieve this, we utilize an AsGa N-channel
transistor to form a common source circuit, as shown in
Fig. 4(a). Two radial stubs are loaded at its drain and source
so that, by tuning the dimension of the radial stubs, the circuit
can operate in the potentially unstable region and achieve
negative Re{Z amp }. A biasing dc resistor is also loaded to the
source to provide a stable dc operating point and forms a
negative feedback path. To avoid potential oscillation at high
frequencies, a capacitor Cgs is also connected between the
gate and the source. Besides, as described in Section II-B,
the gain of the reflection amplifier cannot be too large (the
product S22 amp must remain small for a stable system gain)
nor too small so that the final system gain with phase shifter
insertion loss is negligible. In our design, we set 13 dB as
the target gain of the reflection amplifier, given that S22 is
specified as −25 dB. A matching network is inserted between
the input/output port and the gate to tune the gain to this
desired value.
The reflection amplifier is implemented on a Rogers 4003C
substrate (r = 3.55 and loss tangent = 0.0027) with a
thickness of 0.508 mm. The transistor used is NE3509M04
from CEL, and all the capacitors are from Murata. It should
be noted that inductors with high Q-Factor are required in the
implementation, and the inductors used in the proposed design
are serials 0402HP from Coilcraft. The fabricated circuit is
shown in Fig. 4(b), and its specific parameters are provided
in Fig. 4(c).
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Fig. 6. Phase shifter design. (a) Schematic and (b) geometry where the
dimensions are h 1 = 0.508 mm, l1 = 4.82 mm, l2 = 9.6 mm, l3 = 7.2 mm,
l4 = 8.18 mm, l5 = 4.75 mm, l6 = 8.74 mm, and w1 = 1.1 mm. Gray:
dielectric layer. Brown: copper. (c) Photograph.
Fig. 4.
Reflection amplifier design. (a) Schematic, (b) photograph, and
(c) geometry where the dimensions are h 1 = 0.508 mm, l1 = 5.5 mm,
l2 = 5.34 mm, l3 = 6.57 mm, l4 = 3.72 mm, l5 = 16.5 mm, l6 = 8.1 mm,
w1 = 1.1 mm, w2 = 1.62 mm, w3 = 0.94 mm, w4 = 0.96 mm,
w5 = 0.2 mm, r1 = 16.55 mm, r2 = 11.27 mm, angle1 = 60.7◦ , and
angle2 = 82.7◦ . Gray: dielectric layer. Brown: copper.
Fig. 5. Performance of the fabricated reflection amplifier. (a) Gain measurement for different input power levels and the simulation result when the dc
bias voltage is 1.75 V. (b) Gain measurement for different dc bias voltage
when the input power level is −20 dBm.
A vector network analyzer, Rohde & Schwarz ZVA40, was
used to test the amplifier’s performance. As shown in Fig. 5(a),
the measured and simulated reflection gains for different input
powers are provided. Due to small manufacturing errors, there
is a slight and acceptable difference between simulated and
measured results. The circuit can operate well with stable
performance up to −30-dBm input power. When the input
power is bigger than −30 dBm, the gain starts to wane, and
the reflection amplifier cannot work when the input power is
about 0 dBm. For the application of RIS, the received signal is
usually weak due to the path loss from the transmitter to the
RIS. Thus, the linearity of the proposed reflection amplifier
is sufficient for use in our active RIS. Fig. 5(b) provides the
performance of the circuit under various dc supply voltages.
To realize the desired gain, the dc voltage should be set to at
least 1.75 V.
B. Phase Shifter
As shown in Fig. 3(b)–(d) and elaborated on in Section II-B,
for a reflection amplifier with about 13-dB gain (as in
Section III-A), a phase shifter with performance better than
−25 dB S22 return loss and −2 dB insertion loss is required.
However, most commercial phase shifters and designs reported
in past papers [29], [30], [31], [32], [33], [34], [35], [36], [37]
cannot fulfill these requirements as the insertion loss and return
loss are usually not the first design priority and not usually as
critical to the overall system performance.
Therefore, to reach the desired performance, a design for a
phase shifter based on switched transmission lines is proposed,
as shown in Fig. 6(a). There are two blocks that can achieve
90◦ and 45◦ phase delay separately. In each block, by using
two single-pole double-throw (SPDT) RF switches, the RF
signal can be controlled to pass through either the reference
path or the delay path. The delayed phase can be designed to
achieve the desired value by changing the electrical length of
the delay path while keeping the reference path unchanged.
In the schematic shown, delay path 1 is about 1/4 wavelength
longer than reference path 1 to provide 90◦ phase difference,
while delay path 2 is about 1/8 wavelength longer than
reference path 2 to provide 45◦ phase difference.
The phase shifter presented here is implemented on a Rogers
4003C substrate with a permittivity of 3.55, a loss tangent of
0.0027, and a thickness of 0.508 mm, as shown in Fig. 6(c).
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TABLE I
P HASE S HIFTER P ERFORMANCE C OMPARISON
Fig. 7.
Performance of the fabricated phase shifter. (a) Return loss
measurement for the different states. (b) Insertion loss measurement for the
different states. (c) Raw shifted phases for the different states. (d) RMS phase
error.
A total of four SPDTs from Skyworks for the RF switches
and capacitors from Murata for RF chokes are appropriately
located.
The fabricated circuits are measured by a Rohde &
Schwarz ZVA40 four-port vector network analyzer. Phase,
return loss S22 , and insertion loss are tested for all four states.
The measured return loss over the desired frequency band,
2.2–2.6 GHz, is extremely low, better than −25 dB, as shown
in Fig. 7(a). Fig. 7(b) shows that the proposed phase shifter’s
measured insertion loss or S21 parameter is about 2 dB, and
the variation in the insertion loss over the different states is
less than 0.4 dB. Fig. 7(c) shows the shifted phases for the
four states, which are 0◦ , 45◦ , 90◦ , and 135◦ , and the rms
phase error is found below 5◦ across most frequencies within
the band and reaches the lowest point, of about 0◦ at around
2.33 GHz, as shown in Fig. 7(d).
Table I compares the performance of the proposed phase
shifter with past published related work and commercial phase
shifters. As we can see, the proposed phase shifter presents
the required insertion loss and return loss for application in
active RIS.
C. Phase-Reconfigurable Reflection Amplifier
To verify the performance of the cascaded reflection
amplifier and phase shifter, we combined and fabricated
the reflection amplifier and phase shifter together to obtain
a phase-reconfigurable reflection amplifier, as shown in
Fig. 8(a) and (b). Its performance is also measured by a
calibrated Rohde & Schwarz ZVA40 four-port vector network
analyzer.
The final system can operate in two modes: an active mode
when the reflection amplifier is biased by a 1.75-V dc supply,
and a passive mode when no dc supply is applied.
In the active mode, an average of 8.5-dB reflection gain
is realized, and the maximum gain difference is about 2 dB,
as presented in Fig. 9(a). The rms gain error is smaller than
0.5 dB, as shown in Fig. 9(b). As demonstrated in Fig. 9(c), the
phases of different states can evenly cover 0◦ –360◦ . The rms
phase error is smaller than 5◦ , as shown in Fig. 9(d), which
indicates that the expected tunable phases are well achieved.
In passive mode, the signal will suffer from an average 4.5-dB
insertion loss, as shown in Fig. 10(a), mainly resulting from the
phase shifter. The phases are also well tuned into four states
with small rms phase error, as shown in Fig. 10(c) and (d).
D. Antenna
The antenna is the receiving and reradiating component
in the active RIS. When an incident wave illuminates the
element, the scattered pattern can be decomposed into two
parts: the structural mode part and the antenna mode part [26].
Specifically, the structural mode part is the scattered pattern
when the antenna is conjugate-matched and is a fixed term
that cannot be tuned by the reconfigurable component (phasereconfigurable reflection amplifier in our design). The antenna
mode part is the radiating pattern when the antenna is excited
by the reflected signal from the load and is tunable by changing
the state of the phase shifter. If we use E s (Z ) to denote
the scattered pattern of the antenna loaded by the impedance
of Z , then, for an antenna with small return loss whose input
impedance is near 50 , the scattered pattern when the antenna
is loaded by a circuit or component with impedance Z L can
be expressed as
Γoverall Is E unit
(6)
E s (Z L ) = E s Z ∗A −
2
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Fig. 10.
Performance of the fabricated phase-reconfigurable reflection
amplifier in passive mode (0-V dc bias is applied). (a) Gain measurement
for the different states. (b) RMS gain error. (c) Raw reflection phases for the
different states. (d) RMS phase error.
in Fig. 2, Z L refers to the equivalent impedance of the phasereconfigurable reflection amplifier and is related to Γoverall
in (1) by
Fig. 8.
Overall phase-reconfigurable reflection amplifier. (a) Geometry.
Gray: dielectric layer. Brown: copper. (b) Photograph.
Fig. 9. Performance of the fabricated phase-reconfigurable reflection amplifier in active mode (1.75-V dc bias is applied). (a) Gain measurement for the
different states. (b) RMS gain error. (c) Raw reflection phases for the different
states. (d) RMS phase error.
where Z ∗A is the perfect conjugate match of the antenna
impedance Z A and Is is the current induced by the incident
wave at the antenna’s port when the port is shorted. E unit is
the radiation pattern when the antenna is excited by a unit
current at the input port. In the proposed element architecture
ZL =
Z 0 (Γoverall + 1)
1 − Γoverall
(7)
where Z 0 is the characteristic impedance, usually 50 .
In (6), the first term is the structural mode, and the second
term is the antenna mode. Switching the phase of the phase
shifter, the phase of Γoverall will be tuned accordingly, thus
changing the antenna mode part, while the structural mode
part remains unchanged. Thus, the reconfigurable part of the
scattered pattern is proportional to E unit , which is set by the
antenna’s structure and type. An antenna with wide beamwidth
is preferred to realize wide-angle beamforming, and a patch
antenna and a relatively wide beam are reasonable choices.
Based on the desired antenna requirements, we designed
and fabricated a patch antenna as the scattering element of the
system, as shown in Fig. 11(a) and (b). The designed antenna
consists of two layers, separated by height h 2 , to make the
bandwidth of the proposed antenna wider compared to a thin
single-layer design. The bottom layer is a patch antenna fed
by the probe, and the top layer is a metal sheet slightly bigger
than the patch at the bottom. Both layers are fabricated on a
substrate of Rogers 4003C with (r = 3.55 and loss tangent =
0.0027) and a thickness of 1.524 mm.
The measured and simulated S11 parameter of the fabricated
antenna is shown in Fig. 12(a). There is a frequency shift
of about 50 MHz (2% of the center frequency) between the
simulated and measured results due to the fabrication error.
The bandwidth of the antenna is almost the same as the phasereconfigurable reflection amplifier to prevent oscillation outside of the designed bandwidth. Fig. 12(b) provides the mutual
coupling between the adjacent antenna in the fabricated array,
which is smaller than −20 dB. Fig. 12(c) and (d) provides
a comparison between the simulated and measured radiation
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Fig. 13. Thevenin equivalent of the active RIS illuminated by the external
electromagnetic field.
at different frequencies required by various applications, such
as 2.1 or 3.5 GHz for mobile services.
IV. ACTIVE RIS P ERFORMANCE
Fig. 11.
RIS element antenna design. (a) Structure of the designed
two-layer patch antenna. The dimensions are h 1 = 1.524 mm, h 2 = 12 mm
l1 = 34.5 mm, l2 = 72 mm, l3 = 31.5 mm, d1 = 20.25 mm, and
d2 = 15.15 mm. Gray: dielectric layer. Brown: copper. (b) Photograph of
the fabricated 2 × 2 antenna array.
Fig. 12. Performance of the fabricated antenna. (a) Measured and simulated
input return losses. (b) Measured mutual coupling between the adjacent
antennas in the array. (c) Measured and simulated radiation patterns in the
E-plane (unit: dBi). (d) Measured and simulated radiation pattern in the
H -plane (unit: dBi).
patterns of the antenna. The measured results agree well
with the simulated results, and the 3-dB beamwidth is wide,
reaching about 60◦ .
In terms of the communication frequency bandwidth, the
fabricated active RIS can cover all 14 channels of Wi-Fi
from 2401 to 2495 MHz, while a thin single-layer design
would only be able to cover three channels [38]. It is also
possible to scale the circuit and antenna so that it can operate
Based on the designed components in Section III, an active
RIS with any required size can be formed by using a different number of active elements. In this section, we provide
simulation and measurement results for an active RIS with
2 × 2 elements. As the first step, we first describe how to
perform the active RIS scattered pattern evaluation.
A. Analytical Calculation of the RIS Scattered Pattern
Due to the active properties of the reflection amplifier, electromagnetic solvers, such as CST Microwave Studio, cannot be
used directly to simulate and predict the scattered field pattern
by active RIS. To address this issue, an analytical method
based on the theory of Thevenin equivalent circuits [39]
and the linear superposition of electromagnetic waves in free
space [40] is proposed to calculate the scattered pattern by
active RIS.
As shown in Fig. 13, an active RIS consisting of M elements
in an antenna array is loaded by active loads. The properties of
the antenna array can be expressed as an M × M impedance
matrix Z, where Z mn (m = n) is the mutual impedance
between the mth antenna and the nth antenna, and Z mn (m = n)
is the input impedance of the mth antenna. The impedance
matrix Z can be obtained using standard electromagnetic
solvers.
The voltage and current at the array’s ports are denoted by
vectors v = [v 1 , v 2 , . . . , v M ]T and i = [i 1 , i 2 , . . . , i M ]T where
T
is the transpose operation, and v m and i m refer to the voltage
and current at the mth antenna, respectively. When there are
incident electromagnetic waves, by the theory of the Thevenin
equivalent circuit [39], Z, v and i can be related by
voc = v − Zi
(8)
where voc = [v oc,1 , v oc,2 , . . . , v oc,M ]T with v oc,m denoting the
open-circuit voltage excited at the port of the mth antenna.
We express the impedance of the active load connected
to the mth antenna as Z mL , and a diagonal matrix Z L =
L
diag(Z 1L , Z 2L , Z 3L , . . . , Z M
) can be formed by collecting Z mL
∀m. Z L , v, and i can also be related by
v = −Z L i
(9)
where Z L can be obtained from measured data in
Section III-C.
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Combining (8) and (9) together, the current at each port of
the antenna array excited by external waves can be found as
i = −(Z + Z L )−1 voc .
(10)
The scattered field pattern can then be calculated using the
expression
E s () =
M
i m E m () + E oc ()
(11)
m=1
where E oc () is the scattered pattern when all ports of the
antenna array are open-circuited and E m (θ, ϕ) is the electric
fields radiated by the active RIS when a unit current source is
excited at the port of the mth antenna with all the other ports
open. The currents i m ∀m can be obtained from (10). When the
active RIS contains only one element, (11) can be reconciled
into the form of a scattered mode and antenna mode using (6).
For any combination of the states in the active RIS, its
corresponding i and the scattered pattern can be found from
(10) and (11). For an active RIS with a large number of
elements, the proposed method is very efficient as we only
need to use electromagnetic solvers, such as CST Microwave
Studio, once to obtain Z, E m () ∀m, E oc (), and voc for all
the combinations. It should be noted that the mutual coupling
between antennas is also already considered in the calculation
of the scatted pattern as Z contains the information of mutual
coupling.
In the proposed design, each element has four states that can
provide four scattered fields with different phases. Therefore,
for an active RIS with 2 × 2 elements, a total of 44 = 256
combinations of states are available. Based on the method
in (10) and (11), we can calculate the 256 scattered wave
patterns at 2.4 GHz when the RIS is excited by a plane
wave with vertical incidence, and the reflection amplifier is
in the active mode. The calculated patterns with nine representative directions of the main beam are shown in Fig. 14,
which demonstrates that 3-D beamforming with −25◦ to 25◦
scanning range can be realized with the 2 × 2 active RIS.
The relatively small scanning range is limited by the number
of elements utilized, and a wider range can be realized if a
larger array with more elements is adopted. Fig. 15 shows
a comparison of the normalized scattered patterns between
active and passive modes. It can be found that the RIS in the
active mode can provide reconfigurable scattered patterns with
about 10-dB higher gain compared to the passive version with
the same phase shifters.
Fig. 14. Nine examples of the calculated scattered wave pattern for 2.4 GHz
when the incident wave is a plane wave with vertical incidence, the reflection
amplifier is in active mode, and the main beam is steered to (a) (θ, φ)beam =
(25◦ , 135◦ ), (b) (θ, φ)beam = (25◦ , 90◦ ), (c) (θ, φ)beam = (25◦ , 45◦ ),
(d) (θ, φ)beam = (25◦ , 180◦ ), (e) (θ, φ)beam = (0◦ , 0◦ ), (f) (θ, φ)beam =
(25◦ , 0◦ ), (g) (θ, φ)beam = (25◦ , 225◦ ), (h) (θ, φ)beam = (25◦ , 270◦ ), and
(i) (θ, φ)beam = (25◦ , 315◦ ).
Fig. 15. Calculated scattered pattern in the X O Z plane at 2.4 GHz when
the reflection amplifier is operating in the active mode and the passive mode.
B. Experimental Results
We have also fabricated an active RIS with 2 × 2 elements
to verify the proposed design, and a controlling system is also
designed to configure the states of each element, as shown in
Fig. 16. Each element has one dc supply line, one ground line,
and four controlling lines. A total of 16 controlling lines of
the four elements are connected to the I/O pins (3.3-V output)
of the FPGA, which can configure the states of the elements
to tune the scattered pattern.
As shown in Fig. 17, an experimental testbed was established to measure the scattered patterns by the fabricated
active RIS. Two horn antennas are used: one for the transmitter
to provide an incident plane wave and one for the receiver
to measure the scattered signal by the RIS. Antennas were
placed in the far-field region of the fabricated RIS, and the
distance between RIS and antennas was kept constant, at 3 m,
throughout the measurement. Utilizing a vector network analyzer, Rohde & Schwarz ZVA40, the S21 parameter between
transmitter and receiver was obtained.
The scattered pattern was measured by fixing the transmitter’s position in the direction of normal incidence and
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Fig. 16. Photograph of the prototype of active RIS with 2 × 2 elements and
the controlling system using an FPGA.
Fig. 18. Measured and calculated normalized scattered wave pattern in the
X O Z plane at 2.4 GHz when (a) (θ, φ)beam = (−20◦ , 0◦ ), (b) (θ, φ)beam =
(−10◦ , 0◦ ), (c) (θ, φ)beam = (0◦ , 0◦ ), and (d) (θ, φ)beam = (20◦ , 0◦ ).
is fixed in the z-axis direction. It also provides a comparison of
the normalized scattered patterns between active and passive
modes. It can be observed that the measured pattern generally
agrees with the predicted results using (11), and the small
disagreement is due to the fabrication error, the nonideal wave
excitation, background scattering, and positioning errors. The
fabricated active RIS can realize beam steering from −20◦
to 20◦ in the xoz plane, and the measured average gain is
about 8.5 dB, varying from 7 to 11 dB compared to the
passive version with the same phase shifters. Since the input
impedance of the fabricated antennas is not exactly 50 , the
gain of the scattered pattern is slightly different from the gain
of the reflection amplifier as in Fig. 9. A wider beam steering
range can be realized when active RIS with more elements is
fabricated.
Fig. 17. Photograph of the experimental setup for measuring the scattered
wave pattern of active RIS. The configurations of the coordinate system and
its origin O are also shown.
V. C OMPARISON AND D ISCUSSION
A. Comparison With Related Work
rotating the receiver in 5◦ steps. In addition, to compensate
for the effect of the background scattering and reflections
from the environment, at each position of the receiver, two
S21 parameters were obtained. One is to evaluate the scattered
field by the environment or background, denoted as S21,env (),
which is measured without the RIS. Another is due to both the
environment and the tested RIS, denoted as S21,total (), which
can be obtained by putting the RIS back to the testbed and
keeping the environment unchanged as in the measurement of
S21,env (). Thus, the scattered RIS pattern, S21,scat (), can be
found as
S21,scat () = S21,total () − S21,env ().
(12)
Fig. 18 provides the comparison between the calculated
and measured scattered pattern in the xoz plane with four
representative directions of main beams when the transmitter
There are only a few papers on active RIS design or amplifying reconfigurable reflectarray design [19], [20], [21], [23].
Comparisons of our proposed design with the state-of-the-art
and related works are shown in Table II. In [19], an amplifying surface operating within C-band was proposed, where
a commercial PA was connected to the two ports of a patch
element with two orthogonal hourglass-shaped slots. In [20],
an amplifying reconfigurable reflectarray was proposed and
verified, where a phase shifter and an amplifier were connected
in series between the two ports of a patch antenna to realize
tunable phases and amplification. In [21] and [23], similar
to [19], an amplifier connected between the two ports of an
antenna was proposed at 5.8 and 1.6 GHz, respectively.
All the previous works [19], [20], [21], [23] use two-port
amplifiers and two-port antenna architectures. The polarizations of the two ports of the antenna are also usually orthogonal to ensure low mutual coupling. Thus, this method’s
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TABLE II
C OMPARISON W ITH R ELATED W ORK
limitation is that the scattered wave’s polarization is orthogonal
to the incident wave since the signal is received in one port
and transmitted in another orthogonal port. This reduces the
polarization diversity available to the wireless system and may
reduce performance. On the other hand, for our proposed
architecture, a phase-reconfigurable reflection amplifier with
only one port has been developed. Therefore, the active
RIS element only uses a one-port antenna, and the scatted
wave’s polarization remains unchanged. With our approach,
by utilizing a two-port antenna with two phase-reconfigurable
reflection amplifiers, the proposed active RIS element can be
easily extended into a dual-polarization version with independent control of each polarization, which can amplify and tune
the incident wave with any polarization and, therefore, does
not reduce diversity. In our design, phase reconfiguration is
also included, which is important for beamforming and SNR
enhancement in the active RIS.
B. Complexity and Cost
The proposed active RIS element comprises three main
components: antenna, phase shifter, and reflection amplifier.
From the perspective of these three components, complexity
and cost can be analyzed.
The main complexity of RIS arises from its reconfigurability, which allows the properties of the element to be changed
in real time. In the proposed architecture, the antenna and
reflection amplifier are fixed components with no reconfigurability, and their properties cannot be changed after fabrication.
Therefore, they can be considered standalone components.
The phase shifter, however, is a reconfigurable component.
Its complexity is related to the resolution or control bits of
the phase shifter. More bits require more tunable states, more
control lines, and a more complex controlling system. This
is particularly relevant for RIS size scaling as the control
line routing and controlling system become significantly more
intricate with more elements. In our design, we utilize a
two-bit phase shifter to limit its complexity. A critical aspect is
the complexity of the controller and associated system design.
Channel estimation must also be performed for every element,
and as the surface scales up with more elements, this may also
become prohibitive. Tradeoffs in the system’s performance and
the number of elements are, therefore, required [41], [42].
There are two costs for the proposed active RIS design:
power cost and fabrication cost. For the power cost or power
consumption, the antenna is a completely passive component
consuming no power, and the phase shifter also only consumes negligible power due to the RF switches. Four SPDTs
were used for each element in the phase shifter, and a total
power of 0.066 mW is needed to drive these switches. The
main power consumer is the reflection amplifier, which needs
approximately 17.5 mW for each element. If an active RIS
with 64 elements is implemented, its power consumption is
about 1.12 W.
As an early work implementing an active RIS, our work
focuses more on the method to realize the active RIS element
with less focus on reducing the power consumption of the
elements. Therefore, we believe that the power consumption
of the designed reflection amplifier can be reduced. Instead of
using the self-biasing resistor, a direct dc supply at the source
of the transistor could be used. Alternatively, using lower
biasing voltage to design the circuit can also reduce power
consumption. In terms of the relationship between the power
gain and the power consumption of active RISs, a general rule
is that it is easier to achieve high power gain with a higher
power consumption budget.
Fabrication cost mainly arises from the printed circuit
boards and components (switches, inductors, capacitors, and
transistors). The indicative fabrication cost for each element
is about $45. The component and printed circuit board costs
are indicative only as they are the listed retail prices. If a
large number of elements are fabricated simultaneously, the
fabrication cost can be greatly reduced.
C. Noise and SNR
For the proposed and prototyped active RIS elements, even
though the incident signal can be amplified with around 8.5-dB
power gain, the noise will also be amplified, and extra noise
will also be introduced to the amplified signal.
For a single active RIS element, the SNR of the signal will
become worse due to the additional noise. However, an RIS
usually consists of a large number of elements. At the receiver,
the signals reflected by the different elements can be tuned
in phase and be added constructively, while the phases of
the noise from the elements are random. Therefore, if the
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number of elements is N, the power of the signal received
by the receiver scales by order N 2 , while the noise at the
receiver only scales by order N. Consequently, the SNR of
the received signal will increase approximately by order N
when N is large. This is why the active RIS can still provide
performance improvement to the system and complement the
existing passive RIS when additional noise is introduced [13].
Given the active RIS can enhance SNR, it can outperform
passive RIS in scenarios where the direct link between the
transmitter and receiver is strong [13]. Specifically, for a
passive RIS with the same number of elements as an active
RIS, although the SNR of the signal reflected by it is higher
than that of the active counterpart, the improvement to the total
signal received by the receiver is negligible. This is because
the absolute strength of the signal from the passive RIS is weak
compared to the signal from the direct transmitter-receiver path
due to the “double fading” effect. However, for the active RIS,
it is different due to the amplification. The relatively large
strength of the signal reflected by the active RIS can still
noticeably enhance the SNR of the total signal received by
the receiver. It is shown that, with some realistic assumptions,
the SNR of an active RIS-aided system can outperform that
of passive RIS by 40 dB when N is 256 [12]. In other
words, in these scenarios, many more elements are required
for passive RIS to reach the same performance as active RIS.
D. Tradeoffs and Constraints
As analyzed in Section II-B, S22 gamp , the product of the
phase shifter’s return loss and the reflection amplifier’s gain,
should be close to zero so that the phase adjustment will not
affect the amplification gain. That is, the adjustment of phase
is “decoupled” from effects on the amplifier gain. For a phase
shifter, S22 cannot be very small, and there is always a certain
return loss (usually −15 dB for commercial products and
−25 dB for our designed phase shifter, as shown in Table I).
Therefore, to meet our design criteria, the gain of the reflection
amplifier, gamp , cannot be too large. However, the magnitude
of overall also depends on gamp . If we want to have a relatively
large final gain at the element of RIS, gamp should be as large
as possible so that it can compensate for the double insertion
loss, L PS , of the phase shifter. This is the tradeoff between the
reflection amplifier’s gain and the phase shifter’s loss, and is
also the primary constraint to be considered in the design.
There are also other constraints that need to be carefully
considered. In the design of the reflection amplifier, a capacitor
Cgs is added between the gate and the source of the transistor
to avoid amplification outside the frequency band of interest.
Cgs depends on the desired frequency band and should be
carefully selected. This is because, when the reflection amplifier is connected to the phase shifter, the amplification outside
the working band will lead to potential oscillation, disabling
the whole circuit. The selection of the choke inductor for the
dc voltage supply is also critical. High Q-factor inductors are
preferred to avoid potential oscillation. The final constraint is
that the working bandwidth of the antenna should be similar
to or larger than that of the phase-reconfigurable reflection
amplifier. This is also to avoid signal oscillation outside the
working band of the element. This phenomenon was also
mentioned in [28].
VI. C ONCLUSION
A design for an active RIS with beam-scanning and amplification capability has been described. The active RIS comprises
a number of active elements containing two main components:
a two-layer patch antenna and a phase-reconfigurable reflection amplifier. Theoretical analysis and numerical examples
were provided to quantify the tradeoffs between the reflection
amplifier’s gain and the phase shifter’s loss. Guided by the
analysis, the reflection amplifier was designed to have a gain
of about 13 dB, and the phase shifter was designed with
1.9-dB insertion loss and 25-dB return loss. In addition,
an analytical method has also been described to calculate the
scattered pattern by the active RIS with active devices, such as
transistors. The proposed active RIS with 2 × 2 elements was
also fabricated and measured. Simulation and experimental
results have shown that the proposed RIS could effectively
steer a beam with 8.5-dB gain (compared to the passive version
with the same phase shifters), and the scanning range depends
on the size of the RIS. The advantages of amplification
and reconfigurable phases make the proposed active RIS a
promising candidate in future 6G communication networks for
solving the “double fading” problem faced by conventional
RIS.
R EFERENCES
[1] L. Subrt and P. Pechac, “Intelligent walls as autonomous parts of smart
indoor environments,” IET Commun., vol. 6, no. 8, pp. 1004–1010,
May 2012.
[2] N. Kaina, M. Dupré, G. Lerosey, and M. Fink, “Shaping complex
microwave fields in reverberating media with binary tunable metasurfaces,” Sci. Rep., vol. 4, no. 1, pp. 1–8, Oct. 2014.
[3] S. Hu, F. Rusek, and O. Edfors, “Beyond massive MIMO: The potential
of data transmission with large intelligent surfaces,” IEEE Trans. Signal
Process., vol. 66, no. 10, pp. 2746–2758, May 2018.
[4] E. Basar, M. D. Renzo, J. D. Rosny, M. Debbah, M. Alouini, and
R. Zhang, “Wireless communications through reconfigurable intelligent
surfaces,” IEEE Access, vol. 7, pp. 116753–116773, 2019.
[5] M. D. Renzo et al., “Smart radio environments empowered by reconfigurable intelligent surfaces: How it works, state of research, and the road
ahead,” IEEE J. Sel. Areas Commun., vol. 38, no. 11, pp. 2450–2525,
Nov. 2020.
[6] C. Huang, A. Zappone, G. C. Alexandropoulos, M. Debbah, and
C. Yuen, “Reconfigurable intelligent surfaces for energy efficiency in
wireless communication,” IEEE Trans. Wireless Commun., vol. 18, no. 8,
pp. 4157–4170, Aug. 2019.
[7] B. D. et al., “Hybrid beamforming for reconfigurable intelligent surface based multi-user communications: Achievable rates with limited
discrete phase shifts,” IEEE J. Sel. Areas Commun., vol. 38, no. 8,
pp. 1809–1822, Aug. 2020.
[8] S. Hong, C. Pan, H. Ren, K. Wang, and A. Nallanathan, “Artificial-noiseaided secure MIMO wireless communications via intelligent reflecting surface,” IEEE Trans. Commun., vol. 68, no. 12, pp. 7851–7866,
Sep. 2020.
[9] Y. Ai, F. A. P. de Figueiredo, L. Kong, M. Cheffena, S. Chatzinotas,
and B. Ottersten, “Secure vehicular communications through reconfigurable intelligent surfaces,” IEEE Trans. Veh. Technol., vol. 70, no. 7,
pp. 7272–7276, Jul. 2021.
[10] C. Huang, R. Mo, and Y. Yuen, “Reconfigurable intelligent surface
assisted multiuser MISO systems exploiting deep reinforcement learning,” IEEE J. Sel. Areas Commun., vol. 38, no. 8, pp. 1839–1850,
Jun. 2020.
[11] M. Najafi, V. Jamali, R. Schober, and H. V. Poor, “Physics-based modeling and scalable optimization of large intelligent reflecting surfaces,”
IEEE Trans. Commun., vol. 69, no. 4, pp. 2673–2691, Apr. 2021.
Authorized licensed use limited to: NATIONAL INSTITUTE OF TECHNOLOGY CALICUT. Downloaded on January 11,2024 at 16:21:28 UTC from IEEE Xplore. Restrictions apply.
RAO et al.: ACTIVE RIS UTILIZING PHASE-RECONFIGURABLE REFLECTION AMPLIFIERS
[12] Z. Zhang et al., “Active RIS vs. passive RIS: Which will prevail in 6G?”
2021, arXiv:2103.15154.
[13] R. Long, Y. C. Liang, Y. Pei, and E. G. Larsson, “Active reconfigurable
intelligent surface-aided wireless communications,” IEEE Trans. Wireless Commun., vol. 20, no. 8, pp. 4962–4975, Aug. 2021.
[14] M. H. Khoshafa, T. M. N. Ngatched, M. H. Ahmed, and
A. R. Ndjiongue, “Active reconfigurable intelligent surfaces-aided wireless communication system,” IEEE Commun. Lett., vol. 25, no. 11,
pp. 3699–3703, Nov. 2021.
[15] A. R. Ndjiongue, T. M. N. Ngatched, O. A. Dobre, and H. Haas,
“Design of a power amplifying-RIS for free-space optical communication systems,” IEEE Wireless Commun., vol. 28, no. 6, pp. 152–159,
Dec. 2021.
[16] C. You and R. Zhang, “Wireless communication aided by intelligent
reflecting surface: Active or passive?” IEEE Wireless Commun. Lett.,
vol. 10, no. 12, pp. 2659–2663, Dec. 2021.
[17] S. W. Peters, A. Y. Panah, K. T. Truong, and R. W. Heath, “Relay
architectures for 3GPP LTE-advanced,” EURASIP J. Wireless Commun.
Netw., vol. 2009, no. 1, pp. 1–14, Dec. 2009.
[18] M. D. Renzo et al., “Reconfigurable intelligent surfaces vs. relaying:
Differences, similarities, and performance comparison,” IEEE Open
J. Commun. Soc., vol. 1, pp. 798–807, 2020.
[19] L. Wu et al., “A wideband amplifying reconfigurable intelligent surface,”
IEEE Trans. Antennas Propag., vol. 70, no. 11, pp. 10623–10631,
Nov. 2022.
[20] K. K. Kishor and S. V. Hum, “An amplifying reconfigurable reflectarray
antenna,” IEEE Trans. Antennas Propag., vol. 60, no. 1, pp. 197–205,
Jan. 2012.
[21] X. Yang, S. Xu, F. Yang, M. Li, H. Fang, and Y. Hou, “A distributed power-amplifying reflectarray antenna for EIRP boost applications,” IEEE Antennas Wireless Propag. Lett., vol. 16, pp. 2742–2745,
2017.
[22] M. E. Bialkowski, A. W. Robinson, and H. J. Song, “Design, development, and testing of X-band amplifying reflectarrays,” IEEE Trans.
Antennas Propag., vol. 50, no. 8, pp. 1065–1076, Aug. 2002.
[23] R. W. Clark, G. H. Huff, and J. T. Bernhard, “An integrated active
microstrip reflectarray element with an internal amplifier,” IEEE Trans.
Antennas Propag., vol. 51, no. 5, pp. 993–999, May 2003.
[24] L. Cabria, J. Á. García, J. Gutiérrez-Ríos, A. Tazón, and J. Vassal’lo, “Active reflectors: Possible solutions based on reflectarrays and
Fresnel reflectors,” Int. J. Antennas Propag., vol. 2009, pp. 1–13,
Apr. 2009.
[25] D. M. Pozar, Microwave Engineering. Hoboken, NJ, USA: Wiley,
2011.
[26] R. C. Hansen, “Relationships between antennas as scatterers and as
radiators,” Proc. IEEE, vol. 77, no. 5, pp. 659–662, May 1989.
[27] F. Farzami, S. Khaledian, B. Smida, and D. Erricolo, “Reconfigurable dual-band bidirectional reflection amplifier with applications in
Van Atta array,” IEEE Trans. Microw. Theory Techn., vol. 65, no. 11,
pp. 4198–4207, Nov. 2017.
[28] S.-J. Chung, S.-M. Chen, and Y.-C. Lee, “A novel bi-directional amplifier
with applications in active Van Atta retrodirective arrays,” IEEE Trans.
Microw. Theory Techn., vol. 51, no. 2, pp. 542–547, Feb. 2003.
[29] P.-L. Chi and C.-L. Huang, “Reconfigurable 1.5–2.5-GHz phase shifter
with 360◦ relative phase-shift range and reduced insertion-loss variation,” in IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2017, pp. 897–899.
[30] E. Erkek and S. Demir, “S-band hybrid 4 bit phase shifter,” in Proc.
10th Medit. Microw. Symp., Aug. 2010, pp. 40–43.
[31] M. Meghdadi, M. Azizi, M. Kiani, A. Medi, and M. Atarodi, “A 6-bit
CMOS phase shifter for S-band,” IEEE Trans. Microw. Theory Techn.,
vol. 58, no. 12, pp. 3519–3526, Dec. 2010.
[32] R. Amirkhanzadeh, H. Sjoland, J.-M. Redoute, D. Nobbe, and
M. Faulkner, “L-band 180◦ passive phase shifter employing autotransformer in an SOS process,” in Proc. IEEE Int. Symp. Circuits Syst.
(ISCAS), Jun. 2014, pp. 333–336.
[33] J. C. Wu, T. Y. Chin, S. F. Chang, and C. C. Chang, “2.45-GHz CMOS
reflection-type phase-shifter MMICs with minimal loss variation over
quadrants of phase-shift range,” IEEE Trans. Microw. Theory Techn.,
vol. 56, no. 10, pp. 2180–2189, Oct. 2008.
[34] F. Ellinger, H. Jackel, and W. Bachtold, “Varactor-loaded transmissionline phase shifter at C-band using lumped elements,” IEEE Trans.
Microw. Theory Techn., vol. 51, no. 4, pp. 1135–1140, Apr. 2003.
[35] T. M. Hancock and G. M. Rebeiz, “A 12-GHz SiGe phase shifter with
integrated LNA,” IEEE Trans. Microw. Theory Techn., vol. 53, no. 3,
pp. 977–983, Mar. 2005.
3201
[36] D.-W. Kang, H. D. Lee, C.-H. Kim, and S. Hong, “Ku-band MMIC
phase shifter using a parallel resonator with 0.18-μm CMOS technology,” IEEE Trans. Microw. Theory Techn., vol. 54, no. 1, pp. 294–301,
Jan. 2006.
[37] L.-H. Lu and Y.-T. Liao, “A 4-GHz phase shifter MMIC in 0.18μm CMOS,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 10,
pp. 694–696, Oct. 2005.
[38] IEEE Standard for Information Technology—Telecommunications and
Information Exchange between Systems Local and Metropolitan Area
Networks–Specific Requirements—Part 11: Wireless LAN Medium
Access Control (MAC) and Physical Layer (PHY) Specifications, Standard 802.11-2020, 2021.
[39] J. R. Mautz and R. F. Harrington, “Modal analysis of loaded Nport scatterers,” IEEE Trans. Antennas Propag., vol. AP-21, no. 2,
pp. 188–199, Mar. 1973.
[40] C. A. Balanis, Advanced Engineering Electromagnetics. Hoboken, NJ,
USA: Wiley, 2012.
[41] N. K. Kundu and M. R. McKay, “Channel estimation for reconfigurable
intelligent surface aided MISO communications: From LMMSE to deep
learning solutions,” IEEE Open J. Commun. Soc., vol. 2, pp. 471–487,
2021.
[42] N. K. Kundu, Z. Li, J. Rao, S. Shen, M. R. McKay, and R. Murch,
“Optimal grouping strategy for reconfigurable intelligent surface assisted
wireless communications,” IEEE Wireless Commun. Lett., vol. 11, no. 5,
pp. 1082–1086, May 2022.
Junhui Rao (Graduate Student Member, IEEE)
received the B.Eng. degree in microelectronic science and engineering from the University of Electronic Science and Technology of China, Chengdu,
China, in 2020. He is currently pursuing the Ph.D.
degree at the Department of Electronic and Computer Engineering, The Hong Kong University of
Science and Technology, Hong Kong.
His current research interests include reconfigurable intelligent surfaces, microwave circuits,
multiple-input multiple-output (MIMO) systems,
millimeter waves, and sixth generation (6G).
Yujie Zhang (Member, IEEE) received the bachelor’s degree in optoelectronic information science
and engineering from the Huazhong University of
Science and Technology, Wuhan, China, in 2017,
and the Ph.D. degree in electronic and computer
engineering from The Hong Kong University of
Science and Technology (HKUST), Hong Kong,
in 2021.
He is currently a Post-Doctoral Fellow with
HKUST. His current research interests include
antenna design on Internet-of-Things applications,
reconfigurable antenna and surface, multiple-input multiple-output (MIMO)
systems, millimeter wave, RF energy harvesting, wireless power transmission,
and sixth generation (6G).
Shiwen Tang (Graduate Student Member, IEEE)
received the bachelor’s degree in radio wave propagation and antenna and the master’s degree in circuits and systems from the University of Electronic
Science and Technology of China, Chengdu, China,
in 2016 and 2019, respectively, and the M.Sc. degree
in electronic and electrical engineering from the
University of Strathclyde, Glasgow, U.K., in 2018.
She is currently pursuing the Ph.D. degree at the
Department of Electronic and Computer Engineering, The Hong Kong University of Science and
Technology (HKUST), Hong Kong, China.
Her current research interests include the millimeter-wave antenna design,
multiple-input multiple-output (MIMO) antennas, reconfigurable antenna
design, and sixth generation (6G).
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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 71, NO. 7, JULY 2023
Zan Li (Student Member, IEEE) received the B.Eng.
degree in electromagnetic field and wireless technology from the Huazhong University of Science and
Technology, Wuhan, China, in 2021. He is currently
pursuing the Ph.D. degree at the Department of Electronic and Computer Engineering, The Hong Kong
University of Science and Technology, Hong Kong.
His current research interests include reconfigurable intelligent surfaces, integrated sensing and
radar communication, and RF sensing.
Chi-Yuk Chiu (Senior Member, IEEE) received the
B.Eng. and M.Eng. degrees and the Ph.D. degree in
electronic engineering from the City University of
Hong Kong, Hong Kong, in 2001, 2001, and 2005,
respectively.
In 2005, he joined the Department of Electronic
and Computer Engineering (ECE), The Hong Kong
University of Science and Technology (HKUST),
Hong Kong, as a Research Associate. He worked as
a Senior Antenna Engineer at Sony Mobile Communications, Beijing, China, in 2011. In 2015,
he joined the Department of ECE, HKUST, again as a Research Assistant
Professor. He has published over 100 technical papers and two book chapters.
He holds several patents related to antenna technology. His main research
interests include the design and analysis of small antennas, multiple-input
multiple-output (MIMO) antennas, applications of characteristic modes, and
energy harvesting.
Dr. Chiu is the Vice-Chair of the IEEE Antennas and Propagation Society
(AP-S)/Microwave Theory and Technology Society (MTT-S) Hong Kong
Joint Chapter, a member of the IEEE AP-S Education Committee, the IEEE
AP-S C. J. Reddy Travel Grant Assistant Coordinator, and a Lead Guest
Editor of a special section in IEEE O PEN J OURNAL OF A NTENNAS AND
P ROPAGATION (OJAP).
Ross Murch (Fellow, IEEE) received the
bachelor’s and Ph.D. degrees in electrical and
electronic engineering from the University of
Canterbury, Christchurch, New Zealand.
He was the Department Head with The
Hong Kong University of Science and Technology
(HKUST), Hong Kong, from 2009 to 2015. He is
currently a Chair Professor with the Department
of Electronic and Computer Engineering and a
Senior Fellow with the Institute of Advanced Study,
HKUST. He has been a David Bensted Fellow with
Simon Fraser University, Burnaby, BC, Canada, and an HKTIIT Fellow
with the University of Southampton, Southampton, U.K., and has spent
sabbaticals at the Massachusetts Institute of Technology (MIT), Cambridge,
MA, USA; AT&T, Dallas, TX, USA; Allgon Mobile Communications,
Åkersberga, Sweden; and Imperial College London, London, U.K. His
unique expertise lies in his combination of knowledge from both wireless
communication systems and electromagnetics, and he publishes in both
areas. He has successfully supervised over 50 research graduate students. His
research contributions include more than 175 journal articles and 20 patents.
His current research interests include RF imaging, ambient RF systems,
energy harvesting, multiport antenna systems, the Internet of Things, and
reconfigurable intelligent surfaces.
Dr. Murch is a Fellow of Institution of Engineering and Technology (IET)
and The Hong Kong Institution of Engineers (HKIE). He has won several
awards, including the Computer Simulation Technology (CST) University
Publication Award. He has served IEEE in various positions, including an
IEEE area editor, the technical program chair, a distinguished lecturer, and a
Fellow of the evaluation committee. He enjoys teaching and has won three
teaching awards.
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