sustainability
Review
Isolated DC-DC Power Converters for Simultaneous Charging
of Electric Vehicle Batteries: Research Review, Design,
High-Frequency Transformer Testing, Power Quality Concerns,
and Future
Srinath Belakavadi Sudarshan
and Gopal Arunkumar *
School of Electrical Engineering, Vellore Institute of Technology, Vellore 632014, India
* Correspondence: g.arunkumar@vit.ac.in
Citation: Sudarshan, S.B.;
Arunkumar, G. Isolated DC-DC
Power Converters for Simultaneous
Charging of Electric Vehicle Batteries:
Research Review, Design, HighFrequency Transformer Testing,
Power Quality Concerns, and Future.
Sustainability 2023, 15, 2813. https://
Abstract: The transportation industry is transitioning from conventional Internal Combustion Engine
Vehicles (ICVs) to Electric Vehicles (EVs) due to the depletion of fossil fuels and the rise in nontraditional energy sources. EVs are emerging as the new leaders in the industry. Some essential
requirements necessary for the widespread adoption of EVs include sufficient charging stations with
numerous chargers, less to no wait time before charging, quick charging, and better range. To enable
a quicker transition from ICVs to EVs, commercial organizations and governments would have to put
in a mammoth effort, given the low number of installed chargers in developing nations such as India.
One solution to lower the waiting time is to have multiple vehicles charging simultaneously, which
might involve charging two- and four-wheelers simultaneously, even though their battery voltage
ratings differ. This paper begins by providing the details of the power sources for EV charging,
the charging levels and connector types, along with the specifications of some of the commercial
chargers. The necessity of AC-DC converters in EV charging systems is addressed along with the
power quality concerns due to the increased penetration of EVs. Next, a review of the existing
research and technology of isolated DC-DC converters for simultaneous charging of EV batteries is
provided. Further, several potential isolated DC-DC converter topologies for simultaneous charging
are described with their design and loss estimation. A summary of the existing products and
projects with simultaneous charging features is provided. Finally, insight is given into the future of
simultaneous charging.
Keywords: AC-DC converters; DC-DC converters; EV battery charging; isolated converters;
lithium-ion battery charging; multiple battery charging; power factor correction; power quality;
simultaneous charging
doi.org/10.3390/su15032813
Academic Editor: Noradin Ghadimi
Received: 21 December 2022
Revised: 28 January 2023
Accepted: 30 January 2023
Published: 3 February 2023
Copyright: © 2023 by the authors.
Licensee MDPI, Basel, Switzerland.
This article is an open access article
distributed under the terms and
conditions of the Creative Commons
Attribution (CC BY) license (https://
creativecommons.org/licenses/by/
4.0/).
1. Introduction
The primary mode of transportation in the world is petrol- and diesel-based vehicles.
These vehicles use internal combustion engines (ICEs) to burn the fuel to run the vehicle.
However, there are several issues with the use of ICEs. It has low fuel efficiency, higher
operating cost, and harmful emissions [1]. Due to the emissions of ICEs, the resultant
climate changes have led to the development of hybrid electric vehicles (HEVs) and electric
vehicles (EVs).
1.1. Methods of Obtaining Electricity for Charging
Conventionally, electricity is obtained from thermal, hydro, and nuclear power plants.
While fossil-fuel-based thermal power plants can be used to produce power in megawatt levels, they also produce harmful gases such as carbon dioxide, carbon monoxide,
sulphur dioxide, nitrogen oxide, and ash [2], which has led to a steady increase in the
global temperature [3], and which in turn has caused the degradation of flora and fauna,
Sustainability 2023, 15, 2813. https://doi.org/10.3390/su15032813
https://www.mdpi.com/journal/sustainability
Sustainability 2023, 15, 2813
2 of 71
and the health of humans. In addition, the depletion of fossil fuels is a significant threat
to the continued use of fossil fuel-based power generation. The connection between
emissions and fossil fuel depletion is a significant concern today [4]. While hydro plants
do not lead to such an issue, the main problems with the hydro plant are related to the
submergence of the areas surrounding the dam, the impact on the water resources, and the
rainfall amount. The impact depends mainly on the dimensions of the hydro plant [5].
With nuclear power plants, the main issue is related to resource availability, the complexity
of control, the potential risk of leakage, and waste disposal. Various nations have developed
several policies to prevent nuclear disasters [6]. To prevent the issues resulting from the
conventional sources, non-conventional energy sources have been introduced into the
power generation sector. Among these resources, the solar photovoltaic (PV) system is
being used due to abundant resource availability. Since solar panels can be integrated with
vehicles, solar PV is an attractive option as a source to charge EVs. Nasr et al. [7] have
comprehensively reviewed solar PV based converters for EV battery charging. Several
non-isolated converters suitable for EV battery charging have been reviewed, along with
gird integrated converters. The primary issue that needs to be addressed with the use
of PV systems is the intermittency and variations of solar irradiance and temperature.
The intermittency has been addressed by Jin et al. [8]. Energy optimization is necessary
with the use of PV systems, as addressed by Al-Shahri et al. [9].
EVs, specifically battery electric vehicles (BEVs), have contributed to reducing pollution as they create no gas emissions. However, an important point to note is the power
source for charging [10]. If coal-based power plants are used for power production to
charge the battery, the burden on the system will be dominant. The positive environmental
impact will be significantly higher when renewable energy resources are used [10]. Another
issue that needs to be focused on with the increased use of EVs is the recycling and disposal
of a battery that has been damaged or completed its life cycle. Yu et al. [11] have addressed
the challenges in recycling Li-ion batteries and provided several suggestions to address
recycling. If these issues are addressed effectively, the use of EVs can create a more positive
impact on the environment.
EV research is experiencing a boom in the present day due to a renewed emphasis on
reducing our carbon footprint [12], greenhouse gas emissions, and the depletion of fossil
fuel resources. Several significant difficulties have gained attention due to the growing
interest in EV use. These problems include, but are not limited to, the following:
1.
2.
3.
4.
5.
Power quality concerns due to EV charging;
Low number of charging stations;
Long waiting time before charging;
Higher time for charging;
Lower range of EVs.
Government policies outside the research community’s purview influence the number
of charging stations. The EV range issue is related to the type, C-rate, and battery rating,
which the battery manufacturers and researchers of battery technology usually address.
Electrical engineering research is directly or indirectly interested in the challenges of power
quality issues and reducing the waiting and charging times.
1.2. Charging Techniques for EV Batteries
The technology or method employed for charging depends on the battery chemistry.
Generally, Lithium-ion (Li-ion) batteries are used in EVs due to their high energy density, longer lifetime, and good electrochemical properties [13]. Several techniques for
charging EV batteries have been recommended and reviewed by researchers for EVs. Shahjalal et al. [14] have provided the details of the various charging techniques for EV batteries.
The charging methods could be conventional (slow) charging or fast charging. Conventional charging methods include constant current (CC) charging, constant voltage (CV)
charging, and constant-current constant-voltage (CC-CV) charging. Several researchers
have investigated the idea of reducing charging time by employing fast charging meth-
Sustainability 2023, 15, 2813
3 of 71
ods [15] such as flash charging [16], pulse charging [17,18], negative pulse charging [19],
multistage current control protocol [20], variable current profile charging [21], boost charging [22], and other algorithm-based methods [23,24]. Hemavathi et al. [25] have provided
details of the various conventional and fast charging methods with a qualitative comparison. In addition, the battery modeling has been provided with charging methodologies
specific to various converters. Different charging levels have been described, along with the
charging station architectures and fast charging topologies. The future trend of EV charging
has been described to include smart charging, integration of renewable sources, wireless
charging, and battery swapping. The authors have provided the road map of EV adaption.
Research focus has also been shifted to address the degradation effects on the battery due
to fast charging [26–28], its effect on the existing grid [29–31], and possible solutions [32].
While several of these methods are still at the research level, implementation in real-world
EVs is a possibility that needs to be addressed by the research and development wing of
the manufacturers.
1.3. EV Charging Levels, Connectors, and Standards
The charging levels are based on the power supplied, the source, and the charging
time [25,33].
•
•
•
•
Level-1 charging, also known as AC charging or residential charging, can supply up to
2 kW power from a 120 V, 16 A single-phase source;
Level-2 charging, also known as AC split-phase charging, can supply up to 20 kW
power from a 240 V, 80 A single-phase/split-phase source;
Level-3 charging, also known as DC charging, can supply between 120 kW and 350 kW
power from a variable DC source between voltages of 200 V and 920 V, at a maximum
current of 500 A;
Level-4 charging or Tesla Charging can supply 120 to 250 kW power from a 120 or 240,
or 400 V source. This is applicable only for Tesla EVs.
A level-1 charger will require around 17–20 h to fully charge a battery with a 2 kW
rating, while a level-2 charger will fully charge a 6.6 kW rated battery within 3.5 h. Level3 and level-4 chargers are fast chargers that can charge high-power batteries in under
30 min [25,33]. The Electric Vehicle Supply Equipment (EVSE) for EVs and plug-in hybrid
vehicles (PHEVs) are governed by IEC 61851-1:2017 standards for the conductive charging
system. For India, Bharat EV standards have been proposed by the Government of India
to standardize EV charging in the country. Kumar et al. [34] have reviewed India’s EV
charging station standards. The authors have presented the present scenario of EVs in
India and the different charging levels with the connector specifications. The authors
have also presented the details of the various existing international standards with their
applications. Details of Indian Standards (IS) have been described in detail with the
charging methodologies (CHArge de Move (CHAdeMO), Combined Charging System
(CCS), and Tesla.). Based on the charging standard, the types of connectors have been
standardized. The specifications of such charging connectors are listed below:
•
AC Connector [25]
1.
2.
•
CHAdeMO [25,35]
1.
2.
3.
•
Type-1: up to 7.4 kW; follows SAE J1772 standard;
Type-2: up to 43 kW for public charging; follows IEC 62196-2 standard.
Present: 50 kW, 500 V/125 A rated high power charger;
Proposed: over 100 kW, 500 V/200 A rated high power charger;
ChaoJi (CHAdeMO 3.0): over 500 kW, 600 A rated high power charger for
fast charging.
CCS [25,36]
1.
Type-1: 90 kW, 600 V/150 A (maximum ratings); 13 kW, 250 V (nominal rating);
Sustainability 2023, 15, 2813
4 of 71
2.
3.
•
Type-2: 170 kW, 850 V/200 A (maximum ratings); 44 kW, 230 V (single-phase)/
440 V (three-phase) (nominal rating);
Type-2 and DC connector Combination: 43 kW for AC charging and 100 kW for
DC charging; can go up to 350 kW.
Tesla Supercharger [25]
1.
480 V fast charging technology.
The newer definitions/ratings of chargers are proposed to enable fast charging and
are suitable for high-power vehicles.
2. Commercial EV Charger Specifications: Present Market Status
Several commercial organizations have their own patented EV charging stations with
specific advantages and features for the benefit of the customers. Some companies have
provided their charger specifications to help the public understand their charging system
specifications to enable an informed choice. This section gives a list of some of the electrical
specifications of the chargers.
2.1. Eaton Corporation
2.1.1. AC Charger-Eaton xChargeIn Mobility
The specifications of the Eaton AC Charger-Eaton xChargeIn Mobility are listed below [37].
•
•
•
•
•
Power Range: 3.7 kW to 22 kW;
Connector Type: Type 1 for power levels up to 7.4 kW, and Type 2 for power levels up
to 22 kW;
Input Voltage: 230 V, 50 Hz (1-phase) and 400 V 50 Hz (3-phase);
Input Current: 16 A and 32 A for both 1-phase and 3-phase at respective power levels;
Simultaneous Charging: No (only one vehicle can be charged at any given time).
2.1.2. DC Charger-Green Motion DC 22
The specifications of the DC Charger-Green Motion DC 22 are listed below [38].
•
•
•
•
•
•
•
•
Power: 22 kW;
Input Voltage: 400 V 50 Hz (3-phase);
Power Factor: Greater than 0.99;
Input Current: 32 A;
Output Voltage Range: 50 to 500 V DC;
Output Current: 55 A;
Efficiency: Greater than 96%;
Simultaneous Charging: No (only one vehicle can be charged at any given time).
2.1.3. DC Charger-Green Motion DC 44
The specifications of the Eaton DC Charger-Green Motion DC 44 are listed below [39].
•
•
•
•
•
•
•
•
Power: 44 kW;
Input Voltage: 400 V 50 Hz (3-phase);
Power Factor: Greater than 0.99;
Input Current: 64 A;
Output Voltage Range: 50 to 500 V DC;
Output Current: 110 A;
Efficiency: Greater than 96%;
Simultaneous Charging: No (only one vehicle can be charged at any given time).
2.1.4. DC Charger-Green Motion DC 66
The specifications of the Eaton DC Charger-Green Motion DC 66 are listed below [39].
•
Power: 66 kW;
Sustainability 2023, 15, 2813
5 of 71
•
•
•
•
•
•
•
Input Voltage: 400 V 50 Hz (3-phase);
Power Factor: Greater than 0.99;
Input Current: 96 A;
Output Voltage Range: 50 to 500 V DC;
Output Current: 165 A;
Efficiency: Greater than 96%;
Simultaneous Charging: No (only one vehicle can be charged at any given time).
2.2. Siemens
2.2.1. SICHARGE D for DC Fast Charging
The specifications of the Siemens SICHARGE D are listed below [40].
•
•
•
•
•
Nominal AC Input Voltage: 400 V, 50/60 Hz;
Nominal Input Current: 301 to 515 A, based on output power;
Power Factor: Greater than 0.99 at full load;
DC Output Power: 160 to 300 kW;
DC Output Voltage: 150 V to 1 kV.
2.2.2. VersiCharge for AC Charging
The specifications of the Siemens VersiCharge are listed below [41].
•
•
•
•
Nominal AC Input Voltage: 230 V (1-phase), 230 or 400 V (3-phase); 50/60 Hz;
Nominal Input Current: 10 to 32 A, based on output power;
DC Output Power: up to 7.4 kW (1-phase), up to 22 kW (3-phase);
DC Output Current: 32 A maximum.
2.3. Need for Simultaneous Charging of EV Batteries
A significant issue that needs to be addressed is the reduction in waiting time before
charging. One of the ways to reduce this is by installing more charging stations. While
this process can reduce the waiting time before charging, installing more charging stations
is related to other variables, such as land availability and acquisition and availability of
power supply. Increasing the number of charging stations may not be a very simple or
practical solution. In addition to optimizing the number of charging stations, it would be
of great help if simultaneous charging of multiple EVs with the same or different rating
batteries could be implemented. In a scenario where there are 10 chargers in a charging
station, with each charger capable of charging 1 EV at a time, 10 EVs can be charged at any
given time. This is shown in Figure 1. If it is possible to charge more than 1 battery (say, 2)
in 1 charger, then the charging station described above can charge 20 EVs simultaneously.
This creates a significant reduction in waiting time. This will directly create a positive
impact on the conversion from ICEVs to EVs for mass-use. An added advantage will be
obtained if multiple EVs of different ratings (say, an electric-bike and an electric-car) can
simultaneously be charged by the same charger. This is shown in Figure 2. In addition to
reducing waiting time, simultaneous charging also leads to higher profit for the company
that installs and maintains the charging station since more vehicles can be charged in a given
time-period, leading to an overall increase in the number of vehicles that are being charged
each day. With this, we can conclude that simultaneously charging multiple batteries of the
same and different ratings offers many advantages to customers and companies.
Sustainability 2023, 15, 2813
6 of 71
Figure 1. Conventional EV charging station with one charger charging one EV battery.
Figure 2. EV charging station with one charger charging multiple EV batteries.
Several authors have provided a detailed review of the various converters that can
be used for EV battery charging. Khalid et al. [42] have reviewed the various isolated
and non-isolated converter configurations that suit EV battery charging. In addition,
the authors have provided the configuration of a charging station for fast charging, which
includes the AC-DC multilevel converters. The concept of universal chargers has been
described in work with various charging methods. However, the work does not deal with
simultaneous charging and the corresponding modifications in the converters. In addition,
the required design equations and loss calculation are beyond the scope of the review.
Ghasemi-Marzbali et al. [43] have dealt explicitly with the infrastructure for fast charging of electric vehicles. The work gives details of the studies required for the charging
station design, site selection and size determination, charging time estimation, and modeling of renewable energy resources for charging stations. The demand-side management
Sustainability 2023, 15, 2813
7 of 71
principle has been introduced with risk factors and reliability considerations. The application of machine learning for system design has been described with a particular focus
on fast charging stations. However, the work does not describe any converter topology
required in the charger since the work’s objective is to review fast charging station requirements. Chakraborty et al. [44] have reviewed the various DC-DC converter topologies
with a special focus on EV battery charging. Several isolated and non-isolated DC-DC
converter topologies have been described in the work. Under the isolated converter topologies, full-bridge, current-fed converter, sinusoidal amplitude high voltage bus converter,
and multi-input single output converters have been described. Design formulae for the
memory elements have been provided for the described converters. The architecture of
fast charging stations has been provided with a description of converters for the AC-DC
and DC-DC conversion stages. In the DC-DC stage, full-bridge and phase-shifted fullbridge topologies for single-battery charging have been described. Common equations
for loss estimation have been provided with the comparative efficiency for various switching frequencies. Various performance parameters have been compared for the described
converters. However, the work does not describe loss estimation specific to the converters
and does not include details for simultaneous charging. In addition, only some isolated
converters have been described in previous work.
This work presents a detailed description of the various isolated converters that can
simultaneously charge multiple EV batteries. While high-power converters such as fullbridge and its derived converters can be used to charge the battery of any vehicle, it is
essential to note that a converter will give the desired performance only when it is operated
at or close to the rated or designed conditions [45]. For example, a full-bridge converterbased charger designed for 10 kW can charge an electric car and an electric bike of low
rating (say 250 W, such as the Hero Electric Flash electric bike used in India). However,
not only will the converter be underutilized to deliver very low power for charging the
electric bike, the cost per unit will be high when operated at low power [45]. To avoid
this problem, isolated converters that can deliver low and medium-range power are very
much needed when simultaneous charging infrastructure is planned since the same charger
module should be able to charge vehicles with diverse power ratings. Thus, it is necessary
to consider low and medium-power range isolated converters for EV battery charging
applications, especially with simultaneous charging. This work also presents the detailed
design and loss estimation process for various converters for low, medium, and high-power
charging. In addition, a procedure for testing the high-frequency transformer (HFT) is
provided, which has yet to be dealt with by previous literature.
This review work is organized as follows: Section 3 gives details of the main power
quality issues that need to be addressed with battery charging. Section 4 details the various
research works that have proposed and implemented simultaneous charging, with the
advantages and limitations of each work. Section 5 explains the various isolated DC-DC
converters that can be extended to charge multiple batteries simultaneously. The converters included in this section are the single-switch, two-switch isolated converters and
bridge-type converters, including resonant converters, with their design equations and loss
estimation. Section 6 gives a quantitative explanation of the multi-secondary HFTs required
in the isolated converters for simultaneous charging with their testing procedure. Section 7
explains the trend of EV batteries shifting towards 400 V and 800 V levels. Section 8 describes the various products and projects that have implemented simultaneous charging
features for public use. The paper concludes by describing the future research scope of
simultaneous charging.
3. Power Quality Due to EV Charging: Concerns and Solutions
The supply systems in the world are invariably AC systems. This necessitates using an
AC-DC converter in the battery-charging system. With grid connection, several important
power quality parameters need to be considered. The power systems are usually designed
keeping in mind the increase of load in the future. However, the sudden shift towards EVs
Sustainability 2023, 15, 2813
8 of 71
has drastically increased the power demand, most likely more than the predicted demand.
This has caused a negative impact on the grid and the system in general [46]. In addition, harmonics, low power factor, low voltage profile, and increased losses are causing
significant trouble to the power system [46]. The power quality issues must be addressed
more rigorously when multiple batteries are simultaneously charged. The power factor
and the Total Harmonic Distortion (THD) are the most important parameters that must
be addressed. IEC61000-3-2 [47] governs the Power factor correction (PFC) or harmonic
reduction. These parameters will be controlled using a feedback control system designed
for PFC and harmonic minimization.
3.1. Power Factor Correction
PFC can be done using two [48] methods: (a) passive PFC and (b) active PFC. The passive method uses passive components such as inductors and capacitors to filter out the
harmonics. The passive method is suitable for low-power applications (less than 100 W)
since it leads to lower efficiency, high cost, and weight due to the size of line-frequency
inductors and capacitors. On the other hand, active PFC methods use switching regulators
to correct the wave shape to obtain sinusoidal grid current at unity power factor. This
method is complex but is suited for high power levels. Active PFC can be done using
converters that operate in continuous conduction mode (CCM), critical conduction mode
(CrCM), and discontinuous conduction mode (DCM).
The general requirements of a PFC AC-DC converter topology include a simple power
stage with a low number of components, less distortions in the input current, and the ability
to achieve a near-unity power factor. Kolar et al. [49] and Friedli et al. [50] have provided
an exhaustive review of the various topologies of PFC rectifiers. Conventional Boost PFC
Rectifiers (CBRs) operating in CCM are among the most popular PFC topologies [51].
Integrated PFC controllers such as NCP1650 by ON Semiconductors [52] also work as a
CBR. With Silicon Carbide (SiC) and Gallium Nitride (GaN) MOSFETs and diodes, the issue
of output diode reverse recovery is mitigated [53]. The design of the memory elements of
the CBR is the same as that of a conventional boost converter. In addition to the CBR, dual
boost bridge-less PFC rectifiers [54–59], totem-pole bridge-less PFC rectifiers [60–63], and
interleaved boost PFC rectifiers [64] are more commonly used [65] in several products that
need PFC. On the three-phase system side, Vienna rectifiers have become very popular [66]
since they have inherent PFC capability. Several control techniques can be used to achieve
a unity power factor. Examples of control strategies include hysteresis control [67,68],
average and peak current mode controls [69–73], model predictive control [74], sliding
controller [75], and one-cycle control [76,77].
3.2. Harmonics and THD
EV battery charging systems are made of multiple power electronic converters and
analog electronic systems. These are non-linear loads to the grid and hence cause harmonics
in the grid current. The harmonics will be very significant when multiple vehicles are being
charged at the same time [78]. Not only will the harmonic pollution of the grid current be
higher in that part of the power system, but it will also reflect on all its connected systems
and cause distortion in those parts. The presence of harmonics affects the operation of all the
connected equipment and can also cause failures with considerable financial implications.
EV charging can cause unbalance in the power system. Due to the unbalance, negative
sequence components can be produced, producing a second-order harmonic ripple in the
DC link voltage. This causes distortions in the grid input currents. Equation (1) can be
used to explain the effects of harmonics [79] on the DC-link. The presence of even-order
harmonics on the DC-grid side will cause the generation of odd-harmonics on the gridside, leading to power quality issues. In Equation (1), α represents the negative sequence
components, and Idc represents the DC current.
Sustainability 2023, 15, 2813
9 of 71
3
I2
I
Idc +
sin(4π f t − α2 ) + 4 sin(8π f t − α4 )
4C
4π f
8π f
I6
I8
I
+
sin(12π f t − α6 ) +
sin(16π f t − α8 ) + 12 sin(24π f t − α12 )
12π f
16π f
24π f
Vdc =
(1)
Other effects of harmonics on the various components of the power system are well
known, and their solutions have been investigated for years. PFC converters are to be
controlled such that the harmonics are mitigated well and the THD levels are within
the permissible limits. In addition, strategic placement and optimal sizing of variable
passive filters can also help in harmonic reduction. Alame et al. [80] have comprehensively
analyzed the effects of harmonics due to EV charging on the various components of a
distribution system. The authors have explained transformer loss modeling, temperature
rise modeling, and lifetime modeling. A sample case study on a 1500 kVA distribution
transformer has been provided to show that the percentage of harmonic currents increased
with an increase in the battery state of charge. Further, the impact of EV charging on the
distribution system has been analyzed by considering the IEEE 33-bus system charging
four EVs at different buses using PV-based distributed generation units. The analysis has
been performed using the decoupled harmonic power flow algorithm to obtain the effect
on voltage quality and current THD. A centralized control flow has been proposed as an
optimization problem to mitigate the effects of harmonics.
3.3. Other Detrimental Effects
Due to the diverse charging rates of EVs owing to slow, fast, and ultra-fast chargers,
several negative effects can be observed. These include stability issues, unbalance, and overloading [79]. Dharmakeerthi et al. [81] have concluded that using fast charging stations can
cause issues in the grid since they can significantly reduce the steady state voltage stability
of the grid. In addition, the harmonics and inter-harmonics produced due to charging rates
will also affect the power system’s critical components, such as transformers, breakers,
cables, and meters. Alshareef and Morsi [82] have shown that fast charging stations can
significantly affect and cause voltage flicker in distribution systems.
Despite these issues, EVs will be integral to our system since the advantages outweigh
the limitations. Hence it is necessary to address these issues while designing a charging
station. Nguyen et al. [78] have provided the topology of a photovoltaic inverter used as an
active filter to mitigate power quality issues during simultaneous fast charging of five EV
batteries. The proposed system models a bidirectional DC-DC converter acting as a charger
and a DC-AC converter connected to the AC grid. Control structures for the control of both
converters to implement harmonic mitigation have been explained, and the corresponding
simulation results have been provided to validate the proposed scheme.
Once the AC power has been converted to DC and the power quality is maintained,
a DC-DC converter is required to regulate the voltage, match the battery voltage level,
and implement charging control. These DC-DC converters form a significant research
area and are the focus of the following sections of this review, specific to the simultaneous
charging of multiple batteries.
4. Simultaneous Charging Research: State of the Art
Only a few researchers have taken up experimental validation of simultaneous charging of EV batteries. Although it leads to the many advantages described in Section 2.3,
the research focus has been relatively less compared to the focus on fast charging for charging time reduction. This section describes the various converters proposed by researchers
working on simultaneously charging multiple batteries.
Mahafzah et al. [83] have proposed a synchronized multi-output hybrid buck-boost
converter based on the Ćuk-Flyback topologies for renewable energy systems and electric
vehicle applications. The proposed converter is shown in Figure 3. The design and steady-
Sustainability 2023, 15, 2813
10 of 71
state analysis are provided for the proposed converter. The authors have presented the
simulation results for a 300 W design to obtain a full-load efficiency of 90% at higher
duty cycles close to 80%. This is significantly higher than the conventional Flyback and
Ćuk converters. For EV battery charging applications, the authors have proposed an ACDC full-bridge PFC rectifier connected to the input terminals of the proposed converter.
A simulation of a 1.5 kW system is provided to validate the idea. With the proposed AC-fed
system, a power factor of 0.96 is obtained at the AC input side for an input of 200 V and
0.9401 for an AC input of 110 V. At these two input voltage levels, the THD is 27.69%
and 36.25%, respectively. The work provides a novel multi-output converter fed from a
single source. However, the converter is mainly suited for low-power EV charging at lower
output voltage levels (such as a 48 V Li-ion battery).
Figure 3. Converter proposed in Ref. [83] for EV applications.
Chen et al. [84] have considered the case of simultaneous charging as a constrained
optimization problem to minimize simultaneous charging time. The block diagram of
the solution proposed by the authors is shown in Figure 4. The authors have considered
four objectives: convergence of State-Of-Charge (SOC) deviation, minimization of loss
due to internal resistance, equating the SOCs of different batteries, and minimization
of simultaneous charging time. The constraints are the SOC limitation of the batteries,
the limitation on charging current, and the terminal voltage of the batteries. The authors
have provided the battery pack model and developed the optimal charging strategy where
trade-offs are made between the objective functions since it is impossible to optimize all
objectives simultaneously to obtain a single solution to such problems. To tackle this
problem, the authors have used the Adaptive Momentum-Based Steepest Descent (AMSD)
algorithm and performed a simulation on the PLECS software platform with four battery
packs and four Constant Voltage Current Sources (CVCS). For simulation, 3.1 Ah batteries
have been considered, and the rating of the CVCS is 5 V. The simulation results have shown
that the equilibrium time is reduced by more than 50% compared to the quasi-sliding
mode control. In addition, the convergence of the charging error is shown to be faster with
the proposed algorithm. Experimental validation of the proposed method is provided by
charging NCR 18650 3.1 Ah lithium-ion batteries (considered equivalent to battery packs).
The experimental results have been used to show the superiority of the proposed algorithm
over the battery-assisted charging system. The simultaneous charging time was observed to
be minimum with the proposed algorithm, and the proposed system is said to prevent the
Sustainability 2023, 15, 2813
11 of 71
over-charge and over-discharge of the battery. The work deals with simultaneous charging
as an optimization problem focusing on reducing simultaneous charging time. However,
the proposed system will have an increased component count for high-power EV battery
charging applications, which may lead to lower reliability. If a diode rectifier is used for
AC to DC conversion, there will be no control over the DC link side voltage when the input
voltage changes. If a fully controlled rectifier is employed, the control becomes complex.
With a full-bridge or its derived DC-DC converters, higher output voltage and power can be
obtained, and it will be suitable for higher voltage battery charging, which will also involve
reduced current, thereby reducing the losses in the secondary. Since the DC-DC converters
can be designed based on the requirement of the charging station, the architecture proposed
by the authors is very well suited for the simultaneous charging of vehicles with batteries
of different voltage and power levels.
Figure 4. Block diagram proposed in Ref. [84] for simultaneous battery charging.
Aziz and Oda [13,85] have evaluated a battery-assisted quick-charging strategy for
the simultaneous charging of EVs to reduce the burden on the grid. The proposed system
is represented in Figure 5. The control server is considered to control the energy flow
into all the parts of the community considering the related economics. The authors have
considered the CHAdeMO standard for testing the charging of the quick chargers and the
Li-ion battery for different seasonal conditions. A 50 kW inverter with an input voltage of
450 V and output RMS voltage of 200 V is considered, with the DC-DC converter feeding
a 400 V battery with a maximum current of 150 A. The 64.2 kWh battery with a nominal
voltage of 364.8 V and a maximum charging voltage of 393.6 V is considered. The 50 kW
chargers are taken to have a variable DC output voltage of 50 to 500 V and a maximum
output current of 125 A. The authors have concluded that the proposed battery-assisted
system facilitates an increased charging rate and shorter charging time compared with the
conventional system without battery support. This work describes the system for battery
charging, and can be used for charging batteries at higher power and voltage. Better control
is possible for the AC-DC and DC-DC converters to regulate the DC bus voltage and the
charger output voltage. With full-bridge or dual active bridge (DAB) converter variants,
higher power and voltage charging can be achieved at high efficiency. However, lower
reliability could affect the system, and the control will be relatively complex.
Sustainability 2023, 15, 2813
12 of 71
Figure 5. Simultaneous charging scheme proposed in Ref. [13].
Okon et al. [86] have described a CLLC-based charging circuit, shown in Figure 6,
to charge a Li-ion battery using the CC-CV charging method. The operation is done in two
stages, with the first stage providing control signals in progression and the second stage
controlling the battery’s CC-CV charging. The authors have suggested creating a DC bus
to connect multiple converters in parallel to enable multiple-battery charging. The authors
have proposed a modified combined control system to control the proposed converter.
The authors have provided simulation results obtained from PLECS software to validate
the single battery charging. The authors have claimed that the system can be used for
multiple-battery charging using the DC bus described above. Practical implementation has
not been provided in this work for simultaneous battery charging. The proposed system
is very well suited for high-power, high-efficiency battery charging, even with batteries
with voltages greater than 400 V. On the other hand, the implementation of simultaneous
battery charging will need a higher component count, thereby leading to an increase in the
size and weight of the system.
Figure 6. Converter proposed in Ref. [86] for battery charging.
Li et al. [87] have used two dual active bridge (DAB) converters with a rectifier and
DC voltage divider to charge three Li-ion batteries, as shown in Figure 7. The proposed converter is a two-stage circuit. It uses a single-phase AC-DC converter, a DC voltage divider
(using split capacitors with a neutral inductor and half-bridge configuration), and two parallel DAB converters. Two batteries are connected across the two split capacitors, and the
DAB converter feeds the third battery. Correspondingly, control strategies are provided for
the rectifier, the neutral leg circuit, and the two DAB converters. The rectifier is controlled
using a double loop controller with a PI controller and a PR controller to obtain a constant
DC voltage at the common node of the capacitors C1 and C2 . The neutral leg is controlled
Sustainability 2023, 15, 2813
13 of 71
using another double-loop controller with two PI controllers. The DAB converters are
controlled using the concept of phase shift and feedback compensation. The circuit is fed
from a 400 V, 50 Hz grid to obtain a DC link voltage of 560 V. A resistive load of 60 Ω is used
instead of a battery for testing. Simulation and test results have been provided to validate
the proposed system, considering a switching frequency of 4 kHz. The topologies proposed
in work are well suited for high-power, high-voltage battery charging. The trade-off is
with the higher number of components. The control is very flexible, specifically with
DC-link voltage control. The two batteries that feed at the output terminals of the DC
voltage divider have a common node. It would be preferable to isolate these two batteries
to prevent current flow from one battery to the other.
Figure 7. Converter proposed in Ref. [87] for multiple battery charging.
Ramanathan et al. [88] have proposed a PV- and grid-fed modified impedance source
inverter (ZSI) with transformers and full bridge rectifiers to charge two batteries simultaneously. The block diagram of the proposed circuit is shown in Figure 8. The maximum
shoot-through duty ratio of the ZSI is calculated. Then the inductor and capacitor values
are obtained. The system is configured to operate in four modes. Power flows from the PV
to the grid in the first mode. The PV system charges the battery via the HFT in the second
mode. In the third mode, the battery transfers energy to the grid through the inverter; in
the fourth mode, the grid is used to charge the battery. The authors have considered the
switching frequency of 20 kHz for the ZSI and 40 kHz for the half-bridge converter. A battery power of 200 W is considered. The authors have provided the simulation results from
MATLAB/Simulink to show the simultaneous charging of the two equal-rated batteries.
The system can be used up to a medium power level since it is fed from a single-phase
supply. With the use of controlled rectifiers as proposed by the authors, it is possible to
control the battery charging current for each battery. The system is not well suited for
higher voltage batteries.
Graw and Zimmermann [89] have proposed a boost-flyback converter combination
to charge battery stacks separately from a renewable energy source. The work aims to
simultaneously charge both a high voltage (HV) battery and a low voltage battery from a
single source. The authors have explained the modes of operation of the proposed converter
with the corresponding timing diagrams and the design of the flyback transformer. Overcharging the LV battery is prevented by redirecting the power to the HV battery, leading to
simpler control. The effect of the capacitance between the adjacent transformer windings
has been evaluated, and the effect on efficiency is noted. The authors have also considered
the effect of adding an LC snubber as a method to reduce the MOSFET drain peak voltage.
The proposed system is a combination of an isolated and a non-isolated converter. It can be
used only for power levels below 100 W and is unsuitable for fast charging. The system
has common ground between the secondary and primary due to the combination of the
non-isolated and isolated converters. The circulating current from the HV battery to the LV
battery is possible due to this. It could also lead to EMI issues when etched on a PCB.
Sustainability 2023, 15, 2813
14 of 71
Figure 8. Converter proposed in Ref. [88] for multiple battery charging.
Sun et al. [90] have proposed an isolated multi-port DC charging station as a part of
a smart grid system. The block diagram proposed by the authors is shown in Figure 9.
The HV stage employs a cascaded H-bridge converter, and the isolation stage uses multiple
cascaded DAB converters. The DC side of all the DABs is connected to the DC bus, which
charges multiple batteries simultaneously using separate chargers. Dual-loop control is
used for the rectifier stage, while the voltage control strategy with a current balancing
function controls the DAB stage. Both Grid-to-Vehicle (G2V) and Vehicle-to-Grid (V2G) are
employed in the charging module. The authors have provided the simulation results for the
proposed system. The simulation system comprises 24 rectifier modules, each connected in
series with a DAB operating at a switching frequency of 10 kHz. The three batteries are at
voltage levels of 300 V with capacities of 15 kWh, 15 kWh, and 20 kWh, with different SOCs.
The objective of the work is to validate the proposed topology, which has been done using
simulation. The hardware implementation of the proposed system has not been provided
in the work. The system is very well suited for high voltage batteries as described by the
authors. However, the intermediate conversion stage has a very high number of devices,
which may lead to poor reliability in real-time implementation.
Figure 9. System proposed in Ref. [90] for multiple battery charging.
In Ref. [91], the authors Yesheswini et al. have proposed a DC-DC converter-based
system for charging multiple EVs simultaneously. The system is a solar-fed battery charging
system with a boost converter, backup battery, bidirectional converter, and buck converter to
charge EV batteries. The system proposed by the authors is shown in Figure 10. The authors
have chosen a 36 V, 28 Ah lead-acid battery for simulation. The solar PV source is chosen
to be composed of two 180 W panels connected in parallel. The solar-fed converter is
Sustainability 2023, 15, 2813
15 of 71
controlled using the perturb and observe (P&O) maximum power point tracking (MPPT)
algorithm, while the battery charging is simulated using the CC-CV charging algorithm.
A bidirectional converter connects the station battery with the DC grid, facilitating charging
and discharging based on the requirement. The station battery will support the charging
whenever the solar PV supplies low power (lower than the EV requirement). The station
battery will be charged when the solar power is in excess. The station battery is also used
to maintain a constant DC bus voltage. The hardware prototype is implemented using a
lead-acid battery with the buck converter (with IRFP460 MOSFET) controlled using an STM
controller. The switching frequency is chosen to be 4 kHz. The authors have analyzed the
performance of the converters for EV battery charging. As a future scope, the authors have
suggested optimizing sources such as solar, wind, and biofuels, using various controllers
and machine learning methods to optimize the load on the charging station. The proposed
system has used solar and battery energy storage systems as the two sources. Therefore,
the current rating will be limited, especially considering the intermittency of solar energy.
Since the system does not involve a grid connection, so that power will be limited, it
is hence not very suitable for fast charging.
Figure 10. System proposed in Ref. [91] for simultaneous charging of multiple batteries.
Mishra et al. [92] have proposed a grid and solar-based hybrid charger that is controlled
using adaptive supervisory control (ASC). The topology used by the authors is shown
in Figure 11. The proposed topology uses two isolated converter stages (one DAB and
one resonant converter), one working as a slow charger and the other as a fast charger.
In addition, a PV-fed DC-DC converter with a battery storage unit (BSU) is considered as
shown in the figure. The grid-side converter (GSC) is an active bridge AC to DC converter
used to regulate the DC link voltage and to ensure PFC. It is operated at a frequency
of 10 kHz. A PR controller-based closed loop is used to control the GSC. In the PV-fed
converter, the incremental conductance algorithm is implemented for MPPT. The BSU is the
battery pack with a buck-boost bidirectional converter. The DAB is controlled to have zero
voltage switching (ZVS) with bidirectional power transfer. When only the grid supplies
power, the slow charger is utilized to charge the EV batteries, while slow and fast chargers
are used when both the grid and PV supply power to ensure minimal stress on the grid.
The proposed system is very suitable for simultaneous fast and slow charging. Battery
swapping function is added along with a storage unit that can be used as a backup source.
The proposed system can be used to charge high-power, high-voltage batteries. However,
a higher number of components could pose reliability issues.
Sustainability 2023, 15, 2813
16 of 71
Figure 11. Topology of the converter proposed in Ref. [92] for multi-battery charging.
In Ref. [93], Vu et al. have proposed using a multi-input capacitive coupled wireless
charger for EV battery charging. The block diagram is shown in Figure 12. Capacitive
power transfer (CPT) topologies have high leakage capacitance, which is usually compensated for by using resonant inductors. However, the authors have pointed out that this
method is unsuitable for high-power applications, and that LCL topology is more suitable.
The authors have provided a detailed analysis of the CPT system. A DC input voltage of
250 V is chosen with a variable output of 120 to 270 V. The switching frequency of 1 MHz is
chosen for the system. Cross-coupling is avoided by selecting a distance of 150 cm between
both primary plates. Simulation results are provided for the proposed system for equal
and unequal load conditions of the two converters. With the wireless power transfer of the
capacitive type, the main issue is the distance of separation between the transmitter and
the receiver. In addition, the efficiency of wireless transfer is less. With the use of diode
rectifiers as proposed by the authors, the receiver side circuit also needs to be controlled
from the transmitter side, making the control less flexible.
Figure 12. Capacitive coupled wireless charger for multi-battery charging in Ref. [93].
Fan et al. [94] have proposed a reflex charging technique to charge two batteries
simultaneously. The authors have used ‘N’ sets of interleaved buck-boost converters to
obtain reflex charging. The modes of operation and the corresponding equivalent circuits
have been explained. Phase shift control has been employed to control the converters in this
work. This is also said to prevent over-charging or over-discharging. In addition, a battery
charge-discharge management system has been proposed to maintain the battery voltage.
The authors have validated the system by charging two 12 V batteries from 17 to 24 V input.
The experiment for N = 1, 2, 3, and 4 sets have been provided. Both positive and negative
battery currents have been obtained, validating the two states of operation based on the
power switch operation. Since the proposed system uses buck-boost converters, isolation is
not provided and is only suitable for low-power charging at lower voltages.
Vu et al. [95] have proposed a multi-output inductive wireless power transfer (WPT)
charger for simultaneously charging two batteries. A standard full-bridge voltage source
inverter is used to convert DC input to AC, which is then passed to two compensation
networks of inductors and capacitors with high-frequency transformers, which feed two
Sustainability 2023, 15, 2813
17 of 71
diode rectifiers separately in cascade with synchronous buck converters (SBCs). The output
of the SBCs is used to supply the two batteries, which charge simultaneously. The circuit
diagram proposed by the authors is shown in Figure 13. The AC equivalent circuit of
the inductive power transfer system with its analysis has been provided to calculate the
coupling coefficient. The frequency is kept constant during the operation, and the secondary
side converter functions as the charger controller. The resonant tank circuit components
design has been provided for the proposed converter. A DC input voltage of 400 V is chosen
to obtain a variable output of 250 to 400 V. A switching frequency of 68 kHz was used
for the simulation. Unequal loads have been used to simulate the system to validate the
circuit operation and simultaneous charging of both batteries. With WPT, the two primary
challenges are the separation distance between the two coils and the lower efficiency of
power transfer. The proposed system can be used to charge batteries of unequal ratings.
Synchronous buck converter provides better control possibility but limits the power. Hence
the system is not very suitable for fast charging.
Figure 13. Circuit topology proposed in Ref. [95] to simultaneously charge two batteries.
Chakraborty et al. [96] have proposed a pulse charging-based Li-ion battery charging
converter. The main power supply working as a current source (equivalent) is connected
in a predefined order, in parallel, with different batteries using a set of switches. At any
given time, one battery is being charged. This is decided by turning the selector switch (S1 ,
S2 or S3 ) ON and OFF to connect or disconnect the respective battery. If all batteries are
charged, a reset switch is used for freewheeling the inductor energy. The controlled current
source uses the forward converter topology with a core reset. A number of MOSFETs and
diodes are used to implement the switch selectors. For ‘n’ battery cells connected in series
in each switch selector block, 2n+2 MOSFETs and 2n diodes are required. Thus, if three
battery cells are connected in series for one switch block, say, S1 , then eight MOSFETs and
six diodes are required for making that selector switch. Charge balancing is implemented
along with charging and discharging using the pulse charging technique. A two-output
case is simulated using the parameters of the VALENCE 444 Li-polymer cell. The converter
is fed from a 170 V AC supply with the maximum duty cycle limited to 44%. The switching
frequency selection is flexible, and average current control is used to control the inductor
current. The proposed system uses a higher number of components, especially when
multiple batteries are to be charged simultaneously. In addition, the power limitation leads
to slow charging and is not suited for charging vehicles at higher voltages.
In continuation with the previous work [96], Chakraborty and Mohan [97] have extended the work to a wider range of medium-power AC-DC applications. The suitability
of CCM and DCM has been evaluated. Negative voltage feedback reduces the bus voltage
stress, with a trade-off that includes an increase in the input current distortion. The advantages of the voltage doubler rectifier topology are provided in detail, and the corresponding
simulation waveforms have been provided. The input current THD is around 60% and a
simulation efficiency of 84% has been obtained with an input of 110 V AC.
Table 1 gives a comparison of the various converters described in Section 4.
Sustainability 2023, 15, 2813
18 of 71
Table 1. Comparison of the various converters considered in Section 4.
Ref. No.
Converters Used
Source(s)
Number of
Active Switches
Number
of Diodes
Number of
Inductors
Number of
Capacitors
Isolated or
Non-Isolated
NBS+
Control Methodology
& Technique
Control
Complexity
[83]
1ph diode rectifier with
Hybrid Converter
AC Grid
1
6
1
3
Both
1
PI controller
Simple
[86]
3ph rectifier with CLLC
AC Grid
14
0
6
3
Isolated
1
PI controllers;
Combined Control with
CCCV
Complex
[87]
1ph rectifier, DC
voltage divider with
DAB
AC Grid
22
0
2
6
Isolated
2
PI and PR Controllers
(AC-DC) and PI
controllers (neutral
leg and DAB)
Complex
[88]
ZSI, 1ph inverter with
1ph rectifiers
PV and AC
Grid
17
1
5
5
Isolated
2
NM*
Moderate
[89]
Boost and flyback
converters
PV
1
3
0
1
Isolated, but
common ground
1
NM*
Simple
[90]
1ph inverter with DABs
10 kV AC Grid
12n
0
n
n
Isolated
n
PI controllers
Complex
[91]
Buck, boost and
bidirectional DC-DC
converters
PV and BESS
6
4
5
5
Non-Isolated
3
PID and MMPT with
CCCV
Moderate
[92]
3ph Boost rectifier,
boost, synchronous
buck, DAB and LLC
PV and AC
Grid
21
6
9
7
Isolated
2
PR, phase shift,
and MMPT with CCCV
Complex
[93]
Full bridge,
double-sided LCL
DC
4
8
8
6
Capacitive
Coupling
2
Constant frequency
controller with
charging controller
Moderate
[94]
2 sets of Interleaved
buck-boost
DC
2
12
2
4
Non-Isolated
2
Reflex charging with
phase shift, charge and
discharge management
[95]
Full bridge, LCC-LCC
with buck converters
DC
8
8
6
12
Wireless (IPT)
2
Constant frequency
controller with
charging controller
NM* = Not Mentioned; NBS+ = Number of batteries that can be charged simultaneously.
Moderate
Complex
Sustainability 2023, 15, 2813
19 of 71
5. Potential Extension of Existing Isolated DC-DC Converters for Simultaneous
Charging of Multiple EV Batteries
Simultaneous charging of multiple EV batteries requires multi-output DC-DC converters. For high-power applications such as battery charging, isolated converters are
preferred. While several isolated DC-DC converter topologies exist in the literature, all
the topologies cannot be used to get the same performance characteristics for EV battery
charging applications. This section provides some potential isolated DC-DC converter
configurations for simultaneously charging multiple Li-ion batteries. Galvanic isolation
is required between the different circuits of the charger to ensure safe and reliable operation [42]. A classification of the most common Single-Input Multi-Output (SIMO) Isolated
DC-DC converters is shown in Figure 14.
Figure 14. Classification of common SIMO DC-DC converters.
The choice of a DC-DC converter for EV battery charging depends on the following
factors [98]:
•
•
•
•
•
•
•
•
Power rating and power density;
Isolation requirement;
Charging type (AC/DC charging, slow/fast charging);
Current rating (affects the choice of cable);
Type of switch used (depends on the frequency of switching, which decides the
magnetic component size);
Number of outputs required;
Voltage at each of the output terminals;
Cost, size, and weight of the system.
The appropriate topology is to be chosen based on the trade-offs between the factors.
With the mass usage of EVs, infrastructure needs to be developed to facilitate charging.
The infrastructure in developed countries is better than that in developing nations. During infrastructure development, the land size, availability of power supply, the possibility of
extension for future needs, types of vehicle, charging station layout, cabling cost, distance,
incentives, and other factors need to be considered [99]. Location, cost, waiting time, charging time, and payment options are some of the most critical factors from the consumer’s
point of view [100]. The locations could include workplace parking areas, shopping malls,
and public parking structures [99]. The waiting time needs to be reduced in addition to the
reduction of charging time to attract consumers to a particular charging station. Presently
available infrastructure will significantly affect the further expansion and installation of
new chargers. In this scenario, many of the above-mentioned factors will be predefined
and can pose some limitations during expansion. Care has to be taken, especially regarding
the ratings of the components, protection system, and communication protocols during
expansion. The expansion needs to be planned, modeled, and programmed considering
the present infrastructure along with the pattern of transportation expansion [101]. It
can be taken up as an optimization problem with the existing infrastructure limits as the
constraints [102].
Sustainability 2023, 15, 2813
20 of 71
Sections 5.1–5.5 describe the various converters that can simultaneously charge two or
more batteries. Some converters are limited to low-power applications, while others can
be used for high-power applications. All descriptions related to this are provided in this
section. In addition, power loss estimation (assuming ideal passive elements) is described
for the converters that are more commonly used for battery charging.
5.1. Single-Switch Isolated DC-DC Converters
5.1.1. Single-Switch Multi-Output Flyback Converter
The flyback converter is an isolated buck-boost topology that can be modified to
implement multi-battery charging. The basic flyback converter uses one active switch
(MOSFET/IGBT), diode, capacitor, and transformer. Theoretically, ‘k’ number of batteries
can be charged using a flyback converter if a transformer with ‘k’ number of isolated
secondaries is used [103], as shown in Figure 15. Getting multiple outputs does not involve
much difference in complexity, space, or cost since each additional output requires one
HFT, one diode, and one capacitor. The converter gives relatively good efficiency at a low
cost. The equation for each of the outputs (in CCM) is given by Equation (2), where δ is the
duty cycle. However, this topology has one major limitation. The typical power limitation
of a flyback converter is approximately 60 W [104] (less than 100 W, in general). Hence,
the converter could charge and discharge the low-voltage auxiliary battery of the EV or lowpower vehicles such as e-bicycles or low-power two-wheelers. The other disadvantages
include high stress on the switch, relatively higher conduction losses, and the requirement
of a snubber. The choice of switching frequency and ratings of the various components
depends on the type of source available and the rating of the battery, which will decide
the duty cycle value. The design of the multi-secondary flyback transformer is a critical
design criterion that needs to be considered. The magnetizing inductance value needs to
be greater than a minimum value (Lm,min ) to ensure CCM. This is given by Equation (3).
To ensure CCM operation, the inductance value should be greater than Lm,min value. Since
the inductance chosen should be greater than Lm,min , it could lead to design issues if too
high or too low inductance is chosen. Hence, the required inductance can be calculated
using Equation (4) and then verified if it is greater than Lm,min or not.
vOk =
nsk
δ
·
·V
n p 1 − δ in
Lm,min = k ×
Lm = k ×
(2)
(1 − δ)2 n2sk
·
2 × f sw n2p
(3)
Vin × δ
∆i Lm × f sw
(4)
The capacitance of the capacitor across the load can be calculated using Equation (5),
where R is the equivalent load resistance.
Ck =
vO × δ
∆vO × R × f sw
(5)
A capacitor (Cin ) can be connected across the source if the input is noisy. Such a
capacitance can be calculated using Equation (6).
Cin =
iin × to f f
2 × ∆vcin
(6)
The semiconductor utilization factor (SUF ) is given by Equation (7). A plot of SUF for
duty cycles from 0 to 1 is shown in Figure 16. It can be seen from the graph that the SUF
Sustainability 2023, 15, 2813
21 of 71
is the highest when the converter is operated at a duty cycle of 50% [105]. Therefore, it is
better to operate the flyback converter at or around the duty cycle values of 50%.
SUF = √
δ − δ2
√
δ+ 1−δ
(7)
Figure 15. Single-switch flyback converter with ‘k’ number of secondary windings.
Figure 16. Semiconductor utilization factor of the flyback converter.
The conduction losses in the MOSFET and each diode can be estimated using
Equations (8) and (9), respectively [105].
(
!)
2
iin
∆i2Lm
+δ·
(8)
Pcond,Q = Vth · Iin + r DS,ON ·
δ
3

 iin n p
Pcond,DK = Vth · IO + rONDk (1 − δ)
δnsk
2
+
∆i Lm ·
3
np
nsk
2 


(9)
where r DS,ON is the on-state resistance of the MOSFET, rONDk is the on-state resistance of the
diode, Iin is the average value of the input current, and Vth is the threshold voltage of the
corresponding device (MOSFET or diode). Based on the turn-off energy loss characteristic
(wo f f ,DK , xo f f ,DK and yo f f ,DK ) provided by the manufacturer for the 100FIT test voltage
(V100FIT ; it is the voltage across the device such that there are 100 failures within 109 h of
Sustainability 2023, 15, 2813
22 of 71
operation [106]; FIT is Failures In Time), we can estimate the average value of the switching
loss in the diode [105] using Equation (10).
"
VO
np 2
Iin
δ + ∆vo
Psw,DK =
· wo f f ,DK ·
− ∆i Lm
Tsw V100FIT
δ
nsk
(10)
np
Iin
− ∆i Lm
+ xo f f ,DK ·
+ yo f f ,DK
δ
nsk
Similarly, the average value of the switching loss [105] in the MOSFET (Psw,Q ) is
estimated using Equation (11).
"
n
Vin
2
− ∆vo nskp
I
I
(1− δ )
Psw,Q =
· wo f f ,SW · in + ∆i Lm
+ xo f f ,SW · in + ∆i Lm + yo f f ,SW
Tsw V100FIT
δ
δ
(11)
"
n
Vin
p
2
+
∆v
o
I
I
nsk
(1− δ )
· won,SW · in − ∆i Lm
+
+ xon,SW · in − ∆i Lm + yon,SW
Tsw V100FIT
δ
δ
where wo f f ,SW , xo f f ,SW and yo f f ,SW are the turn-off energy loss characteristic provided by
the manufacturer for V100FIT , and won,SW , xon,SW and yon,SW are the turn-on energy loss
characteristic from the MOSFET data-sheet.
5.1.2. Single-Switch Multi-Output Forward Converter
The other popular isolated converter is the forward converter. It is an isolated version
of the popular buck converter. Compared to a flyback converter, it gives a faster transient
response, better efficiency, and low output ripple. However, the cost is higher when compared to that of a flyback converter. It is generally used whenever a high current output is
required [107]. The major limitation of the forward converter is the duty cycle. Its maximum duty cycle must be about 50%. The forward converter uses a tertiary winding (nter )
and a diode to enable core reset. The high-voltage winding usually provides the core reset.
The time taken to reset is inversely proportional to the winding voltage. Figure 17 shows
the implementation of a forward converter with multiple winding to enable simultaneous
charging of batteries. Similar to a flyback converter, the forward converter also has a power
limitation of 100 W [103]. The output of the kth winding is related to the input as given
by Equation (12). The magnetizing inductance (Lm ), output inductance (Lk ) and output
capacitance (Ck ) can be found using Equations (13), (14) and (15), respectively. If the input
is noisy, a capacitor (Cin ) with a value calculated using Equation (16) can be connected
across the source.
vOk =
nsk
· δ · Vin
np
Lm = k ×
(13)
VO × (1 − δ)
f sw × ∆i Lk
(14)
VO × (1 − δ)
2
8 × ∆VO × Lk × f sw
(15)
( Iin + ∆i Lm ) · To f f
2 · ∆vCin
(16)
Lk =
Ck =
Vin × δ
f sw × ∆i Lm
(12)
Cin =
Sustainability 2023, 15, 2813
23 of 71
Figure 17. Single-switch forward converter with ‘k’ number of secondary windings.
The semiconductor utilization factor (SUF ) for the forward converter [105] is given by
Equation (17). The choice of transformer turns ratio (nsk : n p ) plays a vital role in obtaining
optimal semiconductor utilization factor since the SUF depends on δmax , which depends
on the transformer turns ratio. Optimal SUF leads to optimal converter design. A plot of
duty cycle (δ) vs. SUF can be plotted to obtain the corresponding optimal turns ratio of the
transformer. An example of such a plot is shown in Figure 18 for various δmax between 0.2
and 0.9.
(1 − δmax ) · δ
√
(17)
SUF =
√
(1 + δmax ) · δ + (1 − δmax ) · 1 − δ
Figure 18. Semiconductor utilization factor of the forward converter.
Pcond,Q
The average value of conduction loss of the switch Q, and three diodes D0 , D1k , and
D2k are given by Equations (18), (19), (20), and (21), respectively [105].

h
i 
nsk 2 

2


∆i
+
∆i
Lm
L1
np
n
= Vth · ( Iin + ∆i Lm · δ) + r DS,ON · δ ·
IO · sk + ∆i LM +
(18)


np
3


np δ
·
nter 3
!
2
∆i
= Vth · IO · δ + r DS,ON · δ ∆IO2 + Lk
3
Pcond,D0 = Vth · ∆i Lm · δ + r DS,ON · (2 · ∆i Lm )2 ·
Pcond,D1k
(19)
(20)
Sustainability 2023, 15, 2813
24 of 71
Pcond,D2k = Vth · IO · (1 − δ) + r DS,ON · (1 − δ)
2
∆IO
∆i2
+ Lk
3
!
(21)
Before calculating the switching losses, we note that the switching loss in diode D0 is
negligible since it is turned OFF when the current reaches zero. Therefore, switching losses
must be calculated for switch Q and diodes D1k and D2k . The corresponding switching
losses are estimated using Equations (22), (23) and (24), respectively [105]. The W, x and y
coefficients have the same meaning as described before.
"
n
2
1 + nterp · (Vin − ∆vCin )
n
· wo f f ,SW ( IO + ∆i Lk ) sk + 2 · ∆i Lm
Psw,Q =
Tsw V100FIT
np
n
+ xo f f ,SW ( IO + ∆i Lk ) sk + 2 · ∆i Lm + yo f f ,SW
np
(22)
"
nter
1 + n p · (Vin + ∆vCin )
nsk 2
· won,SW ( IO − ∆i Lk )
+
Tsw V100FIT
np
n
+ xon,SW ( IO − ∆i Lk ) sk + yon,SW
np
Psw,D1k =
Psw,D2k =
nsk
nter
i
· (Vin − ∆vCin ) h
· wo f f ,D1k ( IO + ∆i Lk )2 + xo f f ,D1k ( IO + ∆i Lk ) + yo f f ,D1k
Tsw V100FIT
(23)
nsk
nter
i
· (Vin + ∆vCin ) h
· wo f f ,D2k ( IO − ∆i Lk )2 + xo f f ,D2k ( IO − ∆i Lk ) + yo f f ,D2k
Tsw V100FIT
(24)
5.1.3. Single-Switch Multi-Output Isolated SEPIC
The Single-Ended Primary Inductor Converter (SEPIC) is a popular buck-boostderived converter. A transformer replaces the inductor of the non-isolated SEPIC with
proper dot polarity to obtain the isolated version of the converter. This converter can be
modified to charge multiple batteries simultaneously, as shown in Figure 19. It provides
an output voltage described by Equation (25). The advantage of isolated SEPIC is that the
input current has low ripple and hence is suited very well when the input to the charger is
provided from solar PV panels. Adding multiple outputs does not significantly increase
cost since each additional stage requires only an HFT, a diode, and a capacitor. The input inductance, input capacitance, magnetizing inductance, and output capacitance are
calculated using Equations (26), (27), (28), and (29), respectively [105].
δ
nsk
·
·V
n p 1 − δ in
(25)
Vin
·δ
2 × ∆i L1 × f sw
(26)
IO
n
· sk · δ
2 × ∆vC1 × f sw n p
(27)
Vin
·δ
2 × ∆i Lm × f sw
(28)
vOk =
L1 = k ×
C1 = k ×
Lm = k ×
Ck+1 =
n
IO
· sk · δ
2 × ∆vC1 × f sw n p
(29)
Sustainability 2023, 15, 2813
25 of 71
Figure 19. Single-switch isolated SEPIC with ‘k’ number of secondary windings.
The semiconductor utilization factor of an isolated SEPIC is the same as that of a
flyback converter and is given by Equation (7). The conduction losses in the MOSFET (Q)
and diode (Dk ) can be estimated using Equations (30) and (31), respectively [105].
"
#
2
Iin
(∆i L1 + ∆i Lm )2
(30)
Pcond,Q = Vth Iin + r DSON
+δ
δ
3
(
Pcond,DK = Vth,DK IO + rONDk (1 − δ)
Iin
δ
np
nsk
2
+
np
nsk
2 "
(∆i L1 + ∆i Lm2 )2
3
#)
(31)
where rONDk is the ON-state resistance of the diode and Vth,DK is its threshold voltage.
The average switching loss Psw,Dk in diode DK [105] is given by Equation (32).
Psw,Dk =
Vin
(1− δ )
+ ∆vC1 ·
nsk
np
Tsw · V100FIT
+ ∆vCk
· (W + X + Y )
(32)
where,
2
n
W = wo f f ,Dk · ( Iin − ∆i L1 − ∆i Lm ) · n p + IO
sk
np
X = xo f f ,D1 · ( Iin − ∆i L1 − ∆i Lm ) · n + IO
sk
Y = zo f f ,D1
The average switching loss of the switch Q can similarly be found [105] using
Equation (33).
np
Vin
−
∆v
−
∆v
C1
C (k +1) nsk
(1− δ )
Psw,SW =
(U + V + W )
Tsw · V100FIT
(33)
np
Vin
+
∆v
+
∆v
C1
C
(
k
+
1
)
nsk
(1− δ )
+
[X + Y + Z]
Tsw · V100FIT
where,
h
i2
U = wo f f ,SW · Iin + ∆i L1 + ∆i Lm + IO · nnskp
V = xo f f ,SW · Iin + ∆i L1 + ∆i Lm + IO · nnskp
W = yo f f ,SW
X = won,SW · Iin − ∆i L1 − ∆i Lm + IO ·
nsk
np
2
Sustainability 2023, 15, 2813
26 of 71
Y = xon,SW · Iin − ∆i L1 − ∆i Lm + IO ·
nsk
np
2
Z = yon,Sw
5.1.4. Single-Switch Multi-Output Isolated Ćuk Converter
The Ćuk converter is another popular buck-boost-derived converter. It was invented
and patented by Slobodan M. Ćuk and Robert D. Middlebrook [108]. The non-isolated
Ćuk converter can be modified to obtain the isolated version of the converter. This can
further be modified to charge multiple batteries simultaneously, as shown in Figure 20.
It provides an output voltage that is described by Equation (25), and the design [105]
of the inductance and capacitance of the inductors and capacitors are calculated using
Equations (34) through (39).
L1 = k ×
Vin
·δ
2 × ∆i L1 × f sw
(34)
Lm = k ×
Vin
·δ
2 × ∆i Lm × f sw
(35)
L k +1 =
Vin
2 × ∆i L(k+1) × f sw
·δ
(36)
IO
n
· sk · δ
2 × ∆vC1 × f sw n p
(37)
Ckb =
IO
·δ
2 × ∆vC1 × f sw
(38)
Ck+1 =
∆i L2
8 × ∆vC(k+1) × f sw
(39)
C1a = k ×
Figure 20. Single-switch isolated Ćuk converter with ‘k’ number of secondary windings.
Considering the average value input current represented by Iin , the average conduction
loss in the MOSFET Q [105] is given estimated using Equation (40).
"
Pcond,SW = Vth Iin + r DS,ON
2
Iin
δ
+δ
∆i L1 + ∆i Lm + ∆i L(k+1) ·
3
nsk
np
2 


The average conduction loss in the diode Dk [105] is given by Equation (41).
(40)
Sustainability 2023, 15, 2813
27 of 71
Pcond,Dk = Vth,DK IO + rONDk (1 − δ)


 I


in n p
δ nsk
2

n
p
 ∆i L1 + ∆i Lm ) nsk + ∆i L(k+1)
+
3
2  






The average switching loss in diode Dk [105] is estimated using Equation (42).
nsk
Vin
+
∆v
+
∆v
C1a n p
Ckb
(1− δ )
Psw,Dk =
(W + X + Y )
Tsw V100FIT
(41)
(42)
where,
2
n
W = wo f f ,Dk ( Iin − ∆i L1 − ∆i Lm ) n p + IO − ∆i L(K +1)
sk
n
X = xo f f ,Dk ( Iin − ∆i L1 − ∆i Lm ) n p + IO − ∆i L(k+1)
sk
Y = yo f f ,Dk
The average switching loss of of the switch Q [105] can similarly be estimated using
Equation (43).
np
Vin
−
∆v
−
∆v
·
C1a
Ckb
nsk
(1− δ )
· (U + V + W )
Psw,Q =
Tsw · V100FIT
(43)
n
Vin
+ ∆vC1a + ∆vCkb · nskp
(1− δ )
· [X + Y + Z]
+
Tsw · V100FIT
where,
i2
h
U = wo f f ,SW · ( Iin + ∆i L1 + ∆i Lm + ( IO + ∆i L2 )) nnskp
V = xo f f ,SW Iin + ∆i L1 + ∆i Lm + IO + ∆i L(k+1) nnskp
W = yo f f ,SW
2
X = won,SW Iin − ∆i L1 − ∆i Lm + IO − ∆i L(k+1) nnskp
2
Y = xon,SW Iin − ∆i L1 − ∆i Lm + IO − ∆i L(k+1) nnskp
Z = yon,SW
5.1.5. Single-Switch Multi-Output Isolated Zeta Converter
A Zeta converter is a modified buck-boost converter, and its isolated version uses one
transformer, one inductor, one capacitor, one active switch, and one diode. An isolated Zeta
converter is emerging as an attractive option for EV battery charging as it gives a very stable
response and requires minimal control. The isolated zeta converter modified to charge
multiple batteries is shown in Figure 21. The relationship between the output and input
voltages is given by Equation (25). The inductance Lm of the magnetizing inductance and
the inductor Lk is calculated using Equations (44) and (45), respectively. The capacitance of
the capacitors C1k and C2k are given by Equations (46) and (47), respectively.
Vin × δ
f sw × Lm
(44)
nsk Vin × δ
·
n p f sw × Lk
(45)
vO × δ
f sw × R × ∆vC1k
(46)
Lm = k ×
Lk =
C1k =
Sustainability 2023, 15, 2813
28 of 71
C2k =
vO × ( 1 − δ )
2 × L × ∆v
8 × f sw
k
C2k
(47)
Figure 21. Single-switch isolated zeta converter with ‘k’ number of secondary windings.
5.1.6. Single-Switch Multi-Output Isolated Buck (Fly-Buck) Converter
The single-switch fly-buck converter is shown in Figure 22 for multiple-battery charging. The converter has a capacitor C1 across which the first output can be taken (shown
across C1 as a small battery of voltage Vaux ). This can be used to charge the low-voltage auxiliary battery. The secondary windings can charge other batteries based on the transformer
turns ratio. The equations for output voltages are given by Equations (48) and (49).
v aux = Vin × δ
n
vOk = sk × v aux
np
(48)
(49)
Figure 22. Single-switch isolated fly-buck converter with ‘k’ number of secondary windings.
5.2. Two-Switch Isolated DC-DC Converters
5.2.1. Two-Switch Flyback Converter
The two-switch flyback converter is used to overcome the typical problems of a singleswitch flyback converter. The topology can be used for multi-battery charging, as shown
in Figure 23. It can also be modified into a quasi-resonant flyback converter (a DCM
Flyback having a valley switching turn on). The output voltage equation is the same as the
flyback converter (Equation (2)). The same pulse is provided to both the switches of the
converter. The two-switch flyback converter has better efficiency since the clamping losses
are removed by recycling the leakage energy to the input side through the two diodes. This
also implies reduced thermal stress on the switch. The overall voltage stress is divided
between the two MOSFETs [109]. In this topology, the maximum duty cycle limitation is
0.5 when operating in continuous conduction mode [109].
Sustainability 2023, 15, 2813
29 of 71
Figure 23. Two-switch flyback converter feeding ‘k’ number of batteries.
5.2.2. Two-Switch Forward Converter
The two-switch forward converter feeding multiple batteries is shown in Figure 24.
The output voltage is given by Equation (50). It is usually used between 150 W and
750 W [110]. The circuit has multiple advantages, including no body-diode conduction,
no snubber requirement, and the possibility of handling multiple isolated outputs. It
does not require any dead-time and does not suffer from the problem of a potential shootthrough. However, the frequency of operation is limited due to its inability to switch at zero
voltage [110] and requires a relatively larger transformer since it is a single-ended converter.
The minimum value of inductance Lk is given by Equation (51), and the maximum value of
the duty cycle is limited to 50% [111].
vOk =
nsk
× δ × Vin
np
Lk,min = k ×
VO × (1 − δmin )
2 × f sw × IO,B
(50)
(51)
where IO,B is the output current at the CCM/DCM boundary, given by,
IO,B =
VO
R Load,max
(52)
Figure 24. Two-switch forward converter feeding ‘k’ number of batteries.
The semiconductor utilization factor (SUF ) for the two-switch forward converter is
given by Equation (53). The plot of duty cycle vs. SUF for this converter is presented in
Figure 25.
δ
√
SUF =
(53)
√
(3 · δ ) + 1 − δ
Sustainability 2023, 15, 2813
30 of 71
Figure 25. Semiconductor utilization factor of the two-switch forward converter.
The conduction loss in the diode Dk is given by Equation (54) and that in switch Q is
given by Equation (55).
Pcond,Dk = Vth · ∆i Lm · δ + r DS,ON · (2 · ∆i Lm )2 ·
Pcond,Q = Vth · ( Iin + ∆i Lm · δ) + r DS,ON · δ ·





IO ·
nsk
+ ∆i LM
np
h
2
+
δ
3
(54)
i2

∆i Lm + ∆i L1 nnskp 

3
(55)


The average value of conduction losses in diodes D p1 and D p2 are estimated using
Equations (56) and (57), respectively.
!
2
∆i
2
(56)
Pcond,Dp1 = Vth · IO · δ + r DS,ON · δ ∆IO
+ Lk
3
Pcond,Dp2 = Vth · IO · (1 − δ) + r DS,ON · (1 − δ)
2
∆IO
∆i2
+ Lk
3
!
(57)
Similarly, the average switching losses can be estimated for the MOSFET, and diodes
D p1 and D p2 using Equations (58), (59), and (60), respectively.
"
2
n
(Vin − ∆vCin )
Psw,Q =
· wo f f ,SW ( IO + ∆i Lk ) sk + 2 · ∆i Lm
Tsw V100FIT
np
nsk
+ xo f f ,SW ( IO + ∆i Lk )
+ 2 · ∆i Lm + yo f f ,SW
(58)
np
"
#
(Vin + ∆vCin )
nsk 2
nsk
+
· won,SW ( IO − ∆i Lk )
+ xon,SW ( IO − ∆i Lk )
+ yon,SW
2 · Tsw V100FIT
np
np
Psw,Dp1 =
Psw,Dp2 =
nsk
np
i
· (Vin − ∆vCin ) h
· wo f f ,Dp1 ( IO + ∆i Lk )2 + xo f f ,Dp1 ( IO + ∆i Lk ) + yo f f ,Dp1
Tsw V100FIT
(59)
nsk
np
i
· (Vin + ∆vCin ) h
· wo f f ,Dp2 ( IO − ∆i Lk )2 + .xo f f ,Dp2 ( IO − ∆i Lk ) + yo f f ,Dp2
Tsw V100FIT
(60)
Sustainability 2023, 15, 2813
31 of 71
5.2.3. Push-Pull Converter
The push-pull topology is another isolated converter that can charge multiple batteries
simultaneously. Figure 26 shows the push-pull converter topology used to charge two
batteries. The output voltages of the push-pull converter feeding two batteries are given by
Equations (61) and (62).
ns1
× δ × Vin
n p1
n
= 2 × s2 × δ × Vin
n p1
vO1 = 2 ×
(61)
vO2
(62)
Figure 26. Push-pull converter feeding two batteries.
The semiconductor utilization factor (SUF ) for a push-pull converter is given by
Equation (63), and a plot of duty cycle vs. SUF is provided in Figure 27. From the equation
of SUF , it can be inferred that the SUF would be imaginary for duty cycles greater than
50%. Hence, δ ≤ 50% is the duty cycle limitation for a push-pull converter.
SUF =
(2 ·
√
δ) +
δ
p
1 − (2 · δ )
(63)
Figure 27. Semiconductor utilization factor of a push-pull converter as a function of duty cycle.
The average value of conduction losses in switches Q1 and Q2 are estimated using
Equations (64) and (65), respectively. Similarly, the average conduction losses in the diodes
on the secondary side are estimated using Equations (66) and (67), respectively. We must
Sustainability 2023, 15, 2813
32 of 71
note that the conduction loss in diodes D1 and D2 are equal and that in D3 and D4 are
equal. Hence, the loss equations for D1 and D3 have been presented.

Pcond.Q1 = vth ·
Iin
n

+ r DS,ON · δ ·  IO s1
2
n p1

Pcond.Q2 = vth ·
(
Pcond,D1
I
= vth · O + r DS,ON ·
2
(
Pcond,D3
I
= vth · O + r DS,ON ·
2
Iin
n

+ r DS,ON · δ ·  IO s2
2
n p1
"
(1 + 2δ)
"
(1 + 2δ)
IO
2
2
IO
2
2
1
+
3
1
+
3
∆i L1
2
2 #
∆i L2
2
2 #
!2


3
+
2 
p1
+
!2
∆i Lm + IL1 nns1
∆i Lm + IL2 nns2
2 
p1
3
∆i Lm n p1
+ (1 − 2δ)
2 ns1
2 )
∆i Lm n p1
+ (1 − 2δ)
2 ns2
2 )
(64)


(65)
(66)
(67)
The switching losses of switches Q1 and Q2 are estimated using Equations (68) and (69),
respectively, and that of diodes D1 and D3 are given by Equations (70) and (71), respectively.

!2
(Vin − ∆vCin ) 
ns1
Psw,Q1 =
· wo f f ,SW ( IO + ∆i L1 )
+ ∆i Lm +
Tsw V100FIT
n p1
!
#
ns1
xo f f ,SW ( IO + ∆i L1 )
+ ∆i Lm + yo f f ,SW
n p1

(68)
!2
n
(V + ∆vCin ) 
· won,SW ( IO − ∆i L1 ) s1 − ∆i Lm
+ in
2 · Tsw V100FIT
n p1
!
#
ns1
+ xon,SW ( IO − ∆i L1 )
− ∆i Lm + yon,SW
n p1

!2
(Vin − ∆vCin ) 
ns2
Psw,Q2 =
· wo f f ,SW ( IO + ∆i L2 )
+ ∆i Lm +
Tsw V100FIT
n p1
!
#
ns2
xo f f ,SW ( IO + ∆i L2 )
+ ∆i Lm + yo f f ,SW
n p1

!2
(Vin + ∆vCin ) 
ns2
+
· won,SW ( IO − ∆i L2 )
− ∆i Lm
2 · Tsw V100FIT
n p1
!
#
ns2
+ xon,SW ( IO − ∆i L2 )
− ∆i Lm + yon,SW
n p1
Psw,D1 =
Psw,D3 =
(Vin + ∆vCin ) ·
ns1
n p1
Tsw V100FIT
(Vin + ∆vCin ) ·
Tsw V100FIT
ns2
n p1
( I − ∆i L1 )2
( I − ∆i L1 )
· wo f f ,D1 · O
+ xo f f ,D1 · O
+ yo f f ,D1
4
2
( I − ∆i L2 )
( I − ∆i L2 )2
· wo f f ,D3 · O
+ xo f f ,D3 · O
+ yo f f ,D3
4
2
(69)
(70)
(71)
Sustainability 2023, 15, 2813
33 of 71
5.2.4. Current-Fed Push-Pull Converter
Texas Instruments design review [112] gives the details of the current-fed push-pull
(CFPP) converter. It is also called a push-pull isolated boost converter [105]. It gives
low-noise outputs with good efficiency. Instead of a center-tapped transformer secondary
(CTTS), conventional push-pull and CFPP can use a diode bridge rectifier on the secondary
side [105] to get similar outputs. The maximum duty cycle of the CFPP converter is limited
to 50% [105]. The DC voltage gain is given by Equation (72). The input inductance L1 and
the magnetizing inductance Lm can be calculated using Equations (73) and (74), respectively.
VO1
1
=
Vin
1 − 2δ
VO2
1
=
Vin
1 − 2δ
(72)
Vin
·δ
∆i L1 f sw
(73)
n p1
VO
·
· (1 − 2δ)
4∆i Lm f sw ns1
(74)
L1 =
Lm =
ns1
n p1
n
· s2
n p1
·
The output capacitance of both outputs are calculated using Equations (75) and
(76), respectively.
IO
C1 =
·δ
(75)
2 · ∆vC1 · f sw
C2 =
IO
·δ
2 · ∆vC2 · f sw
(76)
For the CFPP converter, the semiconductor utilization factor is given by Equation (77),
and the corresponding plot of duty cycle vs. SUF is shown in Figure 28. From this curve,
we can observe that the SUF is better at lower duty cycles. Thus, the transformer turns ratio
must be chosen to ensure operation at high SUF .
SUF =
4·
q
1 − (2δ)
q
1− δ
1−2δ
+
2
2
Figure 28. Semiconductor utilization factor of current-fed push-pull converter.
(77)
Sustainability 2023, 15, 2813
34 of 71
For loss estimation, we note that the average conduction losses of both switches are
equal since they conduct the same average current. Similarly, the conduction losses in
diodes D1 and D2 are equal, while the conduction losses in D3 and D4 are also equal.
The conduction loss in switches Q1 and Q2 are estimated using Equation (78), that of diodes
D1 and D2 is estimated using Equation (79), and that of diodes D3 and D4 is estimated
using Equation (80).
! 2 #
∆i2L1
∆i L1
∆i2Lm
Iin
1
2 (1 − δ )
Pcond,Q = Vth
+ r DS,ON Iin
+δ
+
+
−δ
(78)
2
2
2
6
2
3
"
Pcond,D1
I
= Vth O + rON
2
"
Pcond,D3
I
= Vth O + rON
2
n p1
Iin
ns1
2
n p1
Iin
ns2
2
+
+
n p1
ns1
2
n p1
ns2
2
(∆i Lm + ∆i L1 )2
·
3
#
(∆i Lm + ∆i L1 )2
·
3
#
1
−δ
2
1
−δ
2
(79)
(80)
The loss equality mentioned above also remains steady during switching losses. Thus,
the average switching loss in switches Q1 and Q2 are estimated using Equation (81), that in
diodes D1 and D2 using Equation (82), and that in D3 and D4 using Equation (83), respectively.
2(VO − ∆vCO ) nns1
#
( Iin + ∆i L1 + ∆i Lm ) 2
×
wO f f ,SW
+
Psw,Q =
Tsw V100FIT
2
( Iin + ∆i L1 + ∆i Lm )
xO f f ,SW
+ yO f f ,SW
2
("
#
2(VO + ∆vCO ) nns1
( Iin − ∆i L1 − ∆i Lm ) 2
p1
+
×
wO f f ,SW
+
Tsw V100FIT
2
( Iin − ∆i L1 − ∆i Lm )
xO f f ,SW
+ yO f f ,SW
2
(81)
("
#
n p1 2
(VO + ∆vCO )
Psw,D1 =
×
wO f f ,D1 ( Iin − ∆i L1 − ∆i Lm )
2Tsw V100FIT
ns1
n p1
+ xO f f ,D1 ( Iin − ∆i L1 − ∆i Lm )
+ yO f f ,D1
ns1
(82)
("
#
n p1 2
(VO + ∆vCO )
Psw,D3 =
×
wO f f ,D3 ( Iin − ∆i L1 − ∆i Lm )
2Tsw V100FIT
ns2
n p1
+ xO f f ,D3 ( Iin − ∆i L1 − ∆i Lm )
+ yO f f ,D3
ns2
(83)
p1
("
The gating circuit for the push-pull converter is relatively simple. The current-fed
variant has lesser input current ripple and has no flux imbalance [113]. The major disadvantage of the push-pull topology is that the voltage stress across the MOSFETs is more
than the input voltage magnitude [114].
5.2.5. Active Clamp Forward Converter
The active clamp forward converter uses two switches. There are two possible topologies, as shown in Figure 29a,b. The primary difference between the two topologies is the
placement of the clamping switch (Q2 ). An N-Channel MOSFET is used in the high-side
active clamp, while a P-Channel MOSFET is required for the low-side active clamp circuit.
Due to this, the high-side clamp can be used for voltages higher than 500 V, while the
low-side clamp is restricted for voltages below 500 V. The switch stress is (switch Q1 ). The
Sustainability 2023, 15, 2813
35 of 71
high-side clamp can be used for offline applications, while the low-side is restricted to
telecommunication applications. The output voltage as a function of the duty cycle [115] is
given by Equation (84). The minimum inductance value of the transformer magnetizing
inductor (Lm,min ) and the output inductor (Lk,min ) are given by Equations (85) and (86),
respectively, and the minimum value of the capacitance of the output capacitor (Ck,min ) is
given by Equation (87) for the low-side active clamp configuration [116]. The N-MOSFET
is selected based on the maximum drain-source voltage, the transformer primary’s peak
current, and the MOSFET package’s maximum allowable power dissipation. In contrast,
the P-MOSFET should be selected based on the maximum drain-source voltage [117].
For the high-side active clamp, the magnetizing inductance is proportional to the square of
the primary turns [118], as shown in Equation (88).
vOk = Vin × δ ·
(84)
vin × δmin
∆im × f sw
(85)
vO
· (1 − δmin )
∆i Lk × IO × f sw
(86)
Lm,min = k ×
Lk,min = k ×
nsk
np
Ck,min =
0.5 × ∆iO
2 × f sw
Lm = k × n2p
(87)
(88)
Figure 29. Two topologies of active clamp flyback converter: (a) high side active clamp (flyback
clamp), and (b) low side active clamp (boost clamp).
5.3. Suitability for EV Battery Charging
The isolated converters described in Sections 5.1 and 5.2 are converters where the
energy transfer from the primary side to the secondary side depends on the energy stored
in the magnetizing inductance (Lm ) of the transformer during one-half of the switching
cycle. In most high-frequency transformers, the order of Lm will be in µH. This implies that
the energy ( 12 LI 2 ) will be relatively low. Thus, these converters are preferred for low-power
applications, usually between 100 and 500 W. In addition, the output voltage level is also
limited. These converters can be used for charging batteries of low-power and low-voltage.
The present trend for electric cars and electric buses is to use battery voltages of 400 V or
800 V since the current will be lower at these voltages, giving lower losses. The singleand two-switch converters are not very suited for such applications. For high-power
applications, the energy transfer should not depend on stored energy in the magnetizing
inductance of the transformer. Such converters use the full-bridge or its variant topology
with the appropriate transformer. In these circuits, the transformer is not used to store
energy but only for step-up/down action and isolation. These circuits are most suitable for
high-power applications and are described in Section 5.4. These converters can be used for
high-power, high-voltage battery charging and can be employed for fast charging stations.
Sustainability 2023, 15, 2813
36 of 71
Among these, the resonant converters and phase-shifted full bridge variant topologies are
best suited for fast and ultra-fast charging [44,119–121].
5.4. Bridge-Type Isolated DC-DC Converters
5.4.1. Half-Bridge Converter
The half-bridge converter is one of the simplest bridge-type DC-DC converters and
can be used for battery charging applications. Shivaprasad et al. [122] have proposed a
half-bridge converter for charging one battery. The system proposed is used to charge a
48 V battery from 180–270 V input source. The pulses for the switches are generated using
TL494 pulse width modulator (PWM) integrated circuit (IC). The circuit can be modified to
charge multiple batteries to obtain the circuit diagram of the half-bridge converter feeding
two batteries, as shown in Figure 30. The relationship between input and output voltages is
given by Equations (89) and (90). Compared with the push-pull topology, the voltage stress
across the MOSFETs in half-bridge topology is lesser than the input voltage. Compared
with the isolated SEPIC and Ćuk topologies, the half-bridge configuration has lower
conduction and switching losses, leading to better efficiency. Similar to the push-pull
converter described in the previous subsection, the half-bridge converter duty cycle is
limited to 50%. In addition, the center-tapped configuration can be replaced by a diode
bridge rectifier [105].
ns1
× δ × Vin
n p1
n
= s2 × δ × Vin
n p1
vO1 =
(89)
vO2
(90)
Figure 30. A half-bridge converter feeding two batteries.
The magnetizing inductance can be calculated using Equation (91), and the output
inductance can be calculated using Equations (92) and (93), respectively. The capacitance
input side capacitors are usually equal and can be calculated using Equation (94). The capacitance output side capacitors can be calculated using Equation (95).
Vin δ
4∆i Lm f sw
(91)
L1 =
VO (1 − 2δ)
4∆i L1 f sw
(92)
L2 =
VO (1 − 2δ)
4∆i L2 f sw
(93)
Lm =
Sustainability 2023, 15, 2813
37 of 71
Iin ( Tsw − tON )
2∆vC1
I ( Tsw − tON )
C2 = in
2∆vC2
C1 =
where Tsw =
1
f sw
(94)
and tON is the time period of conduction of the switch.
∆i L1
16∆vC3
∆i L2
C4 =
16∆vC4
C3 =
(95)
For a half-bridge converter, the semiconductor utilization factor is the same as that of
the push-pull converter and is given by Equation (63) with its plot in Figure 27. The average
value of conduction losses in switches Q1 and Q2 are the same, and they can be estimated
using Equation (96). Similarly, the average value of conduction losses in diodes D1 and D2
are the same (Equation (97)), and those in D3 and D4 are the same (Equation (98)).

n

Pcond.Q1 = vth · Iin + r DS,ON · δ ·  IO s1
n p1
(
Pcond,D1
I
= vth · O + r D ·
2
(
Pcond,D3
I
= vth · O + r D ·
2
"
(1 + 2δ)
"
(1 + 2δ)
IO
2
2
IO
2
2
1
+
3
1
+
3
∆i L1
2
2 #
∆i L2
2
2 #
!2
+
∆i Lm + IL1 nns1
p1
3
∆i Lm n p1
+ (1 − 2δ)
2 ns1
2 )
∆i Lm n p1
+ (1 − 2δ)
2 ns2
2 )
2 


(96)
(97)
(98)
The average value of switching losses of switches Q1 and Q2 are the same and can be
estimated using by Equations (99) and (100), respectively.

!2
( V2in − ∆vC2 )
ns1

Psw,Q1 =
· wo f f ,SW ( IO + ∆i L1 )
+ ∆i Lm +
Tsw V100FIT
n p1
!
#
ns1
xo f f ,SW ( IO + ∆i L1 )
+ ∆i Lm + yo f f ,SW
n p1

(99)
!2
( V2in + ∆vC2 )
n
· won,SW ( IO − ∆i L1 ) s1 − ∆i Lm
+
Tsw V100FIT
n p1
!
#
ns1
+ xon,SW ( IO − ∆i L1 )
− ∆i Lm + yon,SW
n p1
Sustainability 2023, 15, 2813
38 of 71

!2
( V2in − ∆vC1 )
n
s2
Psw,Q2 =
· wo f f ,SW ( IO + ∆i L1 )
+ ∆i Lm +
Tsw V100FIT
n p1
!
#
ns2
xo f f ,SW ( IO + ∆i L1 )
+ ∆i Lm + yo f f ,SW
n p1

!2
( V2in + ∆vC1 )
n
s2
+
· won,SW ( IO − ∆i L1 )
− ∆i Lm
Tsw V100FIT
n p1
!
#
ns2
+ xon,SW ( IO − ∆i L1 )
− ∆i Lm + yon,SW
n p1
(100)
The average value of switching losses of diodes D1 and D3 are estimated using
Equations (101) and (102), respectively.
Psw,D1 =
Psw,D3 =
( V2in + ∆vCin ) ·
ns1
n p1
Tsw V100FIT
( V2in + ∆vCin ) ·
Tsw V100FIT
ns2
n p1
( I − ∆i L1 )2
( I − ∆i L1 )
· wo f f ,D1 · O
+ xo f f ,D1 · O
+ yo f f ,D1
4
2
(101)
( I − ∆i L2 )2
( I − ∆i L2 )
· wo f f ,D3 · O
+ xo f f ,D3 · O
+ yo f f ,D3
4
2
(102)
A modified half-bridge converter was proposed by Ou et al. [123]. It uses eight diodes,
two switches, three capacitors, and two inductors, and a center-tapped transformer. This
can be modified to charge two batteries (or more) by using transformers with multiple
secondaries. The ratings of the primary side components become critical in such a case.
Hyeon and Cho [124] have proposed a dual half-bridge LLC Resonant Converter with two
outputs (one main output and one sub-output). This could also be used for battery charging
applications, with the main output used to charge the main battery and the sub-output
used to charge the auxiliary battery. Li and Zhang [125] have proposed a modified system
called the isolated voltage type half-bridge three-port converter. It comprises three input
combined half-bridge circuits with a three-winding transformer.
Jain et al. [126] have proposed a bi-directional topology with a half-bridge converter
on the primary side and a current-fed push-pull on the secondary side to charge one battery,
albeit for low-power applications. This can be extended to charge multiple batteries simultaneously.
5.4.2. Half-Bridge Isolated Boost Converter
A half-bridge isolated boost (HBIB) converter is a boost-derived converter [105] that
uses two switches on the primary of the HFT and a diode bridge on the secondary in its basic
topological format. Such a converter can feed two batteries of equal rating and is shown in
Figure 31. Equation (103) gives the relationship between the input and output voltages.
ns1
n p1
n
= s2
n p1
vO1 =
vO2
Vin
1−δ
V
· in
1−δ
·
(103)
Sustainability 2023, 15, 2813
39 of 71
Figure 31. Half-bridge isolated boost converter.
The maximum duty cycle of the converter is limited to 50%. The inductance of
the inductors L1 , L2 , and Lm can be calculated using Equations (104), (105) and (106),
respectively. Similarly, the capacitance of the output capacitors CO1 and CO2 can be found
using Equations (107) and (108), respectively. The capacitance of the input capacitor can be
calculated using Equation (109).
Lm =
L1 =
Vin δ
2 · ∆i L1 · f sw
(104)
L2 =
Vin δ
2 · ∆i L2 · f sw
(105)
n p1
VO
·
· (1 − δ )
2 · ∆i L1 · f sw ns1
(106)
IO
· (2δ − 1)
4 · ∆vCO1 · f sw
(107)
CO1 =
Cin =
IO
· (2δ − 1)
(108)
1
∆i L1
· 2−
16 · ∆vCin · f sw
δ
(109)
CO2 =
4 · ∆vCO2 · f sw
The semiconductor utilization factor (SUF ) of the HBIB converter is given by Equation (110), and the plot of the SUF as a function of the duty cycle is given in Figure 32. It
can be seen that the SUF is better at lower duty cycles. Thus, the transformer turns ratio
must be selected such that the converter operates at a duty cycle close to 50% [105].
SUF =
1−δ
p
(2 1 − δ) + 3 − (2δ)
√
(110)
Sustainability 2023, 15, 2813
40 of 71
Figure 32. Semiconductor utilization factor of the half-bridge isolated boost converter.
The MOSFETs Q1 and Q2 carry the same current, implying equal conduction and
switching losses. Similarly, the diodes in the bridge rectifiers on the secondary have
the same conduction and switching losses. The average value of conduction loss of the
MOSFETs can be estimated [105] using Equations (111) and (112), respectively. Based on
the above loss equality, the conduction loss in each diode (D1 to D4 ) of the bridge rectifier
connected to the first battery can be estimated using Equation (113). Similarly, the average
conduction loss in each of the diodes (D5 to D8 ) of the bridge rectifier feeding the second
battery can be estimated using Equation (114).
Iin
1
3 − (2δ)
2
Pcond,Q1 = vth
+ r DS,ON Iin
+ (∆i Lm · ∆i L1 )
− 2 + ∆i2Lm (2δ − 1)
2
4
δ
#
(111)
∆i2L1
2
5 − − (2δ)
+
3
δ
Iin
3 − (2δ)
1
2
Pcond,Q2 = vth
+ r DS,ON Iin
+ (∆i Lm · ∆i L2 )
− 2 + ∆i2Lm (2δ − 1)
2
4
δ
#
∆i2L2
2
+
5 − − (2δ)
3
δ
"
Pcond,D1
I
= vth O + r D
2
"
Pcond,D5
I
= vth O + r D
2
n p1
Iin
ns1
2
n p1
Iin
ns2
2
+
+
n p1
ns1
2
n p1
ns2
2
(112)
(∆i Lm + ∆i L1 )2
(1 − δ )
3
#
(∆i Lm + ∆i L2 )2
(1 − δ )
3
#
(113)
(114)
For the estimation of switching losses, the loss equality is valid. The losses in the
switches can be calculated using Equations (115) and (116), respectively. The switching
losses in each upper and lower bridge diode can be estimated using Equations (117) and
(118), respectively.
Sustainability 2023, 15, 2813
41 of 71
n
(VO − ∆vO ) np1
s1
"
2
Iin
Psw,Q1 =
· wo f f ,Q1
+ ∆i L1 + ∆i Lm
Tsw V100FIT
2
I
+ xo f f ,Q1 in + ∆i L1 + ∆i Lm + yo f f ,Q1
2
"
n
2
(VO + ∆vO ) np1
Iin
s1
+
· won,Q1
− ∆i L1 − ∆i Lm
Tsw V100FIT
2
I
+ xon,Q1 in − ∆i L1 − ∆i Lm + yon,Q1
2
n
(VO − ∆vO ) np1
s2
"
2
Iin
+ ∆i L2 + ∆i Lm
Tsw V100FIT
2
Iin
+ xo f f ,Q2
+ ∆i L2 + ∆i Lm + yo f f ,Q2
2
"
n
2
(VO + ∆vO ) np1
Iin
s2
+
· won,Q2
− ∆i L2 − ∆i Lm
Tsw V100FIT
2
Iin
+ xon,Q2
− ∆i L2 − ∆i Lm + yon,Q1
2
Psw,Q2 =
(115)
· wo f f ,Q2
(116)
"
n p1 2
Iin
(VO + ∆vO )
· won,D1
− ∆i L2 − ∆i Lm
Psw,D1 =
2Tsw V100FIT
2
ns1
n p1
I
+ xon,D1 in − ∆i L2 − ∆i Lm
+ yon,D1
2
ns1
(117)
"
n p1 2
(VO + ∆vO )
Iin
Psw,D5 =
· won,D5
− ∆i L2 − ∆i Lm
2Tsw V100FIT
2
ns2
n p1
I
+ xon,D5 in − ∆i L2 − ∆i Lm
+ yon,D5
2
ns2
(118)
5.4.3. Voltage-Fed Full Bridge Converter
Full-bridge converter and their variants are one of the most commonly used converters
for medium- and high-power battery charging applications. The circuit diagram of a basic
voltage-fed full-bridge converter (VFFBC) feeding two batteries is shown in Figure 33a.
Equations (89) and (90) give the relationship between input and output voltages. In a
VFFBC, a dead-band is required between the PWM pulses of diagonal and off-diagonal
switches to prevent shoot-through. The dead-band is not required n the case of a current-fed
FBC (CFFBC; described in the following subsection).
In addition, a modulation technique called phase-shifting can be applied to a VFFBC to
get the phase-shifted full-bridge converter (PSFBC). Its pulse pattern is shown in Figure 34.
It is one of the best-suited converters for high-power EV battery charging applications [114].
The PSFBC topology has reduced losses due to zero-voltage-switching. The primary currents are low, and the total flux in the core is utilized. The conversion ratio (output to input
voltages) can be wide, and very high efficiency of the order of 99% can be achieved. Several
works described in Section 4 use full-bridge topology or its variants for these reasons.
Sustainability 2023, 15, 2813
42 of 71
Figure 33. Full-bridge converter (FBC) feeding two batteries: (a) voltage-fed FBC, and (b) currentfed FBC.
Figure 34. Pulse pattern of a phase-shifted FBC.
Sustainability 2023, 15, 2813
43 of 71
The value of the inductance of the output inductors LO1 and LO2 of VFFBC is calculated
using Equation (119).
Vin nns1 − VO
p1
LO1 =
·δ
2∆i LO1 f sw
(119)
Vin nns2 − VO
p1
LO2 =
·δ
2∆i LO2 f sw
The primary leakage inductance can be estimated using Equation (120).
Lσ = VO
Vin
L02
L01
2 = 2
VO
ns1
− n
− nns2
V
p1
in
(120)
p1
The capacitance of the output capacitors can be calculated using Equation (121).
∆i LO1
16 · ∆vCO1 · f sw
∆i LO2
=
16 · ∆vCO2 · f sw
CO1 =
CO2
(121)
The semiconductor utilization factor (SUF ) of a PSFB converter is given by Equation (122),
and the plot of the duty cycle vs. SUF is given in Figure 35. The SUF graph shows that
the SUF will be higher at higher duty cycles. Hence, the transformer turns ratio must be
chosen such that the converter operates at duty cycle values close to 50%.
2·δ
√
√
√
√
2 · ( 2 + δ + 1 − δ) + 2
(122)
Figure 35. Semiconductor utilization factor of the PSFB converter.
For estimation of power losses, it is to be noted that the losses in diagonal MOSFETs
(S1 and S2 ) are the same, and the losses in the off-diagonal MOSFETs (S3 and S4 ) are the
same. This also happens with their respective body diodes (DS1 to DS4 ). The various time
intervals can be seen from the waveform of the current through the leakage inductor [105],
shown in Figure 36.
Sustainability 2023, 15, 2813
44 of 71
Figure 36. Current through the leakage inductance of the PSFB converter described in Ref. [105].
The conduction losses in switches S1 to S4 are calculated using Equations (123), (124),
(125) and (126), respectively.


n p1
T
I
+
∆i
+
I
O
LO1
S ns1
of f
( IO − ∆i LO1 )(to f f − To f f − t a )
ns1 

Pcond,S1 = Vth
I δ +
+
n p1 O e f f
2Tsw
2Tsw
(123)

!2 
ns1 
a
+ r DS,ON 
3Tsw n p1
"
Pcond,S2 = Vth
ns1
I δ
n p1 O e f f

#
( IO − ∆i LO1 )(to f f − To f f − t a )
b
+
+ r DS,ON 
2Tsw
3Tsw
n
To f f IO + ∆i LO2 + IS np1
s2

ns2 
I δ +
n p1 O e f f

!2 
a
ns2 
+ r DS,ON 
3Tsw n p1
Pcond,S3 = Vth
"
Pcond,S4 = Vth
ns2
I δ
n p1 O e f f
2Tsw
!2 
ns1 
n p1
(124)

( IO − ∆i LO2 )(to f f − To f f − t a )

+
2Tsw
(125)

#
( IO − ∆i LO2 )(to f f − To f f − t a )
b
+
+ r DS,ON 
2Tsw
3Tsw
!2 
ns2 
n p1
(126)
The average conduction losses in the body diodes DS1 and DS2 are calculated using
Equations (127) and (128), respectively.
Is t a
I 2 ta
+ r DS,ON s
2Tsw
3Tsw
(127)
Is
( I + ∆i LO1 )
c
t a + To f f 1 + O
+ r DS,ON
2Tsw
Is
3Tsw
(128)
Pcond,DS1 = Vth
Pcond,DS2 = Vth
The average conduction loss in diode D1 is calculated using Equation (129). The losses
in all the other diodes have the same value.
Sustainability 2023, 15, 2813
45 of 71
n p1
I − ∆i LO1
ton IO
ta
Pcond,D1 = Vth (to f f − To f f − t a ) · O
+
+
Is
2Tsw
Tsw
2Tsw ns1
To f f
n p1
+
I + ∆i LO1 + Is
2Tsw O
ns1

n p1 3

3

Is n
− ( IO + ∆i LO1 )

s1
To f f
( I − ∆i LO1 )2
n
+ r D (to f f − To f f − t a ) · O
+

3Tsw
2Tsw

Is np1 − ( IO + ∆i L1 )3

s1
)
2
n p1
(3I 2 + ∆i2LO1 )
ta
+
+ ton O
· Is
3Tsw
ns1
3Tsw
(129)
The body diodes and the secondary diodes do not undergo hard switching. Hence the
losses can be neglected. In addition, the MOSFETs undergo zero voltage switching (ZVS).
Hence, their turn-ON power losses are minimal and can be neglected. Hence, switching
loss equations consider the turn-off characteristics only. The losses are estimated using
Equations (130) and (131).
Psw,S1 =
Psw,S2
Vin
wo f f ,S1 Is2 + xo f f ,S1 Is + yo f f ,S1
Tsw V100FIT


!2
Vin − ∆vcin 
ns1
ns1
=
wo f f ,S2 ( IO + ∆i LO1 )
+ xo f f ,S1 ( IO + ∆i LO1 )
+ yo f f ,S1 
Tsw V100FIT
n p1
n p1
(130)
(131)
If the leakage inductance is negligible, the above expressions can be simplified further.
In the equations related to PSFBC, the terms Is , ta , tb , a, and b are defined as follows:
Is =
ns1
n p1 · IO + ∆i LO1 −
δe f f = δ − Lσ
ta =
tb =
ns1
n p1
Vs ·(0.5−δ)
Lσ f sw
2IO
n
· s1
Vin f sw n p1
!
Is (to f f − To f f )
ns1
n p1
· ( IO − ∆i LO1 ) + Is
Iin + Is
· (to f f − To f f )
· ( IO − ∆i LO1 ) + Is
i
ton h
(2∆i LO1 + IO − ∆i LO1 )3 − ( IO − ∆i LO1 )3
a = (to f f − To f f − t a )( IO − ∆i LO1 )2 +
2∆i LO1
2
n
Is3 np1 − ( IO + ∆i L1 )3 · nns1
s1
p1
h
i
+ To f f
Is − ( IO + ∆i LO1 ) nns1
p1
b = (to f f − To f f − t a )( IO − ∆i LO1 )2 +
i
ton h
(2∆i LO1 + IO − ∆i LO1 )3 − ( IO − ∆i LO1 )3
2∆i LO1
This topology is also well-suited for high-power and fast-charging systems.
5.4.4. Current-Fed Full Bridge Converter
The current-fed full-bridge converter (CFFBC), also known as the isolated-boost
FBC [105] is a unidirectional converter of bridge type with an input inductor, MOSFET
Sustainability 2023, 15, 2813
46 of 71
bridge, HFTs, diode bridge, and output filter capacitor. The CFFBC topology used to charge
two batteries is shown in Figure 33b. The maximum duty cycle of the MOSFETs is limited to
50%, and the relationship between the input and output voltages is given by Equation (132).
Vin ns1
1 − 2δ n p1
Vin ns2
=
1 − 2δ n p1
VO1 =
VO2
(132)
The magnetizing inductance (Lm ) of the transformer and the input inductance (L1 ) can
be calculated using Equations (133) and (134), respectively.
Lm =
VO
n
· s1 · (1 − 2δ)
4∆i Lm f sw n p1
(133)
Vin
·δ
2∆i L1 f sw
(134)
L1 =
Similarly, the output filter capacitance of the two capacitors can be calculated using
Equation (135).
IO
·δ
2∆vCO1 f sw
IO
·δ
=
2∆vCO2 f sw
CO1 =
CO2
(135)
The semiconductor utilization factor for a CFFB converter is the same as that of the
CFPP converter and is given by Equation (77), and its plot as a function of the duty cycle
is shown in Figure 28. The MOSFETs on the input side of the CFFB converter carry the
same current and hence have equal power losses. The same inference is also applicable
to the secondary side diodes. Hence, the average value of conduction loss in MOSFETs is
estimated using Equation (136) and that in diodes D1 and D3 using Equations (137) and
(138), respectively.
! 2 #
∆i2L1
∆i L1
∆i2Lm
Iin
1
2 (1 − δ )
Pcond,S1 = Vth
+ r DS,ON Iin
+δ
+
+
−δ
(136)
2
2
2
6
2
3
"
Pcond,D1
I
= Vth O + rON
2
"
Pcond,D3
I
= Vth O + rON
2
n p1
Iin
ns1
2
n p1
Iin
ns2
2
+
+
n p1
ns1
2
n p1
ns2
2
(∆i Lm + ∆i L1 )2
·
3
#
(∆i Lm + ∆i L1 )2
·
3
#
1
−δ
2
1
−δ
2
(137)
(138)
The average switching loss in each MOSFET is estimated using Equation (139), that in
diodes D1 and D2 using Equation (140), and that in D3 and D4 using Equation (141), respectively.
Sustainability 2023, 15, 2813
47 of 71
s1
(VO − ∆vCO ) nnp1
("
#
( Iin + ∆i L1 + ∆i Lm ) 2
Psw,Q =
×
wO f f ,SW
+
Tsw V100FIT
2
( Iin + ∆i L1 + ∆i Lm )
+ yO f f ,SW
xO f f ,SW
2
("
ns1
#
(VO + ∆vCO ) n p1
( Iin − ∆i L1 − ∆i Lm ) 2
+
×
wO f f ,SW
+
Tsw V100FIT
2
( Iin − ∆i L1 − ∆i Lm )
xO f f ,SW
+ yO f f ,SW
2
(139)
("
#
n p1 2
(VO + ∆vCO )
×
wO f f ,D1 ( Iin − ∆i L1 − ∆i Lm )
Psw,D1 =
2Tsw V100FIT
ns1
n p1
+ xO f f ,D1 ( Iin − ∆i L1 − ∆i Lm )
+ yO f f ,D1
ns1
(140)
("
#
n p1 2
(VO + ∆vCO )
Psw,D3 =
wO f f ,D3 ( Iin − ∆i L1 − ∆i Lm )
×
2Tsw V100FIT
ns2
n p1
+ xO f f ,D3 ( Iin − ∆i L1 − ∆i Lm )
+ yO f f ,D3
ns2
(141)
This topology is also well-suited for high-power and fast-charging systems.
5.4.5. Dual Active Bridge
The Dual Active Bridge (DAB) is a bidirectional buck-boost converter. In a DAB,
the diodes of the conventional VFFBC are replaced by MOSFET-based H-bridges, as shown
in Figure 37. Phase-shifted modulation technique with pulses shown in Figure 38 is
generally used for DAB. The corresponding inductor current is shown in Figure 39.
Figure 37. Dual active bridge converter feeding two batteries.
Sustainability 2023, 15, 2813
48 of 71
Figure 38. Pulses for the phase shifted dual active bridge.
Figure 39. Current through the leakage inductor in a DAB converter.
The semiconductor utilization factor (SUF ) of the DAB is a function of its rated power,
input and output voltages, input and output bridge MOSFET currents, and their corresponding body diode currents [105]. It is also a function of the phase shift (φ) of the pulses
provided to the switches (Figure 38). By plotting the φ vs. SUF waveform and φ vs. Output
Power, we can conclude the following:
•
•
•
SUF is the best at phase- shifts of 0◦ and 180◦ ;
However, at these points of phase-shift, the converter is underutilized (output power
is close to zero);
Hence, despite the low SUF , it is preferred to generate pulses with a phase shift of 90◦
to maximize power transfer.
For the calculation of power losses, it is to be noted that the losses in the input side
MOSFETs are the same as they carry the same current. This argument is valid with the
body diodes and the output side MOSFETs. Hence, power loss is estimated from S1 , S5 ,
DS1 , and DS5 from Equations (142), (143), (144) and (145), respectively.
r DS,ON 2 φ
i1 φ
ta
i1 + imax
φ
ta
Pcond,S1 = Vth
−
+
1−
+
i1
−
2 2π
Tsw
4
π
3
2π
Tsw
(142)
1
φ
2
2
+ (i1 + imax i1 + imax )
−
2 2π
Sustainability 2023, 15, 2813
49 of 71
Pcond,S1 = Vth
i1 n p1
2 ns1
φ
ta
−
2π
Tsw
Pcond,DS1 = Vth
+
n p1 2 φ
r DS,ON
ta
i1
−
3
ns1
2π
Tsw
i 2 ta
imax t a
+ r DS,ON max
2 Tsw
3Tsw
n p1 imax t a
r DS,ON c
r DS,ON n p1 2
φ
i1 + imax
1−
+
·
Pcond,DS5 = Vth
+
+
ns1
2 Tsw
4
π
3
Tsw
3
ns1
φ
ta
1
(i12 + imax i1 + i2max )
−
+ i2max
2 2π
Tsw
(143)
(144)
(145)
The phase-shifted DAB’s main advantage is that the MOSFETs operate under zero
voltage switching, and the body diodes operate under zero current switching conditions.
Therefore, the losses are reduced, and the efficiency is very high, usually above 95%.
Therefore, turn-ON power losses of the MOSFETs and the turn-OFF losses of the body
diodes need not be considered. Thus, the switching losses in the MOSFETs S1 and S5 are
given by Equations (146) and (147), respectively.
Psw,S1 =
Psw,S5
(Vin − ∆vCin ) wo f f ,S1 i2max + xo f f ,S1 imax + yo f f ,S1
Tsw V100FIT
!
n p1 2
n p1
(VO − ∆vCO )
=
· wo f f ,S5 i1
+ xo f f ,S1 i1
+ yo f f ,S1
Tsw V100FIT
ns1
ns1
(146)
(147)
The DAB topology is well suited for high-power charging systems, fast-charging
systems, and systems that charge high-voltage batteries.
5.5. Isolated DC-DC Resonant Full-Bridge Converters
Resonant converters are the most popular type of isolated DC-DC converters used
for high-power battery charging circuits. Among the various types of resonant converters,
the two most commonly used types are the LLC resonant converters and CLLC resonant converters. Half-bridge resonant configurations are possible [127], but the power
level is limited to medium-power applications. Hence, only full-bridge topologies are
considered here.
5.5.1. LLC Resonant Converter
LLC converters are very popular since they can achieve ZVS at the primary and ZCS
on the secondary side [128]. It also offers higher power density due to better utilization of
the transformer core. The current rating is also less, implying a cost reduction. The LLCtype resonant converter, sometimes referred to as the Series Resonant Converter (SRC),
feeding two batteries, is shown in Figure 40. The system consists of a full-bridge converter
whose output is connected to the series combination of a resonant inductor (L R ) and a
resonant capacitor (CR ) and is terminated to the primary of a high-frequency transformer
(HFT) having a magnetizing inductance of Lm . The detailed design steps of the LLC
resonant converter are given in the Infineon application note [129]. Other works [130,131]
have also provided detailed designs and analyses of LLC-type converters. The first step
in the design of the LLC Converter is to decide the basic specifications, including the
range of input voltage (minimum: Vin,min to maximum: Vin,max , and nominal value: Vin ),
range of the output voltage (minimum: vO,min to maximum: vO,max , and nominal value:
vo ), maximum output power (PO,max ) and the resonant frequency (fr ). The switching
frequency (fsw ) choice depends on the trade-off between the magnetics of Lr , and the
Sustainability 2023, 15, 2813
50 of 71
gauge, cost, and losses corresponding to the Litz wire. In addition, the maximum value
of the switching frequency (fsw,max ) is limited by the PWM IC used in the control loop for
driving the MOSFETS on the primary side of the HFT. A general rule to limit the switching
frequency is f sw ≤ (2.5 × f r ) [131]. The next step is the selection of the turn ratio of the
HFT. The transformer turns ratio (nn ) at the resonant frequency is given by Equation (148).
nn =
Vin
2 × (vin,min ) + v AK
(148)
where, vAK is the ON-state voltage across the diode of the bridge rectifier on the secondary.
Next, the minimum value of resonant inductor is calculated using Equation (149) and that
of the resonant capacitor is calculated using Equation (150).
Lr =
Cr =
nn × Vin × vO
8 × PO × f sw,max
(149)
1
(2 × π × f r )2 × Lr
(150)
Figure 40. LLC resonant converter feeding two batteries.
With these values calculated, the characteristic impedance (Zc ), the equivalent load
and rectifier resistance (Req,ac ), and the minimum quality factor (Qmin ) are calculated using
Equations (151), (152) and (153), respectively.
s
Lr
Cr
(151)
2
8 × n2n vO,max
·
PO
π2
(152)
Zc
Req,ac
(153)
Zc =
Req,ac =
Qmin =
Sustainability 2023, 15, 2813
51 of 71
The next value to be obtained is the magnetizing inductance of the transformer. To obtain maximum gain at minimum switching frequency [131], the maximum magnetizing
inductance (Lm,max ) is calculated using Equation (154).
f
Lm,max
r
π 2 f sw,max − 1
= Lr ·
·
4 1− M 1
(154)
dc,max
where Mdc,max is the maximum DC gain. In addition to this, another magnetizing inductance
for zero voltage switching at no-load condition (Lm,ZVS ) is calculated using Equation (155).
1
td × nn × vO,min × 4× f sw,max
− t2d
Lm,ZVS =
(155)
Cstary × Vin,max
where td is the dead time between the pulses, and Cstray is the sum of the stray capacitance
of the PCB, MOSFETs, and inter-winding capacitance of the transformer and inductor.
The final value of the magnetizing inductance is chosen as Lm = min(Lm,max , Lm,ZVS ).
The verification of the choice of total inductance must satisfy the energy balance given by
Equations (156) and (157).
1
1
2
· ( Lm,min + Lr ) Im,pk >
· Cstray · Vin,max
(156)
2
2
Im,pk =
nn × VO,min
4 × Lm × f sw
(157)
Deng et al. [130] have used a slightly modified design approach to obtain two different
resonant frequencies. Infineon [129] has employed a different approach by calculating the
quality factor and gain and then using the same to calculate the resonant component values.
Once the values of the resonant components are calculated, the MOSFETs and diodes must
be carefully selected. The MOSFETs used in the primary side bridge must be selected based
on the current, on-state resistance, dv/dt limits, and reverse recovery time in addition to the
output parasitic capacitance value [132]. The selection of the resonant inductor depends
on its core type, geometry, and current. Metallized Polypropylene (MKP) capacitors are
preferred for resonant capacitors due to higher temperature tolerance, lower cost, and low
effective series resistance (ESR). Its selection mainly depends on the current through and
the peak voltage across it. For the secondary side bridge, Schottky diodes are preferred due
to their low forward voltage drop. Ultrafast diodes can also be chosen for the diode bridge.
The resonant components and transformers in multiple numbers are required for charging
multiple batteries simultaneously, along with diodes, as shown in Figure 40. If a capacitor is
used in the transformer’s secondary, the primary and secondary resonant capacitor values
can be equal [133]. If two batteries of different voltage ratings are to be charged, then the
resonant component values of the two primaries will be different and need to be calculated
separately. Suppose a transformer with one primary and two secondaries is used (instead
of two separate transformers) to charge two batteries. In that case, the design values of
the resonant components need to be changed by considering the energy increase on the
primary side. The main limitation of the LLC resonant converter is the reduced efficiency
and loss of ZCS when the converter is operated away from the resonant frequency [133].
The LLC topology is well suited for high-power charging systems, fast-charging systems,
and systems that charge high-voltage batteries.
5.5.2. CLLC Resonant Converter
The CLLC type converter is an extension of the LLC resonant converter. It uses one
inductor and one capacitor connected in series at the primary and secondary windings.
A CLLC converter feeding two batteries is shown in Figure 41.
Sustainability 2023, 15, 2813
52 of 71
Figure 41. CLLC converter supplying two batteries.
The design of the CLLC resonant tank is based on the LLC resonant tank described
in the previous subsection. The range of input and output voltages and the output power
are considered as the defined specifications to begin the design [134]. The transformer
turns ratio is calculated using Equation (158), where e is the line voltage variation allowed.
Next, the minimum and maximum values of the gain (Mmin and Mmax , respectively) are
calculated using Equations (159) and (160), respectively.
n=
Vin × (1 + e)
vO,min
vout,min × n
Vin
vout,max × n
=
Vin
(158)
Mmin =
(159)
Mmax
(160)
Similar to the LLC converter, the CLLC converter switching frequency selection is an
important step. A high switching frequency reduces the size of the resonant components
and the stress on the resonant capacitors. However, it leads to higher losses. The switching
frequency selection criteria for the CLLC converter are the same as that of the LLC converter.
The maximum value of the magnetizing inductance (Lm,max ) is decided based on the deadtime and the output capacitance (Coss ) of the MOSFET. It is calculated using Equation (161).
Lm,max =
td
16 × f r × Coss
(161)
The primary series resonant inductor (LRP ) is calculated using the normalized inductance value (Ln ). If Ln is small, a larger voltage gain can be obtained, but the circulating
currents will be higher. On the other hand, a large value of Ln will reduce power losses,
but the operating frequency range will be wide to meet the required gain. Thus, a trade-off
is done to find the value of LRP .
Lm
Ln =
(162)
L RP
Sustainability 2023, 15, 2813
53 of 71
Next, the primary series resonant capacitor (CRP ) is calculated using the resonant
frequency according to Equation (163). If the DC gain does not match, the value of Lm
needs to be adjusted, and the calculation of LRP and CRP needs to be revised.
fr =
1
√
2 × π L RP × CRP
(163)
The series resonant capacitor on the secondary (CRS ) needs to be selected based on
four parameters. The details of the same are given in [134]. Firstly, a new parameter
(Ld ) is defined as the ratio of the frequency at the load-independent point to the resonant
frequency. To ensure that the frequency at the load-independent point is close to the
resonant frequency, a higher value of CRS is preferred.
Ld =
f load,independent
fr
(164)
In addition, the selected capacitance should satisfy the gain range. Thus, the value
of the capacitance is limited by the gain requirement. The peak gain is to be selected such
that it is above the maximum gain requirement. This corresponds to the minimum value of
CRS . Based on these, the capacitance CRS is calculated, and the gain is rechecked. Proper
tuning of CRS is essential for proper gain output. A similar method of design is given in
Refs. [135,136]. A circuit-parameter-based design method is proposed and used in Ref. [137]
for CLLC converter design. Ammar et al. [138] have used Equations (165) through (169) to
calculate the resonant tank values.
Ql,max × Rr,max
2 × π × fr
1
=
2 × π × f r × Ql,max × Rr,max
L
= RP
n2
= CRP × n2
L RP =
(165)
CRP
(166)
L RS
CRS
Lm = k × L RP
(167)
(168)
(169)
where n is the transformer turns ratio, k is the normalized switching frequency, Rload is the
load resistance Rr reflected at the rectifier input and referred to the primary side, given
by Equation (170), and Ql,max is the maximum value of the loaded quality factor, and its
nominal value defined by Equation (171).
Rr =
2 × n2 × Rload
π2
q
Ql =
L RP
CRP
Rr
(170)
(171)
The CLLC topology is also well suited for high-power charging systems, fast-charging
systems, and systems that charge high-voltage batteries.
5.5.3. LLC and CLLC DAB
Resonant DAB converters are popular due to their high-efficiency, high-power, and high
power-density [128]. The LLC and CLLC DAB circuits are shown in Figures 42 and 43,
respectively. Texas Instruments design guide [128] details the CLLC DAB converter used
for EV chargers. The design of LLC and CLLC DAB resonant components is similar to
those described in the previous subsections.
Sustainability 2023, 15, 2813
54 of 71
Figure 42. LLC DAB converter feeding two batteries.
Figure 43. CLLC DAB converter feeding two batteries.
A comparison of the various converters described in Section 5 is given in Table 2,
where SS is the soft switching possibility, NL is the number of inductors, NC is the number
of capacitors, Nsw is the number of switches, and Ndio is the number of diodes.
The DAB topology with an integrated resonant tank is well suited for high-power
charging systems, fast-charging systems, and systems that charge high-voltage batteries.
Sustainability 2023, 15, 2813
55 of 71
Table 2. Comparison of various multi-output converters described in Section 5 when feeding four batteries.
SIMO Converter
NL NC Nsw
Ndio
Power
Control
SS [139]
SS Region [140]
Efficiency
Cost
Flyback
0
4
1
4
≤100 W
Simple
No
-
≥95%
[141,142]
Low
Forward
4
4
1
9
50 to 200 W
Simple
No
-
≥94% [142]
Low
Isolated SEPIC
1
5
1
4
50 to 300 W
Simple
No
-
92% [143]
Low
Isolated Ćuk
5
9
1
4
50 to 300 W
Simple
No
-
95%
Low
Isolated Zeta
4
8
1
4
50 to 300 W
Simple
No
-
≥80% [144]
Low
Fly-buck
0
4
1
5
≤100 W
Simple
No
-
≤94% [145]
Low
Two Switch Flyback
0
4
2
4
50 to 200 W
Simple
No
-
≤90% [146]
Low
Two Switch Forward
4
4
2
10
50 to 200 W
Simple
No
-
90% [147]
Low
Push-Pull
4
4
2
8
100 to 500 W
Simple
No
-
92% [148]
Low
CFPP
5
4
3
9
100 to 500 W
Moderate
No
-
≤95% [149]
Low
Flyback Clamp
4
4
2
8
50 to 300 W
Simple
No
-
≤95% [150]
Low
Boost Clamp
4
4
2
8
50 to 300 W
Simple
No
-
≤95%
Low
Half Bridge
4
6
2
8
100 to 500 W
Moderate
No
-
≤94% [151]
Medium
HBIB
2
5
2
16
100 to 500 W
Moderate
No
-
around 94%
Medium
VFFB
4
4
4
16
>500 W
Moderate
Possible
-
>95% [140]
Medium
CFFB
1
4
4
16
>500 W
Moderate
Possible
-
>95% [140]
Medium
DAB
4
4
20
0
kW range
Complex
Possible
Partial Load
>97% [140]
High
LLC FB
6
6
4
16
kW range
Complex
Possible
Full load
>95% [140]
High
CLLC FB
8
8
4
16
kW range
Complex
Possible
Full load
> 95% [140]
High
LLC DAB
6
6
20
0
kW range
Complex
Possible
Partial load
>94% [140]
High
CLLC DAB
8
8
20
0
kW range
Complex
Possible
Partial load
>94% [140]
High
6. High Frequency Transformers in Isolated Converters for Simultaneous Charging
6.1. Relationship Between Switching Frequency and Magnetic Component Size
Isolated converters can be used for higher-wattage systems with the proper design
of magnetic components. If the size of the system is a significant concern (especially with
magnetic components), wide band-gap (WBG) devices such as SiC and GaN devices can
be used. The WBG devices can operate at very high frequencies, SiC in hundreds of kHz
and GaN up to MHz, leading to a drastic reduction in the size of magnetic components.
Figure 44 shows a sample of two inductors, one being a line frequency (50 Hz) inductor
and the other being a high-frequency inductor, having the same inductance value. The size
difference is evident from this.
For any magnetic component, the product of the window area and core area (Aw Ac ) is
inversely proportional to the switching frequency (fsw ) [152] and follows the relationship
expressed by Equation (172).
Aw Ac =
PO × J
k t × Bmax × f sw
(172)
where PO is the output power in W, J is the current density in mA, fsw is in Hz, and kt is the
constant whose value is given by,
Sustainability 2023, 15, 2813
56 of 71

0.00100




0.00140



0.00140
kt =
0.00050





0.00033


0.00025
for Push-Pull Converter
for Half-Bridge Converter
for Full-Bridge Converter
for Forward Converter
for Single-Winding Flyback Converter
for Multi-Winding Flyback Converter
Figure 44. Difference in size of line frequency and high frequency inductors.
Increasing the frequency leads to an increased skin effect. This implies an increase in
the losses due to the AC resistance and an increased temperature. To reduce the skin effect,
a smaller diameter wire (skin depth d equal to half the wire diameter) must be chosen while
ensuring that the wire can carry the required current. Another method to reduce the skin
effect is to use multiple strands of wire, also known as Litz wire. The skin depth can be
calculated using Equation (173).
k
d = pM
(173)
f sw
where k M is the copper constant whose value varies between 7.5 (at 100 ◦ C) and 6.6
(at 20 ◦ C). Along with the skin effect, the proximity effect also exists in inductors and
transformers. The use of Litz wire also helps in reducing the proximity effect. In addition to
these effects, one should evaluate the relationship between the peak-to-peak ripple current
(∆i L ) and switching frequency. As the switching frequency increases, the inductance value
should be proportionately reduced to keep the peak-peak ripple current within the required
limits. For example, if a 100 µH existing inductor operates at 200 kHz, and if the frequency
is increased to 400 kHz, the inductance should be reduced by 53 µH (standard inductance
value) [152]. This is necessitated either due to the frequency roll-off characteristic of the
core material or due to the self-resonance between the winding inductance and its selfcapacitance [153]. Core loss is another component that varies with frequency. To prevent
core losses from increasing when the frequency is increased, the flux density must be
maintained constant. The response of most high-frequency power inductors varies only a
little for a large range of frequencies [153]. Generally, testing is performed at a standard
frequency of 100 kHz.
6.2. Transformers with Multiple Secondary Windings
6.2.1. Transformers with One-Primary and Multiple-Secondary Windings
The converters in the preceding sections need one or more transformers with ‘k’
number of secondary windings. Therefore, it is necessary to design such a transformer.
The window area allocated for each winding wound on the ferrite core depends on the
Sustainability 2023, 15, 2813
57 of 71
power handling capability of each secondary. Such a transformer is shown in Figure 45a. It
has one primary winding and two secondary windings. Each winding takes a fraction of
the total window area (Aw ). This fraction is denoted as ‘k’ for the kth winding [154] and is
given by Equation (174).
A
k= k
(174)
Aw
where Ak represents the winding area of the kth winding. If a wire of length lk is used for
the kth winding, then the wire resistance Rk will be calculated using Equation (175).
Rk =
lk
Awk
(175)
where Awk represents the area of wire used in the kth winding. It is obtained from the wire
gauge table. The total number of turns of the kth winding (nk ) will be inversely proportional
to Awk and the actual length of the wire based on parameter αk (fractional area of the total
winding area) will be estimated using Equation (176).
lk ≡ nk × ( MLT )K
(176)
where MLT is the mean length per turn of the wire. The new resistance can then be
calculated by taking the wire fill factor (ku = 0.3 for Litz wire and 0.9 for foil), as,
Rk =
n2k × ( MLT )k
WA × k u
(177)
For detailed and generic analysis of magnetics for power converters, works [155–157]
can be referred to. The detailed design of transformers with multiple secondaries is out
of the scope of this review. Ref. [154] explains in detail the design and analysis of the
transformer with single primary and multiple secondary windings.
6.2.2. Use of Multiple Transformers to Obtain Equivalent Transformer with Multiple
Secondary Windings
If a transformer described in the previous subsection is unavailable, there is another
way to obtain the same effect. If two separate transformers are available, then the primaries
of both transformers can be connected in parallel to obtain an equivalent transformer with
two isolated secondaries. This is shown in Figure 45b.
Using a single transformer with multiple secondaries has advantages such as lesser
space requirement and relatively smaller size. It weighs less than the setup with multiple
transformers with their primaries in parallel. On the other hand, if a setup with multiple
primary windings is used, each transformer can have a different turns ratio based on
availability (e.g., n p1 : ns1 = 1:2 and n p2 : ns2 = 2:3), thereby enabling charging batteries of
different voltage ratings. Based on the availability and other constraints, either type can be
used in the converters employed to charge multiple batteries.
Figure 45. Multiple secondary transformers: (a) One primary and two isolated secondaries, and
(b) multiple primaries in parallel with isolated secondaries.
Sustainability 2023, 15, 2813
58 of 71
6.3. Testing of High-Frequency Transformers Used in Isolated DC-DC Converters
All the isolated DC-DC Converters use high-frequency transformers. However, researchers and engineers should focus on the following critical points.
1.
2.
The transformer response must be checked for square wave input before it is connected
to any converter;
The transformer response for step-up and step-down operations must be checked.
The response check is vital because the voltage input to the transformer in any DC-DC
converter is a square wave, not a sinusoidal wave. Even in resonant converters such as
full bridge LLC and CLLC types, the voltage is a square while the current is sinusoidal.
The response of the transformer can be quite different when square wave voltage input
is given to it instead of sinusoidal input. The testing needs to be done for both sides
since a confirmation regarding the quality of the response is required for both step-up
and step-down operations. It is customary to assume that a step-up transformer works
as a step-down transformer if the supply and load terminals are interchanged. However,
the frequency range needs to be verified before connecting it as a part of any converter.
A basic step-by-step procedure for high-frequency transformer response testing suggested
by the authors based on their research and testing experience is as follows:
1.
2.
3.
4.
5.
6.
7.
8.
Connect the square wave input source (signal generator) terminals to the transformer’s primary;
Keeping the transformer on no-load, connect the positive and negative terminals of the probe (from the scope where the waveform needs to be seen) to the
secondary terminals;
Set a frequency in the input source;
After checking the transformer ratio (step up or step down, and the peak value of
the voltage increase or decrease), provide the square wave signal of appropriate
magnitude from the source;
Observe the response on the scope;
Change the frequency of the input signal and observe the waveform again;
Repeat the steps for various frequencies and obtain the range of frequencies for which
the transformer gives the best response. The transformer under test (TUT) should be
used only in that frequency range to obtain the best response and efficiency;
Now, interchange the supply and scope terminals (primary to secondary and secondary to
primary) and repeat the steps to obtain the frequency range for this operation.
A setup to perform such testing is shown in Figure 46. The square wave is provided
from the Digilent Analog Discovery 2 processor using its ‘Wavegen’ mode, and the response
is seen in the Keysight IntegraVision Power Analyzer PA2203A. In the waveforms shown
in Figures to, VPrimary represents the voltage across the primary winding, and VSecondary
represents the voltage across the seconday winding of the transformer.
Figure 47 shows the input (5 V, 20 kHz square wave) given to the primary windings of
the 100 V, 40 A/300 V, 12 A high-frequency transformer, and the output voltage across the
secondary. In the output waveform, a slope can be seen. The waveform pattern indicates
that the square wave input does not give square wave output, which is unacceptable in a
converter. Hence the TUT is not suitable for operation at that switching frequency (20 kHz
here). It is seen that the input wave also gets distorted at the frequencies at which the
transformer is not magnetically compatible to work.
Figure 48 show the 5 V, 100 kHz square wave input, and the output voltage. It is
observed here that the output pattern is very close to that of a square wave, implying that
the TUT works well at this frequency (100 kHz here).
Sustainability 2023, 15, 2813
59 of 71
Figure 46. High frequency transformer test setup.
Figure 47. Transformer testing: square wave input of 5 V, 20 kHz given to primary winding and the
corresponding output voltage obtained across the secondary.
Similar tests were performed by providing the input to the terminals labeled ‘secondary’ and output voltage measured across the terminals labeled ‘primary’. This is
done to check the operational frequency range of the transformer for the operation when
the primary and secondary terminals are interchanged. Figure 49 shows the waveforms
for a 50 kHz signal applied to the secondary winding and its response across the primary winding.
Sustainability 2023, 15, 2813
60 of 71
Figure 48. Transformer testing: square wave input of 5 V, 100 kHz given to primary winding and the
corresponding output voltage obtained across the secondary.
Figure 49. Transformer testing: square wave input of 5 V, 5 kHz given to secondary winding and the
corresponding output voltage obtained across the secondary.
Similarly, when a 100 kHz square wave was provided to the secondary and the
response was measured across the primary, the input and the corresponding response
obtained are shown in Figure 50.
Similar tests can be performed for this combination, and the range of frequencies
can be noted. For the TUT shown in Figure 46, the test was carried out, and the range of
frequencies for the step-down operation was found to be between 60 kHz and 270 kHz,
while the range for step-up operation of the same transformer was found to be between
45 kHz and 295 kHz. It can be seen from Figures 47 and 49 that even the voltage across
the primary gets distorted when operated at low frequencies, even when a square wave
Sustainability 2023, 15, 2813
61 of 71
with no disturbance is applied across it. Since the magnetic circuit does not give a proper
response at that frequency, the secondary voltage gets distorted and reflects on the primary
side. This needs to be noted before the HFT is used in any converter.
Figure 50. Transformer testing: square wave input of 5 V, 100 kHz given to secondary winding and
the corresponding output voltage obtained across the secondary.
7. High-Voltage Batteries in Electric Vehicles
The EV charging systems in passenger vehicles could be classified into two main
types [158] based on the battery voltage levels. The systems which operate in the voltage
range of 150 to 450 V are called the 400 V system. Some vehicles in this range include the
Audi e-Tron (396 V), Mercedes Benz (EQC-405 V, EQA and EQB-420 V), Nissan Leaf (350 V),
Tesla S (400 V), Volkswagen (e-Up-374 V) [159] among others. High-end vehicles such as
Porsche Taycan (800 V), Aston Martin Rapide E (800 V), and Kia EV6 (697 V) [159] in which
the battery voltage can reach up to 850 V, come under the 800 V system. With the use of
800 V systems, there are multiple benefits, especially related to fast charging. With the
use of an 800 V system, the size of the cables can be reduced, along with the reduction in
conduction losses, lower cooling requirement (relative to the 400 V systems), and smaller
motor and wire gauge due to lower currents [160].
Jung et al. [158] have described the benefits of higher voltage batteries when used in
EVs. The article discusses the technical constraints, EV layout and design changes, their
standardization, corresponding connectors, and their relation to fast charging. The article
concludes by describing the advantages customers get due to the upgradation of battery
voltages. Aghabali et al. [160] have provided a detailed review of the 800 V vehicle systems
for EV batteries, along with the various challenges and future trends of the system in
commercial use. The researchers have analyzed the performance of the system based on
the battery pack, the cables used (for conventional and fast charging), BMS, the protection
system and the clearance requirements, considering the Tesla Model 3 and Porsche Taycan
Turbo S vehicles which use the 400 V and 800 V battery packs, respectively. The thermal
concerns and risk evaluation with the partial discharge of battery packs have been considered in this work. On the inverter side, the required configurations for the 400 V and
800 V systems have been detailed with case studies, current calculations, and loss analysis.
For the auxiliary power unit (APU), the full-bridge converter with soft switching, DAB,
and full-bridge with current doubler are considered for the 400 V system. At the same time,
Sustainability 2023, 15, 2813
62 of 71
a modular structure is proposed for the 800 V system. Charger configuration has been
provided for the 400 V and 800 V EVs, and the future trends have been explained.
Grazian et al. [159] have proposed a voltage/current doubler circuit for inductive
wireless power transfer for 400 V and 800 V battery applications. The authors have provided
the modeling, component selection, analysis, and economics details. Hardware results
have been provided by considering a 7.7 kW prototype with bidirectional power supplies.
The performance analysis for coil misalignment and coil distance, with the loss analysis,
has been provided. Efficiency greater than 95% has been obtained for all test cases in the
work. Among the isolated converters described in the preceding sections, the full-bridge
topology, phase-shifted full bridge, and resonant topologies are best suited for 400 V and
800 V battery charging applications.
8. Simultaneous Charging Feature: Implementation in Products and Public Projects
ABB has launched a charger called “Terra 360” [161], which is a high-power charger
that supports the combined charging system (CCS, up to 360 kW), CHAdeMO (up to
150 kW) and AC type-2 (up to 22 kW) charging protocols. The DC output voltage ranges
from 200 V to 920 V. It provides two DC outputs and two more options for DC outputs,
thus enabling the simultaneous charging of four vehicles of different ratings.
Electric Vehicle Energy Storage Company (EVESCO) has launched its “EVDC-60NA”
charger [162] of the EVDC series, which has the capability of charging two EVs at a time.
The system can supply a DC power of 60 kW with 200 to 1000 V output. The rated input is
three-phase 480 V, 60 Hz AC. It has an inbuilt AC-DC converter and supports both CCS
and CHAdeMO protocols.
Volkswagen has created a pilot project in Saxony [163] where a mega power bank has
been assembled using EV batteries that are no longer suitable for use in EVs. In this pilot
project, up to eight vehicles can be charged simultaneously with a total output of 75 kW.
The source is a 570 kWh capacity solar panel installed on the site.
Proterra’s Proterra 1440 kW Charging System [164] is a large fleet solution custombuilt to charge 24 vehicles simultaneously and up to 48 vehicles sequentially. It works on a
480 V, 60 Hz AC input at a power factor greater than 0.99 and a total harmonic distortion of
less than 3%. The maximum continuous DC current is 200 A with an optional extension
to 300 A based on the vehicle limitation. The DC output voltage is a large-range output
between 150 V and 1000 V.
The United States Transportation Department (USDOT) has proposed the installation
of fast chargers that can charge four vehicles simultaneously, with each port having a
capacity of 150 kW or higher [165]. The total budget allotted for the proposed project is
around USD 5 billion.
ČEZ public fast chargers at Vestec, Central Bohemia, were tested under a complete
islanding operation for two hours [166] to charge 12 Renault ZOE EVs simultaneously.
The system used a PV source with a capacity of 20 kW and a battery bank with a capacity
of 275 kWh. The ENET VŠB Centre-Technical University of Ostrava, the Brno University of
Technology and ABB supported the project.
City Charger from Delta [167] is a 100 kW EV fast charging system that supports
simultaneous charging outputs with a feature that can optimize the charging rate during
simultaneous charging. It supports CCS and CHAdeMO protocols and operates from a
480 V, 50 Hz AC input at a power factor of 0.99 with an efficiency greater than 94% at
full-load. The current THD satisfies the IEEE 519 standards.
The Firmer Electric DC charging system [168] is a combined charging system with AC
and DC charging features that can charge three vehicles simultaneously. It can dynamically
distribute power optimally to all EVs. The maximum power capacity is 150 kW for the DC
side and 43 kW for the AC side.
While all the products/projects described in Section 8 boast the feature of simultaneous charging, no specific details have been provided in the corresponding references
regarding the number and type of converters used in each charging system and their
Sustainability 2023, 15, 2813
63 of 71
components. By simultaneous charging, if the implication is the installation of ‘k’ number
of charging systems to charge ‘k’ batteries, the system can technically charge multiple
batteries. However, the waiting time will still be high. This scenario should not be confused
with the case where multiple batteries can be charged from one charger, the latter case
being more beneficial than the former from the standpoint of reduction of waiting time.
9. Future Scope of Simultaneous Charging
The future of Electric Vehicles is bright. In this regard, the importance of simultaneous
charging cannot be overstated. Whether the EV charging systems deployment involves
grid-connected or grid-independent systems, the scope is seemingly endless.
1.
2.
The source side challenges involve integrating fuel cells, solar PV systems, and the
grid. Some research has been conducted in this regard, but the implementation is yet
to be done at the product level for mass use. The development of optimized source
combinations for slow and fast charging by integrating conventional and renewable
energy sources is to be taken to reduce the dependency on conventional sources;
With the addition of renewable sources, the intermittency of power availability needs
to be addressed. This poses a significant challenge when grid independence is an
objective. Efficient storage systems need to be integrated to ensure round-the-clock
power availability.
On the converter topology side, some of the research possibilities include the following:
1.
2.
3.
4.
Optimizing existing topologies to reduce the system’s size, weight, and the cost is
a significant research problem. While research works have proposed several new
converters, those topologies have yet to reach implementation level in actual chargers. This gap has to be bridged by optimizing the converter in terms of topology
and performance;
With the advent of SiC and GaN devices, the converters can be switched at very
high frequencies, reducing the size of magnetic components and reducing passive
component losses. The design of appropriate gate driver circuits for SiC and GaN
MOSFETs and protection circuits for these devices is a significant research area with
tremendous untapped potential. The design of proper driving circuits will directly
affect the switching characteristics of the devices, thereby affecting the losses;
Optimization of the layout of PCB with the use of SiC and GaN devices with proper
clearances is another area that has to be addressed. This will be very critical with the
implementation of a simultaneous charging feature;
With HV batteries, topologies used at these voltages must be developed and optimized
for fast and ultra-fast charging.
On the converter control side, some of the research possibilities include the following:
1.
2.
Universal control techniques optimized specifically for battery charging must be
developed and implemented for standard power levels;
Newer techniques that involve deep neural networks and fuzzy-neural systems need
to be employed to make the charger more intelligent.
In addition to these changes, one can look into ways of encouraging the widespread
use of EVs. For this, it is necessary to reduce the waiting time and charging time, along
with deploying sufficient charging systems. This implies that future research should deal
with fast and ultra-fast charging methods and simultaneous charging topologies to address
the time-related concerns. One should look into the possibility of deploying more charging
systems to address the possibility of increasing the number of chargers. Another emerging
field for research in simultaneous charging is wireless charging systems.
1.
Deployment of WPT systems, specifically dynamic charging systems, can be revolutionary. Dynamic charging eliminates the requirement of parking and plugging-in of
vehicles for charging and the associated wiring system. In addition, the range anxiety
of the users can be reduced since charging can be done on the go;
Sustainability 2023, 15, 2813
64 of 71
2.
3.
4.
The two main challenges with wireless power transfer include relatively lower efficiency and coil separation distance limitation. This needs to be addressed by researchers to enable mass use;
Another modification that can be made is to use LLC and CLLC type converters
(full-bridge variants or DAB) with wireless coils instead of transformers. This will
ensure ZVS and ZCS operation, leading to higher efficiency;
If this can be extended to multiple-output chargers to charge multiple batteries of
the same or different ratings simultaneously, then a further boost is provided to
the implementation.
It can be concluded that the implementation of simultaneous charging of EVs will
lead to a greener and cleaner earth (Figure 51), and easier and faster charging with lower
waiting times, which will lead to an overall increase in EV usage.
Figure 51. Pathway of electric vehicle development.
10. Conclusions
This article has presented the state of the art of simultaneously charging batteries for
EV applications. The review has given details of the charging methods, various charging
techniques, standards, and charging levels for EV batteries. The specifications of commercial chargers are presented for both AC and DC charging. Primary concerns related to
power quality with battery charging have been reviewed in this paper. The state of the art
of simultaneous charging of EV batteries has been presented to enable the researchers to
implement newer topologies for the application. Several existing topologies of isolated
DC-DC converters that can be extended to implement simultaneous charging have been
presented with the related design and loss estimation equations. The extensions include
single-switch, two-switch, and bridge-type isolated DC-DC converters, which are suitable
for charging EV batteries that need power at different levels and charging at different
speeds limited by the batteries. The suitability of these converters for EV battery charging
has been described to enable a better choice.
A testing method for HFTs has been provided to enable researchers to decide the
frequency range at which the TUT performs well. The recent advancement in the use of HV
batteries in EVs has been explained with the advantages and details of several EVs in the
market that use HV batteries. The future scope of simultaneous charging has been detailed
for the benefit of the readers to help guide the next phase of research in the domain.
Sustainability 2023, 15, 2813
65 of 71
It can be seen from the review that resonant and phase-shifted full-bridge converters
are best suited for high-power, high-voltage battery charging, and fast charging. Singleswitch and two-switch topologies can be used for low- and medium-power EV charging,
essential in charging stations that implement simultaneous charging. A charging station
that implements simultaneous charging should combine low- and high-power compatible
converters so that vehicles of different power and voltage ratings can be charged simultaneously. This will drive the majority of future research in EV battery charging. An optimal
combination of topologies suitable for low- and high-power EV charging will be the future
of research and implementation for EV charging globally.
Author Contributions: Conceptualization, S.B.S. and G.A.; testing, S.B.S. and G.A.; resources, S.B.S.
and G.A.; data curation, S.B.S. and G.A.; writing—original draft preparation, S.B.S.; writing—review
and editing, S.B.S. and G.A.; visualization, S.B.S.; supervision, G.A.; project administration, G.A. All
authors have read and agreed to the published version of the manuscript.
Funding: This research received no external funding.
Institutional Review Board Statement: Not applicable.
Informed Consent Statement: Not applicable.
Data Availability Statement: Not applicable.
Acknowledgments: The transformer testing shown was performed at the DST-FIST Sponsored Smart
Grid Setup-PHIL Lab at the School of Electrical Engineering, Vellore Institute of Technology, Vellore.
Conflicts of Interest: The authors declare no conflict of interest.
References
1.
2.
3.
4.
5.
6.
7.
8.
9.
10.
11.
12.
13.
14.
15.
16.
Reitz, R.D.; Ogawa, H.; Payri, R.; Fansler, T.; Kokjohn, S.; Moriyoshi, Y.; Agarwal, A.; Arcoumanis, D.; Assanis, D.; Bae, C.; et al.
IJER editorial: The future of the internal combustion engine. Int. J. Engine Res. 2020, 21, 3–10. [CrossRef]
Guttikunda, S.K.; Jawahar, P. Atmospheric emissions and pollution from the coal-fired thermal power plants in India. Atmos.
Environ. 2014, 92, 449–460. [CrossRef]
Holechek, J.L.; Geli, H.M.; Sawalhah, M.N.; Valdez, R. A global assessment: Can renewable energy replace fossil fuels by 2050?
Sustainability 2022, 14, 4792. [CrossRef]
Höök, M.; Tang, X. Depletion of fossil fuels and anthropogenic climate change—A review. Energy Policy 2013, 52, 797–809.
[CrossRef]
Botelho, A.; Ferreira, P.; Lima, F.; Pinto, L.M.C.; Sousa, S. Assessment of the environmental impacts associated with hydropower.
Renew. Sustain. Energy Rev. 2017, 70, 896–904. [CrossRef]
Singh, A.; Singha, N. Environmental impact of nuclear power: Law and policy measures in India. Humanit. Soc. Sci. Rev. 2016,
4, 88–95. [CrossRef]
Nasr Esfahani, F.; Darwish, A.; Williams, B.W. Power converter topologies for grid-tied solar photovoltaic (PV) powered electric
vehicles (EVs)—A comprehensive review. Energies 2022, 15, 4648. [CrossRef]
Jin, C.; Sheng, X.; Ghosh, P. Optimized electric vehicle charging with intermittent renewable energy sources. IEEE J. Sel. Top.
Signal Process. 2014, 8, 1063–1072. [CrossRef]
Al-Shahri, O.A.; Ismail, F.B.; Hannan, M.; Lipu, M.H.; Al-Shetwi, A.Q.; Begum, R.; Al-Muhsen, N.F.; Soujeri, E. Solar photovoltaic
energy optimization methods, challenges and issues: A comprehensive review. J. Clean. Prod. 2021, 284, 125465. [CrossRef]
Notter, D.A.; Gauch, M.; Widmer, R.; Wager, P.; Stamp, A.; Zah, R.; Althaus, H.J. Contribution of Li-ion batteries to the
environmental impact of electric vehicles. Environ. Sci. Technol. 2010, 44, 6550–6556. [CrossRef]
Yu, X.; Li, W.; Gupta, V.; Gao, H.; Tran, D.; Sarwar, S.; Chen, Z. Current Challenges in Efficient Lithium-Ion Batteries’ Recycling: A
Perspective. Glob. Challenges 2022, 6, 2200099. [CrossRef]
Dalala, Z.M.; Alnawafa, M.; Saadeh, O.; Alnawafa, E. Reducing commuter CO2 footprint through transit PV electrification.
Sustainability 2020, 12, 6406. [CrossRef]
Aziz, M.; Oda, T. Simultaneous quick-charging system for electric vehicle. Energy Procedia 2017, 142, 1811–1816. [CrossRef]
Shahjalal, M.; Shams, T.; Tasnim, M.N.; Ahmed, M.R.; Ahsan, M.; Haider, J. A critical review on charging technologies of electric
vehicles. Energies 2022, 15, 8239. [CrossRef]
Guo, S.; Han, Z.; Wei, J.; Guo, S.; Ma, L. A Novel DC-AC Fast Charging Technology for Lithium-Ion Power Battery at LowTemperatures. Sustainability 2022, 14, 6544. [CrossRef]
Shadnam Zarbil, M.; Vahedi, A.; Azizi Moghaddam, H.; Khlyupin, P.A. Design and Sizing of Electric Bus Flash Charger Based on
a Flywheel Energy Storage System: A Case Study. Energies 2022, 15, 8032. [CrossRef]
Sustainability 2023, 15, 2813
17.
18.
19.
20.
21.
22.
23.
24.
25.
26.
27.
28.
29.
30.
31.
32.
33.
34.
35.
36.
37.
38.
39.
40.
41.
42.
43.
66 of 71
Switch-Mode, Linear, and Pulse Charging Techniques for Li+ Battery in Mobile Phones and PDAs. Available online:
https://www.analog.com/en/technical-articles/switchmode-linear-and-pulse-charging-techniques-for-li-battery-in-mobilephones-and-pdas.html (accessed on 15 November 2022).
Vermeer, W.; Stecca, M.; Mouli, G.R.C.; Bauer, P. A Critical Review on The Effects of Pulse Charging of Li-ion Batteries. In
Proceedings of the 2021 IEEE 19th International Power Electronics and Motion Control Conference (PEMC), Gliwice, Poland,
25–29 April 2021; pp. 217–224. [CrossRef]
Huang, X.; Li, Y.; Acharya, A.B.; Sui, X.; Meng, J.; Teodorescu, R.; Stroe, D.I. A review of pulsed current technique for lithium-ion
batteries. Energies 2020, 13, 2458. [CrossRef]
Khan, A.B.; Pham, V.L.; Nguyen, T.T.; Choi, W. Multistage constant-current charging method for Li-ion batteries. In Proceedings
of the 2016 IEEE Transportation Electrification Conference and Expo, Asia-Pacific (ITEC Asia-Pacific), Busan, Republic of Korea,
1–4 June 2016; pp. 381–385.
Liu, P.J.; Chen, T.F.; Yang, H.S. A Li-Ion Battery Charger with Variable Charging Current and Automatic Voltage-Compensation
Controls for Parallel Charging. IEEE J. Emerg. Sel. Top. Power Electron. 2021, 10, 997–1006. [CrossRef]
Notten, P.H.; het Veld, J.O.; Van Beek, J. Boostcharging Li-ion batteries: A challenging new charging concept. J. Power Sources
2005, 145, 89–94. [CrossRef]
Tomaszewska, A.; Chu, Z.; Feng, X.; O’kane, S.; Liu, X.; Chen, J.; Ji, C.; Endler, E.; Li, R.; Liu, L.; et al. Lithium-ion battery fast
charging: A review. ETransportation 2019, 1, 100011. [CrossRef]
Faria, J.P.; Velho, R.L.; Calado, M.R.; Pombo, J.A.; Fermeiro, J.B.; Mariano, S.J. A New Charging Algorithm for Li-Ion Battery
Packs Based on Artificial Neural Networks. Batteries 2022, 8, 18. [CrossRef]
Hemavathi, S.; Shinisha, A. A study on trends and developments in electric vehicle charging technologies. J. Energy Storage 2022,
52, 105013. [CrossRef]
Gao, Z.; Ma, B.; Liu, X.; Chen, S.; Xie, H.; Yu, H. Study on Lithium-ion Battery Degradation Caused by Side Reactions in
Fast-charging Process. Front. Energy Res. 2022, 10, 905710. [CrossRef]
Khalid, A.; Sarwat, A.I. Fast Charging Li-Ion Battery Capacity Fade Prognostic Modeling Using Correlated Parameters’ Decomposition and Recurrent Wavelet Neural Network. In Proceedings of the 2021 IEEE Transportation Electrification Conference &
Expo (ITEC), Chicago, IL, USA, 23–25 June 2021; pp. 27–32.
Al-Saadi, M.; Olmos, J.; Saez-de Ibarra, A.; Van Mierlo, J.; Berecibar, M. Fast Charging Impact on the Lithium-Ion Batteries’
Lifetime and Cost-Effective Battery Sizing in Heavy-Duty Electric Vehicles Applications. Energies 2022, 15, 1278. [CrossRef]
Wang, L.; Qin, Z.; Slangen, T.; Bauer, P.; van Wijk, T. Grid impact of electric vehicle fast charging stations: Trends, standards,
issues and mitigation measures-an overview. IEEE Open J. Power Electron. 2021, 2, 56–74. [CrossRef]
Rwamurangwa, E.; Gonzalez, J.D.; Butare, A. Integration of EV in the Grid Management: The Grid Behavior in Case of
Simultaneous EV Charging-Discharging with the PV Solar Energy Injection. Electricity 2022, 3, 563–585. [CrossRef]
Filip, R.; Püvi, V.; Paar, M.; Lehtonen, M. Analyzing the Impact of EV and BESS Deployment on PV Hosting Capacity of
Distribution Networks. Energies 2022, 15, 7921. [CrossRef]
Alshareef, S.M. Analyzing and Mitigating the Impacts of Integrating Fast-Charging Stations on the Power Quality in Electric
Power Distribution Systems. Sustainability 2022, 14, 5595. [CrossRef]
Chon, S.; Bhardwaj, M.; Nene, H. Maximizing Power for Level 3 EV Charging Stations. 2018. Available online: https://www.ti.
com/lit/wp/sway014/sway014.pdf?ts=1674470563613&ref_url=https%253A%252F%252Fwww.google.com%252F (accessed on
15 November 2022).
Kumar K, J.; Kumar, S.; VS, N. Standards for electric vehicle charging stations in India: A review. Energy Storage 2022, 4, e261.
[CrossRef]
Nakanishi, T.; Zaitsu, H.; Kikuta, T.; Tsuda, S.; Nii, H. CHAdeMO-Conformity High-Power Charger Connector Assembly for
Over 100 kW-Class EV Charge. SEI Tech. Rev. 2019, 88, 49–54.
Jaman, S.; Verbrugge, B.; Garcia, O.H.; Abdel-Monem, M.; Oliver, B.; Geury, T.; Hegazy, O. Development and Validation of V2G
Technology for Electric Vehicle Chargers Using Combo CCS Type 2 Connector Standards. Energies 2022, 15, 7364. [CrossRef]
AC Charger. Eaton xChargeIn Mobility. Available online: https://www.eaton.com/gb/en-gb/catalog/emobility/xchargeinmobility.specifications.html (accessed on 18 December 2022).
DC Charger. Eaton Green Motion DC 22. Available online: https://www.eaton.com/gb/en-gb/catalog/emobility/greenmotion-dc-22.specifications.html (accessed on 18 December 2022).
DC Charger. Eaton Green Motion DC 44/66. Available online: https://www.eaton.com/gb/en-gb/catalog/emobility/greenmotion-dc-44-66.specifications.html (accessed on 18 December 2022).
DC Charger. Siemens SiCharge. Available online: https://new.siemens.com/global/en/products/energy/medium-voltage/
solutions/emobility/sicharge-d.html (accessed on 18 December 2022).
AC Charger. Siemens VersiCharge. Available online: https://new.siemens.com/global/en/products/energy/medium-voltage/
solutions/emobility/versicharge.html (accessed on 18 December 2022).
Khalid, M.; Ahmad, F.; Panigrahi, B.K.; Al-Fagih, L. A comprehensive review on advanced charging topologies and methodologies
for electric vehicle battery. J. Energy Storage 2022, 53, 105084. [CrossRef]
Ghasemi-Marzbali, A. Fast-charging station for electric vehicles, challenges and issues: A comprehensive review. J. Energy Storage
2022, 49, 104136.
Sustainability 2023, 15, 2813
44.
45.
46.
47.
48.
49.
50.
51.
52.
53.
54.
55.
56.
57.
58.
59.
60.
61.
62.
63.
64.
65.
66.
67.
68.
69.
67 of 71
Chakraborty, S.; Vu, H.N.; Hasan, M.M.; Tran, D.D.; Baghdadi, M.E.; Hegazy, O. DC-DC converter topologies for electric vehicles,
plug-in hybrid electric vehicles and fast charging stations: State of the art and future trends. Energies 2019, 12, 1569. [CrossRef]
Agamy, M.S.; Harfman-Todorovic, M.; Elasser, A.; Steigerwald, R.L.; Sabate, J.A.; Chi, S.; McCann, A.J.; Zhang, L.; Mueller, F. A
high efficiency DC-DC converter topology suitable for distributed large commercial and utility scale PV systems. In Proceedings
of the 2012 15th International Power Electronics and Motion Control Conference (EPE/PEMC), Novi Sad, Serbia, 4–6 September
2012; p. LS2d–3.
Karmaker, A.K.; Roy, S.; Ahmed, M.R. Analysis of the impact of electric vehicle charging station on power quality issues. In
Proceedings of the 2019 International Conference on Electrical, Computer and Communication Engineering (ECCE), Cox’s Bazar,
Bangladesh, 7–9 February 2019; pp. 1–6.
IEC61000-3-2; Electromagnetic Compatibility (EMC)-Part 3-2: Limits—Limits for Harmonic Current Emissions (Equipment Input
Current ≤ 16 A per Phase. IEC Standard: Geneva, Switzerland, 2018; pp. 1000–1003.
Monolithic Power. Power Factor Correction (PFC). Available online: https://www.monolithicpower.com/en/power-factorcorrection (accessed on 3 December 2022).
Kolar, J.W.; Friedli, T. The essence of three-phase PFC rectifier systems—Part I. IEEE Trans. Power Electron. 2012, 28, 176–198.
[CrossRef]
Friedli, T.; Hartmann, M.; Kolar, J.W. The essence of three-phase PFC rectifier systems—Part II. IEEE Trans. Power Electron. 2013,
29, 543–560. [CrossRef]
Figueiredo, J.P.M.; Tofoli, F.L.; Silva, B.L.A. A review of single-phase PFC topologies based on the boost converter. In Proceedings
of the 2010 9th IEEE/IAS International Conference on Industry Applications-INDUSCON 2010, São Paulo, Brazil, 8–10 November
2010; pp. 1–6.
Semiconductor, O. Power Factor Correction Handbook; HBD853/D, Rev; Newark Electronics: Chicago, IL, USA, 2007; Volume 3.
Efthymiou, L.; Camuso, G.; Longobardi, G.; Udrea, F.; Lin, E.; Chien, T.; Chen, M. Zero reverse recovery in SiC and GaN Schottky
diodes: A comparison. In Proceedings of the 2016 28th International Symposium on Power Semiconductor Devices and ICs
(ISPSD), Prague, Czech Republic, 12–16 June 2016; pp. 71–74.
Huber, L.; Jang, Y.; Jovanovic, M.M. Performance evaluation of bridgeless PFC boost rectifiers. IEEE Trans. Power Electron. 2008,
23, 1381–1390. [CrossRef]
Jang, Y.; Jovanovic, M.M. A bridgeless PFC boost rectifier with optimized magnetic utilization. IEEE Trans. Power Electron. 2009,
24, 85–93. [CrossRef]
Sharifi, S.; Monfared, M.; Babaei, M. Ferdowsi rectifiers—Single-phase buck-boost bridgeless PFC rectifiers with low semiconductor count. IEEE Trans. Ind. Electron. 2019, 67, 9206–9214. [CrossRef]
Lange, A.D.B.; Soeiro, T.B.; Ortmann, M.S.; Heldwein, M.L. Three-level single-phase bridgeless PFC rectifiers. IEEE Trans. Power
Electron. 2014, 30, 2935–2949. [CrossRef]
Ge, K.; Liu, Q. Research on dual boost semi-bridgeless PFC converter. In Advances in Energy Materials and Environment Engineering;
CRC Press: Boca Raton, FL, USA, 2022; pp. 313–318.
Babaei, M.; Monfared, M. High Step-Down Bridgeless Sepic/Cuk PFC Rectifiers with Improved Efficiency and Reduced Current
Stress. IEEE Trans. Ind. Electron. 2022, 69, 9984–9991. [CrossRef]
Yu, Z.; Xia, Y.; Ayyanar, R. A simple ZVT auxiliary circuit for totem-pole bridgeless PFC rectifier. IEEE Trans. Ind. Appl. 2019,
55, 2868–2878. [CrossRef]
Huang, Q.; Ma, Q.; Liu, P.; Huang, A.Q.; de Rooij, M. 3kW four-level flying capacitor totem-pole bridgeless PFC rectifier with
200V GaN devices. In Proceedings of the 2019 IEEE Energy Conversion Congress and Exposition (ECCE), Baltimore, MA, USA,
29 September–3 October 2019; pp. 81–88.
Su, B.; Zhang, J.; Lu, Z. Totem-pole boost bridgeless PFC rectifier with simple zero-current detection and full-range ZVS operating
at the boundary of DCM/CCM. IEEE Trans. Power Electron. 2010, 26, 427–435. [CrossRef]
Do, N.N.; Huang, B.S.; Phan, N.T.; Nguyen, T.T.; Wu, J.H.; Liu, Y.C.; Chiu, H.J. Design and Implementation of a Control Method
for GaN-Based Totem-Pole Boost-Type PFC Rectifier in Energy Storage Systems. Energies 2020, 13, 6297. [CrossRef]
Kanimozhi, G.; Natrayan, L.; Angalaeswari, S.; Paramasivam, P. An Effective Charger for Plug-In Hybrid Electric Vehicles (PHEV)
with an Enhanced PFC Rectifier and ZVS-ZCS DC/DC High-Frequency Converter. J. Adv. Transp. 2022, 2022, 1–14. [CrossRef]
Power Factor Correction (PFC) Topology Comparison. 2017. Available online: https://training.ti.com/power-factor-correctionpfc-topology-comparison (accessed on 3 December 2022).
Vienna Rectifier-Based, Three-Phase Power Factor Correction (PFC) Reference Design Using C2000 MCU. 2017. Available online:
https://www.ti.com/lit/ug/tiducj0b/tiducj0b.pdf (accessed on 3 December 2022).
Kayisli, K. Hysteresis control of a boost pfc converter circuit. In Proceedings of the FAE Symposium, Lefke, Cyprus, 30
November–1 December 2006.
Lekić-Vervoort, A.; Majstorović, M.; Ristić, L.; Stipanović, D. Hysteresis Control of the Pseudo Boost PFC Converter. In
Proceedings of the 2020 IEEE 29th International Symposium on Industrial Electronics (ISIE), Delft, The Netherlands, 17–19 June
2020; pp. 731–735.
Choudhury, S. Average Current Mode Controlled Power Factor Correction Converter Using TMS320LF2407A; Texas Instruments
Application Note SPRA902A; Texas Instruments: Dallas, TX, USA, 2005; pp. 1–14.
Sustainability 2023, 15, 2813
70.
71.
72.
73.
74.
75.
76.
77.
78.
79.
80.
81.
82.
83.
84.
85.
86.
87.
88.
89.
90.
91.
92.
93.
94.
68 of 71
Frgal, P. Average Current Mode Interleaved PFC Control. 2016. Available online: https://www.nxp.com/docs/en/applicationnote/AN5257.pdf (accessed on 12 December 2022).
Mahmud, K.; Tao, L. Power factor correction by PFC boost topology using average current control method. In Proceedings of the
2013 IEEE Global High Tech Congress on Electronics, Singapore, 11–13 December 2013; pp. 16–20.
Li, L.; Wang, W.; Lyu, D.; Min, R.; Tong, Q.; Peng, H.; Yu, J. Maximum efficiency average current controller based on a
comprehensive charge rate model for DCM boost PFC converter. IEEE Trans. Power Electron. 2020, 36, 6046–6055. [CrossRef]
Abdel-Rahman, S.; Stückler, F.; Siu, K. PFC Boost Converter Design Guide 1200 W Design Example; Application Note; Infineon
Technologies: Neubiberg, Germany, 2016.
Bouafassa, A.; Rahmani, L.; Babes, B.; Bayindir, R. Experimental design of a finite state model predictive control for improving
power factor of boost rectifier. In Proceedings of the 2015 IEEE 15th International Conference on Environment and Electrical
Engineering (EEEIC), Rome, Italy, 10–13 June 2015; pp. 1556–1561.
Bouafassa, A.; Rahmani, L.; Mekhilef, S. Design and real time implementation of single phase boost power factor correction
converter. ISA Trans. 2015, 55, 267–274. [CrossRef]
Brown, R.; Soldano, M. One cycle control IC simplifies PFC designs. In Proceedings of the Twentieth Annual IEEE Applied
Power Electronics Conference and Exposition, Austin, TX, USA, 6–10 March 2005; Volume 2, pp. 825–829.
Brown, R.; Soldano, M. PFC Converter Design with IR1150 One Cycle Control IC; Application Note AN-1077; International Rectifier:
El Segundo, CA, USA, 2005.
Nguyen, V.L.; Tran-Quoc, T.; Bacha, S. Harmonic distortion mitigation for electric vehicle fast charging systems. In Proceedings
of the 2013 IEEE Grenoble Conference, Grenoble, France, 16–20 June 2013; pp. 1–6.
Sharma, G.; Sood, V.K.; Alam, M.S.; Shariff, S.M. Comparison of common DC and AC bus architectures for EV fast charging
stations and impact on power quality. ETransportation 2020, 5, 100066. [CrossRef]
Alame, D.; Azzouz, M.; Kar, N. Assessing and mitigating impacts of electric vehicle harmonic currents on distribution systems.
Energies 2020, 13, 3257. [CrossRef]
Dharmakeerthi, C.; Mithulananthan, N.; Saha, T. Impact of electric vehicle fast charging on power system voltage stability. Int. J.
Electr. Power Energy Syst. 2014, 57, 241–249. [CrossRef]
Alshareef, S.M.; Morsi, W.G. Impact of fast charging stations on the voltage flicker in the electric power distribution systems.
In Proceedings of the 2017 IEEE Electrical Power and Energy Conference (EPEC), Saskatchewan, Canada, 22–25 October 2017;
pp. 1–6.
Mahafzah, K.A.; Obeidat, M.A.; Al-Shetwi, A.Q.; Ustun, T.S. A Novel Synchronized Multiple Output DC-DC Converter Based on
Hybrid Flyback-Cuk Topologies. Batteries 2022, 8, 93. [CrossRef]
Chen, J.; Chen, H.; Zhou, M.; Kumar, L.; Zheng, J. Quadratic Programming-Based Simultaneous Charging Strategy for Battery
Packs of Electric Vehicles. IEEE/ASME Trans. Mech. 2022, 27, 5869–5878. [CrossRef]
Aziz, M.; Oda, T.; Ito, M. Battery-assisted charging system for simultaneous charging of electric vehicles. Energy 2016, 100, 82–90.
[CrossRef]
Okon, Q.; Urquizo, J.; Kondrath, N.; Singh, P. Control Design for 3-Phase Bidirectional Battery Chargers with Multiple Battery
Charging Capabilities for Electric Vehicle Fleet Applications. In Proceedings of the 2021 North American Power Symposium
(NAPS), Virtually, 11–14 April 2021; pp. 01–06.
Li, M.; He, J.; Liang, B.; Han, J. A compact two-stage power converter for flexible multiple-battery charging. In Proceedings of
the 2018 21st International Conference on Electrical Machines and Systems (ICEMS), Jeju, Republic of Korea, 7–10 October 2018;
pp. 2582–2586.
Ramanathan, G.; Bharatiraja, C.; Athikkal, S. Design and Implementation of Modified Z-Source Inverter for Multi-Port Electric
Vehicle Charger. In Proceedings of the 2022 Second International Conference on Power, Control and Computing Technologies
(ICPC2T), Raipur, India, 1–3 March 2022; pp. 1–5.
Graw, J.; Zimmermann, H. Charging multiple batteries using the boost-flyback converter. In Proceedings of the 2012 IEEE
International Energy Conference and Exhibition (ENERGYCON), Florence, Italy, 9–12 September 2012; pp. 963–967.
Sun, F.; Li, S.; Liu, Y.; Xie, C.; Zhao, Q.; Shi, Y. The Power Electronic Transformer based Multi-port DC Charging Station. In
Proceedings of the 2020 12th IEEE PES Asia-Pacific Power and Energy Engineering Conference (APPEEC), Nanjing, China, 20–23
September 2020; pp. 1–5.
Yesheswini, B.P.; Iswarya, S.J.; Amani, B.; Prakash, P.; Sindhu, M. Solar PV charging station for electric vehicles. In Proceedings of
the 2020 International Conference for Emerging Technology (INCET), Belgaum, India, 5–7 June 2020; pp. 1–7.
Mishra, D.; Singh, B.; Panigrahi, B. Implementation of Adaptive Supervisory Control for PV-Integrated Hybrid EV Charger. In
Proceedings of the 2021 National Power Electronics Conference (NPEC), Bhubaneswar, India, 15–17 December 2021; pp. 1–6.
Vu, V.B.; Kamal, L.B.M.; Tay, J.; Pickert, V.; Dahidah, M.; Logenthiran, T.; Phan, V.T. A multi-output capacitive charger for electric
vehicles. In Proceedings of the 2017 IEEE 26th International Symposium on Industrial Electronics (ISIE), Edinburgh, UK, 19–21
June 2017; pp. 565–569.
Fan, S.Y.; Chang, G.K.; Tseng, S.Y. A reflex charger realized by multi-interleaved buck-boost converters. In Proceedings of the
2011 6th IEEE Conference on Industrial Electronics and Applications, Beijing, China, 21–23 June 2011; pp. 1215–1220.
Sustainability 2023, 15, 2813
95.
96.
97.
98.
99.
100.
101.
102.
103.
104.
105.
106.
107.
108.
109.
110.
111.
112.
113.
114.
115.
116.
117.
118.
119.
69 of 71
Vu, V.B.; Phan, V.T.; Nguyen, D.T.; Logenthiran, T.; Naayagi, R. Design and implementation of a multi-output inductive charger
for electric vehicles. In Proceedings of the 2016 IEEE International Conference on Sustainable Energy Technologies (ICSET),
Hanoi, Vietnam, 14–16 November 2016; pp. 414–419.
Chakraborty, S.; Jain, A.K.; Mohan, N. Novel converter topology and algorithm for simultaneous charging and individual cell
balancing of multiple Li-ion batteries. In Proceedings of the INTELEC 2004. 26th Annual International Telecommunications
Energy Conference, Chicago, IL, USA, 19–23 September 2004; pp. 248–253.
Chakraborty, S.; Mohan, N. A comparative study of various single stage PFC converters in implementing novel converter
topology for simultaneous charging and individual cell balancing of multiple Li-ion batteries. In Proceedings of the INTELEC
05-Twenty-Seventh International Telecommunications Conference, Berlin, Germany, 18–22 September 2005; pp. 251–256.
ElMenshawy, M.; Massoud, A. Development of modular DC-DC converters for low-speed electric vehicles fast chargers. Alex.
Eng. J. 2021, 60, 1067–1083. [CrossRef]
Chen, T.; Zhang, X.P.; Wang, J.; Li, J.; Wu, C.; Hu, M.; Bian, H. A review on electric vehicle charging infrastructure development
in the UK. J. Mod. Power Syst. Clean Energy 2020, 8, 193–205. [CrossRef]
Hardman, S.; Jenn, A.; Tal, G.; Axsen, J.; Beard, G.; Daina, N.; Figenbaum, E.; Jakobsson, N.; Jochem, P.; Kinnear, N.; et al. A
review of consumer preferences of and interactions with electric vehicle charging infrastructure. Transp. Res. Part D Transp.
Environ. 2018, 62, 508–523. [CrossRef]
Zhang, Y.; Chen, J.; Cai, L.; Pan, J. Expanding EV charging networks considering transportation pattern and power supply limit.
IEEE Trans. Smart Grid 2019, 10, 6332–6342. [CrossRef]
Monikandan, A.; Chellaswamy, C.; Geetha, T.; Sivaraju, S. Optimized Convolutional Neural Network-Based Capacity Expansion
Framework for Electric Vehicle Charging Station. Int. Trans. Electr. Energy Syst. 2022, 2022, 2915910. [CrossRef]
Litrán, S.P.; Durán, E.; Semião, J.; Díaz-Martín, C. Multiple-Output DC–DC Converters: Applications and Solutions. Electronics
2022, 11, 1258. [CrossRef]
Dostal, F. When the Flyback Converter Reaches Its Limits. Available online: https://www.analog.com/en/technical-articles/
when-the-flyback-converter-reaches-its-limits.html (accessed on 7 November 2022).
Barrenetxea, M.; Baraia-Etxaburu, I.; Larrazabal Bengoetxea, I.; Zubimendi Azaceta, I. Power Electronic Converter Design Handbook;
Mondragon Unibertsitatea: Gipuzkoa, Spain, 2018.
Sau-Bassols, J.; Morel, F.; Sellé, T.; Poullain, S.; Jacquier, F. Methodology to obtain the specifications and perform the sizing of a
power flow controller for meshed HVDC grids. In Proceedings of the 2021 23rd European Conference on Power Electronics and
Applications (EPE’21 ECCE Europe), Ghent, Belgium, 6–10 September 2021.
Advantages of Forward converter, Disadvantages of Forward Converter. Available online: https://www.rfwireless-world.com/
Terminology/Advantages-and-Disadvantages-of-Forward-converter.html (accessed on 11 November 2022).
Cuk, S.M.; Middlebrook, R.D. Dc-to-dc Switching Converter. USA Patent US4184197A, 15 January 1980.
STMicroelectronics. Two Switch Flyback Converter. Available online: https://www.st.com/en/applications/power-suppliesand-converters/two-switch-flyback-converter.html (accessed on 12 November 2022).
Two-Switch Forward Converter: Operation, FOM, and MOSFET Selection Guide. Available online: https://www.vishay.com/
doc/?91616=#:~:text=The%20two%2Dswitch%20forward%20converter%20is%20quite%20popular%20with%20ATX,having%
20no%20body%20diode%20conduction (accessed on 7 December 2022).
Choragudi, V.S.A.K. Analysis and Design of Pulse-Width Modulated Two-Switch Forward DC-DC Converter for Universal
Laptop Adapter. Master’s Thesis, Wright State Univeristy, Dayton, OH, USA, 2011.
Balogh, L. Design Review: 140W, Multiple Output High Density DC/DC Converter. 1997. Available online: https://www.ti.
com/seclit/ml/slup117/slup117.pdf (accessed on 15 December 2022).
Advantages of Push Pull converter and disadvantages of Push Pull Converter. Available online: https://www.rfwirelessworld.com/Terminology/Advantages-and-Disadvantages-of-Push-Pull-converter.html#:~:text=Following%20are%20the%20
drawbacks%20or,one%20of%20the%20main%20disadvantages (accessed on 17 December 2022).
Habib, S.; Khan, M.M.; Abbas, F.; Ali, A.; Faiz, M.T.; Ehsan, F.; Tang, H. Contemporary trends in power electronics converters for
charging solutions of electric vehicles. CSEE J. Power Energy Syst. 2020, 6, 911–929.
AN1348: Si34071 Active Clamp Forward Transformer Design Principles. Available online: https://www.silabs.com/documents/
public/application-notes/an1348-si34071-active-clamp-fwd-xformer-design.pdf (accessed on 17 December 2022).
Understanding and Designing an Active Clamp Current Mode Controlled Converter Using the UCC2897A. Available online: https:
//www.ti.com/lit/an/slua535a/slua535a.pdf?ts=1665210749510&ref_url=https%253A%252F%252Fwww.google.com%252F (accessed on 17 December 2022).
Designing Active-Clamp Forward Converters Using Peak-Current-Mode Controllers; Analog Devices: Wilmington, MA, USA, 2014.
Jinno, M.; Sheen, J.C.; Chen, P.Y. Effects of magnetizing inductance on active-clamped forward converters. In Proceedings of the
25th International Telecommunications Energy Conference, Yokohama, Japan, 23 October 2003; pp. 636–642.
Hassanzadeh, N.; Yazdani, F.; Haghbin, S.; Thiringer, T. Design of a 50 kw phase-shifted full-bridge converter used for fast
charging applications. In Proceedings of the 2017 IEEE Vehicle Power and Propulsion Conference (VPPC), Belfort, France, 14–17
December 2017; pp. 1–5.
Sustainability 2023, 15, 2813
70 of 71
120. Jin, F.; Nabih, A.; Li, Q.; Lee, F.C. A Three Phase CLLC Converter with Improved Planar Integrated Transformer for Fast Charger
Applications. In Proceedings of the 2021 IEEE Fourth International Conference on DC Microgrids (ICDCM), Arlington, VA, USA,
18–21 July 2021; pp. 1–5.
121. Cittanti, D.; Vico, E.; Gregorio, M.; Mandrile, F.; Bojoi, R. Iterative design of a 60 kW all-Si modular LLC converter for electric
vehicle ultra-fast charging. In Proceedings of the 2020 AEIT International Conference of Electrical and Electronic Technologies for
Automotive (AEIT AUTOMOTIVE), Turin, Italy, 18–20 November 2020; pp. 1–6.
122. Sivaprasad, A.; Deepa, K.; Mathew, K. Half bridge converter for battery charging application. Int. J. Eng. Res. Appl 2012,
2, 994–999.
123. Ou, S.Y.; Hsiao, H.P.; Tien, C.H. Analysis and design of a prototype single-stage half-bridge power converter. In Proceedings of
the 2010 5th IEEE Conference on Industrial Electronics and Applications, Taichung, Taiwan, 15–17 June 2010; pp. 1168–1173.
124. Hyeon, B.C.; Cho, B.H. Multiple output of dual half bridge LLC resonant converter using PFM-PD control. In Proceedings of the
2009 IEEE Energy Conversion Congress and Exposition, Detroit, MI, USA, 9–13 October 2009; pp. 1133–1140.
125. Li, J.; Zhang, J. Steady-state output characteristics of a three-port bidirectional DC-DC converter with dead time. IOP Conf. Ser.
Earth Environ. Sci. 2019, 354, 012105. [CrossRef]
126. Jain, M.; Daniele, M.; Jain, P.K. A bidirectional DC-DC converter topology for low power application. IEEE Trans. Power Electron.
2000, 15, 595–606. [CrossRef]
127. Huang, H. Designing an LLC Resonant Half-Bridge Power Converter; Texas Instruments: Dallas, TX, USA, 2010; Volume 3,
pp. 2010–2011.
128. Bhardwaj, M.; Yu, S. Bidirectional CLLLC Resonant Dual Active Bridge (DAB) Reference Design for HEV/EV Onboard Charger.
2020. Available online: https://www.ti.com/lit/ug/tidueg2c/tidueg2c.pdf?ts=1674544714197&ref_url=https%253A%252F%25
2Fwww.google.com%252F (accessed on 15 December 2022).
129. Abdel-Rahman, S. Resonant LLC converter: Operation and design. Infineon Technol. N. Am. (IFNA) Corp 2012, 19, 14.
130. Deng, J.; Li, S.; Hu, S.; Mi, C.C.; Ma, R. Design methodology of LLC resonant converters for electric vehicle battery chargers.
IEEE Trans. Veh. Technol. 2013, 63, 1581–1592. [CrossRef]
131. Musavi, F.; Craciun, M.; Gautam, D.S.; Eberle, W.; Dunford, W.G. An LLC resonant DC–DC converter for wide output voltage
range battery charging applications. IEEE Trans. Power Electron. 2013, 28, 5437–5445. [CrossRef]
132. Ramakrishnan, H. Bi-Directional, Dual Active Bridge Reference Design for Level 3 Electric Vehicle Charging Stations. Des. Guide
June 2019, 2019, 1–51.
133. Kundu, U.; Pant, B.; Sikder, S.; Kumar, A.; Sensarma, P. Frequency domain analysis and optimal design of isolated bidirectional
series resonant converter. IEEE Trans. Ind. Appl. 2017, 54, 356–366. [CrossRef]
134. Liu, C. Analysis, Design and Control of DC-DC Resonant Converter for On-board Bidirectional Battery Charger in Electric
Vehicles. Ph.D. Thesis, University of Sheffield, Sheffield, UK, 2017.
135. Jung, J.H.; Kim, H.S.; Ryu, M.H.; Baek, J.W. Design methodology of bidirectional CLLC resonant converter for high-frequency
isolation of DC distribution systems. IEEE Trans. Power Electron. 2012, 28, 1741–1755. [CrossRef]
136. Dhakar, A.K.; Soni, A.; Saini, V.; Chandel, S. Design of Bi-directional CLLC Resonant Converter with Planar Transformer and
Synchronous Rectification for Energy Storage Systems. In Proceedings of the 2020 IEEE International Conference on Power
Electronics, Drives and Energy Systems (PEDES), Jaipur, India, 6–19 December 2020; pp. 1–8.
137. Zhao, B.; Zhang, X.; Huang, J. Design of CLLC resonant converters for the hybrid AC/DC Microgrid applications. In Proceedings
of the 2018 IEEE International Power Electronics and Application Conference and Exposition (PEAC), Shenzhen, China, 4–7
November 2018; pp. 1–5.
138. Ammar, A.M.; Ali, K.; Rogers, D.J. A bidirectional GaN-based CLLC converter for plug-in electric vehicles on-board chargers. In
Proceedings of the IECON 2020 The 46th Annual Conference of the IEEE Industrial Electronics Society, Singapore, 18-21 October
2020; pp. 1129–1135.
139. Ruan, X.; Yan, Y. Soft-switching techniques for PWM full bridge converters. In Proceedings of the 2000 IEEE 31st Annual Power
Electronics Specialists Conference. Conference Proceedings (Cat. No. 00CH37018), Galway, Ireland, 23 June 2000; Volume 2,
pp. 634–639.
140. He, P.; Khaligh, A. Comprehensive analyses and comparison of 1 kW isolated DC–DC converters for bidirectional EV charging
systems. IEEE Trans. Transp. Electrif. 2016, 3, 147–156. [CrossRef]
141. Granello, P.; Soeiro, T.B.; Van der Blij, N.H.; Bauer, P. Revisiting the Partial Power Processing Concept: Case Study of a 5 kW
99.11% Efficient Flyback Converter-Based Battery Charger. IEEE Trans. Transp. Electrif. 2022, 8, 3934–3945. [CrossRef]
142. Martis, J.; Vorel, P.; Cipin, R.; Prochazka, P.; Toman, M. Compact High-Efficiency Li-Ion Fast-Charger. ECS Trans. 2016, 74, 17.
[CrossRef]
143. Kushwaha, R.; Singh, B. An improved SEPIC PFC converter for electric vehicle battery charger. In Proceedings of the 2019 IEEE
Industry Applications Society Annual Meeting, Baltimore, MD, USA, 29 September–3 October 2019; pp. 1–8.
144. Kushwaha, R.; Singh, B. UPF-isolated zeta converter-based battery charger for electric vehicle. IET Electr. Syst. Transp. 2019,
9, 103–112. [CrossRef]
145. Lee, C.G.; Park, J.H.; Park, J.H. Buck-flyback (fly-buck) stand-alone photovoltaic system for charge balancing with differential
power processor circuit. J. Power Electron. 2019, 19, 1011–1019.
Sustainability 2023, 15, 2813
71 of 71
146. Murthy-Bellur, D.; Kazimierczuk, M.K. Two-switch flyback PWM DC-DC converter in continuous-conduction mode. Int. J.
Circuit Theory Appl. 2011, 39, 1145–1160. [CrossRef]
147. Remes, C.L.; Rosa, M.B.; Oliveira, S.V.G. A Two-Switch Forward Converter application for battery charging. In Proceedings
of the 2015 IEEE 13th Brazilian Power Electronics Conference and 1st Southern Power Electronics Conference (COBEP/SPEC),
Fortaleza, Brazil, 29 November–2 December 2015; pp. 1–6.
148. Ye, M.; Song, P.; Li, S.; Xiao, Y. Voltage-Fed Push-Pull PWM Converter Featuring Wide ZVS Range and Low Circulating Loss with
Simple Auxiliary Circuit. J. Power Electron. 2018, 18, 965–974.
149. Rabello, A.L.; Marcio, C.; Sousa, G.; Vieira, J. A fully protected push-pull current-fed DC-DC converter. In Proceedings of the
IECON’97 23rd International Conference on Industrial Electronics, Control, and Instrumentation (Cat. No. 97CH36066), New
Orleans, LA, USA, 14 November 1997; Volume 2, pp. 587–592.
150. Baek, J.; Youn, H.S. Full-bridge active-clamp forward-flyback converter with an integrated transformer for high-performance and
low cost low-voltage DC converter of vehicle applications. Energies 2020, 13, 863. [CrossRef]
151. Dow, Y.; Son, H.; Lee, H.D. A study on half bridge LLC resonant converter for battery charger on board. In Proceedings of the 8th
International Conference on Power Electronics-ECCE Asia, Jeju, Republic of Korea, 30 May–3 June 2011; pp. 2694–2698.
152. Effects of Increasing Frequency on Transformers and Inductors. 2021. Available online: https://shreejee.co/blog/knowledge/
effects-of-increasing-frequency-on-transformers-and-inductors/ (accessed on 27 November 2022).
153. Crane, L. Selecting the Best Inductor for Your DC-DC Converter. Available online: https://www.mouser.com/pdfDocs/doc469
_selecting_inductors.pdf (accessed on 27 November 2022).
154. High Frequency Transformer. Available online: https://www.engr.colostate.edu/ECE562/98lectures/l34.pdf (accessed on 27
November 2022).
155. McLyman, W. Designing Magnetic Components for High Frequency DC-DC Converters; Kg Magnetics: Idyllwild, CA, USA, 1993.
156. Umanand, L.; Bhat, S. Design of Magnetic Components for Switched Mode Power Converters; New Age International (P) Limited:
New Delhi, India, 1992.
157. Umanand, L. Power Electronics: Essentials and Applications; Wiley India Pvt. Limited: Noida, India, 2009.
158. Jung, C. Power Up with 800-V Systems: The benefits of upgrading voltage power for battery-electric passenger vehicles. IEEE
Electrif. Mag. 2017, 5, 53–58. [CrossRef]
159. Grazian, F.; Soeiro, T.B.; Bauer, P. Voltage/Current Doubler Converter for an Efficient Wireless Charging of Electric Vehicles with
400 V and 800 V Battery Voltages. IEEE Trans. Ind. Electron. 2022. [CrossRef]
160. Aghabali, I.; Bauman, J.; Kollmeyer, P.J.; Wang, Y.; Bilgin, B.; Emadi, A. 800-V electric vehicle powertrains: Review and analysis of
benefits, challenges, and future trends. IEEE Trans. Transp. Electrif. 2020, 7, 927–948. [CrossRef]
161. ABB. Terra 360, the High-Power Charger for Everyone. Available online: https://new.abb.com/ev-charging/terra-360 (accessed
on 18 December 2022).
162. EVESCO. EVDC-60NA-60kW DC Fast Charger, Simultaneous Charging. Available online: https://www.power-sonic.com/
product/evdc-60na/ (accessed on 18 December 2022).
163. Schmidt, B. Volkswagen to Reuse ID.3 and ID.4 Batteries in EV Charging Park. 2022. Available online: https://thedriven.io/2022
/07/15/volkswagen-to-reuse-id-3-and-id-4-batteries-in-ev-charging-park/ (accessed on 14 November 2022).
164. Proterra Charging Systems. 2019. Available online: https://www.proterra.com/products/charging-infrastructure/ (accessed on
14 November 2022).
165. Shepardson, D. U.S. proposes Standards for Fast Electric Vehicle Charging Projects. 2022. Available online: https://www.reuters.
com/business/autos-transportation/us-propose-standards-government-funded-ev-charging-projects-2022-06-09/ (accessed on
14 November 2022).
166. Schreier, M. Twelve Renaults Surrounded a Trio of Fast Chargers at the D0 Motorway for Two Hours. The Battery System with
Photovoltaics at the Back has Withstood the Onslaught! 15 September 2020. Available online: http://www.cez.cz/en/media/
press-releases/twelve-renaults-surrounded-a-trio-of-fast-chargers-at-the-d0-motorway-for-two-hours.-the-battery-system-withphotovoltaics-at-the-back-has-withstood-the-onslaug-141330 (accessed on 14 November 2022).
167. Delta Electronics. City Charger 100 kW. Available online: https://www.deltaww.com/en-US/products/EV-Charging/4864/
(accessed on 12 December 2022).
168. Saur News Bureau. Italy’s Fimer to Showcase Charging Solutions at World’s Largest Mobility Trade Fair. 28 September 2022.
Available online: https://www.saurenergy.com/solar-energy-news/italys-fimer-to-showcase-charging-solutions-at-worldslargest-mobility-trade-fair (accessed on 14 December 2022).
Disclaimer/Publisher’s Note: The statements, opinions and data contained in all publications are solely those of the individual
author(s) and contributor(s) and not of MDPI and/or the editor(s). MDPI and/or the editor(s) disclaim responsibility for any injury to
people or property resulting from any ideas, methods, instructions or products referred to in the content.