RF APPLICATION NOTE:
PUSH-PULL CIRCUITS
and
WIDEBAND
TRANSFORMERS
Push-Pull Transistors
Semelab plc produces a wide range of push-pull MOSFETs and this application note is intended
as a guide to some circuit design principles which are particularly appropriate when using these
devices. Two examples of Semelab push-pull MOSFETs are shown in figure 1.
Fundamentally, a push-pull circuit uses a pair of effectively separate transistors operating 180
O
out of phase with one another. If good amplitude and phase balance is maintained between the
signals in each half of the device, then an RF ground will exist at the midpoint. This approach
leads to several advantages over single-ended designs:
° Input and Output Impedances Doubled
° Reduced Even Order Harmonics
° Increased Output Power
° Higher Bandwidths Possible
Figure 1 – Semelab MOSFETS
° Reduced Effect of Common Lead Inductance
Perhaps the most significant disadvantages are the need for differential RF excitation and the fact
that excellent symmetry is required in both the matching circuit and the device itself. Semelab
resolves the latter of these issues by individually selecting each side of their push-pull MOSFETs
and rigorously testing them to ensure exceptional conformity between the threshold voltage and
transconductance parameters. A generic push-pull circuit configuration is shown in figure 2.
DC Bias
RF In
RF Out
Balun
50•
Impedance
Impedance
Matching
Matching
Intermediate Z
ZSopt
ZLopt
Balun
Intermediate Z
50•
Figure 2 – Generic Push-Pull Amplifier Schematic
2
Balanced to Unbalanced Transformations
As has already been mentioned, push-pull transistors require splitting the RF signal at the input
(along with the reciprocal operation i.e. the need to recombine the output). Circuit elements that
provide this function are referred to as baluns (from balanced to unbalanced). The ideal balun
o
would split the signal into halves of equal amplitude without loss, along with providing an 180
differential phase shift across all frequencies. Although “ideal” baluns are only theoretical devices,
several approaches can be used to build practical baluns with excellent performance over wide
bandwidths. Principle methods include:
° Conventional Coil Transformers
° Transmission Line Transformers
° Microstrip Structures (Wilkinson Divider; Line and Ring Hybrids)
The choice of method is largely dependant on frequency. For HF (up to 30MHz) magnetically
coupled coil transformers are suitable. Above these frequencies, leakage inductance and
parasitic capacitance degrades the performance making them a poor choice. VHF and UHF
circuits (30MHz – 1GHz) most commonly employ transmission line balun structures, built from
coaxial cable or twisted pairs. Microstrip structures such as Wilkinson dividers also offer very
good performance, but these are limited by their physical size (in the order of λ/2) and are
therefore only practical at higher frequencies. Substrates with high dielectric constants can
mitigate this problem to some extent.
After the balun, each half of the matching network is usually calculated by conventional means
*
treating each half of the transistor as if it were a single ended device . One important point to note
is the difference between balanced (i.e. drain to drain) impedance and unbalanced (drain to
ground) impedance. In a well matched circuit, the drain to drain impedance should be twice that
measured from drain to ground.
*
Optimum source and load impedances for Semelab transistors can be found at
http://www.semelab.co.uk/rf/zopt_index.php
3
Coupled Coil Transformers
Conventional transformers operate principally by coupling signals between primary and
secondary windings through a suitable magnetic material. From Faradays’ law it can be shown
that, for an ideal transformer, the voltage per turn is the same on each side. The relationship
between impedances on the primary and secondary side can then be found as:
ZIN = ZOUT * (nIN / nOUT)
2
This implies that transformers are ideally suited to impedance transformations since they can be
scaled by any factor, depending only on the turns ratio. When the parasitics are negligible (i.e. at
low frequencies – up to 30MHz) coupled coil transformers are often used where wideband
resistance transformation is required. Various winding topologies are possible, and are chosen
depending on whether DC isolation and/or balanced signals are required. Figure 3 below shows
the simplest configurations. The autotransformer in 3a has a tapped continuous winding and
provides a DC short between input and output. Figure 3b shows a more conventional transformer
with DC isolation between primary and secondary windings. Other arrangements can allow for
balanced-unbalanced operation. The centre tapped secondary transformer in 3c is a balanced
2
signal splitter which also provides an N /2 impedance transformation.
2
(N .ZIN )/2
2
N .ZIN
ZIN
2
N .ZIN
ZIN
ZIN
2
(N .ZIN )/2
nOUT /nIN = N
nOUT /nIN = N
Fig 3a – Autotransformer
nOUT /nIN = N
Fig 3b – Conventional Transformer
Fig 3c – Centre Tapped Secondary
Higher permeability (•) ferrite material is usually preferred for better coupling, although it should
be noted that due to the increased flux density, high • cores will saturate more easily than those
with lower •. Permeability can also vary as a function of current level, frequency and temperature.
4
Two-hole balun cores and toroids are the most commonly used ferrite structures (see Appendix C
for suppliers). The choice of wire also affects performance and is essentially a trade off. Heavier
wire will increase the coupling, but it will also increase the parasitic winding capacitance.
In practice, all transformers suffer from parasitic effects that cause their behaviour to deviate from
the ideal. The schematic below shows an equivalent circuit for modeling the transformer.
CW
RWP
LLEAK_P
LLEAK_S
RWS
ZOUT
ZIN
CP
LMAG
CS
RCORE
nIN : nOUT
Figure 4 – Conventional Transformer Equivalent Circuit
The ideal nIN:nOUT transformer is represented inside the dashed line. Series resistances RWI and
RWO are the resistances of the primary and secondary windings. Although these are negligible at
DC, the skin effect will increase the resistance in proportion to the frequency. RCORE models the
loss of energy in the transformer’s core due to eddy currents and hysterisis effects. Leakage
inductances represented by LLEAK_P and LLEAK_S model the magnetic flux which passes outside of the
core and therefore is not coupled between the input and output ports. LMAG limits the low
frequency response of the transformer and it has its physical origins in the finite magnetizing
inductance in the coils. At higher frequencies, the capacitive effect between windings (CW) and
between the turns of each winding (CP and CS) also become important.
Single winding transformers are the type most commonly used at RF. The primary winding
consists of a single turn of hollow metal tubing threaded through suitable ferrite material shorted
at one end via a copper strap or PCB. Insulated wire is then passed inside this tube to form the
2
secondary. Since the primary is limited to a single turn, the impedance ratios are limited to 1:n ;
where n is the integer number of turns in the secondary. The principle advantage of this
construction method is the close proximity of the windings and ferrites, leading to minimal leakage
5
inductance and improved performance at high frequency. A 1:4 transformer is shown below in
figure 5.
Figure 5 – Single Winding Transformer Implementation
These types of transformer can be easily constructed from coaxial cable. By using the centre
conductor as the secondary, the cable can be wound to the required number of turns. The outer
conductor is then removed at the open circuit end and soldered together along its remaining
length. Figure 6 below shows a range of these coaxial transformers and the associated “E” and “I”
magnetic cores.
Figure 6 – Practical Single Winding RF Transformers
6
Generally, conventional transformers will not perform as well as transmission line transformers in
terms of bandwidth, losses and power handling. They do, however, offer a wider variety of
possible impedance ratios and DC isolation from primary to secondary windings.
Transmission Line Transformers
These types of transformers rely on a large variation in impedance being presented to the
differential (balanced) mode and common (unbalanced) mode currents on the transmission line.
This is readily achieved at high frequencies, and the low frequency performance can be extended
*
using a suitable ferrite material around the transmission line . It is important to note that unlike
conventional transformers, the ferrite is not the medium being used to transfer RF power between
ports. For this reason much larger bandwidths and greater efficiencies are achievable using the
transmission line transformer. Degraded performance at high frequency is caused by the length
of the line (max •/8), by ground plane effects, and by non-optimum characteristic impedance of
the chosen transmission line.
I
I
o
V
o
50• (0 )
25• (0 )
2V
V
o
25• (180 )
I
Figure 7 – Guanella Transmission Line Balun
Figure 7 shows the basic “building block” of transmission line transformers – the 1:1 unbalancedto-balanced transformer first introduced by Guanella in 1944. Note that this provides no
impedance transformation, but each side of the balanced load has an impedance half that seen at
the input. This is effectively an RF transmission line equivalent of the centre-tapped transformer
used at lower frequencies.
The choice of transmission line largely depends on the characteristic impedance as dictated by
the choice of balun or transformer. Coaxial cables of fixed values are freely available, and these
*
Differential mode currents produce no magnetic field external to the transmission line and therefore remain unaffected by
the addition of ferrites.
7
can be combined to make transmission lines of varying impedances. As with most UHF/VHF
transmission line transformers, Teflon insulated coaxial cable is preferred – see appendix C for a
list of suppliers. The power handling capability is generally limited by the maximum allowable
temperature, which itself is a function of dielectric material and cable diameter. The table in
appendix B provides a rough guide. Twisted pairs are useful since a limitless range of
characteristic impedances can be realised. Below are some guidelines for designing twisted pairs
of specific ZO.
° Increasing the number of twists per unit length decreases impedance
° Increasing the wire diameter will reduce the impedance
° Very low impedances are achieved by using multiple pairs in parallel
° 50• wire can be made using AWG-22 enameled wire with 2.5 twists per cm
The impedance will also be affected by the wire separation, the dielectric material between them
as well as the proximity of any ground plane.
The simplest type of transmission line transformer is the •/4 line. Since the impedance match is
dependant on the presence of a quarter wavelength standing wave, these transformers are
inherently narrowband. They also require a transmission line of characteristic impedance which is
the geometric mean of the input and output impedances to be matched i.e.:
ZO = ¥ ZIN * ZOUT)
This can limit the practicality of this technique since many types of transmission line are only
available in a limited range of impedances. However, cables can be combined to achieve ZO
variations. For example, two paralleled 25• cables will produce a transmission line with a
characteristic impedance of 12.5•.
As well as the impedance matching function, the •/4 line can be used as a balun in the same way
as above – by grounding one conductor on the unbalanced side and connecting each conductor
to one half of a balanced load on the other side. The diagram below shows an implementation of
this using coaxial cable.
8
/ •
=,
=2
¥ =,1*=287
½ =2
½ =2
Figure 8 – Coaxial Quarter-Wavelength Balun
As well as the narrowband λ/4 transformer, a single transmission line can be used as an
unbalanced to unbalanced (unun) transformer. This is capable of providing wideband 1:4
impedance transformation ratios by connecting it in the so called ‘bootstrap’ configuration as
shown in figure 9. Here, the two conductors which constitute the transmission line are used as the
primary and secondary windings in a similar way to the conventional autotransformer.
2I
Ri
I
I
V
V
RO
I
I
V
Figure 9 – Ruthroff 1:4 Unun
If a voltage V is present across the input, the same voltage will be impressed across the lower
conductor of the transmission line. Charge conservation demands that the same voltage must be
present across the upper conductor also. The voltage across Ro is therefore the sum of these
voltages - 2V. If the current I is to flow through the load, it must also flow through the upper
conductor. But, again since both conductors have the same voltage across them, the currents
through each must be identical. Therefore since Ro = 2V/I and Ri = V/2I then:
Ri = Ro/4
Calculations show that maximum power transfer occurs for this transformer with the optimum
transmission line impedance ZO = 2*Ri. For best performance the transmission line should be kept
as short as possible. Figure 10 below shows two practical implementations of the 1:4 unun using
9
coaxial cable and wire-wrapped toroid configurations. Obviously, these devices are bilateral and
simply reversing the ports will provide 4:1 step-down transformations.
5,1
=2
5287
2*R,1
5,1
5287
Figure 10 – 1:4 Unun Coaxial and Toroidal Implementations
This technique can be extended to other transformation ratios. By adding extra transmission
2
lines, other 1:n transformers can easily be achieved. Figure 11 shows a 1:9 transmission line
schematic alongside an implementation using coaxial cable. 1:16 transformations can be realised
by adding a third conductor pair and connecting them in the same way as the second.
,
5/
5/
9
,
9
Figure 11 – 1:9 Ruthroff Unun
One of the prime factors limiting high frequency performance is the phase error caused by the
arbitrary length of the transformers’ interconnections. If these connections were made using a
transmission line of the same length, velocity factor and impedance as the transformer line itself,
then the phase error would be eliminated. An additional advantage is that the physical shape of
the transformer is also no longer restricted by the need to bring connecting points close together.
Devices of this type are referred to as equal delay transformers. Figure 12 shows how the
principle is applied to the 1:4 Ruthroff Unun seen before in figure 9.
10
,QWHUFRQQHFWLQJ/LQH
5287
5287
0DLQ7UDQVIRUPHU
)HUULWH6OHHYH
Figure 12 – 1:4 Equal Delay Ruthroff Unun
Note the absence of any magnetic material on the interconnecting line
In push-pull circuits, impedance transformations between a balanced source and balanced load
are often required. By combining a number of basic building blocks in a range of parallel/series
2
combinations balanced to balanced impedance ratios of 1:n can be achieved. The diagrams
below show 1:4 and 1:9 Guanella transmission line transformers based on this principle. Like the
quarter-wavelength transformer, the optimum transmission line impedance is the geometric mean
of input and output; however, small deviations from this can be permitted if some bandwidth
degradation is acceptable.
Parallel (Low Z)
Series (High Z)
Parallel (Low Z)
3I
I
Series (High Z)
I
2I
2V
V
3V
V
2I
I
I
Fig 13a – 1:4 Balanced-Balanced Transformer
3I
Figure 13b – 1:9 Balanced-Balanced Transformer
These transformers are often constructed of coaxial cable, and it is good practice to form the
cable into a suitable shape to keep the interconnects as short as possible. The limitation of
squared integer impedance ratios can be avoided to some extent by combining the conductors in
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more complex arrangements, but the benefits of doing this must be weighed against the
practicality of the design and the possible loss of bandwidth.
Appendix A – AWG Chart
2
O
AWG
Diameter (mm)
Diameter (mil)
Area (mm )
•/km (@20 C)
0
8.25
325
53.49
0.322
2
6.54
258
33.62
0.513
4
5.19
204
21.15
0.815
6
4.12
162
13.30
1.30
8
3.26
129
8.367
2.10
10
2.59
102
5.261
3.30
12
2.05
80.7
3.310
5.25
14
1.63
64.2
2.081
8.34
16
1.29
50.8
1.309
13.2
18
1.02
40.2
0.823
20.9
20
0.81
31.9
0.518
33.3
22
0.64
25.2
0.326
53.0
24
0.51
20.1
0.205
84.2
26
0.40
15.7
0.129
134
28
0.32
12.6
0.081
213
30
0.25
9.84
0.051
339
32
0.20
7.87
0.032
539
34
0.16
6.30
0.020
856
36
0.13
5.12
0.013
1361
38
0.10
3.94
0.008
2164
40
0.08
3.15
0.005
3340
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Appendix B – Approximate Coaxial Cable Power Handling
Outer Diameter
Cable Type
1MHz
10MHz
100MHz
500MHz
1.7mm
Flexible
1kW
300W
90W
-
2.8mm
Flexible
1kW
800W
250W
-
1.1mm
Semi-Rigid
-
-
68W
32W
2.2mm
Semi-Rigid
-
-
330W
140W
6.4mm
Semi-Rigid
-
-
1.2kW
515W
Appendix C
C1 - Coaxial Cable Suppliers
Micro-Coax Pottstown, PA (+1-800-223-2629)
http://www.micro-coax.com
Gigalink Rotterdam, The Netherlands (+31-10-789-11-22)
http://www.gigalink-mce.com
NaF Technology Hawsung-City, Kyunggi-Do, Korea (+82-31-352-6671)
http://www.naftech.co.kr
Storm Products Santa Monica, CA (+1-310-319-5388)
http://www.stormproducts.com
C2 - Ferrite Suppliers
Eastern Components W. Conshohocken, PA (+1-610-825-8610)
http://www.eastern-components.com
Fair-Rite Wallkill, NY (+1-845-895-2055)
http://www.fair-rite.com
Magnetics Pittsburgh, PA (+1-412-696-1333)
http://www.mag-inc.com
Magnetics Group Bethlehem, PA (+1-610-867-7600)
http://www.magneticsgroup.com
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C3 – Balun/Transformer Suppliers
XFMRS Camby, IN (+1-317-834-1066)
http://www.xfmrs.com
Coilcraft Europe Cumbernauld, Scotland (+44-123-673-0627)
http://www.xfmrs.com
Bibliography
1. S.C.Cripps; “RF Power Amplifiers for Wireless Communications”; Artech House 1999; ISBN:0-89006989-1
2. P.L.D. Arbie; “RF and Microwave Amplifiers and Oscillators” - Ch 6; Artech House 1999; ISBN:0-89000679-7X
3. R.J.Weber; “Introduction to Microwave Circuits” - Ch 12; IEEE Press 2001; ISBN:0-7803-4704-8
4. N.Dye & H. Granberg; “RF Transistors – Principles and Practical Applications” - Ch 10; Butterworth
Heinemann 1993; ISBN: 0-7506-9059
5. J. Sevick; “Transmission Line Transformers”; ARRL1990; ISBN 0-8725-9296-0
6. G. Guanella; “Novel Matching Systems for High Frequencies”; Brown-Boveri Review Sept 1944
Application Note written by S.J.McCarthy with contributions from P.Smith, N.Padfield and Dr J.Walker.
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