2022 IEEE International Power Electronics and Application Conference and Exposition (PEAC) | 978-1-6654-9141-9/22/$31.00 ©2022 IEEE | DOI: 10.1109/PEAC56338.2022.9959484
Design of a 300 kW Partial Power Processing
Based DC-DC Converter for Electric Vehicles
Extreme Fast Charging Stations
Bin Guo
College of Electrical Engineering
Zhejiang University
Hangzhou, China
binguo_cr7@zju.edu.cn
Xin Zhang
College of Electrical Engineering
Zhejiang University
Hangzhou, China
zhangxin_ieee@zju.edu.cn
Zhengqing Zhang*
Wuhan Second Ship Design
and Research Institute
Wuhan, China
zhangzhengq219@163.com
Hao Ma
College of Electrical Engineering
Zhejiang University
Hangzhou, China
mahao@zju.edu.cn
Xiang Jin
Wuhan Second Ship Design
and Research Institute
Wuhan, China
1085955753@qq.com
Yongmao Wang
Wuhan Second Ship Design
and Research Institute
Wuhan, China
ymwang2023@126.com
Abstract—This paper proposes a design approach of a
partial power processing (PPP) based dc-dc converter for
electric vehicle extreme fast charging stations. The partial
power converter is made up of two stages: the front-end stage is
an input-series output-parallel (ISOP) LLC converter, which
acts as a dc-dc transformer (DCX). The rear-end stage is a threephase interleaved buck converter to realize flexible constant
current/voltage (CC/CV) charging control. With the proposed
topology structure, only less than one fifth of the total charging
power is processed. Thus, compared with full rated charging
converters, its efficiency and power density are significantly
improved. The operation principles with modulation strategy is
presented in depth. In addition, a simple control strategy is
proposed to realize current-sharing control of the three-phase
interleaved buck converter. The system design approach is
provided in detail. Finally, simulation and experimental tests on
a 300-kW prototype validate its appropriate performance.
Keywords—Partial power processing, electric vehicle, extreme
fast charging station, dc-dc converter
I. INTRODUCTION
Due to the reduced reliance on conventional energy
resources such as oil and gas [1], electric vehicles (EVs) are
regarded as an environmental friendly automotive technology
towards sustainable road transportation [2]. However, analysis
indicates that the prolonged charging time of EVs will lead to
driving range anxiety. Thus, a charging station with extreme
fast charging capability is necessary to eliminate range anxiety
and improve the market penetration of EVs in longer distance
journey [3].
DC link
PCC
ac
dc
MV grid
Line frequency
transformer
EVs
dc
Inverter
dc
dc
dc
Partial power
converter
Fig.1 Structure of extreme fast charging station.
As shown in Fig.1, in the extreme fast charging station,
EVs is usually connected to a medium voltage (MV) grid by a
This work was supported in part by the National Nature Science
Foundation of China under Grant 52177198, in part by the National Key
Research and Development Program under Grant 2021YFB2500600, and in
part by the Delta Environmental and Education Foundation under Grant
DERG2021002. (Corresponding author: Zhengqing Zhang.)
Iin
Vin
Iin
Io
dc
dc
Vo
Io
dc
Vin
dc
Vo
Vin
(a) Type Ⅰ input-parallel output-series (b) Type Ⅱ input-series output-parallel
Fig.2 Two typical PPP-based dc-dc converter structures.
line frequency transformer, frond-end ac-dc converter and
rear-end dc-dc converter. For the rear-end dc-dc converter,
usually a full rated converter, such as buck-boost converter [4],
LLC converter [5] and dual active bridge (DAB) [6] converter
is adopted. Though the abovementioned converters can realize
soft-switching, the system efficient is still low since the
converters need to handle full power. What’s more, to increase
power capacity, multiple converters are usually required to be
connected in parallel, which results in bulky size and complex
control. To solve this issue, the concept of partial power
processing (PPP) has been successfully employed to
photovoltaic system [7], traction power trains [8] and battery
storage systems [9]. In this case, the partial power converter is
only rated less than 30% of its system power, yet it provides
flexible voltage and current control.
As shown in Fig. 2, generally, there are two typical PPP
structures produced by isolated dc-dc converter, including the
input-series output-parallel (ISOP) structure and inputparallel output-series (IPOS) structure. Based on the two
typical structures, many PPP-based converters have been
provided by using DAB converter [10], LLC converter [11],
flyback converter [12] and multi-active bridge converter [13].
However, from the above, the existing PPP converters are
only rated at few kilowatt, which still has a gap for direct
application in extreme fast charging station.
To fill this gap, this paper thus presents a detailed design
approach of a 300-kW PPP-based dc-dc converter for extreme
fast charging station. The system configuration, operation
principle and control strategy are presented in Section Ⅱ. The
system parameters design is shown in Section Ⅲ. In section
Ⅳ, various simulation and experimental tests are performed
on a 300-kW prototype to verify the performance. Finally,
Section Ⅴ gives the conclusion.
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1
1
3
Q1
3
Q3
D1
n:1
o
k
p1
1
m
s1
s
A
2
3
2
4
2
in
4
S1
S3
S5
LLC-DCX # 1
5
S2
6
S4
S6
b
3
Q2
Three-phase interleaved
buck converter
k
p2
3
C1 Vin1
1
2
5
6
m
s2
s
ip1
B
7
8
C
D
C3
D2
Vo1
C2
D4
LLC-DCX # 1
(a) [t0, t1]
3
7
i
vs s1
Q4
o
4
Lk
Lm
D3
8
LLC-DCX # 2
(a) System structure of the proposed 300 kW DC-DC converter
Q1
A
ip
I
Q3
ip1
C1 Vin1
t
iLm
D1
n:1
B
Q2
Lk
Lm
is1
D3
C
vs
D
C3
D2
Q4
Vo1
C2
D4
LLC-DCX # 1
VCr
(b) [t1, t2]
t
vCr
Q1
Vgs
Q1&Q4
t0
A
Q2 & Q3
t1 t2 t3
Q3
t4 t5 t6
t
(b) Main working waveforms of LLC-DCX1
Fig.3 System structure of the proposed DC-DC converter and the main
working waveforms of the frond-end LLC-DCX
II. PROPOSED PPP-BASED DC-DC CONVERTER
A. System Configuration and Operation Principle
The structure of the proposed PPP-based dc-dc converter
is shown in Fig. 3(a), where it is a type Ⅰ (IPOS) structure, thus,
it is a step-up converter. For the proposed 300 kW dc-dc
converter, the input voltage range is 800-900 V and its output
voltage range is 900-960 V. To realize such a high voltage and
high power output, the dc-dc converter is made up of two
stages. The front-end stage is a ISOP LLC converter, which
acts as a DCX. By using the ISOP structure, it is beneficial to
achieve input-side neutral point voltage balance. The rear-end
stage is a three-phase interleaved buck converter, which is
used to realize constant current (CC)/voltage (CV) charging.
As shown in Fig. 3(a), since the proposed dc-dc converter is a
ISOP structure, the output voltage of the interleaved buck
converter is the difference between the output voltage Vo and
the input voltage Vin.
As shown in Fig. 3(a), for the frond-end stage converter,
since the operation principle of LLC-DCX1 and LLC-DCX2
is the same, taking the LLC-DCX1 as an example. Fig. 3(b)
shows the main working waveforms of LLC-DCX1 during
each time interval, where ip1 and iLm are resonant current and
magnetic inductor current, respectively, vCr is the resonant
capacitor voltage. Fig. 4 gives the equivalent circuits during
each time interval at positive-half cycle. The detailed analysis
during each time interval is shown as follow.
Mode Ⅰ [t0, t1]: During this mode, switches Q1 and Q4 are
turn on and thus the primary side bridge output voltage VAB is
equal to Vin1. The magnetic inductor current is increased
linearly and the secondary side voltage of transformer is
ip1
C1 Vin1
Q2
D1
n:1
B
Lk
Lm
i
vs s1
D3
C
D
C3
Q4
D2
Vo1
C2
D4
LLC-DCX # 1
(c) [t2, t3]
Fig.4 Equivalent circuit during each time interval.
clamed to nVo1. The resonant inductor Lk and capacitor C3
form a resonant network, its voltage is Vin1-nVo1. In this mode,
the state-space model of resonant inductor current ip1,
magnetic inductor current iLm and resonant capacitor voltage
vcr can be derived as:
 dvcr
C3 dt  i p1

 dip1
 Vin1  nVo1  vcr
 Lk
(1)
 dt
 diLm
 nVo1
 Lm
dt

Mode Ⅱ [t1, t2]: As shown in Fig. 3(b) and Fig. 4(b), during
this mode, the resonant inductor current is equal to magnetic
inductor current, thus, the primary and secondary side current
of transformer are zero. The diode D1 and D4 are turned off
with zero current. In this mode, the resonant inductor Lk and
magnetic inductor Lm jointly participate in the resonant
process. The following equations are derived:
 dvcr
C3 dt  i p1

di p1

 Vin1  vcr
( Lk  Lm )
(2)
dt

diLm

 Vin1  vcr
( Lk  Lm )
dt

Mode Ⅲ [t2, t3]: during this mode, switches Q1 and Q4 are
turned off, as shown in Fig. 4(c). The junction capacitor of Q1,
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Vref
CV
L1
L2
S4
Vb
(a) Three-phase interleaved buck converter
Ts
0
S1
S3
0
0
S5
iL1
iL2
iL3
(b) Main waveforms of driving signal and inductor current
Fig. 5 Topology and main waveforms of the three-phase interleaved buck
converter.
Start
Input voltage measurement
Voltage error
Verr >Vth?
N
Y
Variable frequency control
If fs ∈[fL, fH]?
Y
N
Output voltage error Verr
Output voltage error Verr
Variable frequency
control
Phase-shifted control
iL1+iL2+iL3
Gi1(s)
ref
di
Δdi
iLi
Gi2(s)
1/3
S1...S6
vtri
Current sharing control
L3
S6
ref
Gv(s)
iL
Vbin
S2
CC iL
Vbout
S5
S3
S1
Fig. 6 Control logic diagram of the proposed hybrid phase-shifted and variable
frequency control strategy.
Q4 are charged and Q2, Q3 are discharged by resonant current
ip1. At the time of t3, the body diode of Q2 and Q3 are turned
on. Therefore, switches Q2 and Q3 will be turned on with zero
voltage switching (ZVS) in the next mode. In this mode, the
state-space model is the same as (2).
For the rear-end stage of the proposed dc-dc converter,
buck converter bears low voltage and high current. To reduce
the filter inductor and output capacitor, three-phase
interleaved structure is adopted. Fig. 5 gives the interleaved
buck converter and its main working waveforms. As shown in
Fig .5(b), the driving signal of each phase leg differ by 120
degrees and the upper and lower switches of each phase leg
are switched on complementary.
B. Control Strategy
As shown in Fig. 3, the LLC resonant converter is working
as LLC-DCX, therefore, in steady-state, the duty cycle of Q1
Pulsewidth
modulator
Fig. 7 Control block diagram of the three-phase interleaved buck converter.
to Q8 is fixed at 0.5. However, considering the inconsistency
of system parameters, the input side voltage of LLC converter
with the ISOP structure may suffer from imbalance, which
will degrade system reliability. To solve this issue, a hybrid
phase-shifted and variable frequency control strategy is
proposed. Fig. 6 gives the control logic diagram, as can be
seen in Fig. 6, when the input voltage error Verr of LLC1 and
LLC2 is larger than the threshold voltage Vth, LLC converter
will enter variable frequency control. However, if the
switching frequency fs reaches its upper and lower boundaries
while the input voltage error still larger than Vth, LLC
converter will perform phase-shifted control.
While for the three-phase interleaved buck converter,
since it needs to realize system CC and CV charging, a
cascade control structure is adopted, where the outer loop is
voltage loop and inner loop is current loop. The detailed
control block diagram of buck converter is shown in Fig. 7,
where iLn and dn (n = 1,2,3) indicate the n-th phase-leg inductor
current and duty cycle, respectively. Notably, the voltage
reference of buck converter Vbref comes from the output
voltage reference of PPC minus its input voltage value, i.e.,
Vbref =Voref -Vin. To realize current sharing of buck converter, a
simple current sharing control loop is added. Due to the
limited space, the detailed voltage and current loop control
design are not provided here.
III. SYSTEM PARAMETERS DESIGN
A. LLC-DCX Converter
For the front-end LLC-DCX, considering that the large
switching loss under high voltage and high current, the steadystate switching frequency fs of LLC-DCX is set as 10 kHz in
this paper. According to Fig. 3(a), to ensure the normal
operation of system, the input/output voltage range of LLCDCX1 and LLC-DCX2 are set as: Vin1min = Vin2min = 400 V,
Vo1min = Vo2min =240V.
k:1
k:1
Fig. 8 Schematic diagram of transformer structure
(1) Design of Transformer
To reduce the difficulty of transformer design and increase
its heat dissipation, as shown in Fig. 8, the transformer of
LLC-DCX1 and LLC-DCX2 adopts the ISOP structure. The
turns ratio of each transformer can be calculated as
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Vin1min
(3)
2Vo1min
Select the ferrite core with model LP3A, its maximum
magnetic induction intensity is Bmax = 0.5T. Set the variation
of magnetic flux density as ΔB = 0.4, then the area product of
magnetic core can be deduced as
2  Po  D
AP 
(4)
  B  ku  J  f s
where Po is output power, η is efficiency, ku is window
utilization, J is current density and fs is operating frequency.
According to the effective cross-section Ae of the selected
magnetic core, the primary turns can be derived as
Vin1
NP 
(5)
4Bf s Ae
Then, the secondary turns is calculated as
Np
Ns 
(6)
k
Since the steady-state switching frequency of LLC-DCX
is 10 kHz, there has skin effect on transformer wire, the
effective wire diameter can be derived as
2

(7)
2 f s 0
where ζ is constant and its value is 58×106, μ0 is permeability.
(2) Design of Resonant Circuit
For the resonant circuit parameters design of LLC-DCX,
the following two equations are obtained:
k
Q
fr 
1
Lk / C3
(8)
Re
,

Then, the inductance value of each buck converter is
obtained as
Ln 
3Vbin (Vbin  Vb )
 Po _ max f s 2
(12)
where fs2 is switch frequency of buck converter.
TABLE Ⅰ
SYSTEM PARAMETERS
Parameter Description
Rated Power (Pn)
Input Voltage (Vin)
Output Voltage (Vo)
Turns Ratio of Each Transformer (n:1)
LLC Switching Frequency (fs)
Buck Converter Switching Frequency (fs2)
LLC Resonant Capacitor (C3)
LLC Resonant Inductor (Lk)
Buck Converter Filter Inductor (L1, L2, L3)
Buck Converter Output Capacitor (Cb)
Value
300 kW
800-900 V
900-960 V
5:3
10 kHz
5 kHz
25.3 uF
10 uH
200 uH
2.2 mF
Fig. 9 Steady-state simulation results of LLC converter.
Lm
Lk
(9)
2 Lk C3
where Q is power quality factor, Re is equivalent load
resistance, fr is resonant frequency and λ is ratio of
magnetizing inductance to resonant inductance. When the
gain curve of LLC converter is drawn, the properly value of Q
and λ can be decided, then the parameters Lk, C3 and Lm can be
obtained.
B. Three-Phase Interleaved Buck Converter
For the three-phase interleaved buck converter, the
average current of each inductor is derived as
I L _ ave 
Pb _ max
Fig. 10 Steady-state simulation results of the proposed PPP-based dc-dc
converter under CV charging state.
(10)
3Vb
where Pb_max is the maximum power that processed by buck
converter, Vb is the output voltage of buck converter.
Thus, the current ripple in a switching cycle can be derived
as
iL 
Vbin  Vb  Vb
Ln
Vbin
Ts  I L _ ave 
Po _ max
3Vb

(11)
where Ln (n = 1, 2, 3) is n-th phase buck converter inductance,
γ is the ripple coefficient of inductor current, it is usually
selected as 15% ～ 30%. Ts is switching cycle of buck
converter.
Fig. 11 Dynamic simulation results of the proposed PPP-based dc-dc
converter under CC charging state.
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Iout (50A/div)
Interleaved
Buck Converter
Control Board
Vout (200V/div)
Vin (200V/div)
LLC-DCX
Output
Fig. 15 Dynamic experimental results of the proposed dc-dc converter under
CV charging state.
Input
Fig. 12 Experimental Setup
Vin1 (250 V/div)
Vin2 (250 V/div)
ip1 (25A/div)
ip1 (25A/div)
Fig. 13 Steady-state experimental results of LLC converter.
Vout (200V/div)
Vin (200V/div)
Iout (50A/div)
Fig. 14 Steady-state experimental results of the proposed dc-dc converter
under CC charging state.
IV. SIMULATION AND EXPERIMENTAL RESULTS
A. Simulation Results
To verify the feasibility of the proposed PPP-based 300kW dc-dc converter, a simulation prototype was built in PSIM.
The key system parameters are listed in Table Ⅰ.
Fig. 9 gives the steady-state simulation results of LLC
converter, where ip1 and ip2 are the resonant current of LLCDCX1 and LLC-DCX2, respectively, Vo1 is the output voltage
of LLC-DCX1. As can be seen in Fig. 9, the resonant current
is almost sinusoidal, which means ZVS of LLC converter can
be well realized. In addition, the output voltage Vo1 of LLCDCX1 is 240 V with the total input voltage of LLC converter
is 800 V, this verifies that the front-end stage LLC converter
acts as a DCX. Fig. 10 shows the steady-state simulation
results of the proposed PPP-based dc-dc converter under CV
charging state. It can be seen from Fig. 10 that the output
voltage of the dc-dc converter is 960 V with 290 A output
current. In addition, the currents of three-phase buck converter
differ by 120 degrees from each other and the currents are well
shared.
To demonstrate the good dynamic performance of the
presented system, the simulation is carried out in the case of
the charging current is suddenly jumps from half load to full
load. The dynamic simulation result is shown in Fig. 10. As it
is observed in Fig. 10, the output current of dc-dc converter
has a very short tracking time with nearly less than 1 ms and
without any overshoot. This result verified the robust
performance of the proposed system.
B. Experimental Results
To further verify the feasibility of the proposed PPP-based
dc-dc converter, a 300-kW experimental prototype was built
in laboratory, as shown in Fig. 12. The control algorithms are
implemented in digital signal processor (DSP)
TMS320F28335. The key system parameters are the same as
simulation and also shown as Table Ⅰ.
Fig. 13 gives the steady-state experimental results of the
front-end stage LLC converter. As can be seen in Fig. 13, the
input voltage of LLC-DCX1 and LLC-DCX2 are well shared
with the proposed control strategy. Fig. 14 gives the steadystate experimental results of the proposed dc-dc converter
under CC charging state, it can be seen that the system is stable
with the charging current is 210 A. Fig. 15 shows the dynamic
experimental results under CV charging state. As it can be
observed in Fig. 15, when the load current step changes from
120 A to 150 A, the output voltage of dc-dc converter has very
short tracking time and nearly without any overshoot. These
steady-state and dynamic results verified very good
performance of the proposed system.
V. CONCLUSIONS
This paper presents a PPP-based 300 kW dc-dc converter
for extreme fast charging station. The system configuration,
operation principle, control and parameters design are
provided in detail, various steady-state and dynamic
simulation and experimental results verified the feasibility of
the proposed PPP-based dc-dc converter. Form the above
simulation and experimental results, it can be concluded that
the PPP-based dc-dc converters are very suitable for high
power and non-isolated applications.
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