IEEE SOLID-STATE CIRCUITS LETTERS, VOL. 2, NO. 6, JUNE 2019
41
Low-Noise Integrated Potentiostat for Affinity-Free Protein Detection With
12 nV/rt-Hz at 30 Hz and 1.8 pArms Resolution
Sean Fischer , Student Member, IEEE, Dante Muratore, Member, IEEE, Stephen Weinreich, Student Member, IEEE,
Aldo Peña-Perez, Member, IEEE, Ross M. Walker , Member, IEEE, Chaitanya Gupta , Member, IEEE,
Roger T. Howe, Fellow, IEEE, and Boris Murmann , Fellow, IEEE
Abstract—This letter presents a low-noise integrated potentiostat
for affinity-free molecular detection in applications for personalized
medicine. The affinity-free sensing technique uses a digital classifier to
identify molecules through unique vibrational signatures. The sensing
mechanism relies on coherent interference of electron wave functions
at the interface between a nanoscale working electrode and a liquid
electrolyte. Coherence at the sensing interface is enabled by low-noise
feedback, which reduces the effective temperature of the electrons. The
described three-channel potentiostat IC uses chopping and correlated
double sampling to achieve an input-referred voltage noise of 12 nV/rt-Hz
at 30 Hz and a current resolution of 1.8 pArms with 0.5-s averaging time.
Each channel consumes 5 mW and occupies 0.41 mm2 in 65-nm CMOS.
Index Terms—Bio-sensing, chopping, correlated double sampling
(CDS), personalized medicine, potentiostat.
I. I NTRODUCTION
Blood protein concentrations are important for the diagnosis and
treatment of disease. Conventional methods of detection [1]–[3] use
specially designed affinity probes (see Fig. 1) that bind to the target protein, separating it from the sample background and labeling
it with an observable tag (optical, magnetic, electronic, etc.). The
implementation of these chemical domain filters is time consuming
and expensive.
A novel method of protein detection described in [4] addresses this
problem by moving the filtering step into the digital domain, eliminating the need for chemical affinity (see Fig. 1). In this approach,
target proteins are isolated from the sample background using a digitally implemented pattern classification algorithm. Since the target
is isolated in the digital domain, a single measurement apparatus
can serve as a general purpose platform for a wide variety of target
molecules. The digital algorithm processes the conductance spectrum of a nanoscale working electrode (WE), defined as (IS )
versus VS , where IS is the dc current through the WE and VS is
the dc potential across the WE/electrolyte interface (see Fig. 2).
Manuscript received April 17, 2019; revised May 29, 2019 and June 24,
2019; accepted June 26, 2019. Date of publication July 3, 2019; date of
current version July 19, 2019. This paper was approved by Associate Editor
Patrick P. Mercier. This work was supported in part by the Defense Advanced
Research Projects Agency through the Mesodynamic Architectures Program
(Dr. Jeffrey L. Rogers, Ph.D., Program Manager) under Grant N66001-111-4111, and through the Dialysis Like Therapeutics Program (Dr. Matthew
Hepburn, M.D., Program Manager) under Grant HR0011-15-C-0131, and in
part by the Maxim Integrated through Stanford’s SystemX Fellow-MentorAdvisor Program. (Corresponding author: Sean Fischer.)
S. Fischer, D. Muratore, S. Weinreich, R. T. Howe, and B. Murmann are
with the Department of Electrical Engineering, Stanford University, Stanford,
CA 94305 USA (e-mail: seanrf@stanford.edu).
A. Peña-Perez is with the Integrated Circuit Department, Advanced
Instrumentation for Research Division, SLAC National Accelerator
Laboratory, Menlo Park, CA 94025 USA.
R. M. Walker is with the Department of Electrical and Computer
Engineering, University of Utah, Salt Lake City, UT 84112 USA.
C. Gupta is with ProbiusDx Inc., El Cerrito, CA 94530 USA.
Digital Object Identifier 10.1109/LSSC.2019.2926644
Fig. 1.
Conventional [1]–[3] versus affinity-free [4] molecular sensing.
The spectrum is sensitive to the vibrational energy of molecules in
the sample. Vibrational energy states modulate the rate of charge
transfer across the WE/electrolyte interface, similar to a solid-state
technique reported in [5]. Vibrational transduction is lost if the power
spectral density of VS is too high. In this letter, we demonstrate an
integrated potentiostat circuit which enables vibrational transduction
using the nanoscale WE by achieving an input-referred noise density
√
of 12 nV/ Hz at f = 30 Hz.
Section II of this letter explains the sensing concept in more
detail. Section III describes the circuit implementation. Section IV
summarizes measured results using a proof-of-concept IC in 65-nm
CMOS. Section V provides a concluding summary.
II. S ENSING M ECHANISM
Charge transfer occurs continuously at the WE/electrolyte interface
(see Fig. 2), even if the net movement of charge is zero. Electron
transfer is facilitated by vibrating molecules in the electrolyte, which
exchange energy with tunneling electrons in quanta of ω when
energy conservation is satisfied, i.e., qVS = ω. This creates a spectrum analyzer for the vibrational signatures of molecules, where high
|(IS )| at VS indicates that a vibrational frequency ω = qVS /
exists in the sample. The scan of VS is implemented with a staircase potential waveform where VS ∝ ω (see Fig. 3). Using
VS ∼ 100 μV implies ω/2π ∼ 30 GHz, which is sufficient
for detecting cytokines in blood. It can be shown that the SNR of
each spectral peak |(IS )| is approximately
(1)
SNRdB ≈ 20 log10 (IS ) − 20 log10 σ (IOUT ) − 7.8
when VS /σ (VS ) > 100, which implies σ (VS ) ∼ 1 μVrms . Typical
transducer |(IS )| = 10 pA, so resolution σ (IOUT ) = 2.5 pArms
implies SNR = 4.2 dB.
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42
IEEE SOLID-STATE CIRCUITS LETTERS, VOL. 2, NO. 6, JUNE 2019
Fig. 4.
WE readout.
Fig. 5.
Amplifier with CDS.
Fig. 2.
Molecular vibrations modulate charge transfer at the transducer
interface and create spikes in conductance spectrum [4].
Fig. 3.
Arrayed measurement implementation.
The motion of molecular oscillators in the sample approximately follows a second-order response. The transducer gain ω →
|(IS )| is highest when the molecular oscillators are underdamped.
From the fluctuation-dissipation theorem, electronic noise added to
VS increases the damping γ seen by the oscillator
v2s
= 4kTγ
f
(2)
where k is the Boltzmann constant, T is the temperature, and the LHS
is the power spectral density of VS . The first-principle analysis and
supporting experimental work in [4] indicate that a power spectral
√
density of 15 nV/ Hz yields sufficient transducer gain. To achieve
σ (VS ) ∼ 1 μVrms , the bandwidth of VS should be fu < 2.8 kHz (see
Fig. 4, 2π · fu ∼ ao /Ro Co ).
III. C IRCUIT I MPLEMENTATION
√
Achieving 15-nV/ Hz power spectral density in fu < 2.8 kHz
while maintaining sufficient current resolution σ (IOUT ) is challenging in advanced CMOS processes due to 1/f noise in active devices.
The power spectral density added to VS is
v2
i2
v2s
≈ 2 in + in
f
f
f
Ro 2
ao
(3)
where vin and iin are defined in Fig. 4. Chopper modulation upconverts the 1/f noise of the amplifier in Fig. 4 to 240 kHz, where it
is removed from VS by the low-pass response of the feedback loop.
The amplifier also employs correlated double sampling (CDS) prior
to chopper modulation, in order to keep chopper ripple caused by
the upconverted offset voltage of the amplifier less than the peak-topeak thermal noise added to VS . Chopping and CDS add switched
capacitance and charge injection at the amplifier input, increasing
the current noise iin and forcing longer averaging time to achieve
σ (IOUT ). The amplifier with CDS is implemented using a two-sample
integrator and a boxcar sampler (see Fig. 5), which minimizes the
switching capacitance at the amplifier input relative to alternative
CDS implementations where sample subtraction is implemented with
a switched capacitor at the amplifier input. However, the output of the
two-sample integrator must handle the amplified offset of the boxcar
sampler during the offset-sampling phase. We address this problem
with a 3-bit offset tuning DAC at the output of the Gm -cell, which
prevents integrator saturation by keeping VOS < 1 mV prior to CDS.
Switches at the amplifier input are bootstrapped [6] to maintain constant thermal noise over the input range 1.0 V < VWE <
1.6 V. Switching the bootstrap capacitors connected to the inverting
amplifier input in Fig. 4 increases the current noise iin . To address
this problem, the bottom plates of the bootstrap capacitors are connected to the noninverting amplifier input, where higher current noise
is acceptable. The negative feedback minimizes the voltage difference between the amplifier inputs, so the switch resistance remains
approximately independent of VWE .
The first-order feedback loop in Fig. 4 quantizes IS . Firstorder modulators suffer from low-frequency limit cycles, which
would corrupt current measurements that use long averaging to
achieve low σ (IOUT ). We address this problem by dithering an
8-bit quantizer with the amplifier thermal noise, enabling sufficiently
low σ (IOUT ) despite increased iin from chopping and CDS. Static
nonlinearity from the 8-bit feedback DAC is corrected through startup
calibration.
FISCHER et al.: LOW-NOISE INTEGRATED POTENTIOSTAT FOR AFFINITY-FREE PROTEIN DETECTION
Fig. 6.
Fig. 7.
Measured noise density of readout referred to WE.
Fig. 8.
Measured error of current readout.
43
Measurement setup for validation of integrated readout.
IV. M EASURED R ESULTS
The three-channel IC and the measurement setup for validation is
shown in Fig. 6. Proof-of-concept electrochemical measurements are
collected with single electrochemical cells. The input-referred voltage
noise of the integrated readout is shown in Fig. 7. The measured spot
√
noise at 30 Hz is 12 nV/ Hz. 1/f noise appearing in the measured
spectrum for f < 20 Hz is from an LT1793 buffer (see Fig. 4), which
is used only for noise characterization in Fig. 7. The buffer allows the
integrated SAR ADC to measure the low-frequency voltage noise of
the amplifier, which would otherwise be dominated by iin · Ro . Tones
appearing at 60 Hz and the first 13 harmonics are attributed to mains
power, since none of these tones match clock frequencies used on
chip. The high frequency tone at the Nyquist frequency is from the
upconverted residual amplifier VOS after CDS. This tone represents a
0.44 μVpp ripple, which is negligible compared to 3.4 μVpp variation
from noise alone.
A 300-M resistor is used to validate the performance of the
current readout. The resistor voltage is scanned differentially from
−300 mV to +300 mV and the current is recorded. The measured
error after gain and offset correction is shown in Fig. 8(a). The standard deviation of the total error current is 4.4 pArms . The second
difference of the error current is shown in Fig. 8(b). The standard deviation of the total error current after a second difference is
4.5 pArms , which is dominated by the thermal noise of the readout.
Picoampere current linearity is achieved through startup-calibration
of the R2R DAC. Calibration is implemented using an external readout, which measures DAC voltages for all 256 codes. A look-up-table
stored in an external FPGA translates the 8-bit SAR codes into 16-bit
codes representing the calibrated R2R output voltage (see Fig. 4).
The standard deviation of the error current due to noise alone in
0.5 s averaging time is 1.8 pArms , which corresponds to an input
√
referred current noise density of 1.27 pA/ Hz. This is larger than
the current noise introduced by Ro = 1 M used in this measurement (see Fig. 4). The extra current noise is attributed to charge
injection from the chopping/CDS switches [7]. An external CDS circuit (see Fig. 3) corrects for signal-dependent input bias current
IB ≈ 100 pA at the negative input terminal of the IC, caused by
a shared 2.5-V supply between the R2R DAC and the bootstrap
capacitors used in the switches at the amplifier input. The residual error after the CDS operation has a component IB due to the
nonlinear small-signal transducer resistance (rs > 1 M), which
changes the virtual ground resistance of the current readout during the signal-sampling phase, i.e., rvg ≈ Ro /ao ||rs . The sensitivity
(rvg /rs )/(rvg /rs ) is less than 0.05% (Ro /ao ≈ 500 ). This
translates to IB ≈ IB · (rvg /rs )/(rvg /rs ) on the order of 50 fA,
which is negligible compared to the 1.8 pArms error due to thermal
noise alone. Repeated current-voltage measurements of the nanoscale
WE in phosphate buffer solution are shown in Fig. 9. The associated
spectra (IS ) versus VS are shown in Fig. 10. The background of
the integrated readout after differentiation is 40 pApp . Spectral peaks
outside of this region represent high SNR vibrational information
which may be useful for molecular sensing.
Table I compares this letter to readouts reported in literature.
Algorithm development for molecular sensing with the transducer
in [4] is ongoing, so assay sensitivity is not reported. Fair comparison of electronic readouts is challenging in this area, since application
requirements vary dramatically. Typical electrochemical readouts do
not emphasize or report input-referred low-frequency voltage noise,
since it is divided by the large transimpedance gain of the readout
when referred to iin . For the application in [4], however, voltage noise
is critical since the transducer gain degrades as the power spectral
density of VS increases. The readout in this letter achieves the best
reported low-frequency voltage noise in Table I, enabling molecular
sensing using the transducer in [4]. The readouts in Table I achieve
better σ (IOUT ) than the readout in this letter, but their higher voltage
noise would render the measured current meaningless for molecular
sensing with the transducer in [4]. Low voltage noise performance is
enabled by chopping and CDS. These techniques increase σ (IOUT )
due to switched capacitance and charge injection, which is minimized
by the choice to implement CDS at the output of the boxcar sampler
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IEEE SOLID-STATE CIRCUITS LETTERS, VOL. 2, NO. 6, JUNE 2019
TABLE I
P ERFORMANCE C OMPARISON
identify molecules through unique vibrational signatures. Vibrational
energy of the molecules in the sample modulate the rate of charge
transfer across the WE/electrolyte interface. Vibrational transduction
is enabled by the low-noise potentiostat in this letter, which achieves
√
12-nV/ Hz noise density at 30 Hz and 1.8 pArms current resolution.
ACKNOWLEDGMENT
The authors would like to thank Mentor Graphics for automated
layout of the amplifier, TSMC University Shuttle Program for chip
fabrication, and ProbiusDx for providing sensors for IC validation.
R EFERENCES
Fig. 9.
Fig. 10.
Measured IV characteristic of nanoscale WE.
Measured conductance spectra of nanoscale WE.
and the alternative bootstrap connection for the input switches. Low
voltage noise performance also comes at the expense of higher power
consumption, which precludes the integration of a high number of
electrodes with this readout. However, only three to five electrodes are
needed for the affinity-free sensing described in [4], since the measured output of one transducer provides high-dimensional data from
which the digital algorithm can detect a variety of molecules.
V. C ONCLUSION
The molecular sensing technique described in [4] detects
molecules without affinity-probes by using a digital classifier to
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