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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 6, JUNE 2012
Dual-Band Long-Range Passive RFID Tag Antenna
Using an AMC Ground Plane
Dongho Kim, Member, IEEE, and Junho Yeo, Member, IEEE
Abstract—A dual-band passive radio frequency identification
(RFID) tag antenna applicable for a recessed cavity in metallic
objects such as heavy equipment, vehicles, aircraft, and containers
with long read range is proposed by using an artificial magnetic
conductor (AMC) ground plane. The proposed tag antenna consists of a bowtie antenna and a recessed cavity with the AMC
ground plane installed on the bottom side of the cavity. The AMC
ground plane is utilized to provide dual-band operation at European (869.5 869.7 MHz) and Korean (910 914 MHz) passive
UHF RFID bands by replacing the bottom side of the metallic
cavity of a PEC-like behavior and, therefore, changing the reflection phase of the ground plane. It is worthwhile to mention that
the European and the Korean UHF RFID bands are allocated very
closely, and the frequency separation ratio between the two bands
is just about 0.045, which is very small. It is demonstrated by
experiment that the maximum reading distance of the proposed
tag antenna with optimized dimensions can be improved more
than 3.1 times at the two RFID bands compared to a commercial
RFID tag.
Index Terms—Antenna, artificial magnetic conductor (AMC),
cavity, dual-band, radio frequency identification (RFID), tag.
I. INTRODUCTION
W
ITH a widespread adaptation of radio frequency identification (RFID) systems, possibilities of RFID tags
being attached on various types of metallic objects have been
increased and many researchers have extensively investigated
for RFID tag antenna design for metallic objects in recent
years. Although a simple design method for a low-cost tag
antenna using a double-folded dipole antenna with T-matching
and a foam spacer has been used in the past [1], most designs
are based on a microstrip-patch-type antenna or a planar inverted-F-type antenna that has its own ground plane [2]–[5].
Later, a somewhat different approach using a dipole antenna
and an artificial magnetic conductor (AMC)-type ground
plane has been studied by the authors for low-profile and
platform-tolerant RFID tags [6], [7]. For multiband operation,
dual-band planar inverted-F antenna (PIFA)-type RFID tag
antennas mountable on metallic surfaces operating in the two
frequency bands over 860 960 MHz passive UHF bands have
Manuscript received November 17, 2010; revised October 21, 2011; accepted
November 26, 2011. Date of publication April 12, 2012; date of current version
May 29, 2012. This work was supported by the Daegu University Research
Grant.
D. Kim is with Department of Electronic Engineering, Sejong University,
Seoul 143-747, Korea (e-mail: dongkim@sejong.ac.kr).
J. Yeo is with School of Computer and Communication Engineering, Daegu
University, Gyeongbuk 712-714, Korea (e-mail: jyeo@daegu.ac.kr).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2012.2194638
also been proposed [8], [9]. However, the read range of all
the tag designs for metallic objects is limited to a few meters,
which is similar to that of a common dipole-like label tag.
In metallic objects such as vehicles, aircraft, ships, heavy
equipment, and metallic containers, there exist various types
of recessed volumes, and these recessed volumes can be used
as “cavities” to form a cavity-backed antenna with a common
label-type RFID tag. This type of RFID tag antenna can increase the read range considerably. By using this concept, an
embedded circular patch-type RFID tag antenna in UHF band
using ceramic material for identifying metallic objects has been
proposed, and the effect of various positions of the embedded
tag in a cavity of a metallic object on the maximum reading distance has been analyzed [10]. However, the read range has not
been considerably enhanced in this case. Recently, both a longrange passive RFID tag antenna consisting of a bowtie-type antenna in a recessed rectangular cavity in metallic objects and an
impedance matching technique for this configuration have been
proposed by the authors [11]. Instead of modifying antenna geometry itself, the coupling between the bowtie antenna and the
cavity has been used to match the antenna’s input impedance to
the tag’s chip impedance. The maximum read range has been
improved about 3.2 times compared to a commercial label-type
RFID tag in single band.
In this paper, we propose a dual-band passive RFID tag antenna applicable for a recessed cavity in metallic objects with
long read range by using an AMC ground plane. The proposed
tag antenna is comprises a bowtie antenna and a recessed cavity
with the AMC ground plane installed on the bottom side of
the cavity. The AMC ground plane is utilized to provide dualband operation at European (869.5 869.7 MHz) and Korean
(910 914 MHz) passive UHF RFID bands, which are allocated very closely, and the frequency separation ratio between
the two bands is about 4.5%. By simply replacing the bottom
side of the metallic cavity with the AMC ground plane, we can
change the reflection phase of the ground plane, and therefore,
we can obtain double resonance at the desired two frequency
bands.
The maximum reading distance of the proposed tag antenna
with optimized dimensions is compared to that of a commercial ALN-9540-WR RFID tag from Alien Technology Co. [12],
which is flexible and cheap and designed to provide performance in a general standalone environment. All simulation data
are obtained using CST Microwave Studio [13].
II. ANTENNA DESIGN AND RESULTS
A. Antenna Design With an AMC Ground Plane
The geometries of the proposed RFID tag antenna and the
AMC ground plane are shown in Figs. 1 and 2, which have
0018-926X/$31.00 © 2012 IEEE
KIM AND YEO: DUAL-BAND LONG-RANGE PASSIVE RFID TAG ANTENNA USING AMC GROUND PLANE
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Fig. 1. Geometry of the proposed dual-band bowtie-type RFID tag antenna
embedded in a recessed volume of a metallic cavity.
Fig. 3. Photograph of (a) the fabricated cavity with the AMC ground plane and
(b) bowtie tag antenna with the attached Higgs-2 chip.
Fig. 2. Geometry of the AMC substrate with
. The radius of the via is 0.5 mm.
mm,
mm, and
optimized design parameters of
mm,
mm,
mm,
mm,
mm,
mm,
mm,
mm,
mm,
mm,
mm,
mm,
,
mm,
mm,
mm, and
mm. Basically, our tag is
a dipole-type antenna. Each arm of the tag is composed of a
modified bowtie-shaped loop. An RFID chip is attached in between the bowtie loops. The tag is etched only on the bottom
side of a Taconic TRF-45 dielectric laminate. The thickness
and the relative permittivity of the substrate is 1.63 mm and
4.5, respectively. In this paper, to show the easiness and effectiveness of impedance matching by using a coupling effect between the tag and the metallic cavity, we intentionally chose
the bowtie-shaped tag antenna, which is very simple and provides little variety in its design parameters for a fine impedance
tuning.
All directions of the rectangular metallic cavity are enclosed
with metallic walls except the top opening (see Fig. 1). The
cavity measures , , and , respectively. To easily attach our
tag on various platform structures, we intentionally included the
metallic wing in our tag antenna design, which entirely encloses
the top opening of the air-filled recessed volume of the cavity.
This wing can alleviate the tag’s input impedance variation from
the possible change of a variety of shapes and material properties of candidate platform structures.
The bowtie dipole antenna is placed inside the rectangular
metallic cavity with a spacing of from the open top aperture
of the cavity. As shown in Fig. 1, to prevent possible damage
caused by any physical impact or weathering, we deliberately
have placed the antenna and the chip facing the bottom of the
cavity.
For the tag chip, we have selected a strap-type Higgs-2 chip
from Alien Technology Co., which is much easier to directly
attach on the bowtie dipole antenna than a bare-type chip
without additional strap. The input impedance of the chip is
about 11-j130 at 910 MHz.
The top view of a compact AMC ground plane in the cavity
consisting of an
array of an -directed narrow rectangular patch pair with offset vias is depicted in Fig. 2. The total
number of AMC unit cells used for the proposed tag is 23. Each
narrow rectangular copper patch is etched on 1.63-mm-thick
Taconic TRF-45 dielectric laminate. The offset via posts are
connecting the rectangular AMC patches to the metallic bottom
side of the cavity, and by using these posts, a high-impedance
frequency band of the AMC can be considerably lowered maintaining the same length of unit cell as indicated in [6].
Fig. 3 presents the fabricated cavity with the AMC ground
plane and the bowtie antenna.
To examine the effect of a tag’s depth
from the top
opening of the cavity on the variations of the input reflection
coefficient ( ) and the induced power of the tag chip, we
have changed from 6 to 30 mm, which is shown in Fig. 4.
To compute the input reflection coefficient, we have replaced
the RFID chip with a discrete port and two identical capacitors.
Two capacitors of 2.68 pF are connected at the two ends of the
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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 6, JUNE 2012
TABLE I
PERFORMANCE COMPARISON BETWEEN THE ALIEN’S ALN-9540 SQUIGGLE TAG AND THE PROPOSED BOWTIE TAG EMBEDDED IN THE CAVITY
Maximum realized gain
Maximum induced power
Maximum increased reading distance
ALN-9540 shows an omnidirectional beam pattern in an H-plane.
the antenna are most well matched to complex conjugate values
of the chip impedance at both frequency bands as shown in
Fig. 4(a) and (b), and these optimized values are used for the
final design and fabrication.
The realized gain of the proposed antenna is 6.74 dBi at 869
MHz and 6.46 dBi at 913 MHz, respectively, and these are
more than 4.36 dB higher compared to that of a commercial
tag ALN-9540 from Alien Technology Co. [12]. Maximally induced power on the chip is 1.35 dBm at 869 MHz and 1.05 dBm
at 913 MHz, respectively, which is more than 4.72 dB larger
than that of ALN-9540 (see Table I).
For the comparison of RFID tags’ recognition performance,
the maximum reading distance is one important parameter.
Thus, we have compared the reading distance between the
proposed tag and the commercial ALN-9540 tag, which can be
computed by
(1)
Fig. 4. Effect of the distance on an antenna performance (a)
and (b)
power induced in the RFID chip of the tag embedded inside the cavity ( -polarized 1-V/m plane-wave incidence is assumed).
port of 11 , which gives impedance of about 11-j130
at
910 MHz.
To calculate the induced power on the chip, we need a reader
antenna. Of course, we can model the reader antenna and include it in our computer simulation together with the proposed
tag antenna. However, this kind of approach demands extremely
large computation cost not only for computer memories, but for
a simulation time. Hence, instead of directly model the reader
antenna, we have supposed that an -polarized plane wave is an
incoming signal toward the tag from the reader antenna. We assume that the plane wave has an electric field strength of 1 V/m.
It is observed that two different resonant frequency bands can
be produced by placing the bowtie antenna inside the recessed
cavity with the AMC ground plane, and the locations of the two
frequency bands and the matching characteristic of the antenna
can be adjusted by changing the spacing of the antenna inside
the cavity. In fact, as the spacing decreases, the first resonant
frequency moves toward higher frequencies, while the second
one shifts toward lower frequencies, and the antenna’s matching
characteristic improves. At
mm, the input impedances of
where
is a difference of maximum reading distances between tag1 and tag2,
is the induced power at each chip of
RFID tag antennas, and
is antenna gain of each tag. The detailed derivation of (1) is given in [11]. Based on the computed
gain and induced power, we could determine that our antenna
shows much better performance in the reading distance than the
commercial ALN-9540 squiggle tag, which is more than at least
2.9 times longer at both frequency bands. The detailed comparison between the proposed antenna and ALN-9540 is summarized in Table I.
Now, we show Fig. 5 to explain how dual-band operation can
be achieved from a single-band cavity antenna simply by installing the AMC structure. Fig. 5(a) shows the
characteristic of the proposed antenna without the AMC ground plane.
In this case, we see that single resonant frequency band is produced at around 890 MHz, and the input reflection coefficient
of 20 dB and the realized gain of 6.9 dB are achieved. Note
that these values are very similar to those obtained at dual bands
when the AMC ground plane is used. Fig. 5(b) shows the total
phase behaviors of total fields computed at
mm (at the
open top of the cavity) and at
mm (near the bottom of
the cavity) with and without the AMC ground plane. First, when
we do not install the AMC ground plane, the total phase at the
resonant frequency of 890 MHz is 90 . In other words, the phase
of about 90 can be considered as a best phase value resulting
in the best impedance matching and the highest realized gain
characteristics for the proposed tag environment. At this point,
it is worth noting that the key idea of impedance matching in
KIM AND YEO: DUAL-BAND LONG-RANGE PASSIVE RFID TAG ANTENNA USING AMC GROUND PLANE
Fig. 5. (a)
and realized gain of the proposed RFID tag antenna with and
without the AMC ground plane. (b) Total phase responses of total fields with
and without the AMC ground plane.
this paper is none other than utilizing coupling effect between
the tag and cavity. Therefore, we can come to a conclusion that
the phase of 90 is the result of an optimized effect of coupling
for the best impedance matching. A similar phase response of
an AMC ground plane was also reported in [14].
Next, we have examined the phase value with the AMC
ground plane. When the AMC ground plane is installed on the
bottom side of the cavity, we can find the total phase of 90
at two frequencies of 869 and 913 MHz, which is different
from the resonant frequency without the AMC. The reason for
this can be explained by observing the total phase at the top
opening of the cavity
mm with the AMC ground plane.
At 869 MHz, the total phase at
mm and
mm
are almost the same to be about 90 . However, the total phase
at
mm is about
at 913 MHz, and this phase
value is complemented by the pass difference of 48 mm from
the AMC to the cavity open top ( -plane at
). Therefore,
the total phase at
mm again becomes around 90 at
913 MHz. Consequently, we can say that the first resonance
at around 869 MHz comes from the cavity structure itself,
while the second resonance at 913 MHz is from the negative
reflection phase of the AMC ground plane.
To further investigate the characteristics of the compact AMC
ground plane, the variation of the reflection phase of the unit
cell of the AMC ground plane when the position of the offset
via post changes is plotted in Fig. 6(a). For this purpose, the
distance
between the via post and the center of the unit
cell is varied from 40.3 to 42.3 mm. We observe that the zero
reflection phase frequency position of the AMC unit cell moves
toward lower frequency as the distance
increases, which
means the offset in the via posts is increased. Fig. 6(b) shows
the effect of varying
on the input reflection coefficient of
the antenna. It is seen that the two resonant frequency bands can
be controlled by the offset distance of the AMC ground plane.
Fig. 6. Effects of a via position
and (b) input reflection coefficients (
Fig. 7. Effect of a cavity size
properties.
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on (a) reflection phase of the AMC
) of the tag antenna.
on antenna’s impedance matching
Similar to the behavior of the reflection phase characteristic of
the AMC unit cell, the input reflection coefficient of the antenna
shifts toward lower frequency as the offset distance
increases. It is important to note that any two impedance-matched
frequency bands for each value of
correspond to the frequencies of total phase of about 90 , which is a starting point of
the proposed dual-band tag antenna design.
Next, we have investigated the effect of a cavity size on an
input reflection coefficient of our antenna, which is shown in
Fig. 7. In the figure,
and
denote the length and width of
the recessed rectangular cavity (see Fig. 1). From the figure, we
can find a resonant frequency variation tendency of the input
reflection coefficient caused by changing the dimension of the
cavity, which is another important antenna tuning parameter together with the via offset distance
shown in Fig. 6(b).
There are two ways to move resonant frequency bands toward
higher frequencies: One is to increase , and the other is to
decrease . That is to say, increasing or decreasing of
and
results in an opposite effect, which can be explained by analyzing induced surface current distribution on the antenna and
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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 6, JUNE 2012
Fig. 8. Equivalent circuit to model the proposed RFID tag antenna shown in
Fig. 1.
the internal walls of the cavity. The detailed explanation for the
reason of variation in the input reflection coefficient shown in
Fig. 7 can be found in [11].
B. Equivalent Circuit Representation and Experimental
Results
Fig. 8 shows an equivalent-circuit model of our antenna,
which consists of some lumped elements [15], [16]. To model
the reactance of the bowtie antenna at a very low-frequency
region, a series capacitor
is introduced, and simply to
negate the effect of
at higher frequencies,
is connected.
A parallel
resonance network of
, , and
creates
resonance at frequencies lower than , where
denotes a
radiation resistance. To describe the parasitic coupling effect
between the tag and the cavity, the shunt capacitance
and
inductance
are also inserted [11]. After that, the antenna’s
input impedance
seen from an antenna feed point can be
written as
Fig. 9. Comparison of antenna’s input impedance (
in Fig. 8) computed
from 3-D simulation of the antenna structure and from an equivalent circuit
shown in Fig. 8. (a) Resistance. (b) Reactance.
(2)
Fig. 10. Surface current density distribution induced on the AMC substrate at
913 MHz.
To complete circuit representation, we computed best values
for each lumped element by using a hybrid genetic algorithm
(HGA) with a population of 12. The target fitness function that
should be minimized is set by
the HGA optimization, each circuit component value is determined like the following:
pF,
nH,
pF,
nH,
pF,
nH,
and
k . We can see that the HGA result agrees very
well with the simulated antenna’s input impedance, which verifies that the proposed equivalent-circuit model and the computed lumped elements values are fairly accurate.
Fig. 10 presents the vector surface current distribution induced on the AMC ground plane at 913 MHz. It is clearly observed that most energy is concentrated between the via posts
on the AMC, and this makes the offset via lower the zero reflection phase frequency of the AMC.
The input impedance variations for some RFID tag antennas
are depicted in a Smith chart normalized with 50 , which is
given in Fig. 11. In the figure, the solid line with square symbols
denoted as
corresponds to the proposed tag antenna, and
and
are for the same antenna without the AMC ground plane
and the standalone bowtie antenna in free space, respectively.
(3)
where
is the total number of frequency points within a target
frequency band,
and
are real and imaginary parts of the input impedance computed by CST MWS, and
and
are real and imaginary parts of the
input impedance obtained from the circuit elements by using the
HGA.
We compared the simulated input impedance calculated from
a three-dimensional field computation with the optimized circuit impedance
(see Fig. 8), which is given in Fig. 9. After
KIM AND YEO: DUAL-BAND LONG-RANGE PASSIVE RFID TAG ANTENNA USING AMC GROUND PLANE
Fig. 11. Comparison of the impedance matching behaviors for some tags:
the proposed tag antenna embedded in the cavity with the AMC substrate;
the same antenna used in
without the AMC substrate;
the same
in free space, where
.
bowtie antenna used in
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Center, Incheon, Korea. During the experiment, we set the transmission power of the reader antenna as 36 dBm. Compared
to ALN-9540, we can clearly see the dual-band characteristic
of the proposed tag, and the minimum tag sensitivity is about
10 dB smaller at both around 869 and 913 MHz. The maximum
reading distances are more than 3.1 times longer at both frequency bands.
Finally, we summarize the theoretical and experimental
performance of the proposed tag and the ALN-9540, which is
compared in Table I. The measured maximum reading distance
of the proposed tag is 22.75 m at 864 MHz and 23.74 m at
910 MHz, respectively, while it is 7.23 m at 869 MHz and
6.54 m at 910 MHz, respectively, for ALN-9540. Note that the
frequencies chosen in Table I are those that provide maximum
reading distances. Therefore, the distance increase at both
bands is more than 3.1 times, which very well agrees with
the theoretical reading distance derived by using (1). This is a
result of trading off a 3-dB gain bandwidth because ALN-9540
is omnidirectional on the H-plane and its 3-dB gain bandwidth
is broader than the proposed tag covering 860–940 MHz.
Consequently, we can confirm that the proposed design method
is greatly successful to improve the tag performance for various
platform materials.
III. CONCLUSION
Fig. 12. Radiation pattern (realized gain) of the proposed RFID bowtie-tag anMHz.
tenna embedded in the large metallic body at
Fig. 13. Comparison of measured sensitivity and reading distances for the proposed and the ALN-9540 tag.
The radiated realized gain properties of the proposed antenna
on E- and H-planes at 913 MHz are plotted in Fig. 12. We intentionally, omitted the radiation pattern at 869 MHz because it
is very similar to that shown in Fig. 12. The 3-dB beamwidths
in E- and H-planes are 81 and 93 at 869 MHz, and 77 and
90 , respectively, at 913 MHz.
Fig. 13 shows the measured minimum tag sensitivity and
the maximum reading distance of the proposed tag from 860
to 940 MHz. Minimum tag sensitivity, which is the same as
a minimum reader transmission power enabling communication with the tag, and the maximum reading distance of the
proposed tag are measured by using a commercial TESCOM
TC-2600A RFID tester with ALR-9800 reader. The experiment
was carried out in a fully anechoic chamber at the RFID/USN
We have proposed a long-range dual-band passive RFID tag
antenna applicable for a recessed volume in metallic objects
such as heavy equipment, vehicles, aircraft, and large-sized
containers by using an AMC ground plane. The proposed tag
antenna consists of a bowtie antenna and a recessed cavity
with the AMC ground plane installed on the bottom side of the
cavity. The AMC ground plane is utilized to provide dual-band
operation at European (869.5 869.7 MHz) and Korean
(910 914 MHz) passive UHF RFID bands by replacing the
bottom side of the metallic cavity with a PEC-like behavior and,
therefore, changing the reflection phase of the ground plane.
The frequency separation ratio between the European and the
Korean UHF RFID bands is very small, and the dual-band operation for this kind of very low-frequency separation ratio can
be achieved by using the AMC structure. In general, RFID tags
covering both UHF frequency bands (European and Korean)
have good impedance matching property from 869 to 920 MHz
including non-RFID frequency band in between the two RFID
bands. However, our antenna shows good impedance matching
behavior only in the two RFID bands. Therefore, we can say
that our antenna itself is equipped with a kind of a guard-band
filter, which filters out signals existing in non-RFID frequencies, and is therefore helpful to prevent possible hazardous
external interference.
Although the aperture size of our tag is
with
a cavity height of
, which is larger than the commercial
ALN-9540 RFID tag (
with negligible height),
the main purpose of the proposed tag is on special identification
of large metallic bodies such as heavy vehicles, containers, etc.,
on which our tag can be considered relatively small in comparison to the size of those platform bodies. Consequently, in spite
of not using any kind of active power source (like batteries),
we can obtain more than 3.1 times longer reading distance than
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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 6, JUNE 2012
the commercial ALN-9540 RFID tag at the target RFID bands,
which is fairly desirable to identify large-sized metallic bodies.
This proves the validity and usefulness of our design approach
from the theoretical and practical points of view. For the future
work, we are focusing on a miniaturization of the proposed tag
for a smaller size tag for long-distance applications.
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Dongho Kim (M’08) received the B.S. and M.S. degrees in electronic engineering from Kyungpook National University, Daegu, Korea, in 1998 and 2000,
respectively, and the Ph.D. degree in electrical and
electronic engineering from Korea Advanced Institute of Science and Technology (KAIST), Daejeon,
Korea, in 2006.
From 2000 to 2011, he was a Senior Researcher
with the Electronics and Telecommunications Research Institute (ETRI), Daejeon, Korea, where
he was involved with the development of various
antennas including RFID, mobile communication and high-gain Fabry–Perot
resonance antennas, and artificially engineered structures such as electromagnetic band-gap (EBG) structures, frequency selective surfaces (FSSs), and
artificial magnetic conductors (AMCs), etc. In 2011, he joined the Department
of Electronic Engineering, Sejong University, Seoul, Korea, where he is now
an Assistant Professor. His research interests include advanced electromagnetic
wave theory and applications, design of highly efficient and miniaturized antennas using artificially engineered materials, design of EBG structures, FSSs,
and AMCs, platform-tolerant special RFID antenna design, and development
of a variety of metamaterials with negative permittivity and permeability.
Prof. Kim is a life-member of the Korean Institute of Electromagnetic Engineering and Science (KIEES).
Junho Yeo (S’01–M’08) received the Bachelor’s
and Master’s degrees in electronics engineering from
Kyungpook National University, Daegu, Korea, in
1992 and 1994, respectively, and the Ph.D. degree
in electrical engineering from Pennsylvania State
University, University Park, in 2003.
During 1994 and 1999, he was a Researcher with
the Republic of Korea Agency for Defense Development (ROKADD), Daejeon, Korea, where he was
involved with the development of missile telemetry
systems, especially the design and fabrication of lowprofile transmitting and ground-station receiving antennas. From 1999 to 2003,
he was a Graduate Research Assistant with the Electromagnetic Communication Laboratory (ECL), Pennsylvania State University. From September 2003
to June 2004, he was a Postdoctoral Research Scholar in the same laboratory.
In August 2004, he joined Radio Frequency Identification (RFID) technology
research team, Electronics and Telecommunications Research Institute (ETRI),
Daejeon, Korea, as a Senior Researcher. Since March 2007, he has been an Assistant Professor with the School of Computer and Communication Engineering,
Daegu University, Gyeongsan, Korea. His research interests include computational electromagnetics, design of a class of antennas using electromagnetic
band-gap (EBG) and artificial magnetic conductor (AMC) structures for RFID
and mobile applications, portable wideband directive antenna design, and development of RFID sensor tags and long-range passive RFID tags.
Prof. Yeo is a member of the IEEE Wave Propagation Standards Committee
and a reviewer for the IEEE TRANSACTION ON ANTENNAS AND PROPAGATION,
IET Microwaves, Antennas and Propagation, Progress in Electromagnetic Research, and the ETRI Journal.