Designing a Microstrip Patch
Antenna in the WiFi Region
Abstract- This report describes
simulations done in CST Microwave Studio to
investigate the properties of microstrip patch
antennas. It is divided into several parts. The
first part of the report outlines the basic design
of the patch antenna and the tools in CST which
allow for the investigation into its properties.
The second part discusses ways to increase the
bandwidth of antenna operation by adjusting
the patch parameters. The third part describes
what happens when the aspect ratio is changed.
this narrow bandwidth with varying results. Due to
the small separation between the patch and the
ground plane, the patch antenna can only handle
relatively lower radio frequency (RF) power levels
(a few tens of Watts). It also suffers from relatively
larger ohmic losses compared to other types of
antennas of similar aperture size which normally
occur at the dielectric substrate and the metal
conductor in the feed.
Keywords: patch, antenna, bandwidth, gain,
1. Introduction
Printed antennas are a type of antenna that is
usually manufactured using printed circuit board
(PCB) technology onto a substrate or board. The
most common type of printed antenna is the
microstrip patch antenna. This is made up of a
ground plane, a substrate and the patch antenna. An
example of this antenna is shown in figure 1. In this
configuration, the electric field is concentrated
within the substrate and the charge is mainly
distributed on the lower surface of the patch.
Fringing at the edges means that there is a field
distribution in the air region around the surface of
the patch. This will be more obvious from the Efield and surface current analysis in the following
sections.
Microstrip patch antennas carry many advantages
that make it an attractive design choice. Firstly, it is
a planar structure and can be made to fit the shape
of the surface it is placed on. It does not need to
have a large footprint allowing it to be integrated
easily with other circuit elements. It has a low radar
cross-section which makes it attractive in
applications relating to stealth. As mentioned
before, they can be manufactured using PCB
technology and be made to be quite durable. This
also means that they can easily be made into an
array which opens the possibility of it being multifunctional.
However, the patch antenna has several
disadvantages. In its basic form, the patch antenna
has a narrow bandwidth (usually less than 5%).
Various methods have been devised to overcome
Figure 1 Microstrip line fed patch antenna
2. Basic Design of the Microstrip Patch
Antenna
A basic microstrip patch is designed and its
properties are investigated.
Firstly, the material used for each component of the
microstrip antenna was considered. Normally, thin
copper foil is used for the metallic patch and
ground due to the requirement of a good conductor.
The substrate mainly serves as mechanical support
for the patch and to maintain the required spacing
between the patch and the ground plane. The
choice of dielectric mainly depends on cost and its
dielectric constant. For good bandwidth (wider)
performance, lower dielectric constants are
desirable. Many PCBs use FR-4 as the substrate
because of its low dielectric constant and low cost
(admittedly for the lower grade ones). Hence, FR-4
was used for the basic design.
For the patch to be able to radiate, there needs to
be a feed which will supply the signal to the patch.
There are a few methods of doing this; using a coax
feed through the ground and substrate, using a
microstrip line feed, proximity coupled feed, and
aperture coupled feed. Due to the complexities
associated with multilayer fabrication, the aperture
and proximity coupled feeds are not considered.
The coax feed and microstrip line feed are both
easy to match by controlling the insert position of
the feed. A coax feed is more straightforward when
it comes to changing the feed position. Therefore,
the coax feed was used in the basic design.
The most common patch element is the rectangular
patch probably due to ease of manufacture.
Therefore a rectangle with dimensions a×b shape
was used in the basic design.
Next, the desired operating frequency of the patch
is around the 2.0-2.5 GHz range which corresponds
to the WiFi region. The expression used to calculate
the operating frequency is
𝑓𝑟 =
Eq. 1
2(𝑎+ℎ)√𝜀𝑒𝑓𝑓
1
𝜀𝑒𝑓𝑓 =
2
+
The location of the feed is an important parameter
that determines the nature of the radiation of the
patch as well as its impedance. The parameter y in
figure 2 denotes the vertical distance of the centre
of the feed from the centre of the patch. For now,
only the vertical distance will be varied and the
horizontal position will remain constant at the halfway point. It is desirable that the impedance of the
patch is matched to that of the coax feed (about 50
Ω) to minimise reflections that occur at the
interface.
3. Simulations
𝑐
where c is the speed of light, a is the width of the
rectangular patch, h is the thickness of the
substrate, and εeff is the effective permittivity of the
structure which is found using
𝜀𝑟 +1
accordingly based on Eq. 1 to obtain the desired
frequency. The rectangle then would have an aspect
ratio of 1.5:1 making a = 45 mm.
𝜀𝑟 −1
2
(1 +
10ℎ −2
)
𝑏
Eq. 2
CST Microwave Studio is a simulation tool used to
simulate the behaviour of material in the presence
of an electromagnetic field. The simulation is based
on the finite integration technique solving in either
the time domain or frequency domain. In this
report, all the simulations carried out were done
using the Time Domain Solver.
The set up of the patch antenna is shown in figure 2.
where εr is the dielectric constant (which is 4.5 for
FR-4) and b is the length of the patch.
The designer needs to select values for a, b, and h
that would result in the desired frequency. For h,
the designer’s hand may be forced. This is because
companies that produce the substrates already
have a set of thicknesses to choose from. For
example, the lab supplies boards with thicknesses
of 1.6 mm. This was chosen for the basic design.
There is another expression to predict the resonant
frequency of a certain mode and it depends on a
and/or b
𝑓𝑚𝑛 =
𝑘𝑚𝑛 𝑐
Eq. 3
2𝜋 √𝜀𝑟
𝑘 2 𝑚𝑛 = (
𝑚𝜋 2
𝑎
𝑛𝜋 2
) +(𝑏)
Eq. 4
We can set m = 0 and n = 1 to obtain the resonant
frequency of the TM01 mode. For a resonant
frequency of 2.5 GHz, b = 30 mm. It should be noted
also that Eq. 3 and Eq. 4 neglect the effects of the
fringing fields and two-layer dielectric media so is
not accurate. Having said that, it is a good starting
point and the dimensions can be adjusted
Figure 2 the left panel shows the set up of the patch antenna. The
panel on the right shows a cross-sectional view of the microstrip
patch. Here, a = 45 mm, b = 30 mm, l = 60 mm, w = 90 mm
The thickness of the copper foil for both the patch
and the ground was 0.1 mm. The dimensions for a,
b, and h were as described in the previous section.
Initially, y = -7.7 mm. A useful way of knowing
whether there is matching is through observing the
S11 parameters, which represent the reflection loss
in antenna.
Figure 3 shows a plot of the S11 parameters for the
first resonance at 2.26 GHz.
Figure 3 shows the S11 parameters for the basic design at 2.26
GHz. The measure lines intersectthe curve at -10 dB
Another way to check if the resonance is indeed the
actual resonance is by looking at the imaginary
impedance plot. From RLC circuit theory, resonance
generally occurs when the reactive component is
zero. More specifically, this is when the reactive
impedance crosses from negative to positive. Figure
4 and 5 show the real and imaginary impedance
profiles respectively around the 2.26 GHz
resonance.
called the impedance bandwidth. A commonly used
boundary for acceptable operation is the -10 dB
level in the S11 parameters plot. This equates to
reflection coefficients of 0.1 or less. For the first
resonance, the impedance bandwidth was 0.0781
GHz or 3.46% of the resonant frequency. There may
also be a radiation pattern bandwidth which may
not be the same as the impedance bandwidth.
Generally, the radiation pattern of an antenna does
not change radically at a certain resonance.
The full S-parameter results are shown in figure 6.
The multiple resonances indicate multiple modes of
operation. A way to differentiate the modes is to
look at the E-field distribution at the different
modes. The E-field along the z-component is
expressed as
𝐸𝑧 = 𝐸0 𝑐𝑜𝑠 𝑐𝑜𝑠
𝑚𝜋𝑥
𝑎
𝑐𝑜𝑠 𝑐𝑜𝑠
𝑛𝜋𝑦
𝑏
Eq. 5
Eq. 5 indicates that at particular values of m and n,
the E-field may or may not have a cosine profile. For
example if n = 1, we can expect half a cosine cycle
along the y-axis of the patch. This means that the
amplitude will cross zero at the mid-point of the
vertical length patch. If m = 1, the same would be
expected along the x-axis.
Figure 4 shows the imaginary impedance profile. The reactive
component at resonance is very close to 0
Figure 6 shows the full S11 parameters. There exist two other
resonances at 3.08 GHz and 3.9 GHz
Figures 7, 8, and 9 show the z-component E-field
distributions for the modes at 2.26, 3.08 and 3.9
GHz respectively.
Figure 5 shows the real impedance profile around the first
resonance. The impedance is very close to 50 at resonance
indicating matching
At this point, we can see that the resistance varies
as a function of frequency meaning that the antenna
will be matched to the feed for a limited range of
frequencies around resonance. The range of
frequencies at which the antenna is matched is
or not. Figure 10 shows the E-field distribution of
the patch antenna at the TM01 mode resonance
when y = -0.5. At this value, the impedance was not
matched; therefore the mode was not excited. This
is reflected in the E-field distribution where there
were no zero crossings along the y-axis of the patch.
Figure 7 shows the E-field along the z-component for the TM01
mode
Figure 10 shows the E-field along the z-component for the
unmatched case. Notice that the TM01 is not excited
The directivity of an antenna can be expressed as
𝐷=
Figure 8 shows the E-field along the z-component for the TM20
mode
4𝜋𝑈𝑚𝑎𝑥
𝑃𝑇
Eq. 6
where PT is the total power radiated in all
directions and Umax is the maximum radiation
intensity which occurs at some direction.
Directivity is expressed in units of dBi because the
value is compared to the power radiated by an
isotropic antenna.
Each mode has different radiation patterns because
of this difference in E-field distribution. Figures 11,
12 and 13 show the directivity plots for the TM01,
TM20 and TM21 modes respectively. For the TM20
and TM21 modes, the feed was adjusted so that
impedance was less mismatched.
Figure 9 shows the E-field along the z-component for the TM21
mode
Looking at figure 7, the amplitude of the E-field
along the horizontal edge of the patch does not
change (i.e. there are no zero crossings). There was
however a zero crossing at the mid-way point on
the vertical edge. Thus this mode is known as the
TM01 mode. In the same way, the resonances at 3.08
and 3.9 are TM20 and TM21 modes respectively.
Different modes can have different impedances.
This determines whether the mode will be excited
broadside were more for the TM21 mode. It is also
important to note that the magnitude of the main
lobe for TM01 was higher than that of TM20. This
makes sense since if we assume that at both modes
the patch receives the same power, the magnitude
of a single main lobe would be higher than the
magnitude of one of the main lobes in modes with
multiple main lobes.
Directivity does not take into account ohmic and
dielectric losses. Therefore, term gain (G) is
introduced. Gain is simply expressed as
Eq. 7
𝐺 = 𝑒𝐷
Figure 11 shows the directivity plot for the TM01 mode. The main
lobe is at broadside
where e is efficiency. Since losses result in efficiency
less than 1, G is always less than directivity. Beyond
that, everything else is the same. In figure 12, the
main lobe magnitude is 7.3 dBi. Figure 14 shows
the realised gain for the TM01 mode with lower
main lobe magnitude of 2.8 dB and the same
radiation pattern.
Figure 12 shows the directivity plot for the TM20 mode. There are
two main lobes pointing away from broadside
Figure 14 shows the gain plot for the TM01 mode. The radiation
pattern is similar to the directivity
Another thing to note is that G does not take into
account impedance mismatch at the antenna
terminals. So, if 50% of the power was reflected due
to mismatch G describes how the 50% that was
accepted by the antenna is radiated into space.
Figure 13 shows the directivity plot for the TM21 mode. There are
two main lobes pointing away from broadside
The TM01 mode had its main lobe at broadside
(direction the antenna faces). The TM20 had two
main lobes at angles to the broadside which is
similar to TM21 except that the angles away from
A useful parameter used in antenna theory is the Qfactor. Q-factor is defined as a ratio of the total
energy stored in the patch and the energy
dissipated by it. Its mathematical form is
𝑄= 𝜔
𝑊𝑇
𝑃𝑟 +𝑃𝑑 +𝑃𝑐
Eq. 8
where WT energy stored, Pr is the radiated power,
Pd is the dielectric loss, and Pc is the copper loss.
Since we minimise Pd and Pc by carefully selecting
the materials for the patch and substrate, the main
term in the denominator is Pr. This means that a
high Q indicates that most of the power received by
the antenna is not being radiated. Having a low Q
also means that the operating bandwidth is large.
Therefore, it is desirable to have a low Q factor.
Unfortunately, one of them main disadvantages of
patch antennas is that they have relatively high Q
factors. Hence, there has always been an effort to
lower the Q-factor and several methods are
discussed in this report. In the following sections
most of the analysis will focus on the fundamental
mode because this mode is the easiest to match
with the impedance of the feed.
4. Widening the Impedance Bandwidth
This section discusses a few methods to increase
the bandwidth of the patch antenna. The results
presented are all in the matched case.
The first method to widen the bandwidth is to
increase the value of h by decreasing the Q factor.
However, from Eq. 1, increasing h will decrease the
resonant frequency. Figure 15 shows the S11
parameters for h = 2.6 mm. As predicted, the
resonance red-shifted (decrease in frequency) to
2.215 GHz. Changing h also changed the input
impedance resulting in a slight mismatch. The value
of y for which impedance was matched was 7.8 mm.
The impedance bandwidth obtained was 0.10087
GHz (4.55%) which is about twice as large as the
one obtained in the basic design. Figure 16 shows a
comparison S11 parameter results for h = 3.6 mm
where the bandwidth attained is 0.1226 GHz
(5.6%) with a resonance at 2.188 GHz.
Figure 16 shows the S-parameters for h = 3.6 mm. There is a
continued improvement of bandwidth with h
As mentioned before, increasing h had the effect of
increasing the feed inductance which increased the
imaginary impedance. Generally, above a certain
fraction of the resonant wavelength, the inductance
becomes significant enough to cause impedance
mismatch. Figure 17 shows the S11 parameter
results for h = 7 mm. Here, the first resonance
barely goes under the -10 dB mark. Also, the
bandwidth improvement obtained is very small
compared to that in figure 16.
Figure 17 shows the S-parameters for h = 7 mm.
Figure 15 shows the S-parameters for h = 2.6 mm. There is an
improvement in the bandwidth
Another parameter that affects the bandwidth is εr.
Generally, lower εr results in higher bandwidth.
Again from Eq. 1, decreasing εr causes the
resonance to blue-shift (increase in frequency).
Figure 18 and 19 show the S11 parameters for εr =
2.1 and 6 (referring to Teflon and porcelain). For
Teflon, the bandwidth was 0.08682 GHz and for
porcelain, the bandwidth was 0.02682 GHz which
follows from our predictions. Obviously the
minimum εr possible is unity. It also depends on the
mechanical support the substrate can offer to the
patch.
Figure 18 shows the S-parameters for when the substrate was
Teflon (εr = 2.1). The bandwidth achieved was 0.0862 GHz
Figure 20 shows the S11 parameters for the 1:1 patch
Besides that, the feed position to achieve
impedance matching was closer to the centre which
indicates that a patch of this ratio may be easier to
match when certain parameters are changed.
6. Conclusion
In this report we have presented a basic design of a
patch antenna to operate in the WiFi region. The
properties of this antenna were investigated using
the tools in CST Microwave Studio.
Figure 19 shows the S-paramters for when the substrate was
porcelain (εr =6). The bandwidth achieved was 0.0268 GHz
Additionally, the feed position to achieve
impedance matching did not follow a pattern. For
Teflon, y = -6 mm and for FR-4, y = -7.7 mm.
However, for porcelain the required y = -4.5 mm. So
it was not clear how the feed position is related to
εr .
5. Changing the Aspect Ratio
For the basic design, the aspect ratio was 1.5:1. In
this section the aspect ratio is changed to 1:1 (30
mm × 30 mm) to observe the effect of the aspect
ratio on the antenna operation.
The S11 parameters are shown in figure 20. The
first thing to notice is that the second resonance
was shifted further away from the first resonance.
The higher modes also appear to be closer than for
the 1.5:1 case.
It was shown that increasing h increased the
bandwidth up to maximum value. Beyond this
value, the bandwidth improvement was not
substantial and the inductance introduced caused
impedance mismatch. It was also shown that
decreasing εr improved the bandwidth.
Changing the aspect ratio did not have much of an
effect on the resonant frequency but had the effect
of red-shifting the higher order modes.
7. References
[1] H. J. Visser, Antenna Theory and Applications,
Wiley, 2012