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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
1
Ultra-Wideband Scanning Antenna Array
With Rotman Lens
Amin Darvazehban, Omid Manoochehri, Student Member, IEEE, Mohammad Ali Salari,
Parisa Dehkhoda, Member, IEEE, and Ahad Tavakoli
Abstract— An ultra-wideband (6–18 GHz) phased-array
antenna with a beam scanning angle of ±28° is proposed.
A step-by-step design procedure consisting of beamforming
network (BFN), end-launcher feed adapter, and the radiating
element is presented. Microstrip Rotman lens has been designed
to act as the BFN, and optimized to achieve minimum phaseerror over the whole frequency range. In order to satisfy the
condition needed for avoiding grating lobes, as well as achieving
a wide radiation bandwidth and a high power handling capability,
an E-plane double-ridged horn antenna is used as the radiating
element. A novel wideband end-launcher coaxial to double-ridged
waveguide transition has also been developed for connecting the
BFN to the antenna array. Extensive optimization procedures
have been applied to the end-launched adapter together with
the antenna to achieve the best return loss over the frequency
band of operation. The whole system has been simulated using
CST full-wave simulator. An excellent agreement between the
measurements of the fabricated system and the simulated results
is observed.
Index Terms— Antenna arrays, coaxial to waveguide transition, phased arrays, radar systems, Rotman lens, ultrawideband (UWB) antennas, waveguide components.
I. I NTRODUCTION
L
OW profile ultra-wideband (UWB) phased-array antennas, capable of scanning a large area with high gain
and low-grating lobes, are useful to many ground-based and
airborne communication and radar systems [1]. Various solutions have already been proposed in order to increase the
coverage of broadband multibeam systems. Popular beamforming solutions have been designed based on the Butler
matrix, Rotman lens, and Luneburg lens [2]. A beamforming
network (BFN) based on an N × N Butler matrix can be used
to generate N beams. From the array theory, the optimum
sidelobe level (SLL) for a uniform array is about −13 dB.
To reduce the SLL, a larger number of elements can be
used, however, the complexity of the Butler matrix increases
significantly [3].
Manuscript received September 30, 2016; revised December 9, 2016 and
January 31, 2017; accepted February 6, 2017.
A. Darvazehban, P. Dehkhoda, and A. Tavakoli are with the Electrical
Engineering Department, Amirkabir University of Technology, Tehran
15914, Iran (e-mail: amin.darvazehban@gmail.com; pdehkhoda@aut.ac.ir;
tavakoli@aut.ac.ir).
O. Manoochehri is with the Department of Electrical and Computer
Engineering, University of Illinois at Chicago, Chicago, IL 60607 USA
(e-mail: omanoo2@uic.edu).
M. A. Salari is with the Department of Physics, RWTH-Aachen University,
52074 Aachen, Germany (e-mail: salari.mohammadali@gmail.com).
Color versions of one or more of the figures in this paper are available
online at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TMTT.2017.2666810
The Luneburg lens is a spherical lens designed to have a
gradual variation of permittivity from 2 at the core of the lens,
to 1 on its surface. By placing several feeds around the lens,
it is possible to create a high-gain multibeam antenna with
several beams working independently of each other. The main
drawbacks of the Luneburg lens are high costs, construction
problems, and bulk [4]. In contrast, the Rotman lens has the
advantages of being low cost and wideband; however, it shows
significant losses due to the reflections from sidewalls [5].
In this paper, a new UWB (6–18 GHz) phased-array antenna
is presented for direction finding and radar applications. Three
parts of the proposed antenna including the BFN, the endlaunched feed adapter, and the radiating elements will be
discussed. The main novelty and focus of this paper is the
new optimized microstrip Rotman lens as the BFN. In general,
lens beam-forming networks suffer from a relatively high
phase-error, which causes some difficulties in beam scanning
especially in wide-band applications [6]. Here, we use a
genetic algorithm (GA) to optimize the position of the input
and output ports of the lens in order to have minimum phaseerror over 6–18 GHz.
Since the antenna may be used in high-power applications,
the radiating elements should be capable of handling high
powers if needed. To meet this need, an UWB double-ridge
waveguide antenna is designed for the radiating elements.
The distance between the elements should be small to avoid
emerging of the grating lobes. The transition is end-launched
to reduce the element spacing. To this end, a novel UWB endlauncher coaxial to the double-ridged waveguide adapter is
proposed to connect the Rotman lens to the antennas. Without
using the end-launched adapter, the elements should be fed
from the antenna side that leads to large space between the
array elements. Our proposed adapter consists of a mode
converter which converts the TEM mode of the coaxial line
to TE10 mode of the double-ridged waveguide and a tapered
transmission line matching section in order to match the
50- characteristic impedance of the coaxial line to input
impedance of the antenna in the whole frequency band.
The designed antenna system scans 56° of space by 8 distinct beams which have 1-dB overlapping level of the adjacent
beams at 6 GHz and 3-dB overlapping level of the adjacent
beams at 18 GHz. Its 3-dB beamwidth radiation pattern is 20°
at 6 GHz and 7° at 18 GHz. The return loss of the whole
structure consisting of the Rotman lens BFN, end-lunched
adapter, and the antennas is below −10 dB at 6–18 GHz.
Such an antenna system can be widely used in radars, direction
finding systems, and satellite communications.
0018-9480 © 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.
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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
Fig. 3.
Optimized positions for input and output ports of the lens.
Fig. 1. Top view of the microstrip Rotman lens with a ten-element array of
E-plane double-ridged horn antennas.
Fig. 2.
Rotman lens configuration and design parameters.
This paper is organized as follows. Section II describes
the microstrip Rotman lens configuration and the optimization
results. Section III deals with the design of the radiating
elements and the end-launched transition. The antenna system
is simulated in this section by CST Microwave Studio, and the
results are presented. Section IV shows the fabricated antenna
and the measurements.
II. D ESIGN OF A M ICROSTRIP ROTMAN L ENS
Generating appropriate phase shifts is essential for the
performance of a phased-array system. Common phase shifters
are not capable of providing appropriate phase shifts over a
wide range of frequencies. A wideband alternative to common
phase shifters is the Rotman lens, which has several other
advantages. These advantages include having multiple beams
with no need for switches or phase shifters, a wide scanning angle and pattern switching at high rates [7]. Another
important issue is its capability of operating at X, K u, and
millimeter-wave frequency bands, whereas active digital phase
shifters can be lossy at these frequencies [8].
In a Rotman lens, the required phase distribution on the
antenna ports is achieved by shaping the path, through which
a wavefront travels to create the needed time delay and
Fig. 4. Comparison of the normalized path length error ((L 2 − L 1 )/ f 1 ).
(a) Reference [15]. (b) Our optimization procedure.
consequently the appropriate phase distribution over the end
ports. The Rotman lens is called a true time-delay (TTD)
device meaning that it maintains a constant time delay over
a broadband frequency range of operation (linear phase progression). Waveguide, microstrip, stripline, and surface wave
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DARVAZEHBAN et al.: UWB SCANNING ANTENNA ARRAY WITH ROTMAN LENS
3
Fig. 5. (a) Front view of the E-plane double-ridged horn antenna. (b) Back
view of the E-plane double-ridged horn antenna.
Fig. 7. Simulated results for the (a) return loss of the adapter in the operation
range of frequencies and (b) return loss of the antenna with the end-launcher
coaxial to waveguide adapter.
Fig. 6. (a) Side view of the horn antenna with the end-launcher coaxial to
waveguide adapter. (b) Dimensions of the end-launched adapter.
transmission line implementations of Rotman lenses have been
reported [9], [10].
The schematic of a microstrip Rotman lens is shown
in Fig. 1. There are three types of ports in a Rotman lens:
input, dummy, and output ports. Dummy ports, which are
terminated to matched loads, are essential for reducing the
reflections from side walls as well as increasing the adjacent
beam port isolation [10]. As in Fig. 2, input and output
ports are located on the geometrical beam and inner receiver
contours, respectively.
These ports are connected to the lens central structure by
tapering the microstrip trace lines to minimize the reflections.
Please note that, the number of input ports (called beam
ports) determines the steps of scanning, and the number of
output ports is determined by the desired gain. Output ports
are connected to the phased-array elements via transmission
lines (see Figs. 1 and 2). In our design, a Rotman lens
consisting of eight beam ports and ten array output ports
offers a scanning angle of −28° to +28° with eight steps over
6–18-GHz frequency band. In addition, we have chosen highpower E-plane double-ridged horn antennas as the radiating
array elements (see Fig. 1).
As shown in Fig. 2, the Rotman lens has three focal
points, F1 , F2 , and F3 , that produce theoretically perfect
beams [7]–[10]. However, in practical purposes usually more
than three ports are excited. Locating the beam ports at points
between the focal points produce the phase fronts with phaseerrors at the array element locations. Phase-error between the
array elements may be significant for lenses having a large
number of array feeds with wide angle scanning capability.
Phase-error could cause beam steering errors as well as
increasing the beamwidth and SLL.
In the conventional Rotman’s paper [7], the beam and the
inner receiving contours are assumed to be circular. Hansen [6]
introduced an elliptical beam contour in order to reduce the
path length error. Further optimization methods of this kind
are limited to altering the positions of the nonfocal ports
by changing the eccentricity of the ellipse [11], [12] or
beam port perturbation. A refocusing method for phase error
minimization of a Rotman lens without significant changes
in the shape of the beam and array curves was introduced
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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
Fig. 8. Far-field directivity for azimuth plane, pattern generated by the array
and lens at 18-GHz exciting. (a) First port. (b) Second port. (c) Third port.
(d) Fourth port.
in [13]. In [14], a new method is presented to reduce the
phase error by determining the beam’s location following the
ray optics theory. Another method is based on having three
positions on the radiating array with no phase error for each
feed point on the curve [15]. In this paper, however, we do not
limit our design to a selected number of focal points (beam
ports), where they generate no phase error. Instead, the lens
is designed to achieve minimum average phase error for all
the ports rather than achieving no errors for a limited number
of beam ports. To overcome the problem of generated phase
error, we propose a lens structure in which the locations of
the both beam ports on the beam contour and the array ports
on the inner receiver contour are optimized.
According to the phase error relation reported in [6], we
define an objective function as
o=
M
N |L 2 − L 1 |
(1)
i=1 j =1
where N and M are the number of beam and array ports on the
respective contours. L 1 and L 2 are the electrical path length
from i th beam port to the central phase array element (on the
x-axis; (see Fig. 2) and from i th beam port to j th phase array
element, respectively, as
√
√
L 1 = (Fi O) × εr + W0 × εe
√
√
L 2 = (Fi P j ) × εr + W j × εe + d × sinψa
(2)
where Fi and P j are the position of i th beam port on the
beam contour and the position of j th array port on the inner
receiver contour, respectively. O is the center coordination of
the inner receiver contour that is the reference point of the
lens topology in Fig. 2. W0 is the electrical length of the
transmission line that connects O to the central phased array
Fig. 9. Measured patterns generated by exciting all four ports at (a) 6 GHz,
(b) 10 GHz, (c) 14 GHz, and (d) 18 GHz.
element, and W j is the electrical length of the transmission line
that connects j th array port on the inner receiver contour to the
corresponding radiating element.εr and εe are the permittivity
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DARVAZEHBAN et al.: UWB SCANNING ANTENNA ARRAY WITH ROTMAN LENS
5
Fig. 10.
Measurement setup. (a) Lens and array in anechoic chamber.
(b) Return loss measurements by network analyzer.
TABLE I
B EAMWIDTH AND G RATING L OBE A NGLE
constants of the Rotman lens and the transmission lines,
respectively. In addition, d is the distance between the phasedarray radiating elements, and ψa is the maximum scanning
angle. For this paper, we utilize the GA toolbox in MATLAB
software to optimize Fi , for i = 1, 2, 3, . . . , 8 and P j , for
j = 1, 2, . . . , 10. Please note that the initial population of
the GA is set to the input and output port locations of the
conventional Rotman lens. The optimized ports locations are
demonstrated in Fig. 3.
To verify the validity of our optimization procedure, we
have applied our procedure to the designed lens presented
in [15] and compared the results. The path length error is
normalized to focal length ((L 2 − L 1 )/ f 1 ), and it can be seen
from Fig. 4, the proposed method improves it by two orders
of magnitude.
III. D ESIGN OF THE A NTENNA A RRAY
A. Design of an E-Plane Double-Ridged Horn Antenna
In order to avoid grating lobes with a beam scanning angle
of ±28°, the distance between the array elements must satisfy
the following condition [16]:
d<
λmin
1 + |sinθmax |
θmax =28°
⇒ d < 1.11 cm
(3)
where λmin is the minimum wavelength (wavelength
at 18 GHz), and θmax is the maximum scanning angle.
Decreasing the distance between the array elements also
decreases the directivity, therefore, there is a tradeoff between
the gain of the array and emergence of grating lobes. Table I
shows the beamwidth and the emergence of grating lobes for
Fig. 11. Verification of the TTD property of the designed lens. The locations
of the peaks remain almost unchanged by exciting the (a) first port, (b) second
port, (c) third port, and (d) fourth port.
the 6- and 18-GHz frequencies. This table tells us that grating
lobes, which occur at 18 GHz, are out of our field-of-view,
therefore, they are not problematic. In addition to having a
wide bandwidth, the antenna must be capable of handling high
power, which can easily be handled by waveguide antennas
like the horn antenna.
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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
TABLE II
M EASURED AND S IMULATED B EAMWIDTH FOR D IFFERENT S CAN A NGLES AT VARIOUS F REQUENCIES
Microstrip Vivaldi antennas are wideband, however, they are
not able to handle such a high RF power particularly at high
frequencies. For example, a Vivaldi antenna with a substrate
thickness of 62 mils and a permittivity of 2.2 is able to carry
a maximum power of 100.08 W at 18 GHz [17]. Reducing
the substrate thickness, reduces the power handling capability.
Accordingly, a horn antenna is used as the radiating element.
In order to meet the criterion for avoiding the grating lobes,
the spacing between the horn antennas must be kept as small
as possible. Therefore, the aperture cannot be arbitrarily big
and E-plane horn antennas are used. Using ridges in horn
antenna causes the impedance of the antenna to be matched
gradually to the impedance of the free space, which makes the
antenna wideband. In order to provide a better matching over
the frequency range of operation, ridges are tapered along the
length as well as the width of the antenna, which is shown in
Fig. 5(a).
The back view of the double-ridged horn antenna can be
seen in Fig. 5(b). Although reducing the distance between the
elements causes an increase in the mutual coupling between
them, but also it has the effect of decreasing the beamwidth
variations over the frequency range of operation. This is
another reason for keeping the distance between the elements
as small as possible. However, this small interelement spacing
makes it hard to feed the elements. A horn antenna is typically
fed by an SMA connector attached to its side wall. For a
ridged-horn antenna, the same thing is true with the difference
that the central conductor of the feed goes through one ridge
and gets connected to the other ridge [18]. A waveguide
transition with a coaxial to single-ridged waveguide is presented in [19], where a five-section matching network is
proposed to achieve a bandwidth ratio of 2.5:1. Another
coaxial to ridged waveguide transition using quarter wavelength Chebyshev impedance transformer with a bandwidth
of 18–40 GHz is proposed in [20]. In these methods, the
transition is excited from the side wall of the waveguide.
However, due to space limitations mentioned earlier, sidefeeding is not feasible. We have devised a way for feeding
antennas from the waveguide cross section, which enables a
direct connection between the output port of the Rotman lens
and the E-plane double-ridged horn antenna. For this purpose,
the so-called end-launcher coaxial to waveguide adapter has
been developed [21]–[23]. In these cases, the bandwidth is less
than 40%, which has improved to 100% in our design.
At the input of the end-launched adapter, there is a mode
convertor in order to convert the coaxial TEM-mode to the
TE-mode of horn antennas [24]. The rest of the adapter
provides the impedance matching. Fig. 6 shows the antenna
and the adapter with optimized dimensions. An optimization
process is applied to the adapter in order to minimize the return
loss. The return loss of the adapter in the frequency range of
operation is shown in Fig. 7(a). After achieving an appropriate
adapter with desired specifications, we place it in the back of
the antenna, and the whole structure is optimized once more to
achieve the best performance. The return loss of the antenna
combined with the adapter can be seen in Fig. 7(b).
B. Radiation Pattern
In this section, the generated radiation patterns for
6 and 18 GHz are shown. Due to the symmetry of the input
ports, just the first four ports are excited to show the radiation
patterns and also the beam scanning up to 28°. Fig. 8 shows
the radiation patterns at 18 GHz for different port excitations.
Table II compares the simulated and measured beamwidths as
well as the measured gain.
Table II shows that there is a reasonable agreement between
the measurements and the simulations. In the phased arrays
implemented by a Rotman lens, all the beams can be used
simultaneously. Fig. 9 shows the generated patterns at different
frequencies, when all four ports are excited.
IV. FABRICATION AND M EASUREMENT
In this section, the measurement results of the fabricated
phased-array system are presented. The designed antenna array
together with the lens is fabricated and tested. Fig. 10 shows
the fabricated phased-array system. At each test stage, one
of the input ports is excited, and all the other ports are
terminated to a matched load. Since the system is symmetric,
only the four input ports are tested (see Fig. 9). In the
following, the measurement results of the array pattern in
terms of different frequencies and input ports are given. The
position of the peak of the beam should not change as we scan
through frequency [25]. The measured patterns generated by
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DARVAZEHBAN et al.: UWB SCANNING ANTENNA ARRAY WITH ROTMAN LENS
exciting all four ports at different frequencies are indicated
in Fig. 9. By exciting each of the input ports, a distinct beam
is created in the desired direction. Due to the design and
spacing between the elements, there are no grating lobes at
all, when the first port is excited. Grating lobes occur just at
18 GHz for the patterns with scanning angle of 20° and larger
(ports 3 and 4), which are out of our field-of-view. Moreover,
when the grating lobe is about to emerge at 18 GHz by exciting
port 3 and port 4, the gain of antenna decreases. The ideal
uniform amplitude distribution results in a SLL of 13 dB,
however, in practice, there are some deviations from this ideal
value. As can be seen from Figs. 9 and 11, the SLL varies from
9 to 13 dB. The degradation of SLL is due to the reflections
from side walls as well as imperfections of the measurements
such as the imperfect properties of anechoic chamber, which
could affect the SLL by an amount of 0.5–1 dB.
In order to verify that the designed lens is a TTD device,
the received power for different frequencies versus the angle
has been measured. As can be seen from Fig. 11, the location
of the power peak remains almost unchanged as we scan the
frequency. This shows that the Rotman lens has a very small
phase-error.
V. C ONCLUSION
In this paper, an UWB phased-array system with a total
scanning angle of 56° in the frequency range of 6–18 GHz
has been proposed. A microstrip Rotman lens with eight input
ports and ten output ports has been designed, optimized, and
fabricated to achieve the minimum phase-error. Considering
the properties such as power handling capability, high bandwidth, and small interelement spacing, an E-plane doubleridged horn antenna is designed as the radiating element. Due
to the small interelement spacing, it is required to feed the
horn antenna from the waveguide cross section. Therefore,
an optimized wideband end-launcher coaxial to waveguide
adapter has also been designed. This phased-array system has
been comprehensively simulated, optimized, fabricated, and
measured. A very good agreement between the measured and
simulated results has been achieved.
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Amin Darvazehban was born in Tehran, Iran,
in 1988. He received the B.S. degree in electrical
engineering from Shahid Beheshti University,
Tehran, in 2011, and the M.S. degree in electrical
engineering from the Amirkabir University of
Technology, Tehran, in 2013.
He is currently with the Electromagnetic and
Nondestructive Testing Laboratory, Amirkabir
University of Technology and Telecommunication
Research Center. His current research interests
include microstrip antennas, beam forming
networks, wideband transceivers, and multiple-beam phased-array antenna
systems.
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8
Omid Manoochehri (S’07) was born in
Kermanshah, Iran, in 1985. He received the B.S.
degree in electrical engineering from the Shiraz
University of Technology, Shiraz, Iran, in 2007,
and the M.S. degree in electrical engineering from
Tarbiat Modares University, Tehran, Iran, in 2011.
He has been pursuing the Ph.D. degree at the
University of Illinois at Chicago, Chicago, IL,
USA, since 2015.
His current research interests include miniaturized
frequency-selective surfaces, phase-array design,
scattering of electromagnetic waves, and beam-forming systems.
Mohammad Ali Salari received the B.S. and
M.S. degrees in electrical engineering from the Iran
University of Science and Technology, Tehran, Iran,
the B.S. degree in physics from Bonn University,
Bonn, Germany, and the M.S. degree in physics from
RWTH Aachen University, Aachen, Germany.
He was with the Physical Institute, Bonn
University, where he was involved in ATLAS pixel
detectors as well as ultracold quantum gases. He was
with the Multiphysics Department, Fraunhofer Institute for Scientific Computing and Algorithms,
Sankt Augustin, Germany. He was with RWTH-Aachen University, where
he was involved with the research group of Prof. D. DiVincenzo involved
with quantum information processing with surface acoustic waves. His current
research interests include microwave devices and circuits, microstrip antennas,
and the theory of electromagnetics and quantum information processing.
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
Parisa Dehkhoda (M’12) was born in Tabriz, Iran, in 1978. She received the
Ph.D. degree from the Amirkabir University of Technology, Tehran, Iran.
She is currently an Assistant Professor with the Electrical Engineering
Department, Amirkabir University of Technology. Her current research interests include electromagnetic compatibility, scattering, inverse scattering, and
microstrip antennas.
Ahad Tavakoli was born in Tehran, Iran, in 1959.
He received the B.S. and M.S. degrees in electrical engineering from the University of Kansas,
Lawrence, KS, USA, in 1982 and 1984, respectively,
and the Ph.D. degree in electrical engineering from
the University of Michigan, Ann Arbor, MI, USA,
in 1991.
He is currently a Professor with the Department
of Electrical Engineering, Amirkabir University of
Technology, Tehran. His current research interests
include electromagnetic compatibility, scattering of
electromagnetic waves, and microstrip antennas.