This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES 1 Ultra-Wideband Scanning Antenna Array With Rotman Lens Amin Darvazehban, Omid Manoochehri, Student Member, IEEE, Mohammad Ali Salari, Parisa Dehkhoda, Member, IEEE, and Ahad Tavakoli Abstract— An ultra-wideband (6–18 GHz) phased-array antenna with a beam scanning angle of ±28° is proposed. A step-by-step design procedure consisting of beamforming network (BFN), end-launcher feed adapter, and the radiating element is presented. Microstrip Rotman lens has been designed to act as the BFN, and optimized to achieve minimum phaseerror over the whole frequency range. In order to satisfy the condition needed for avoiding grating lobes, as well as achieving a wide radiation bandwidth and a high power handling capability, an E-plane double-ridged horn antenna is used as the radiating element. A novel wideband end-launcher coaxial to double-ridged waveguide transition has also been developed for connecting the BFN to the antenna array. Extensive optimization procedures have been applied to the end-launched adapter together with the antenna to achieve the best return loss over the frequency band of operation. The whole system has been simulated using CST full-wave simulator. An excellent agreement between the measurements of the fabricated system and the simulated results is observed. Index Terms— Antenna arrays, coaxial to waveguide transition, phased arrays, radar systems, Rotman lens, ultrawideband (UWB) antennas, waveguide components. I. I NTRODUCTION L OW profile ultra-wideband (UWB) phased-array antennas, capable of scanning a large area with high gain and low-grating lobes, are useful to many ground-based and airborne communication and radar systems . Various solutions have already been proposed in order to increase the coverage of broadband multibeam systems. Popular beamforming solutions have been designed based on the Butler matrix, Rotman lens, and Luneburg lens . A beamforming network (BFN) based on an N × N Butler matrix can be used to generate N beams. From the array theory, the optimum sidelobe level (SLL) for a uniform array is about −13 dB. To reduce the SLL, a larger number of elements can be used, however, the complexity of the Butler matrix increases significantly . Manuscript received September 30, 2016; revised December 9, 2016 and January 31, 2017; accepted February 6, 2017. A. Darvazehban, P. Dehkhoda, and A. Tavakoli are with the Electrical Engineering Department, Amirkabir University of Technology, Tehran 15914, Iran (e-mail: firstname.lastname@example.org; email@example.com; firstname.lastname@example.org). O. Manoochehri is with the Department of Electrical and Computer Engineering, University of Illinois at Chicago, Chicago, IL 60607 USA (e-mail: email@example.com). M. A. Salari is with the Department of Physics, RWTH-Aachen University, 52074 Aachen, Germany (e-mail: firstname.lastname@example.org). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2017.2666810 The Luneburg lens is a spherical lens designed to have a gradual variation of permittivity from 2 at the core of the lens, to 1 on its surface. By placing several feeds around the lens, it is possible to create a high-gain multibeam antenna with several beams working independently of each other. The main drawbacks of the Luneburg lens are high costs, construction problems, and bulk . In contrast, the Rotman lens has the advantages of being low cost and wideband; however, it shows significant losses due to the reflections from sidewalls . In this paper, a new UWB (6–18 GHz) phased-array antenna is presented for direction finding and radar applications. Three parts of the proposed antenna including the BFN, the endlaunched feed adapter, and the radiating elements will be discussed. The main novelty and focus of this paper is the new optimized microstrip Rotman lens as the BFN. In general, lens beam-forming networks suffer from a relatively high phase-error, which causes some difficulties in beam scanning especially in wide-band applications . Here, we use a genetic algorithm (GA) to optimize the position of the input and output ports of the lens in order to have minimum phaseerror over 6–18 GHz. Since the antenna may be used in high-power applications, the radiating elements should be capable of handling high powers if needed. To meet this need, an UWB double-ridge waveguide antenna is designed for the radiating elements. The distance between the elements should be small to avoid emerging of the grating lobes. The transition is end-launched to reduce the element spacing. To this end, a novel UWB endlauncher coaxial to the double-ridged waveguide adapter is proposed to connect the Rotman lens to the antennas. Without using the end-launched adapter, the elements should be fed from the antenna side that leads to large space between the array elements. Our proposed adapter consists of a mode converter which converts the TEM mode of the coaxial line to TE10 mode of the double-ridged waveguide and a tapered transmission line matching section in order to match the 50- characteristic impedance of the coaxial line to input impedance of the antenna in the whole frequency band. The designed antenna system scans 56° of space by 8 distinct beams which have 1-dB overlapping level of the adjacent beams at 6 GHz and 3-dB overlapping level of the adjacent beams at 18 GHz. Its 3-dB beamwidth radiation pattern is 20° at 6 GHz and 7° at 18 GHz. The return loss of the whole structure consisting of the Rotman lens BFN, end-lunched adapter, and the antennas is below −10 dB at 6–18 GHz. Such an antenna system can be widely used in radars, direction finding systems, and satellite communications. 0018-9480 © 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. 2 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES Fig. 3. Optimized positions for input and output ports of the lens. Fig. 1. Top view of the microstrip Rotman lens with a ten-element array of E-plane double-ridged horn antennas. Fig. 2. Rotman lens configuration and design parameters. This paper is organized as follows. Section II describes the microstrip Rotman lens configuration and the optimization results. Section III deals with the design of the radiating elements and the end-launched transition. The antenna system is simulated in this section by CST Microwave Studio, and the results are presented. Section IV shows the fabricated antenna and the measurements. II. D ESIGN OF A M ICROSTRIP ROTMAN L ENS Generating appropriate phase shifts is essential for the performance of a phased-array system. Common phase shifters are not capable of providing appropriate phase shifts over a wide range of frequencies. A wideband alternative to common phase shifters is the Rotman lens, which has several other advantages. These advantages include having multiple beams with no need for switches or phase shifters, a wide scanning angle and pattern switching at high rates . Another important issue is its capability of operating at X, K u, and millimeter-wave frequency bands, whereas active digital phase shifters can be lossy at these frequencies . In a Rotman lens, the required phase distribution on the antenna ports is achieved by shaping the path, through which a wavefront travels to create the needed time delay and Fig. 4. Comparison of the normalized path length error ((L 2 − L 1 )/ f 1 ). (a) Reference . (b) Our optimization procedure. consequently the appropriate phase distribution over the end ports. The Rotman lens is called a true time-delay (TTD) device meaning that it maintains a constant time delay over a broadband frequency range of operation (linear phase progression). Waveguide, microstrip, stripline, and surface wave This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. DARVAZEHBAN et al.: UWB SCANNING ANTENNA ARRAY WITH ROTMAN LENS 3 Fig. 5. (a) Front view of the E-plane double-ridged horn antenna. (b) Back view of the E-plane double-ridged horn antenna. Fig. 7. Simulated results for the (a) return loss of the adapter in the operation range of frequencies and (b) return loss of the antenna with the end-launcher coaxial to waveguide adapter. Fig. 6. (a) Side view of the horn antenna with the end-launcher coaxial to waveguide adapter. (b) Dimensions of the end-launched adapter. transmission line implementations of Rotman lenses have been reported , . The schematic of a microstrip Rotman lens is shown in Fig. 1. There are three types of ports in a Rotman lens: input, dummy, and output ports. Dummy ports, which are terminated to matched loads, are essential for reducing the reflections from side walls as well as increasing the adjacent beam port isolation . As in Fig. 2, input and output ports are located on the geometrical beam and inner receiver contours, respectively. These ports are connected to the lens central structure by tapering the microstrip trace lines to minimize the reflections. Please note that, the number of input ports (called beam ports) determines the steps of scanning, and the number of output ports is determined by the desired gain. Output ports are connected to the phased-array elements via transmission lines (see Figs. 1 and 2). In our design, a Rotman lens consisting of eight beam ports and ten array output ports offers a scanning angle of −28° to +28° with eight steps over 6–18-GHz frequency band. In addition, we have chosen highpower E-plane double-ridged horn antennas as the radiating array elements (see Fig. 1). As shown in Fig. 2, the Rotman lens has three focal points, F1 , F2 , and F3 , that produce theoretically perfect beams –. However, in practical purposes usually more than three ports are excited. Locating the beam ports at points between the focal points produce the phase fronts with phaseerrors at the array element locations. Phase-error between the array elements may be significant for lenses having a large number of array feeds with wide angle scanning capability. Phase-error could cause beam steering errors as well as increasing the beamwidth and SLL. In the conventional Rotman’s paper , the beam and the inner receiving contours are assumed to be circular. Hansen  introduced an elliptical beam contour in order to reduce the path length error. Further optimization methods of this kind are limited to altering the positions of the nonfocal ports by changing the eccentricity of the ellipse ,  or beam port perturbation. A refocusing method for phase error minimization of a Rotman lens without significant changes in the shape of the beam and array curves was introduced This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. 4 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES Fig. 8. Far-field directivity for azimuth plane, pattern generated by the array and lens at 18-GHz exciting. (a) First port. (b) Second port. (c) Third port. (d) Fourth port. in . In , a new method is presented to reduce the phase error by determining the beam’s location following the ray optics theory. Another method is based on having three positions on the radiating array with no phase error for each feed point on the curve . In this paper, however, we do not limit our design to a selected number of focal points (beam ports), where they generate no phase error. Instead, the lens is designed to achieve minimum average phase error for all the ports rather than achieving no errors for a limited number of beam ports. To overcome the problem of generated phase error, we propose a lens structure in which the locations of the both beam ports on the beam contour and the array ports on the inner receiver contour are optimized. According to the phase error relation reported in , we define an objective function as o= M N |L 2 − L 1 | (1) i=1 j =1 where N and M are the number of beam and array ports on the respective contours. L 1 and L 2 are the electrical path length from i th beam port to the central phase array element (on the x-axis; (see Fig. 2) and from i th beam port to j th phase array element, respectively, as √ √ L 1 = (Fi O) × εr + W0 × εe √ √ L 2 = (Fi P j ) × εr + W j × εe + d × sinψa (2) where Fi and P j are the position of i th beam port on the beam contour and the position of j th array port on the inner receiver contour, respectively. O is the center coordination of the inner receiver contour that is the reference point of the lens topology in Fig. 2. W0 is the electrical length of the transmission line that connects O to the central phased array Fig. 9. Measured patterns generated by exciting all four ports at (a) 6 GHz, (b) 10 GHz, (c) 14 GHz, and (d) 18 GHz. element, and W j is the electrical length of the transmission line that connects j th array port on the inner receiver contour to the corresponding radiating element.εr and εe are the permittivity This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. DARVAZEHBAN et al.: UWB SCANNING ANTENNA ARRAY WITH ROTMAN LENS 5 Fig. 10. Measurement setup. (a) Lens and array in anechoic chamber. (b) Return loss measurements by network analyzer. TABLE I B EAMWIDTH AND G RATING L OBE A NGLE constants of the Rotman lens and the transmission lines, respectively. In addition, d is the distance between the phasedarray radiating elements, and ψa is the maximum scanning angle. For this paper, we utilize the GA toolbox in MATLAB software to optimize Fi , for i = 1, 2, 3, . . . , 8 and P j , for j = 1, 2, . . . , 10. Please note that the initial population of the GA is set to the input and output port locations of the conventional Rotman lens. The optimized ports locations are demonstrated in Fig. 3. To verify the validity of our optimization procedure, we have applied our procedure to the designed lens presented in  and compared the results. The path length error is normalized to focal length ((L 2 − L 1 )/ f 1 ), and it can be seen from Fig. 4, the proposed method improves it by two orders of magnitude. III. D ESIGN OF THE A NTENNA A RRAY A. Design of an E-Plane Double-Ridged Horn Antenna In order to avoid grating lobes with a beam scanning angle of ±28°, the distance between the array elements must satisfy the following condition : d< λmin 1 + |sinθmax | θmax =28° ⇒ d < 1.11 cm (3) where λmin is the minimum wavelength (wavelength at 18 GHz), and θmax is the maximum scanning angle. Decreasing the distance between the array elements also decreases the directivity, therefore, there is a tradeoff between the gain of the array and emergence of grating lobes. Table I shows the beamwidth and the emergence of grating lobes for Fig. 11. Verification of the TTD property of the designed lens. The locations of the peaks remain almost unchanged by exciting the (a) first port, (b) second port, (c) third port, and (d) fourth port. the 6- and 18-GHz frequencies. This table tells us that grating lobes, which occur at 18 GHz, are out of our field-of-view, therefore, they are not problematic. In addition to having a wide bandwidth, the antenna must be capable of handling high power, which can easily be handled by waveguide antennas like the horn antenna. This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. 6 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES TABLE II M EASURED AND S IMULATED B EAMWIDTH FOR D IFFERENT S CAN A NGLES AT VARIOUS F REQUENCIES Microstrip Vivaldi antennas are wideband, however, they are not able to handle such a high RF power particularly at high frequencies. For example, a Vivaldi antenna with a substrate thickness of 62 mils and a permittivity of 2.2 is able to carry a maximum power of 100.08 W at 18 GHz . Reducing the substrate thickness, reduces the power handling capability. Accordingly, a horn antenna is used as the radiating element. In order to meet the criterion for avoiding the grating lobes, the spacing between the horn antennas must be kept as small as possible. Therefore, the aperture cannot be arbitrarily big and E-plane horn antennas are used. Using ridges in horn antenna causes the impedance of the antenna to be matched gradually to the impedance of the free space, which makes the antenna wideband. In order to provide a better matching over the frequency range of operation, ridges are tapered along the length as well as the width of the antenna, which is shown in Fig. 5(a). The back view of the double-ridged horn antenna can be seen in Fig. 5(b). Although reducing the distance between the elements causes an increase in the mutual coupling between them, but also it has the effect of decreasing the beamwidth variations over the frequency range of operation. This is another reason for keeping the distance between the elements as small as possible. However, this small interelement spacing makes it hard to feed the elements. A horn antenna is typically fed by an SMA connector attached to its side wall. For a ridged-horn antenna, the same thing is true with the difference that the central conductor of the feed goes through one ridge and gets connected to the other ridge . A waveguide transition with a coaxial to single-ridged waveguide is presented in , where a five-section matching network is proposed to achieve a bandwidth ratio of 2.5:1. Another coaxial to ridged waveguide transition using quarter wavelength Chebyshev impedance transformer with a bandwidth of 18–40 GHz is proposed in . In these methods, the transition is excited from the side wall of the waveguide. However, due to space limitations mentioned earlier, sidefeeding is not feasible. We have devised a way for feeding antennas from the waveguide cross section, which enables a direct connection between the output port of the Rotman lens and the E-plane double-ridged horn antenna. For this purpose, the so-called end-launcher coaxial to waveguide adapter has been developed –. In these cases, the bandwidth is less than 40%, which has improved to 100% in our design. At the input of the end-launched adapter, there is a mode convertor in order to convert the coaxial TEM-mode to the TE-mode of horn antennas . The rest of the adapter provides the impedance matching. Fig. 6 shows the antenna and the adapter with optimized dimensions. An optimization process is applied to the adapter in order to minimize the return loss. The return loss of the adapter in the frequency range of operation is shown in Fig. 7(a). After achieving an appropriate adapter with desired specifications, we place it in the back of the antenna, and the whole structure is optimized once more to achieve the best performance. The return loss of the antenna combined with the adapter can be seen in Fig. 7(b). B. Radiation Pattern In this section, the generated radiation patterns for 6 and 18 GHz are shown. Due to the symmetry of the input ports, just the first four ports are excited to show the radiation patterns and also the beam scanning up to 28°. Fig. 8 shows the radiation patterns at 18 GHz for different port excitations. Table II compares the simulated and measured beamwidths as well as the measured gain. Table II shows that there is a reasonable agreement between the measurements and the simulations. In the phased arrays implemented by a Rotman lens, all the beams can be used simultaneously. Fig. 9 shows the generated patterns at different frequencies, when all four ports are excited. IV. FABRICATION AND M EASUREMENT In this section, the measurement results of the fabricated phased-array system are presented. The designed antenna array together with the lens is fabricated and tested. Fig. 10 shows the fabricated phased-array system. At each test stage, one of the input ports is excited, and all the other ports are terminated to a matched load. Since the system is symmetric, only the four input ports are tested (see Fig. 9). In the following, the measurement results of the array pattern in terms of different frequencies and input ports are given. The position of the peak of the beam should not change as we scan through frequency . The measured patterns generated by This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. DARVAZEHBAN et al.: UWB SCANNING ANTENNA ARRAY WITH ROTMAN LENS exciting all four ports at different frequencies are indicated in Fig. 9. By exciting each of the input ports, a distinct beam is created in the desired direction. Due to the design and spacing between the elements, there are no grating lobes at all, when the first port is excited. Grating lobes occur just at 18 GHz for the patterns with scanning angle of 20° and larger (ports 3 and 4), which are out of our field-of-view. Moreover, when the grating lobe is about to emerge at 18 GHz by exciting port 3 and port 4, the gain of antenna decreases. The ideal uniform amplitude distribution results in a SLL of 13 dB, however, in practice, there are some deviations from this ideal value. As can be seen from Figs. 9 and 11, the SLL varies from 9 to 13 dB. The degradation of SLL is due to the reflections from side walls as well as imperfections of the measurements such as the imperfect properties of anechoic chamber, which could affect the SLL by an amount of 0.5–1 dB. In order to verify that the designed lens is a TTD device, the received power for different frequencies versus the angle has been measured. As can be seen from Fig. 11, the location of the power peak remains almost unchanged as we scan the frequency. This shows that the Rotman lens has a very small phase-error. V. C ONCLUSION In this paper, an UWB phased-array system with a total scanning angle of 56° in the frequency range of 6–18 GHz has been proposed. A microstrip Rotman lens with eight input ports and ten output ports has been designed, optimized, and fabricated to achieve the minimum phase-error. Considering the properties such as power handling capability, high bandwidth, and small interelement spacing, an E-plane doubleridged horn antenna is designed as the radiating element. Due to the small interelement spacing, it is required to feed the horn antenna from the waveguide cross section. 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IEEE Antennas Propag. Soc. Int. Symp., vol. 3. Jun. 1998, pp. 1390–1393.  H. Ma, B. Liang, Y. Shi, and J. Miao, “Analysis of coaxial-to-waveguide transitions in end launcher type,” in Proc. Int. Conf. Microw. Millim. Wave Technol. (ICMMT), Shenzhen, China, May 2012, pp. 1–4.  M. Durga, S. Tomar, S. Singh, and L. Suthar, “Millimeter wave inline coaxial-to-rectangular waveguide transition,” in Proc. IEEE Appl. Electromagn. Conf. (AEMC), Kolkata, India, Dec. 2011, pp. 1–3.  C. W. Yuan, Q. X. Liu, H. H. Zhong, and B. L. Qian, “A novel TEM— TE11 mode converter,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 8, pp. 513–515, Aug. 2005.  M. Longbrake, “True time-delay beamsteering for radar,” in Proc. IEEE Nat. Aerosp. Electron. Conf. (NAECON), Dayton, OH, USA, Jul. 2012, pp. 246–249. Amin Darvazehban was born in Tehran, Iran, in 1988. He received the B.S. degree in electrical engineering from Shahid Beheshti University, Tehran, in 2011, and the M.S. degree in electrical engineering from the Amirkabir University of Technology, Tehran, in 2013. He is currently with the Electromagnetic and Nondestructive Testing Laboratory, Amirkabir University of Technology and Telecommunication Research Center. His current research interests include microstrip antennas, beam forming networks, wideband transceivers, and multiple-beam phased-array antenna systems. This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination. 8 Omid Manoochehri (S’07) was born in Kermanshah, Iran, in 1985. He received the B.S. degree in electrical engineering from the Shiraz University of Technology, Shiraz, Iran, in 2007, and the M.S. degree in electrical engineering from Tarbiat Modares University, Tehran, Iran, in 2011. He has been pursuing the Ph.D. degree at the University of Illinois at Chicago, Chicago, IL, USA, since 2015. His current research interests include miniaturized frequency-selective surfaces, phase-array design, scattering of electromagnetic waves, and beam-forming systems. Mohammad Ali Salari received the B.S. and M.S. degrees in electrical engineering from the Iran University of Science and Technology, Tehran, Iran, the B.S. degree in physics from Bonn University, Bonn, Germany, and the M.S. degree in physics from RWTH Aachen University, Aachen, Germany. He was with the Physical Institute, Bonn University, where he was involved in ATLAS pixel detectors as well as ultracold quantum gases. He was with the Multiphysics Department, Fraunhofer Institute for Scientific Computing and Algorithms, Sankt Augustin, Germany. He was with RWTH-Aachen University, where he was involved with the research group of Prof. D. DiVincenzo involved with quantum information processing with surface acoustic waves. His current research interests include microwave devices and circuits, microstrip antennas, and the theory of electromagnetics and quantum information processing. IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES Parisa Dehkhoda (M’12) was born in Tabriz, Iran, in 1978. She received the Ph.D. degree from the Amirkabir University of Technology, Tehran, Iran. She is currently an Assistant Professor with the Electrical Engineering Department, Amirkabir University of Technology. Her current research interests include electromagnetic compatibility, scattering, inverse scattering, and microstrip antennas. Ahad Tavakoli was born in Tehran, Iran, in 1959. He received the B.S. and M.S. degrees in electrical engineering from the University of Kansas, Lawrence, KS, USA, in 1982 and 1984, respectively, and the Ph.D. degree in electrical engineering from the University of Michigan, Ann Arbor, MI, USA, in 1991. He is currently a Professor with the Department of Electrical Engineering, Amirkabir University of Technology, Tehran. His current research interests include electromagnetic compatibility, scattering of electromagnetic waves, and microstrip antennas.