Proceedings of ASME Turbo Expo 2015: Turbine Technical Conference and Exposition GT2015 June 15 – 19, 2015, Montréal, Canada GT2015-42895 Real-Time, Advanced Electrical Filtering for Pressure Transducer Frequency Response Correction Adam M. Hurst1 Steve Carter1 Doug Firth2 Alan Szary2 Joe VanDeWeert1 1 Kulite® Semiconductor Products Inc. One Willow Tree Road Leonia, New Jersey, USA Contact Author Email: adamh@kulite.com 2 Precision Filters Inc. 240 Cherry Street Ithaca, New York, USA response in applications such as shock and blast testing. A theoretical model of the frequency response correction methodology is presented. We additionally present temperature dependent experimental results that compare the frequency response with and without the correction scheme. These results demonstrate that the usable bandwidth of pressure transducers can be increased when real time, analog frequency response correction is applied. This work shows that if the frequency response of a transducer is well characterized, advanced signal conditioning can be implemented to substantially extend the flat bandwidth of the transducer without changes to the sensor, packaging or mounting. ABSTRACT The frequency response of a pressure transducer is influenced by the natural resonance of the sensor structure, the spatial resolution of the sensor due to its diaphragm size, the sensor packaging, signal conditioning and mounting at the measurement location. The resonance of the sensor and aerodynamically-driven resonances related to the sensor packaging and/or mounting, specifically, can distort dynamic pressure measurements within the range of greatest interest (10Hz-20kHz), typically resulting in erroneous amplification. Historically, correcting for such errors within the frequency response of a pressure transducer or measurement system has been challenging, because such errors are hard to quantify with unknown resonant frequencies and damping factors (quality factors). However, with the ability to fully characterize resonant frequencies that lie within 10Hz - 50kHz using a previously demonstrated dynamic pressure characterization methodology, it is possible to apply electrical filtering to substantially extend the flat (0±2dB) frequency response of a transducer before any digital signal conversion. In this work, we present a real-time frequency response compensation scheme that uses electrical filtering to correct for aerodynamically driven packaging or mounting related resonances while at the same time preventing signal distortion caused by the sensor resonances. The compensation extends the useable, flat amplitude bandwidth of the transducer while also correcting the phase response to maintain constant time delay over the extended bandwidth. This real-time frequency response correction scheme can be similarly used to compensate for chip resonances, which can limit the frequency INTRODUCTION The measurement of unsteady or dynamic pressures is critical for gas turbine health monitoring, design validation, research and active control. Unsteady pressures exist throughout rotating machinery that can lead to phenomena such as rotating stall and surge in the compressor, flow instabilities in the combustor, low and high cycle fatigue, accelerated component wear and noise pollution. Through accurate measurement and characterization of wideband pressure levels within gas turbines, engineers can redesign turbines to reduce the presence of undesirable dynamic pressures. Accurate, realtime dynamic pressure measurements on production engines can be used for engine health monitoring as well as active control to prevent potentially catastrophic events such as compressor surge and stall [1, 2]. Dynamic pressure measurements are also critical to understanding and mitigating screech and rumble in gas turbine augmentors [3]. Since these 1 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME time delay errors in the measurement. This problem has historically been addressed by modifying the packaging of the sensor to achieve the highest measurement bandwidth. This results in a sensor design having no protective screen and all cavities around the sensing element filled with an epoxy or a room temperature vulcanizing silicone (referred to herein as RTV) [4, 5]. These package modifications are often at the cost of sensor protection, as high speed, small particles in a wind tunnel or gas turbine can impact the diaphragm of the sensor, catastrophically destroying it. The tradeoff between robust, protective packaging and frequency response, specifically for MEMS sensors (pressure, acoustics, acceleration, etc.) has been a significant challenge. measurements provide such a vital role in both the design and operation of gas turbines, it is important for instrumentation engineers to understand and quantify the accuracy of unsteady pressure measurements. The frequency response of the pressure sensors employed in such measurements along with the mounting within the turbine determines the measurement accuracy. Unfortunately, there are limited published standards for dynamic pressure measurement in gas turbines and sensor calibration as well as no known commercially available, in-situ dynamic pressure calibration tools [2, 4, 5]. Therefore, instrumentation engineers often attempt to characterize the response of a pressure sensor mounted in a test apparatus using the same mounting configuration employed within the turbine [2, 6]. Due to sensor temperature limitations, dynamic pressure sensors are often recessed away from the high temperatures within gas turbines at the cost of frequency response. Historically, the loss of frequency response has been compensated through data postprocessing [7]. This work presents a real-time, analog solution to extend the flat bandwidth of a dynamic pressure sensor and its mounting configuration. Historically, to achieve the widest bandwidth, pressure transducers are mounted flush with the flow field at the desired measurement location [8, 9]. Even with an ideal flush mounting configuration, the natural resonance or size of the pressure sensing diaphragm as well as the packaging have been demonstrated to limit the frequency response [5, 8, 10]. All pressure transducers have a force collecting diaphragm or suspended membrane that deflects under pressure. Diaphragm deflection changes the strain in the diaphragm, which, for Si MEMS piezoresistive pressure transducers, is measured by integrated resistors on or diffused within the surface of the sensor’s Si diaphragm. The Si diaphragm of a typical MEMS pressure sensor is ~500-1000μm with a thickness of several microns to hundreds of microns [11, 12]. This micro-scale geometry and the high stiffness of Si results in a typically natural diaphragm resonance of 100+kHz to several MHz, depending on pressure range. The high natural resonance of the diaphragm and small size of MEMS piezoresistive pressure sensors results in the capability of accurate dynamic pressure measurements over the range of greatest interest 10Hz to 20kHz at the chip level [5]. When the sensing element is packaged into a full transducer housing, there is some volume surrounding the chip and often a protective screen covering the chip, as illustrated in Figure 1. This volume cavity and screen results in aerodynamically-driven resonances, modeled with the modified Helmholtz equation, Eq. 1 [5, 10]. n, Holes of Diameter, a r Protective B-Screen Volume Cavity, VCavity MEMS Pressure Sensing Element Pressure Transducer T Housing Figure 1: Common packaging methodology for MEMS pressure transducers. Similar to packaging, there is again a tradeoff between mounting of sensors at the location of measurement and the achievable frequency response. In practice, close coupling the pressure transducer to the flow is not always possible due to the transducer limited operating temperature range, sensor size, acceleration at the measurement location or other constraints. In such instances, engineers install the transducer at the termination of a recess tube. As illustrated in Figure 2, a transducer at the end of a recess tube results in the well-known organ pipe resonance, Eq. 2 [13]. The organ pipe resonance will amplify dynamic pressure signals by factors as high as ~10+X at the organ pipe resonant frequency depending on tube length, diameter, surface roughness, etc. [14, 15]. As a general rule, the frequency response of a recessed diaphragm system will be useable from static conditions to approximately 30% of the 1st harmonic of the organ pipe resonant frequency where a signal increase of approximately 10% (1 dB) is present. (1) These aerodynamically-driven resonances can be on the order of 25-100+kHz for air which overlaps with the range of measured pressure signals. The resonances cause amplification and/or attenuation of the pressure signal as well as phase or 2 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME substantially improve the accuracy of high-frequency dynamic pressure measurements. The resulting hardware solution will be of significant benefit to instrumentation engineers interested in high frequency measurements and will ultimately result in lower overall program costs. All data reported herein is with air as the media. (2) THEORETICAL BACKGROUND While the resonant frequencies of these packaging and recess mounting aerodynamic resonances have been successfully modeled and are now easily predicted, the quality factor, Q, is not easily predicted. The quality factor is also referred to as the damping factor, ξ, for under-damped second order systems and can be equated using Figure 2: Pressure transducer recessed length, L, from flow field. A 2.54cm (1in) long recess tube terminated with a pressure transducer for example, will exhibit an organ pipe resonance at ~3.36kHz for air at room temperature, yielding a potentially useable flat bandwidth to ~1kHz (± 1dB), (damping or quality factor dependent). This large resonant peak located so close to the desired alias free pressure data bandwidth creates the possibility for clipping on pressure signals located at the resonance as well as an aliasing problem for a data acquisition system. Advanced signal conditioning and filtering techniques are required to address these problems [14]. We present an electronic filtering algorithm and signal conditioning solution (referred to as KSC-2) that solves this historic problem of aerodynamic resonances from packaging and/or recess mounting. The filtering algorithm presented in this work corrects the frequency response in the analog domain, enabling for example the use of robust packaging methods, such as the B-screen illustrated in Figure 1 without a significant loss in frequency response. For example, a Kulite XCS-062 Bscreen configuration without any RTV around the chip exhibits a bandwidth of DC-16kHz (± 2dB), but with the signal conditioning algorithm presented herein the same configuration exhibits a flat frequency response of DC-~45kHz (± 2dB), a ~280% improvement. This solution reduces the tradeoff between robust protective packaging or constraints on recess mounting and the achievable frequency response. We demonstrate that this realtime frequency response correction scheme typically extends the flat frequency response (defined as 0dB ±2dB or less) [16, 17] of a pressure transducer from ~30% of the mounting or packaging resonance to 100% of the resonance value (a 200300% improvement in the flat, useable bandwidth). We further present frequency response data over a range of temperatures and the coefficients to correct the frequency response over temperature. This electrical filter frequency response correction (also referred to as FRC) scheme has been designed and built into a signal conditioning unit that enables easy modification of both the signal conditioning employed (excitation, gain, pre and post amplification, filtering) and frequency response correction parameters for any measurement. The application of the adjustable electronic filter with various pressure transducer packaging schemes and recessed installations will be shown to [18]. Experimental advancements have made these resonances and Q factors easier to accurately measure. A theoretical example of this capability is shown in Figure 3, and an experimental example is shown with the magnitude and phase plot of the Kulite P/N: XCQ-062 B-screen without RTV transducer in Figure 8. The experimental data exhibit a clear resonance peak at ~37kHz with a Q of ~6, which can either be extracted visually or by curve fitting the data to the secondorder system transfer function equation below [19]. (3) Knowing the packaging resonance and its Q, we take the reciprocal of the transfer function, G(s), to create an equal and opposite transfer function which can be used to correct for the resonance of the packaging, extending the useable frequency range. The complementary transfer function to correct a sensor having a cavity resonance as modeled by a second order system is given by: (4) where ωr is the resonant frequency of the cavity in units of rad/sec and Q is the resonance quality factor. The sensor Q and ωr may be estimated or determined experimentally and will vary for different sensor types and packages. It is therefore highly desirable to implement such a frequency response correction approach where Q and ωr are programmable values. By combining Equations 3 and 4, a characterized, unwanted resonant frequency, ωr, can be removed, providing an improved frequency response. As Figure 3 demonstrates, the resulting combined transfer function now exhibits a flat frequency response (0dB). The amplitude response is flattened to the extent that the ωr and Q are known for a particular sensor. The overall phase of the sensor is compensated as well. 3 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME 20 15 second order transfer function, the frequency response correction algorithm utilizes a double integrator with appropriate feedback to realize the denominator of the transfer function. The numerator is realized by weighting of the second order high-pass, band-pass and low-pass outputs, each multiplied by their respective constants, KHP, KBP and KLP. The three outputs are then summed to form the final transfer function. Sensor Response (Uncompensated) 10 Gain (dB) 5 0 Compensated Response –5 –10 KSC-2 Compensator –5.0 –20 3k 5k Figure 3: algorithm. 7k 10k 20k 30k 50k 70k 100k Frequency (Hz) Example frequency 200k 300k response correction The effectiveness of the correction relies on the sensor adhering to a second order response model and the accurate characterization of a particular sensor’s Q and ωr. For sensor Q >> 1, the overall correction of the transfer function is less sensitive to the Q setting of the frequency response correction algorithm. Referring to Figure 4, a sensor with packaging resonances characterized by a Q of 10 shows excellent amplitude response correction with the Q set between 8 and 12.5 (-20%/+25% error). The sensitivity of the frequency response correction algorithm to ωr is much more dramatic, particularly if the compensator ωr is adjusted too low. When the correction frequency is set to 0.8 ωr, the amplitude response is over compensated so that instead of response error due to peaking, there is dip in the frequency response. If the correction frequency is set too high at 1.25ωr the amplitude response, while not optimally corrected, shows significant improvement. Figure 5: Block diagram of sensor frequency response correction algorithm. Having defined the frequency response correction methodology, a signal conditioner may be implemented in hardware to include appropriate excitation to the sensor, amplification and low-pass filtering. Figure 6 shows the block diagram of the KSC-2 Signal Conditioner as pictured in Figure 7. The KSC-2 is a compact, dual-channel high precision amplifier/filter with programmable constant voltage excitation optimized for conditioning pressure sensors, accelerometers and microphones. Bipolar \ Unipolar Excitation Supply Reg Power 20.0 Uncompensated Sensor Q = 10 f/Fr = 1 15.0 10.0 Gain (dB) Post-Filter Gain Compensated Response fcomp = Fr Qcomp = 8 5.0 Input 2-6 Wire w/Shield Amp Resonance Compensator Cal –5.0 Compensated Response fcomp = 0.8 Fr –10.0 Qcomp = 10 –20.0 0.1 Compensator In Pre-Filter Gain 0.0 –15.0 Output Overload Input Overload Compensated Response fcomp = Fr Qcomp = 12.5 0.2 0.3 Input Compensated Response fcomp = 1.25 Fr Qcomp = 10 0.5 0.7 1 2 Normalized Frequency (f/Fr) 3 5 7 Auto Balance/ Zero Suppress Compensator Out LP6F or LP6P Programmable 6-Pole Filter In Prog. Buffered Amp Filter Out Auto Calibrate Gain & Offset Input Short Figure 6: Block diagram of amplification and filtering in KSC-2. 10 Figure 4: Sensitivity of frequency response correction to Q and ωr , (Fr ). The block diagram for the implementation of the frequency response correction is shown in Figure 5. To implement a 4 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME Output Front (a) ï Back ï (b) Figure 7: KSC-2 Signal conditioner (a) Front, (b) Back. The KSC-2 contains an analog, hardware implementation of the frequency response correction technology presented in this work that extends the useable frequency response of sensors with packaging (cavity), recessed pressure sensors or seismic resonances in real-time with no need for data postprocessing. Based on user entry of sensor ωr and Q, the frequency response correction tool extends the usable sensor bandwidth. As the block diagram in Figure 6 illustrates, a programmable pre-filter gain is first applied to the input signal in order to preserve signal-to-noise ratio of the in-band sensor signal while allowing for headroom for out-of-band signals. Next the frequency response correction, FRC, filter is applied to remove unwanted resonant frequencies of the sensor or mounting configuration. A programmable 6-pole low-pass precision filter is then applied to the signal to eliminate outband energy and prevent aliasing. Finally, the signal is passed through a post gain stage to ensure use of the full range of the A/D after removal of the resonance characteristics [20]. Overload detectors alert the user to pre-filter overload conditions that could otherwise be masked by the filter. In addition, output overload detectors are provided with programmable threshold voltages. Alternative to this frequency response correction method, it is possible to correct for undesired resonant frequencies by post-processing test results [7]. However, post-processing data after digitization can result in poor signal-to-noise ratios because the A/D input must accommodate both the in-band signal of interest and the sensor resonance. Allowing for transducer packaging or mounting resonance headroom results in the user not maximizing the amplification for the small inband signal above the self-noise of the signal conditioner and A/D. This results in less than ideal signal-to-noise ratios regardless of the resolution of the A/D [20]. With this in mind, this real-time, analog frequency response correction approach provides superior performance by maximizing signal-to-noise ratios. With the frequency response correction applied to the output of a transducer using the KSC-2, a flat frequency response beyond the original resonance (~45kHz) is easily achieved, as shown in Figure 8. ï !"#ï$ ï% & '* +,- ./* +" ï 0%"ï "*1 * 2345 #34 ï "*1 6171& +" ï Figure 8: Overview of frequency response correction. This advanced amplification and higher-order filtering scheme is carefully designed to have a linear phase response and thus constant time delay. Figure 9 illustrates the phase shift plotted with a linear frequency scale, making the linear phase properties more apparent. "ï$ ï+!!'& ( //( " "ï +!*.!' "(++!-%(' "+34&& 34 "(++!-! .-).-& " $,! !#+!!, ï ï ï ï 4 4 4 4 +!*.!' 4 0 Figure 9: Phase of frequency response correction system, KSC-2. A linear phase shift corresponds with a constant time delay at all frequencies [21]. This can be quantitatively understood by using Equation 5, below. 5 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use (5) Copyright © 2015 by ASME Applying Equation 5, the time delay of the frequency response correction algorithm is quantified over frequency and displayed in Figure 10. The time delay is 4±0.5μs from 1kHz-10kHz with a gradual increase in the delay to 7μs at 50kHz. Figure 10 displays the phase shift as a time delay associated with the non-frequency response corrected XCQ-062 B-screen transducer without RTV around the chip. As the data show, the delay is approximately 0s until the resonance (33.5kHz) is approached at which point the time delay grows rapidly to ~10μs. preamplifier adding a gain of 128X, the 6-pole, flat anti-alias filter set at 75 kHz (-3dB) to remove out-of-band energy and the post-filter amplifier gain set to 16X. Signals were additionally AC coupled to remove any DC content. These filter parameters were experimentally characterized using a HP 3562A Dynamic Signal Analyzer the results of which are displayed in Figure 11. 2 0 Delay [μs] -2 -4 -6 -8 XCQ-062 B-Screen, No RTV w/o Freq. Resp. Correction With Freq. Resp. Correction -10 -12 10 3 10 4 Frequency [Hz] Figure 10: Time delay of frequency response correction, KSC-2. EXPERIMENTAL SETUP The periodic dynamic pressure calibration tool employed to quantify the resonant frequencies of various sensor packages as well as recess mounted transducer configurations has previously been reported [5]; however a brief description is provided for clarity. The dynamic calibration system uses a speaker and waveguide. The speaker generates large amplitude acoustic waves with peak-to-peak pressures ranging from 10Pa1300Pa over the frequency range of 1-50kHz. A flush mounted Kulite XCQ-062 with no screen and an RTV filled volume cavity is used as a reference sensor. This sensor configuration was previously demonstrated to have a flat frequency response of DC to >50kHz [5]. Test transducer configurations are mounted directly across from this reference sensor. Sensors are evaluated by sweeping the dynamic pressure frequency from 1kHz to 50kHz over a time period of 25s. The output voltages from each sensor are conditioned using the KSC-2 and presented herein with the Figure 11: KSC-2 signal conditioner filter characterization with a 6 pole, flat anti-aliasing 75kHz (-3dB) low-pass filter. *Signals were AC coupled at 0.25Hz for all measurements herein. When employed, the frequency response correction algorithm provides the desired attenuation at the resonant frequency. Post signal conditioning and frequency response correction, all dynamic pressure data presented herein were collected using a 16-bit NI USB-6366 X-Series DAQ with data sampled at a rate of 250kHz. SENSOR PACKAGING RESULTS AND DISCUSSION: As discussed in the Theoretical Background section, the resonant frequency, ωr, and quality factor, Q, of the package or 6 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME The experimental results in Figure 12 confirm theoretical predictions, showing an increase in ωr with temperature for the XCQ-062-35kPa with a B-screen and without RTV around the chip. Figure 13a shows good agreement between the observed increase in ωr with temperature to that predicted theoretically using the change in the speed of sound with temperature (Eq. 6) and the modified Helmholtz equation (Eq. 1). In addition, the inverse relationship of Q with temperature is anticipated as the viscosity of air increases with temperature which contributes to higher damping [22]. Figure 13b provides a plot of Q versus temperature. recess related resonance of a transducer needs to be accurately quantified in order to implement the active frequency response correction in the KSC-2 signal conditioner. Using the dynamic pressure calibration tool, the frequency response was characterized for the following MEMS piezoresistive sensor packaging configurations: 1) 2) 3) 4) 5) 6) XCQ-062-35kPa with B screen XCQ-062-35kPa with A screen XCQ-062-35kPa without a screen XTEL-190-35kPa with B screen XTEL-190-35kPa with M screen XTEL-190-35kPa without a screen To determine the package related resonance and Q, data were taken without the KSC-2 frequency response hardware correction algorithm. The data were fit to Equation 3 to determine ωr and Q. These values were then input into the freqeucny response correction (FRC) stage of the KSC-2 and the frequency response was retested using the periodic dynamic pressure calibration tool. Aside from quantifying ωr and Q for each MEMS package configuration, there are two additional critical areas of study required: a.) Change in ωr and Q over temperature b.) Variation in ωr and Q for a batch of transducers with identical packaging or mounting. Beginning with the impact of temperature, theory predicts that ωr will increase with temperature, as the speed of sound in air is directly dependent on temperature [22]. 46 (a) Model XCQ−062 B−Screen, No RTV 44 Exp. Results Resonant Frequency,fr [kHz] Linear Fit to Exp. Results 42 fr(T)=0.0506*T+34.742 40 38 36 34 0 50 100 Temperature [deg. C] 150 200 (6) (b) 20 Q(T)=−0.0022*T+5.9054 5.9 5.8 10 5 Quality Factor, Q Gain [dB] 15 0 -5 3 10 10 4 5.6 5.5 0 Phase [Degrees] 5.7 -50 Exp. Results 5.4 Linear Fit -100 T = 24°C T = 100°C T = 150°C T = 200°C -150 -200 3 10 5.3 10 0 50 100 Temperature [deg. C] 150 200 Figure 13: Plots of XCQ-062-35kPa B-Screen, No RTV fr and Q coefficients over temperature. 4 Frequency [Hz] Figure 12: Frequency response of XCQ-062-35kPa with Bscreen, no RTV at four temperatures. In Figure 13, the equations for the linear fit to the experimental data are included which can be extrapolated to provide ωr and 7 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME Q for this frequency response correction algorithm outside the test temperature range or at a higher resolution within the range. These equations can further be combined with Equation 4 to create a frequency response correction transfer function (Eq. 7) as a function of temperature and frequency with only the quality factor being estimated from experimental data, which is shown below: B-screen or an XCQ-062 with a B-screen. If the same coefficients can be used then it is not necessary to measure each transducer. However differences in sizing due to normal manufacturing variability creates slightly different resonant frequencies for each transducer. Figure 15 shows three different XCQ-062-35kPa differential transducers with Bscreens, Figure 16 shows the same transducers each corrected with individual coefficients of ωr and Q as listed in Table 1. The variance in Q is small for these transducers but there is a 5 kHz variance in the resonant frequency. (7) !" Equation 7 can be used to develop an active adjustment of the frequency response correction algorithm if temperature is measured, which can be achieved via the change in overall bridge resistance [23]. The equations for ω r and Q were then used to calculate the required coefficients, and experimental tests were then conducted at each temperature point applying the calculated coefficients of ωr and Q. The resulting experimental frequency response corrected transfer functions are illustrated in Figure 14. This XCQ-062 transducer packaged with a B-screen and without RTV now exhibits an achievable flat bandwidth from DC to approximately 40kHz; a 200-300% improvement depending on temperature. All transducer packaging configurations and multiple types of transducers were characterized in this manner over temperature with the goal of producing a set of linear equations that provide ωr and Q coefficients that can be used universally. ï & %& ï ï ï ï%"& # & ï ï %$'"( ) Figure 15: Three XCQ-062 B-screen, no RTV sensors with no frequency response correction. 0 Gain [dB] 5 -5 103 104 ï 0 -50 -100 -150 -200 103 T = 24°C T = 100°C T = 150°C T = 200°C ï Phase [Degrees] 104 Frequency [Hz] ï ï Figure 14: Frequency response of XCQ-062-35kPa with Bscreen, no RTV at four temperatures with frequency response correction. ï ! Figure 16: Three XCQ-062 B-screen, no RTV sensors with individual frequency response correction coefficients. It would be advantageous to be able to use the same correction coefficients for an entire family of transducers with identical packaging, such as a Kulite series XTEL-190 with a 8 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME The average resonance and Q were then used to correct all three transducers as shown in Figure 17. Using the averaged correction values, a 150% improvement in the flat (+/- 2dB) frequency response was obtained. As shown in Figure 16, the application of individual correction values for transducers can result in as much as a 300% improvement in frequency response (DC to ~40 kHz). Therefore, individual or average correction coefficients should be used depending upon the frequency response needs of the application. In many instances, a 150% improvement in frequency response is sufficient and average correction coefficients can be applied without the need to calibrate and correct each transducer individually. frequency response correction technique will work at any temperature. "# ï ï !' &' "# ï ï ï ï&# ï ï&# ï '$ &# ï ï ï&# ï ï&# ï '$ &# ï &%(#) * !' &' Figure 18: Frequency response corrected output for various transducer series and packaging configurations. ï ï ï&#& $ & ï ï &%(#) * Figure 17: Three XCQ-062 B-screen, no RTV with average correction coefficients applied to all sensors. Similar tests were conducted on several identical transducers for multiple packaging configurations. Figure 18 shows various transducer packaging configurations with frequency correction applied. Table 1 provides a complete set of experimentally determined ωr and Q coefficients for each transducer over temperature (24, 100, 150, 200°C) and the achieved flat frequency response with and without the correction algorithm applied. The 200°C upper bound was chosen due to the temperature limitation of the test setup. This 9 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME Table 1: Correction Coefficients and Frequency Response Improvement: No Frequency Response Correction Type XCQ-062 Screen B Serial Number: 651 XCQ-062 B 650 XCQ-062 B 133 XCQ-062 A 598 XCQ-062 None 598 XTEL-190 B 114 XTEL-190 XTEL-190 M None 114 114 Temp [°C] 24 100 150 200 24 100 150 200 24 100 150 200 24 100 150 200 24 100 150 200 24 100 150 200 24 24 100 150 200 fr [kHz] 36.6 40 43.5 45.4 33 37.6 39 41.7 38.2 41.5 45.3 47 >50 >50 >50 >50 >50 >50 >50 >50 37.7 42.3 45.1 >50 47.6 >50 >50 >50 >50 ±2 dB [kHz] 16 15 14.3 13.6 14.1 17.4 19 20.5 17.6 19.2 20.3 21.3 22.7 27 29.5 31.6 29.7 33.65 37.54 40.9 18.63 18.9 19.3 19.7 26.2 36.9 40.5 42 47 With Frequency Response Correction Bandwidth fr Improvement [kHz] Q ± 2 dB [kHz] [%] 36.5 6 41 256 40 6 44 293 43.5 6 44.5 311 45.5 6 44 324 33 5.5 42.3 300 37.5 5.5 43.5 250 39 5 48 253 41.5 5 46.9 229 38 6 48.9 278 41.5 5.8 48.2 251 45.5 5.5 45 222 47 5.5 45 211 50 10 1.0 dB* >220 57 10 1.6 dB* >185 61 10 1.7 dB* >169 65 10 1.6 dB* >158 61 10 1.1 dB* >168 69 10 1.0 dB* >149 77 10 1.4 dB* >133 84 9.5 1.5 dB* >122 38 10 36 193 42 8 41 217 45 6 44 228 47 6 45 228 50 5 48.5 185 75 10 1.0 dB >136 83 9.5 0.8 dB >123 86 9.5 < 1.0 dB >119 95 9 < 1.0 dB >106 *2 dB point beyond measurement range. Error at 50 kHz presented. Bandwidth approximation based on 50 kHz. **All test results apply to air or gaseous media with equivalent properties only. the termination of the pipe. These configurations were examined to simulate a possible mounting in a hightemperature location on a gas turbine where flush mounting the transducer is not possible due to the operating temperature limitations of the sensor and some dynamic pressure content up to the highest achievable flat bandwidth is desired [7]. Typically such applications are for static and low-frequency dynamic pressure applications because the low, organ pipe resonance of the standoff tube distorts any frequency above 1kHz (2.54cm standoff tube). Figure 19 shows the uncompensated frequency response of a 2.54cm (1in) standoff SENSOR MOUNTING RESULTS AND DISCUSSION: Sensor mounting typically has the greatest impact on the achievable flat frequency response of dynamic pressure measurements. We experimentally evaluated the frequency response correction algorithm (KSC-2) presented herein on two recess mounting configurations. The recess mounting configurations included a 2.54cm (1in) and a 7.62cm (3in) standoff pipe with an XCS-190 dynamic pressure transducer at 10 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME tube. After the coefficients are determined for the organ pipe resonance, the frequency response correction algorithm is applied to extend the frequency response by 220% to a flat (±2dB) bandwidth of DC-2.83kHz, which is a significant extension of the useable frequency response again while retaining a linear phase response. The current frequency response correction algorithm built into the test KSC-2 signal conditioning system limits the resolution of fr, (ωr), to 500Hz increments. At the lower resonant frequencies of such longer standoff pipes, this resolution limits the effectiveness of the correction algorithm. With greater resolution of fr, (ωr), better frequency response correction can be achieved. To demonstrate the potential improvement of the system with better resolution, we applied the model frequency response correction transfer function using the exact value of fr, (ωr), which is indicated as the best-fit correction curve in the plots below. Gain [dB] 30 20 also address varying length to diameter ratios as well as an array of different lengths of tubing for various applications. 40 7.62cm (3in) Recess Tube, No FRC Gain [dB] 10 102 Phase [Degrees] 2.54cm (1in) Recess Tube, No FRC Recess Tube w/ Exp. FRC Recess Tube w/ Best-Fit FRC 0 -100 2 3 10 10 10 4 Frequency [Hz] Figure 20: Plot of XCS-190 at the termination of a 7.62cm (3in) standoff tube (3.81mm diameter) without and with frequency response correction. 103 104 Table 2: Standoff Tube Correction Coefficients and Frequency Response Improvement at 24°C 100 0 -100 102 104 100 -200 200 -200 103 200 0 Phase [Degrees] Recess Tube w/ Best-Fit FRC 20 0 10 102 Recess Tube w/ Exp. FRC 30 103 104 Frequency [Hz] Figure 19: Plot of XCS-190 at the termination of a 2.54cm (1in) standoff tube (3.81mm diameter) without and with frequency response correction. The 7.62cm (3in) standoff tube exhibits a significantly lower first resonance of ~1kHz, as Figure 20 shows. This standoff tube’s harmonics, occurring at odd integers of the first resonance, are also accurately measured. Again applying frequency response correction, we demonstrate a substantial improvement in the achievable flat bandwidth of 235% from 400Hz to 1kHz for this mounting configuration. While such a standoff is still inappropriate for higher frequency dynamic pressure measurements, this correction methodology improves the flat frequency response of a recessmounted transducer by 200%. Table 2 provides values for both the experimental and best-fit frequency response correction for the 2.54cm and 7.62cm standoff tubes. Further experiments are required to determine the effects of temperature and thermal gradients within standoff tubing. Additional experiments will Standoff Pipe Length Uncorrected ±2 ftube dB [kHz] [kHz] ftube [kHz] Corrected ±2 dB Q [kHz] Improv ement [%] 2.54cm, Exp. 3.164 1.278 3 15 2.83 220 2.54cm,Best-Fit 3.164 1.278 3.021 10.8 3.08 240 7.62cm, Exp. 1.076 0.4 1 9.4 0.94 235 7.62cm,Best-Fit 1.076 0.4 1.006 7.6 1.04 260 CONCLUSION This work presents a real-time, analog frequency response correction algorithm that extends the bandwidth of dynamic pressure measurements by 150-300%. A theoretical and experimental characterization of this frequency response correction algorithm, as implemented in the KSC-2 signal conditioning system, has been undertaken. We demonstrate the performance of the algorithm by utilizing the filtering and amplification features of the KSC-2 signal conditioner to correct the frequency response of pressure transducers manufactured in several different configurations. This frequency response correction should be implemented whenever the transducer, sensing element and/or package, or mounting configuration limit the bandwidth of the 11 Downloaded From: https://proceedings.asmedigitalcollection.asme.org on 06/30/2019 Terms of Use: http://www.asme.org/about-asme/terms-of-use Copyright © 2015 by ASME ωr measurement by generating predictable resonances. This correction enables the use of transducers with a screen or recess mount when making wide bandwidth measurements, which historically would require a less protected, flush mounted sensor. This system can specifically be used to extend the useable bandwidth of transducers that are recess mounted on gas turbines such that the temperature seen by the transducer is within its operating range. With better unsteady pressure measurement accuracy, engineers can make informed design changes, ensure safe system operation and reduce system down time through longer component life and predictive maintenance. This frequency response correction system has been developed into a commercially available analog signal conditioner such that the frequency response limitations of a transducer or mounting configuration can easily be overcome in real time without the need for post-processing data. In the future, we plan to perform in-situ testing of this frequency response correction algorithm in an operational gas turbine. In such a test, a recessed transducer utilizing the frequency response correction algorithm will be compared to a flush mounted transducer, a recessed transducer without correction and a transducer utilizing a semi-infinite tube configuration. s R M γ Δt T dB REFERENCES [1] Andronenko, S., Stiharu, I., Packirisamy, M., Moustapha, H., and Dionne, P., 2005, "The Use of Microelectromechanical Systems for Surge Detection in Gas Turbine Engines," International Conference on MEMS, NANO and Smart Systems, pp. 355-358. [2] Kobayashi, H., Leger, T., and Wolff, J. 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ACKNOWLEDGEMENTS We would like to acknowledge and thank Andrew Bemis and Lou DeRosa for their oversight, guidance and support of this work. We would also to thank John P. Hilton and Andrew Bemis for their technical feedback and careful review of this work. Finally, we would like to thank Mrs. Nora Kurtz for funding and supporting this effort. NOMENCLATURE i G(s) f Tc(s), Tc(s,T) d fScreen-Vol. n a VCavity fTube. L c m Q Q(T) fr. 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