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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 20, NO. 6, NOVEMBER 2005
Active-Clamp Snubbers for Isolated Half-Bridge
DC–DC Converters
Hong Mao, Songquan Deng, Jaber Abu-Qahouq, Member, IEEE, and Issa Batarseh, Senior Member, IEEE
Abstract—In conventional isolated half-bridge dc–dc converters,
the leakage-inductance-related losses degrade converter efficiency
and limit the ability to increase the converters’ switching frequencies. In this paper, a novel active-clamp snubber circuit for halfbridge dc–dc converters is proposed to recycle the energy stored
in the leakage inductance by transferring this energy to a capacitor with zero-voltage zero-current-switching switched auxiliary
switches, such that body-diode conduction of primary-side main
switches are prevented and primary side ringing are attenuated
resulting in improved converter efficiency. Principles of operation
and simulation analysis are presented and supported by experimental results that show significant improvement in efficiency.
Index Terms—Active-clamp, converter, dc–dc, half-bridge,
isolation, leakage inductance, reverse recovery, snubber, topology,
transformer.
I. INTRODUCTION
I
N ISOLATED dc–dc converters, the isolation transformer
leakage inductance is an important factor that affects the
performance of converters. A variety of topologies and control
methods are proposed to improve the converter performance by
utilizing the transformer leakage inductance [1]–[7]. The phaseshifted full-bridge [1] and the active-clamp forward [2] dc–dc
converters are good examples to utilize the transformer leakage
inductance to achieve zero-voltage-switching (ZVS) and further
reduce electromagnetic interference (EMI) noise.
There are two conventional control schemes for half-bridge
dc–dc converters, namely, asymmetric control [3]–[5] and
symmetric control [6]–[8]. Symmetric controlled half-bridge
dc–dc converter [6]–[10] has simple configuration and operates with symmetric components stresses. However, the two
primary switches operate at hard switching condition and there
exist leakage-inductance-related ringing losses and problems.
To damp such ringing, usually, dissipative snubber circuits
are employed across switches, resulting in leakage inductance
energy dissipated in the snubber circuits. Consequently, the
efficiency is degraded and the power level is limited.
Asymmetric control scheme applies two complementary signals to the two half-bridge switches. Due to the small dead time
Manuscript received September 17, 2004; revised March 24, 2005. Recommended by Associate Editor C. K. Tse.
H. Mao is with the Astec Power Advanced Technology Division, Emerson
Network Power, Andover, MA 01810 USA (e-mail: hongmao@astec.com).
S. Deng is with Synqor, Inc., Boxboro, MA 01719 USA.
J. Abu-Qahouq was with the Department of Electrical and Computer Engineering, University of Central Florida, Orlando, FL 32816 USA. He is now with
Intel Corporation, Hillsboro, OR 97124 USA.
I. Batarseh is with the Department of Electrical and Computer Engineering,
University of Central Florida, Orlando, FL 32816 USA.
Digital Object Identifier 10.1109/TPEL.2005.857529
between two switches, the primary-side ringing problem is eliminated and ZVS for both switches can be achieved with the help
of the transformer leakage inductance [3]–[5]. However, asymmetric half-bridge converter suffers from the asymmetric components stresses distribution in the corresponding components
and a dc bias in the transformer. Therefore, it is not suitable
for applications with wide input-voltage range [3], [5]. Furthermore, the dc gain of the asymmetric-controlled half-bridge converter is nonlinear [5], resulting in lower duty cycle at high line
voltage compared to the symmetric-controlled half-bridge converter, which results in degrading the converter performance at
high line input.
An active current clamping method was proposed in [6],
[7] to achieve ZVS of switches and to attenuate the ringing.
During the off-time interval (when both switches are turned
off), the leakage inductance current freewheels through the auxiliary circuits. Before turning on the main switches, auxiliary
switches interrupt the freewheeling path, such that the energy in
the leakage inductance is released to create ZVS condition for
main switches. However, this method has two main demerits:
The first is that the value of the leakage inductance should be
high enough, and the other is that high circulating current exists
especially at low duty cycle. This means that this scheme is not
suited for applications with wide-range of input voltage. In [8]
and [10], duty-cycle-shift (DCS) controlled ZVS half-bridge
topologies are proposed. However, it has similar disadvantage,
which is that circulating conduction loss increases significantly
at high line input due to long freewheeling time.
In this paper, a new active-clamp snubber circuit is proposed
to clamp the leakage inductance current and damp the ringing
during the off-time interval. In the proposed circuit, the energy
in the transformer leakage inductance is transferred to a capacitor during the off-time interval, hence, the ringing is eliminated
and the switches’ body diodes never conduct. As a result, there
is no body diodes’ reverse recovery and the ringing losses are
minimized. As the current-transferring interval accounts for a
small part of the switching period, the conduction loss during
the off-time interval is reduced significantly.
The next section presents the proposed active-clamp snubber
topology and discusses the modes of operation. The main features and design considerations of the converter are provided in
Section III. Section IV gives simulation and experimental verification, and conclusions are given in Section V.
II. PROPOSED ACTIVE-CLAMP LOSSLESS SNUBBER
A. The Ringing Issue and the Principle of Snubber Circuits
Fig. 1 shows the conventional half-bridge dc–dc converter
is the transformer
with current doubler rectification, where
leakage inductance and
and
are junction capacitances
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MAO et al.: ACTIVE-CLAMP SNUBBERS
Fig. 1.
1295
Half-bridge isolated dc–dc converter.
Fig. 3. Proposed active-clamp snubber circuits in half-bridge dc–dc
converters: (a) topology A and (b) topology B.
termined by the junction capacitances and input voltage as follows:
(1)
Fig. 2. Equivalent circuits during the freewheeling period: (a) without snubber
and (b) with clamp circuit.
of metal-oxide semiconductor field-effect transistor (MOSFET)
switches
and , respectively.
and
During the freewheeling period, when both switches
are off, the transformer secondary is shorted by the two
conducting diodes. By neglecting other parasitic capacitances
and stray inductances, the equivalent circuit of the freewheeling
mode is shown in Fig. 2(a). It should be noted that the switches’
junction capacitance values are smaller than
and
capacitance values, and the voltages across
and
can be
assumed constant during the modes. If the two body diodes are
ideal, the energy in the leakage inductance will be recycled to
the dc bus through the two body diodes with undamped oscillation between the transformer leakage inductance and the junction capacitances. The energy involved in the oscillation is de-
where
is the individual MOSFET’s junction capacitance.
However, body diodes of the MOSFETs have undesirable reverse-recovery characteristics, especially for high voltage rating
MOSFETs, which results in more energy involved in the oscillation. As a result, the ringing is more severe and there are
more reverse-recovery and ringing losses due to the nonideality
of MOSFETs’ body diodes. Moreover, the ringing and reverse
recovery may lead to EMI problems.
Usually, snubber circuits are added to damp such ringing. The
) series snubber across a switch or transresistor–capacitor (
former primary windings is the most common snubber circuit.
Depending on the size of the snubbers, the energy remained in
the leakage inductance may be partly recycled to the dc bus,
or totally dissipated in the snubbers. For a small snubber, body
diodes conduct to partly recycle the energy with part of energy
dissipated in the snubber. For a large snubber, the energy in the
leakage inductance may be fully dissipated in the
snubber
without recycling through MOSFETs’ body diodes. In the later
case, the power losses increase significantly with the increase
of primary-side peak current, input voltage and switching frequency.
B. Proposed Active-Clamp Snubber
A clamping concept is proposed as shown in the dashedline frame of Fig. 2(b). With the proposed clamping snubber,
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 20, NO. 6, NOVEMBER 2005
and
is blocked. Considering the possibility to
through
use Schottky diode for
, the diode reverse recovery can be
eliminated. However, with zero current leakage inductance current and
0, the circuit does not reach steady state and
the voltage across
will cause the voltage
to oscillate
until
. Since the voltage across
is much smaller
than half of the input voltage, the MOSFETs’ body diodes will
not be involved in the oscillation, and thus, the ringing is negligible. Therefore, there are no body-diode-related losses and
, since both the voltage
the ringing loss is very small. At
across and the current through
are zero,
can be turned off
with ZVZCS.
Mode 3
:
is turned on at
causing
the leakage inductance current to start charging from zero at the
following slope:
(2)
Fig. 4. Key theoretical waveforms of the proposed topology.
the energy in the leakage inductance can be recycled into the
voltage sources. It is important to note that the diodes used in
the snubber circuits have better characteristics than body diodes
of the MOSFETs. Employing this clamping concept, a variety of
practical active-clamp snubber circuits are proposed as shown in
Fig. 3, where the capacitor
acts as a voltage source to be used
to absorb the energy in the leakage inductance and damp the
ringing. In Fig. 3(a), the diodes
and
are external fast-recovery diodes, to simplify the circuit, body diodes of switch
and
can replace diodes
and
as shown in Fig. 3(b).
Both topologies have the same drive timing and principle of
operation. Basically, during the freewheeling period when both
switches are off, since the transformer secondary side is shorted,
the leakage inductance energy cannot be delivered to output, and
the middle branch provide paths to transfer the leakage induc.
tance energy to the capacitor
The topology of Fig. 3(a) is taken here as an example to analyze the principle of operation, with the key operational waveforms shown in Fig. 4. To simplify the analysis of the operation
modes, components are considered ideal except otherwise indicated. The main equivalent operation modes are shown in Fig. 5,
and described as follows.
Mode 1
: Initially, it is assumed that
was conducting and
was turned on at zero-voltage-zero-current-switching (ZVZCS). At
,
is turned off, causing
the primary current
to charge the junction capacitance
and discharge
. When the voltage across
is charged to a
half of the input voltage V , the leakage inductance current will
flow through , , and . Considering the fact that the transformer secondary is shorted and there is a dc voltage across
,
the leakage current continues to charge the capacitor
until
the current resets to zero at . At the end of this mode, all the
energy in the leakage inductance is transferred to the capacitor
.
: Since the diode
blocks any poMode 2
tential reverse current in its branch, the reverse oscillation path
where
is the voltage across the capacitor
. The smaller
the leakage inductance is, the faster its current charge-up will be,
until it becomes equal to the reflected inductor current, causing
the diode
to be blocked and the converter to start transferring
energy from the primary-side to the secondary-side. This current
will continue to charge at a slope of
(3)
where
is the secondary inductor,
and
are the transformer primary number of turns and secondary number of turns,
respectively, and
is the output voltage. During this interval,
the inductor current is charged and the capacitor
is discharged. The polarity of the voltage across the capacitor
will
reverse during this interval. The auxiliary switch is turned on
with ZVZCS since the diode
block the current.
During this mode, the capacitor
is charged linearly by
the reflected output current. Assuming that the output current
is constant during this mode, the reflected current through the
is given by
capacitor
(4)
The capacitor voltage changes during this mode and is given
by
(5)
where
;
and
are transformer primary and
secondary number of winding turns, respectively. Ignoring the
capacitor
voltage change due to resetting leakage inductance
current, the capacitor
voltage can be described as follows:
(6)
Mode 4
: This mode is similar to mode 1.
is turned off at
, causing the primary current
to
charge
and to discharge
. When the voltage across
is
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MAO et al.: ACTIVE-CLAMP SNUBBERS
1297
Fig. 5. Modes of operation: equivalent circuits.
charged to half of the input voltage, the leakage inductance current flows through
,
, and . Considering the fact thatthe
transformer is shorted and there is a dc voltage across
, the
until the
leakage current continues to charge the capacitor
current resets to zero at . At the end of this mode, all the energy in the leakage inductance is transferred to the capacitor .
At the end of Mode 3, the capacitor
voltage is
: This mode is similar to mode 2. After
Mode 5
minor oscillation, this mode will end with
(7)
(10)
is large enough and the
Assuming that the capacitor
leakage inductance current is linearly reset to zero, the capacitor
voltage stays almost constant during this mode. The
leakage inductance current at
is equal to the reflected
output current and is given by
(11)
(8)
The duration time of this mode
can be obtained as
(9)
0. At
, the switch
can be turned off
where
with ZVZCS in the same manner described in mode 2 for .
will be derived in Section III.
: This mode is similar to mode 3.
Mode 6
The switch
is turned on and the converter starts to deliver
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 20, NO. 6, NOVEMBER 2005
energy from the primary-side to the secondary-side. At
is turned on with ZVZCS.
the auxiliary switch
,
III. MAIN FEATURES AND DESIGN CONSIDERATIONS
A. Main Features
As discussed earlier, with the proposed active-clamp snubber,
the energy in the leakage inductance is transferred to the capacitor instead of being dissipated in dissipative snubbers. Furthermore, the body diodes of main switches are not involved in
the oscillations, such that the switches’ body-diodes conduction
losses and reverse-recovery losses are eliminated.
and
are turned on with ZVZCS
The auxiliary switches
and turned off with ZVZCS. Therefore, the switching losses are
minimized. Since the conduction intervals of the auxiliary circuit only account for a very short portion of the whole period,
the auxiliary switches and diodes conduction losses are negligible. In addition, all components’ voltage stresses in the middle
branches are half of the input voltage. Therefore, lower voltage
and current rated components with lower gate charge can be
chosen for auxiliary components. The auxiliary circuits losses
are limited compared to the leakage inductance ringing losses.
When the main switches are turned off, the leakage inductance current starts to reset toward zero as shown in Fig. 6(a),
where
is reset time of leakage inductance current, and
and
are the on-time and off-time of the two main
, the operation is the same
switches, respectively. If
as described earlier, and the primary voltage and current are as
shown in Fig. 6(a). However, if
, there is not enough
time for the leakage inductance to transfer the energy to the
capacitor
. When the auxiliary switches are turned off, the
remained leakage inductance energy is used to charge/discharge
the main switches junction capacitance. In this case, ZVS may
be achieved for the main switches, and the corresponding
waveforms of the primary voltages and currents are shown in
Fig. 6(b).
In other words, depending on the duty cycle width, the proposed snubber topology has two possible operation cases: The
small-duty-cycle case, where the converter operates with the active-clamp mode to reduce ringing problem, and the large-dutycycle case, where the active-clamp interval is reduced, such that
the energy in the leakage inductance will be directly used to
achieve ZVS for the main switches instead of being transferred
into the capacitor . Therefore, this topology is suited for wide
input voltage range. At high line, the converter reduces the energy circulation conduction losses, and at low line, the converter
reduces the switching losses.
can be written as
The reset time
Fig. 6. Two possible operation schemes for the proposed converter: (a)
and (b)
.
T
T <T
T >
where
is the converter output current.
value can be designed to approximately achieve V
10
20 V at full
load.
can be removed. At this case,
As a matter of fact, capacitor
the leakage inductance energy is trapped in the middle branch
and the main switches have higher possibilities to achieve ZVS
using the trapped energy compared to the first case with the
capacitor
. However, conduction loss may be significant at
low duty cycle operation. In order to determine if the capacitor
is better to be added or not, an appropriate loss comparison
can be made based on this following analysis.
, assuming the leakage inFor the case without capacitor
ductance current keeps constant during the freewheeling times
(1-2D) T, the conduction loss can be estimated as follows:
(14)
(12)
is the duty cycle that
where is the switching period and
satisfies
. As shown in (12),
is independent of the
load, which simplifies the design for the capacitance
. The
voltage stress in
is given by
(13)
is the leakage inductance current value during the
where
freewheeling mode when two main switches are off,
is
the on-resistance of Switch
or , V is forward voltage
drop across body diodes of Switch
or . By substituting (8)
into (14), the conduction loss can be written as
(15)
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MAO et al.: ACTIVE-CLAMP SNUBBERS
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In the proposed converter, when
, the main switches
operate at hard switching, and the switching loss is calculated
as follows:
(16)
is the transition time of switches. The total
where
switching loss of the two switches can be approximated by
(17)
and
values at certain converter
Depending on
operating points, the tradeoff between using
or not can be
made accordingly under the guidance of the above equations.
It should be noted that it is difficult to estimate the switching
loss accurately and the equations given above are only for rough
estimation.
Compared to the phase-shift full bridge dc–dc converter,
in the proposed converter stand only half of
Switches and
the input voltage versus the full input voltage in the full bridge
converter, and they act as auxiliary switches and carry smaller
RMS currents. Moreover, both switches
and
operate
at soft switching. In addition, the proposed half-bridge dc–dc
converter has options to make design tradeoff of choosing
various capacitor
value which provide design flexibility
according to specific applications.
B. Converter dc Voltage Gain
In the previous analysis, the voltage increase across capacitor
is ignored in Mode 1 and Mode 4, and the converter dc
voltage gain is assumed to be the same as it is for the conventional half-bridge dc–dc converter, which is given by
(18)
However, the voltage increase has effect on the converter dc
voltage gain. In Mode 4, assuming all the energy stored in the
leakage inductance at the end of Mode 3 is transferred to the
without additional loss and voltage change across
capacitor
the capacitor
in Mode 4 is
, the energy balance equation
is expressed by
Fig. 7. Simulation waveforms comparison (top traces: voltages V ; bottom
traces: transformer primary currents): (a) without snubber, (b) with conventional
dissipative RC series snubber (R = 30 , C = 2 nF), and (c) with active-clamp
snubber (C = 3 F).
Ignoring the duty cycle loss due to the leakage inductance, the
transformer primary voltage in the power delivery modes can be
described as
(19)
where
is the voltage value at the end of Mode 3
which is given by (7). Substituting (7) into (19) yields
,
(20)
(22)
where V
is the voltage change in power delivery modes that
is expressed in (5). In power delivery modes, transformer primary voltage is reflected to the secondary side to charge the inductors. Applying the volt-seconds balance across the inductors,
we have
The capacitor voltage value at the end of Mode 4 and 5 is
(23)
(21)
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 20, NO. 6, NOVEMBER 2005
Fig. 8. ZVZCS waveforms of auxiliary switches.
Fig. 10. Experimental waveforms of transformer primary voltage and current
(D = 0.37; top trace Vab: 20 V/div; bottom trace i : 5 A/div): (a) with the
conventional RC series snubber and (b) with the proposed active-clamp snubber
(C = 3 F).
From (24), it can be observed that the proposed converter
has higher dc gain than the conventional half-bridge dc–dc converter. The increase of dc gain is proportional to the load current.
For typical applications, numerical analysis shows that the dc
addition is small and ignorable compared with output voltage.
IV. SIMULATION AND EXPERIMENTAL VERIFICATION
Fig. 9. Waveforms under the condition of T
< T : (a) primary voltage
V
and transformer current and (b) ZVS waveforms of the switch S and S ,
and (c) switching waveforms of switch S and S .
Substituting (5) into (22) and then substituting (22) into (23),
and solving for
, yield
(24)
The proposed topology with the active-clamp snubber of
Fig. 3(a) was first simulated with Pspice using Spice models of
Si7456DP and Si7892DP for primary-side and secondary-side
MOSFETs, respectively. Experimental prototype is built and
36
75 V,
tested with the following specifications: V
V
3.3 V,
25 A, switching frequency of 200 kHz,
isolation transformer turns ratio of 4:2 with leakage inductance
of 220 nH as reflected to the primary-side,
value of 3 F,
and output inductances values of 2.3 H.
In both the simulation and the experimental work, the conventional
series snubber and the proposed active-clamp snubber
werecomparedwiththesamespecificationsandconditionsforthe
half-bridge dc–dc converter. Figs. 7 and 8 show simulation wave48 V, V
3.3 V and
forms at the operating point of
25 A. Fig. 7 shows primary voltage V and transformer
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MAO et al.: ACTIVE-CLAMP SNUBBERS
1301
Fig. 12. Experimental waveforms of transformer primary voltage and current
(C = 6.6 F; top trace Vab: 40 V/div; bottom trace i : 5 A/div).
Fig. 13. Efficiency comparison between the proposed active-clamp snubber
and the conventional RC snubber.
Fig. 11. Experimental waveforms of transformer primary voltage and current
(D = 0.28; top trace Vab: 20 V/div; bottom trace i : 5 A/div): (a) with the
conventional R/C series snubber and (b) with the proposed active-clamp snubber
(C = 3 F).
primary current waveforms for different snubber circuit cases. If
thereisnoanysnubberontheprimaryside,theleakageinductance
and MOSFET junction capacitanceoscillateduring the interval of
freewheeling when both switches are off as analyzed in Section II.
The oscillation involves body-diode reverse recovery and leads to
EMI issues. The corresponding oscillation waveforms are shown
snubber across transformer primary
in Fig. 7(a). With series
side winding,the oscillation isdamped as shownin Fig. 7(b), however, leakage inductance energy is dissipated which degrades efficiency. With the proposed active-clamp snubber, the ringing is
eliminated and clean waveforms are observed in Fig. 7(c).
As analyzed in Section II, both auxiliary switches
and
turn on and turn off under ZVZCS conditions. Fig. 8 shows the
waveforms of both switches drive voltages, drain-to-source voltages, and drain currents. It is clearly observed that both switches
turn on with zero voltage and zero current, and turn off with zero
voltage and zero current. Thus, the switching loss of the auxiliary switches is nearly zero.
Moreover, at low line input, the duty cycle is maximized
and the freewheeling period is minimized such that the energy trapped in the transformer leakage inductance cannot be
completely transferred to the capacitor. With the turn-off of
auxiliary switches
and , the remained leakage inductance
energy will be released to charge/discharge MOSFETs junction
capacitance, such that ZVS may be achieved for the switches
and , as shown in the waveforms of Fig. 9. The primary
voltage VAB and transformer current waveforms are shown in
Fig. 9(a), it can be seen that no ringing occurs during the freewheeling time. The ZVS switching waveforms of the switch
and
are shown in Fig. 9(b).
and
switching waveforms
are shown in Fig. 9(c), where
and
turn on with ZVZCS.
In the experimental prototype, three MOSFETs (Si4420DY)
are paralleled in each of the two channels of synchronous
rectifiers in the current-doubler secondary side. IRFS59N10D
MOSFETs are used for the main primary-side switches
and , 30CTQ060S Schottky diodes are selected for
and
, and Si4470EY MOSFETs are used for
and . The
MOSFET drive IC HIP2100 is used to drive the switch
and . LTC4440 high-side driver can be used for driving the
and . However, as an alternative driving scheme
switch
in the prototype, a simple self-driven circuitry based on the
transformer windings is used to drive switches
and
without adding additional IC driver.
Experimental waveforms at two different duty cycles are
shown in Figs. 10 and 11 compared with conventional
snubber. It can be observed that the ringing is attenuated with
the help of the proposed active-clamp snubber circuit and
the body diodes of the main switches was not involved in
the operation. Compared with the conventional
snubber,
the body-diodes reverse recovery and
snubber losses are
eliminated. As discussed in Section III, depending on the
capacitance value of
, the freewheeling time is adjustable.
For
6.6 F, the experimental waveforms of transformer
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voltage and current are shown in Fig. 12. Efficiency comparison curves are shown in Fig. 13 with three cases of snubber
snubber, active-clamp snubber with
3 F,
circuits:
and active-clamp snubber with
short circuit. It is shown
that higher overall efficiencies are achieved with the proposed
active-clamp converter.
V. CONCLUSION
An active-clamp snubber circuit is presented for isolated halfbridge dc–dc converters. Theoretical analysis, simulation, and
experimental results show that the proposed snubber attenuates
transformer-leakage-inductance-related ringing and eliminates
MOSFETs’ body-diode reverse recovery, resulting in improved
converter efficiencies compared to the conventional
series
snubber circuit. The half-bridge dc–dc converter with the proposed active-clamp snubber topology is a strong candidate for
applications requiring a wide range of input voltage.
ACKNOWLEDGMENT
The authors would like to thank G. Potter and B. Higgins, Astec
Power, for their thoughtful insight and helpful discussions.
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[7]
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auxiliary of a transformer,” in Proc. IEEE PESC’98, 1998, pp. 942–947.
[8] H. Mao, J. Abu-Qahouq, S. Luo, and I. Batarseh, “Zero-voltageswitching half-bridge dc–dc converter with modified PWM control
method,” IEEE Trans. Power Electron., vol. 19, no. 4, pp. 947–958, Jul.
2004.
[9] H. Mao, S. Deng, and I. Batarseh, “An active-clamp snubber for isolated
half-bridge dc–dc converters,” in Proc. 29th Annu. Conf. IEEE Industrial
Electronics Soc. IECON’03, 2003, pp. 42–48.
[10] H. Mao, J. Abu-Qahouq, S. Luo, and I. Batarseh, “New zero-voltageswitching half-bridge dc–dc converter and PWM control method,”
in Proc. 18th Annu. IEEE Applied Power Electronics Conf. Expo
(APEC’03), vol. 2, Feb. 2003, pp. 635–640.
Hong Mao received the B.S. degree from the Sichuan
University of Science and Technology, Chengdu,
China, in 1992, M.S. degree from Chongqing
University, Chongqing, China, in 1997, and Ph.D.
degree from Zhejiang University, Hangzhou, China,
in 2000, all in electrical engineering.
He is currently a Senior Design Engineer with
Astec Power, Andover, MA. From 1992 to 1994,
he was an Assistant Professor with the Department
of Electrical Engineering, Sichuan University of
Science and Technology. From 1999 to 2000, he was
a Project Leader with Zhongxing (ZTE) Telecom Corporation, China. From
2000 to 2001, he was a Senior Engineer and Division Manager with PI
Electronics, China, where he led a team to develop adaptors for SONY laptop
computers. From 2001 to 2002, he was with the Center for Power Electronics
Systems, Virginia Polytechnic Institute and State University, Blacksburg,
focusing on high-efficiency front-end dc–dc converters. In 2002, he joined the
University of Central Florida, Orlando, working on low-profile low-voltage
dc–dc converters. He was an Assistant Director with the Florida Power Electronics Center, University of Central Florida. His current research interests
include soft-switching dc–dc power conversion, power factor correction, and
modeling of power electronics systems.
Songquan Deng received the B.S. degree in mechanical engineering from Xihua University, Chengdu,
China, in 1998, the M.S. degree in electrical engineering from Chongqing University, Chongqing,
China, in 2001, and is currently pursuing the Ph.D.
degree in electrical engineering at the Florida Power
Electronics Center, University of Central Florida,
Orlando.
He is a Design Engineer with Synqor, Inc.,
Boxboro, MA. His research interests include
low-voltage high-current dc–dc converters, soft
switching techniques, and power converter topology, and control schemes.
Jaber Abu-Qahouq (M’98) received the B.Sc. degree (with first class honors) from Princess Sumaya
University/Royal Scientific Society, Amman, Jordan,
in 1998, and the M.S. and Ph.D. degrees from the
University of Central Florida (UCF), Orlando,
in 2000 and 2003, respectively, all in electrical
engineering/electronics.
He is currently a Senior Power Electronics/Power
Management Engineer with Intel Corporation, Hillsboro, OR. From January 2004 to August 2005, he was
a Faculty Member of the School of Electrical Engineering and Computer Science, UCF, and from January 2002 to December 2003,
and was a member of the Adjunct Faculty. He was a Research Assistant/Associate with the Florida Power Electronics Center, UCF, from 1999 to 2003.
From 1998 to 1999, he was with the Royal Scientific Society (RSS), Electronic
Services and Training Center (ESTC), Amman. He led and worked on several projects funded by the Florida Space Consortium, NASA, NSF, ASTEC
Power, INTEL, and the University of Central Florida. His research interests
include soft-switching power conversion, low-profile high-density low-voltage
high-current fast-transient dc–dc converters, power factor correction converters,
digital control in power electronics, and dc–ac inverters.
Dr. Abu-Qahouq has received several awards and is a member of Eta Kappa
Nu and Phi Kappa Phi.
Issa Batarseh (SM’91) received the B.S. degree
in computer engineering and the M.S. and Ph.D.
degrees in electrical engineering from the University
of Illinois, Chicago, in 1983, 1985, and 1990,
respectively.
He is a Professor and Chair with the Department
of Electrical and Computer Engineering, University
of Central Florida (UCF), Orlando. He was a Visiting Assistant Professor of Electrical Engineering at
Purdue University, Calumet City, IN, from 1989 to
1990 before joining the Department of Electrical and
Computer Engineering, UCF, in 1991. He has more than 11 U.S. patents, and
more than 50 refereed journal and 200 conference publications. His research
work has been sponsored by Federal agencies and private sector. He published
Power Electronic Circuits (New York: Wiley, 2003). His major research interests are power electronics, focusing on high frequency dc–dc conversion,
soft-switching and dynamic modeling of dc-to-dc converters, harmonic analysis, and power factor correction.
Dr. Batarseh received many national and international teaching, research, and
service awards. He served as Session Chair and Committee Member for APEC
and PESC conferences. He will be the general chair for PESC’07 conference in
Orlando. He has served as a Chairman of the IEEE Orlando Power Engineering
Chapter, Chairman of the IEEE Orlando Section, and Faculty Advisor for the
IEEE Student Branch, and Eta Kappa Nu. He is a Registered Professional Engineer in Florida.
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