Soft Switched Coupled Inductor based High Step Up
Converter for Distributed Energy Resources
Shelas Sathyan, Student member IEEE, H M Suryawanshi, Senior member IEEE, and A B Shitole
Electrical Engineering Department
Visvesvaraya National Institute of Technology, Nagpur, India 440010
Email: shelassathyan@yahoo.co.in, hms 1963@rediffmail.com, amardeep.shitole@gmail.com
Abstract—In this paper coupled inductor based high step up
converter for dc micro grid is proposed. Switched capacitor based
improved voltage extension cell is utilized to obtain high voltage
gain. By using extended and improved voltage doubler cell, the
voltage stress on the main switch and output diode are reduced.
So active switches of low RDS(on) can be used and thus overall
efficiency of the system is improved. Active clamp technique is
incorporated to limit the voltage excursion on main switch and
also to obtain zero voltage switching for both main and auxiliary
switches which further improve the efficiency and power density.
Design and analysis of the proposed converter is carried out and
the result are verified by using PSIM Simulation package. Finally,
a 500 W experimental prototype is build to verify the theoretical
and simulation results.
keywords- Coupled Inductor; DC-DC converter; High step
up; Zero voltage switching; Active clamp.
I.
I NTRODUCTION
Increasing demand of the electrical energy and the concern
of environmental pollution leads to the development of renewable energy based power generation system. Distributed generation systems using renewable and non renewable sources are
found to be a better alternative for traditional power generation
system [1]. Renewable energy sources like solar photovoltaic,
wind, wave and tidal energy systems are highly intermitted in
nature and its continuous and steady power generation depend
on various factors like environmental conditions, geographical
area and time (month) of operation etc. Thus reliable and
efficient utilization of these sources require power electronics
interfaces [2], [3]. Low voltage dc sources like solar PV and
fuel cells require to be connected in series and parallel to
obtain sufficient power and voltage level. This will increase the
overall system cost. Mismatches due to shading and unhealthy
modules reduces the available power and voltage level. It is
found that parallel connection of solar PV modules are less
sensitive to current mismatches. Since current mismatch causes
higher impact on energy yield, parallel connection of solar
modules or modular or multi string structure are preferred over
a large series strings of PV modules due to less mismatch
losses and better tracking of maximum power [4], [5]. So for
better utilization of these sources we require individual and
dedicated power converters and these converters has to step
up the energy sources output voltage to the required utility
voltage level.
Conventional boost converter can not provide high voltage
gain due to its increased conduction losses and voltage stress
at high duty ratio operation [6]. Transformer isolated converter
k,(((
can easily obtain high step up . But the stress on the switches,
weight, leakage energy problem and the power density are
limiting factors. In [7] non isolated converter using voltage
multiplier cell is introduced. But to obtain high conversion
ratio multiple such cell has to be used. coupled inductor
based converters are an alternative solution for high step
up applications. Voltage stress due to leakage inductance in
coupled inductor based converter can reduced by using active
or passive clamp technique. Voltage extension cell can be
utilized to improve the voltage conversion ratio and thus
reduce the turn’s ratio of the coupled inductor. In [8], passive
clamp coupled inductor converter is introduced for high step
up application. But this converter suffer from high voltage
stress on output diode. A switched capacitor based coupled
inductor converter is introduced in [9]. Here switched capacitor
cell charges in parallel and discharges in series to obtain
high step up. To get higher power density, converters has to
operate at high switching frequency where switching losses
dominate the conduction losses. So soft switched converters
are preferred over passive clamp converters. In [10] active
clamp soft switched coupled inductor converter is presented.
But the converters output diode voltage stress are higher. A
soft switched extended voltage doubler circuit is presented in
[11]. This converter has higher voltage conversion ratio and
the voltage stress on output diode is equal to output voltage of
the converter. In this paper improved switched capacitor based
coupled inductor converter is proposed. Proposed converter
has higher voltage gain and less voltage stress on the active
switches and output diode. So switches of low RDS(on) can
be utilized and thus improve the efficiency. By adding active
clamp circuit the voltage stress on the main switch is reduced
and zero voltage switching of both main and auxiliary switches
are obtained.
II.
PROPOSED CONVERTER AND OPERATIONAL PRINCIPLE
The proposed converter is shown in Fig.1. It consists of a
coupled inductor and improved voltage extension cell. Voltage
extension cell consisting of two switched capacitors C1 and
C2 which charges in parallel and discharges in series as given
in [9]. The diodes D1 and D2 forms the charging path for the
switched capacitors . Active clamp circuit consists of clamp
capacitor Cc and auxiliary switch SAU X . Output diode Do
is similar to the output diode of a normal boost converter.
The coupled inductor is replaced by its equivalent transformer
model with magnetizing inductance Lm and effective leakage
inductance Lk referred to primary. N1 and N2 represents
number of turns of primary windings and secondary windings
Fig.1 Proposed converter
of the coupled inductor respectively. Turn’s ratio of the coupled
2
inductor is given by N, where N = N
N1 . All switches and other
elements are considered to be ideal except leakage inductance
of coupled inductor and output capacitance of MOSFETs.
Capacitance Cr includes parallel combination of output capacitance of MOSFET switches and winding capacitance of
coupled inductor.
III.
I NTRODUCTION
There are eight operating modes in one switching cycle
of the proposed converter. Key wave form of the converter
is given in Fig.2. In the key wave forms, Vg1 and Vg-aux are
the gate signals for main and auxiliary switches. ILK , ILm ,
Is1 , and Icc are current through leakage inductance, magnetizing inductance, main MOSFET switch and clamp capacitor
respectively. The currents through output diode and charging
diodes of switched capacitors are given by IDo , ID1 and ID2
respectively. The eight operating modes of the converter are
given in Fig.3.
Mode 1 (t0 − t1 ): In this mode, the main switch S1 is
conducting and the auxiliary switch is turned off. Magnetizing
current iLm and leakage current iLk are increases linearly. The
output diode Do is reversed biased. The switched capacitors
C1 and C2 charges in parallel through the secondary winding
of the coupled inductor and diodes D1 and D2 respectively.
Load is supplied by output capacitor C0 . Current through
magnetizing inductance and leakage inductance are given by
iLm (t) = iLm (t0 ) +
iin (t) = iLk (t) = iLk (t0 ) +
kVin
(t − t0 )
Lm
(1 − k)Vin
(t − t0 )
Lk
(1)
(2)
Vin − Vc1N(t)
(t − t0 )
Lk
where k is the coupling coefficient. And k is given by
(3)
1
Cr (Lm +Lk )
and Z1 =
Lm +Lk
.
Cr
Since Cr is
very small ,this rise in voltage can be approximated as linear
increase.
Mode 3 (t2 − t3 ): At t2 , Vds (t) = Vcr (t) becomes equal to
Vin +Vcc . Since Cc is much larger than Cr almost all charging
currents now flow through the body diode of the auxiliary
switch SAUX thus limit voltage excursion on main switch.
Leakage inductance and clamp capacitor forms a resonant
circuit and the resonant current through Cc is given by
VLk (t)
sin(ω2 (t−t2 ))
Z2
(6)
Lm +Lk
and Z2 =
. Voltage across
Cc
where ω2 = √
1
Cc (Lm +Lk )
the magnetizing inductance is given by
Lm
(4)
Lm + LK
Mode 2 (t1 − t2 ): At time t1 , main switch S1 is turned
off. The leakage inductor current and magnetizing current start
to charge parasitic capacitor Cr of the main switch. Voltage
across Cr increases from 0V to Vin +Vcc . Voltage across main
switch is given by
k=
Vds (t) = Vcr (t) = Vin (1 − cos ω1 (t − t1 ))
+ iLk (t1 )Z1 sin(ω1 (t − t1 ))
where ω1 = √
iCc (t) = iLk (t) = iLk (t2 )cos(ω2 (t−t2 ))+
which can also be expressed as
iLk (t) = iLk (t0 ) +
Fig.2 Key waveforms of proposed converter
(5)
VLm ≈ −VCc
Lm
Lm + Lk
(7)
So iLm start to decrease . This mode ends at t3 when the
voltage across output diode Do become equal to zero and start
to conduct.
Mode 4 (t3 − t4 ): At t3 , output diode Do start to conduct.
Input voltage Vin , magnetizing inductance Lm , switching
capacitors C1 and C2 discharge its energy to the load in
series. Leakage inductance Lk continue to resonate with clamp
capacitor Cc . In order to get ZVS for auxiliary switch, it
Fig. 1.
(a) Mode 1 (to - t1 )
(b) Mode 2 (t1 - t2 )
(c) Mode 3 (t2 - t3 )
(d) Mode 4 (t3 - t4 )
(e) Mode 5 (t4 - t5 )
(f) Mode 6 (t5 - t6 )
(g) Mode 7 (t6 - t7 )
(h) Mode 8 (t7 - t8 )
Different modes of operation of proposed converter.
should be turned on before clamp capacitor current reverse
its direction.
Mode 5 (t4 − t5 ): At t4 , gate signal is applied to auxiliary
switch. Prior to the application of gate signal, body diode of
the switch was in conduction. So Auxiliary switch (SAU X )
turned on with ZVS. Input voltage Vin , magnetizing inductance
Lm , switching capacitors C1 and C2 continue to discharge its
energy to the load through output diode Do . Resonant current
iLk (t) reverse its direction and flow through the auxiliary
switch as shown in Fig.3.
Mode 6 (t5 − t6 ): This mode start at t5 ,when the auxiliary
switch is turned off. Now leakage inductance Lk and parasitic
capacitance of Cr forms new resonance circuit and start to
discharge the parasitic capacitance voltage Vcr (t). Since Cr
is small, the voltage across main switch decreases rapidly.
This modes end at t6 , when the parasitic capacitance voltage
becomes to zero. In order to achieve ZVS for main switch S1 ,
energy stored in leakage inductance at this mode should be
greater than the energy stored in parasitic capacitor as given
in [12] and [13]. So condition for ZVS of the main switch can
TABLE I.
be expressed as
Lk =
2
(t5 )
Cr Vcr
2
iLk (t5 )
Parameters
(8)
Power
Mode 7 (t6 − t7 ): This modes start at t6 , when the parasitic
capacitor is fully discharged. Body diode of the main switch
S1 start to conduct the current iLk . To achieve ZVS for main
switch, Gate pulse Vg1 should applied during this mode.
Mode 8 (t7 − t8 ): At time t7 , main switch turns on
with ZVS. Leakage inductor current and magnetizing current
increases linearly. Output diode current start to decrease. This
mode end at t8 , when the output diode currents equal to zero
and reverse bias. Here after switched capacitors c1 and c2 start
to charge in parallel through secondary of the coupled inductor
and cycle repeats.
IV.
PARAMETERS OF THE PROPOSED CONVERTER
Values
500 W
Turns Ratio
1:2
Vin
48 V
Vo
400 V
fs
100 KHz
Lm
132μH
Lk
2.5μH
Cc
3μF
C1 , C 2
4.7μF
Co
200μF
Cr
1.5 nF
STEADY STATE ANALYSIS
In order to simplify the analysis, coupling coefficient of
the coupled inductor is considered to one. ie, no leakage
inductance. Only mode 1 and mode 5 are considered for the
analysis. Voltage across the capacitors are considered to be
constant over one switching cycle. During mode 1, main switch
is turned on. Voltage across the magnetizing inductance is
given by
on
= Vin
(9)
VLm
and reflected voltage across secondary is given by
on
= N.Vin
Vsec
and
(10)
Fig.4 Volatge gain comparison of proposed converter
Vc1 = Vc2 = N.Vin
(11)
During the switch turned off duration (Mode 5), voltage across
Lm is given by
Vin + Vc1 + Vc2 − Vo
(12)
1+N
By volt-second balance of magnetizing inductance, voltage
gain of the proposed converter is obtained as
of f
VLm
=
Vo
1 + 2N − N D
=
Vin
1−D
(13)
Which is higher than that of the converters proposed in [10]
and [11]. Voltage stress on the clamp capacitor is given by
D.Vin
(14)
1−D
Where D is the duty ratio of the converter. Voltage stress on
the output diode is less than the output voltage and is given
by
(1 + N )Vo
VDo =
(15)
1 + 2N − N D
Voltage stress of the main switch is clamped to Vcc + Vin .
Which is given by
Vin
Vds =
(16)
1−D
Actual voltage gain of the converter will be slightly less than
that of the derived one. It is due to the the fact that presence
of leakage inductance causes duty cycle loss which slightly
reduces the actual voltage gain. Since coupling coefficient of
Vcc =
the converter is taken as unity for the steady state analysis,
effects due to this duty cycle loss is not considered for the
derivation of voltage gain. Comparison of voltage gain of
proposed converter with respect to that of [11] is shown in
Fig.4 for different duty ratio and turn’s ratio.
V.
SIMULATION AND EXPERIMENTAL RESULTS
To verify and to analysis the effectiveness of the proposed
converter, A 500 W prototype is designed and simulation
of the converter is done in PSIM simulation package. Later
experimental prototype of the converter is build and results
are verified. Different parameters of the proposed converters
are given in Table.1.
In Fig.5, simulation results of magnetizing current (iLm )
and leakage currents (iLk ) are given. Where Vg1 and Vg−aux
represents the gate signal for main and auxiliary switch respectively. In Fig.6, Simulation result of drain to source voltage of
the main switch S1 and current through the switch are shown.
The voltage of the main switch is clamped to a voltage below
125 V at full load condition. Zero voltage turn on of the switch
is also shown in Fig.6. To get ZVS for the main switch, there
should be a time delay between turn off instant of the auxiliary
switch and turn on instant of the main switch. The optimum
time delay (td ) between turned off of the auxiliary switch and
turned on of the main switch is given by
√
π Lr Cr
td =
(17)
2
Fig.5 Simulation result of magnetizing current and leakge
inductor current
Fig.8 Auxiliary switch voltage, auxiliary switch current and
switched capacitor (C1 ) current.
Fig.6 Simulation result of volatge across switch S1 and
current through S1
Fig.9 Output voltage, Switched capacitor volatge (Vc1 ) and
clamp capacitor (Vcc ) voltage.
Fig.7 Clamp capacitor current iCc , Diode D1 current iD1 ,
and output diode current iDo
Fig.10 Experimental wave form of volatge across switch S1
with out active clamp, Scale (100V/Div and 5μs/Div)
In Fig.7, current through the clamp capacitor (iCc ), diode D1 ,
and output diode Do at full load is given. Proposed converter
is also achieved zero voltage switching for auxiliary switch. In
Fig.8, voltage across auxiliary switch and current through the
switch are shown. Zero voltage turn on of the auxiliary switch
can achieve, if the delay between turn off of the main switch
S1 and turn on of the auxiliary switch is adjusted properly.
This delay is also taken as td .
In Fig.9, Simulation result of output voltage of the converter is shown along with clamp capacitor voltage (Vcc ) and
switched capacitor voltage (VC2 ). The ripple in output voltage
is less than 1% of the output voltage.
A 500 W experimental prototype is tested to verify the
design. Parameters for the experimental prototypes are given
in table 1. In proposed converter, coupled inductor is different
Fig.11 Experimental wave form of volatge across switch S1 ,
Scale (50V/Div and 2.5μs/Div)
Fig.12 Experimental wave form of current through switch
S1 . ( Scale 20A/Div and 2.5μs/Div)
Fig.14 Converter efficiency as a function of output power
VI.
Fig.13 Experimental wave form of volatge across switch S1
and current through the switch. ( Scale:50V/Div, 20A/ Div )
from flyback inductor. Here coupled inductor transfer energy
to the output during both turn on and turn off period of the
main switch. So unlike flyback converter, it does not require
large air gap to store energy. This help to choose coupled
inductor of small size. For the proposed converter, Arnold
Micrometals’s sendust core of part number MS-184125-2 is
selected as coupled inductor. Main and auxiliary MOSFET
switches are IRFP4227pbF. The clamp capacitor has to carry
high impulse current. Polypropylene capacitors are selected for
this purpose. Output filter capacitor and switched capacitors
are electrolytic type. Leakage inductance of the converter is
found to be 2.5μH. To measure exact leakage inductance using
π model of two winding transformer, the method given in [14]
is utilized. Parasitic capacitance of the MOSFET is taken as
Coss ef f and is equal to 360 pF. Control circuit is developed
in low cost C2000 Piccolo LaunchPad from Texas instruments.
In Fig.10, experimental wave form of the voltage across the
main switch (S1 ) without active clamp circuit is given . In this
case, voltage spike due to the leakage inductor increases the
stress on the switch and also induces EMI. We require switches
of higher voltage rating to sustain this spike. In Fig.11 drain to
source voltage of the main switch with active clamp is shown.
Active clamp circuit effectively clamp the voltage excursion
on the switch. So switches of low voltage rating can be used.
This improve the efficiency since low voltage rated switch
has low Rds(on) . Main switch current iS1 is shown in Fig.12.
Zero voltage turn on of main switch S1 is shown in Fig.13.
Measured maximum efficiency of the converter is 95.92%. In
Fig.14, efficency of the proposed converter for different power
level is given. Efficiency of the proposed converter remains in
the range of 95% for most of the load condition.
C ONCLUSION
In this paper coupled inductor based high step up active
clamp converter is proposed for low voltage renewable and
non renewable energy sources. In order to validate the design,
simulation and experimental result of the converter is given.
Improved and extended voltage doubler cell is utilized to increase the voltage gain of the converter. This reduces the output
diode stress compared to the conventional boost and other
coupled inductor converters. The voltage stress on the active
switches is eliminated by using active clamp technique. So
low voltage switches can be utilized to reduce the conduction
losses. It also helps to turn on both main and auxiliary switches
with zero voltage. Soft switching of the active switches help
to operate this converter at higher switching frequencies and
thus improve the power density and efficiency of the overall
system.
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Shelas Sathyan received B.Tech degree in electrical and electronics engineering from Government College of Engineering Kannur, Kerala, India in
2009 and M.Tech degree in Power electronics and Drives from Visvesvaraya
National Institute of Technology Nagpur, India in 2012. Currently he is
working towards PhD degree in electrical engineering at Visvesvaraya National
Institute of Technology Nagpur, India. His research interest includes DC-DC
converters for renewable energy sources and power factor correction.
H. M. Suryawanshi (SM’06) was born in Nagpur, India, on January 1,
1963. He received the B.E. degree in electrical engineering from Walchand
College of Engineering, Sangli, India, in 1988, the M.E. degree in electrical
engineering from the Indian Institute of Science, Bangalore, India, in 1994,
and the Ph.D. degree from Nagpur University, Nagpur, in 1999. He is currently
a Professor with the Department of Electrical Engineering, Visvesvaraya
National Institute of Technology, Nagpur. He is currently an associate editor
of IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS. His research
interests include the field of power electronics, emphasizing developmental
work in the area of resonant converters, power factor correctors, active
power filters, FACTS devices, multilevel converters, and electric drives. Dr.
Suryawanshi is fellow of IE(I) and IETE (India).
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A.B. Shitole received B.E degree in electrical engineering from Government
College of Engineering Karad, Maharashtra, India in 2008 and M.Tech degree
in Power Systems from College of Engineering Pune,India in 2012. Currently
he is working towards PhD degree in electrical engineering at Visvesvaraya
National Institute of Technology Nagpur, India. His research interest includes
control of power converters, multilevel inverters and renewable energy integration.