IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 5, MAY 2015
3021
Industrial Electronics for Electric Transportation:
Current State-of-the-Art and Future Challenges
Sheldon S. Williamson, Senior Member, IEEE, Akshay K. Rathore, Senior Member, IEEE, and
Fariborz Musavi, Senior Member, IEEE
Abstract—This paper presents the current research
trends and future issues for industrial electronics related to
transportation electrification. Specific emphasis is placed
on electric and plug-in hybrid electric vehicles (EVs/PHEVs)
and their critical drivetrain components. The paper deals
with industry related EV energy storage system issues, EV
charging issues, as well as power electronics and traction
motor drives issues. The importance of battery cell voltage equalization for series-connected lithium-ion (Li-ion)
batteries for extended life time is presented. Furthermore,
a comprehensive overview of EV/PHEV battery charger
classification, standards, and requirements is presented.
Several conventional EV/PHEV front-end ac/dc charger converter topologies as well as isolated DC/DC topologies are
reviewed. Finally, this paper reviews various EV propulsion
system architectures and efficient bidirectional DC/DC converter topologies. Novel DC/AC inverter modulation techniques for EVs are also presented. The architectures are
based on the battery voltage, capacity, and driving range.
transportation sector will rely more heavily on electricity and
related infrastructure needed for storage and distribution. All
the above issues point toward the realization of public and
private facilities to generate electricity locally, to recharge EVs
and PHEVs.
This paper aims to review and present the current status
and future research trends for transportation electrification from
an industrial electronics standpoint. EV/PHEV charging infrastructure issues and related industrial electronics solutions
are presented. The role of power electronics for on-board EV
battery energy storage and management is presented. Finally,
the paper reviews various EV propulsion system architectures
and efficient bidirectional DC/DC converter as well as novel
DC/AC traction inverter topologies.
Index Terms—Batteries, chargers, electric propulsion,
energy storage, inductive energy storage, power electronics, transportation.
In this section, a detailed charger classification, standards
and requirements are presented. Several conventional plugin hybrid electric vehicle charger front end ac/dc converter
topologies and isolated DC/DC topologies are reviewed. Considerations to improve the efficiency and performance, which
is critical to minimize the charger size, charging time, and
the amount and cost of electricity drawn from the utility, are
discussed.
I. I NTRODUCTION
C
ONVENTIONAL vehicles (CVs), which use petroleum as the only source of energy, represent the majority
of the existing worldwide on-road vehicles today. As shortage
of petroleum is considered as one of the most critical worldwide
issues, costly fuel becomes a major challenge for CV users.
Moreover, CVs emit green house gases (GHG), thus making
it harder to satisfy stringent environmental regulations. A practical and realistic solution lies in electric and plug-in hybrid
electric vehicles (EVs/PHEVs). However, lithium-ion battery
storage issues and charging issues have yet to be sorted out.
Moreover, traction inverter and electric machine issues also
need a serious thought. The bottleneck for EVs is the high
voltage battery pack, which utilizes most of the space and
increases the weight of the vehicle. At the same time, the large
II. B ATTERY C HARGERS FOR EV S /PHEV S
Charger Classification and Standards
A PHEV is a hybrid electric vehicle with a storage system
that can be recharged by connecting a plug to an external
electric power source through an AC or DC charging system.
The AC charging system is commonly an on-board charger
mounted inside the vehicle and is connected to the grid. The DC
charging system is commonly an off-board charger mounted at
fixed locations supplying required regulated DC power directly
to the batteries inside the vehicle.
A. AC Charging Systems
Manuscript received May 30, 2014; revised September 9, 2014;
accepted October 17, 2014. Date of publication March 5, 2015; date
of current version April 8, 2015.
S. S. Williamson is with the Department of Electrical and Computer
Engineering, Concordia University, Montreal, QC H3G 1M8, Canada
(e-mail: sheldon.williamson@uoit.ca).
A. K. Rathore is with the Department of Electrical and Computer
Engineering, National University of Singapore, Singapore 129788.
F. Musavi is with CUI Inc., Portland, OR 97062 USA.
Color versions of one or more of the figures in this paper are available
online at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TIE.2015.2409052
The charging AC outlet inevitably needs an on-board
AC–DC charger with a power factor correction. Table I illustrates Charge Method Electrical Ratings according to SAE EV
AC Charging Power Levels. These chargers are classified by the
level of power they can provide to the battery pack [1].
Level 1: Common household circuit, rated up to 120 V-AC
and up to 16 A. These chargers use the standard
three-prong household connection, and they are usually considered portable equipment.
0278-0046 © 2015 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.
See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
3022
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 5, MAY 2015
TABLE I
SAE EV AC C HARGING P OWER L EVELS
TABLE II
SAE EV DC C HARGING P OWER L EVELS
C. Charger Requirements
Several considerations and regulatory standards must be met.
The charger must comply with the following standards for
safety.
Level 2: Permanently wired electric vehicle supply equipment used especially for electric vehicle charging;
rated up to 240 V-AC, up to 60 A, and up to
14.4 kW.
Level 3: Permanently wired electric vehicle supply equipment used especially for electric vehicle charging;
rated greater than 14.4 kW. Fast chargers are rated
as Level 3, but not all Level 3 chargers are fast
chargers. This designation depends on the size of
the battery pack to be charged and how much time is
required to charge the battery pack. A charger can be
considered a fast charger if it can charge an average
electric vehicle battery pack in 30 minutes or less.
B. DC Charging Systems
The DC charging systems are mounted at fixed locations, like
the garage or dedicated charging stations. Built with dedicated
wiring, these chargers can handle much more power and charge
the batteries more quickly. However, as the output of these
chargers is DC, each battery system requires the output to be
changed for that car. Modern charging stations have a system
for identifying the voltage of the battery pack and adjusting
accordingly. Table II illustrates Charge Method Electrical Ratings according to SAE EV DC Charging Power Levels. These
chargers are classified by the level of power they can provide to
the battery pack [1].
1) Level 1: Permanently wired electric vehicle supply equipment (EVSE) includes the charger; rated 200–450 V DC,
up to 80 A, and up to 36 kW.
2) Level 2: Permanently wired electric vehicle supply equipment (EVSE) includes the charger; rated 200–450 V DC,
up to 200 A, and up to 90 kW.
3) Level 3: Permanently wired electric vehicle supply equipment (EVSE) includes the charger; rated 200–600 V DC,
up to 400 A, and up to 240 kW.
• UL 2202: Electric Vehicle (EV) Charging System
Equipment.
• IEC 60950: Safety of Information Technology
Equipment.
• IEC 61851-21: Electric Vehicle Conductive Charging
System—Part 21: Electric Vehicle Requirements for Conductive Connection to an ac/dc Supply.
• IEC 61000: Electromagnetic compatibility (EMC).
• ECE R100: Protection against Electric Shock.
• ISO
6469-3:
Electric
Road
Vehicles—Safety
Specifications—Part 3: Protection of Persons Against
Electric Hazards.
• ISO 26262: Road Vehicles—Functional safety.
• SAE J2929: Electric and Hybrid Vehicle Propulsion Battery System Safety Standard.
• FCC Part 15 Class B: The Federal Code of Regulation
(CFR) FCC Part 15 for EMC Emission Measurement Services for Information Technology Equipment. In addition,
it may be affected by high temperatures, vibration, dust,
and other parameters which comprise the operating environment. Therefore, the charger must meet the following
Operating Environment:
• IP6K9K, IP6K7 protection class.
• −40 ◦ C to 105 ◦ C ambient air temperature.
III. C ONVERTER TOPOLOGIES FOR L EVEL -1 AND
L EVEL -2 AC C HARGERS
The front-end AC–DC converter is a key component of the
charger system. A variety of circuit topologies and control
methods have been developed for the PFC application [2], [3].
The single-phase active PFC techniques can be divided into
two categories: the single-stage approach and the two-stage
approach. The single-stage approach is suitable for low power
applications. In addition, due to large low frequency ripple
in the output current, only lead acid batteries are chargeable.
Galvanic isolation in onboard battery chargers is one way of
addressing the double fault protection requirements for the
safety of the users of PHEV. Therefore, the two-stage approach
is the proper candidate for PHEV battery chargers, where the
power rating is relatively high, and lithium-ion batteries are
WILLIAMSON et al.: INDUSTRIAL ELECTRONICS FOR ELECTRIC TRANSPORTATION
3023
Fig. 1. Simplified system block diagram of a universal on-board twostage battery charger.
Fig. 2.
Fig. 3.
Bridgeless/dual/semi-bridgeless PFC boost converter.
Fig. 4.
Bridgeless interleaved PFC boost converter.
Interleaved PFC boost converter.
used as the main energy storage system. The front end PFC
section is then followed by a DC–DC section to complete the
charger system. Fig. 1 illustrates a simplified block diagram
of a universal input two-stage battery charger used for PHEVs
and EVs.
The PFC stage rectifies the input AC voltage and transfers it
into a regulated intermediate DC link bus. At the same time,
power factor correction function is achieved. The following
DC–DC stage then converts the DC bus voltage into a regulated
output DC voltage for charging batteries, which is required to
meet the regulation and transient requirements.
A. Front End AC–DC Converter Topologies
As a key component of a charger system, the front-end
AC–DC converter must achieve high efficiency and high power
density. Additionally, to meet the efficiency and power factor
requirements and regulatory standards for the AC supply mains,
power factor correction is essential. The conventional boost
topology is the most popular topology for PFC applications. It
uses a dedicated diode bridge to rectify the ac input voltage to
dc, which is then followed by the boost section.
1) Interleaved Boost PFC Converter: The interleaved
boost converter, illustrated in Fig. 2, consists of two boost
converters in parallel operating 180◦ out of phase [4]–[6].
The input current is the sum of the two input inductor currents. Because the inductors’ ripple currents are out of phase,
they tend to cancel each other and reduce the input ripple
current caused by the boost switching action. The interleaved
boost converter has the advantage of paralleled semiconductors. Furthermore, by switching 180◦ out of phase, it doubles
the effective switching frequency and introduces smaller input
current ripple, so the input EMI filter is relatively small [7], [8].
With ripple cancellation at the output, it also reduces stress on
output capacitors.
2) Bridgeless/Dual Boost PFC Converter: The bridgeless boost topology, illustrated in Fig. 3, is the second topology
considered for this application. The gates of the power-train
switches are tied together, so the gating signals are identical.
In dual boost topology, the MOSFET gates are not decoupled,
enabling one of the switches to remain on and operate as a
synchronous MOSFET for half line-cycle. It avoids the need
for the rectifier input bridge yet maintains the classic boost
topology [9]–[12].
The semi-bridgeless configuration, shown in Fig. 4, includes the conventional bridgeless topology, with two additional slow diodes, Da and Db that connect the input to the PFC
ground [13].
3) Bridgeless Interleaved Boost PFC Converter: The
bridgeless interleaved topology, shown in Fig. 4, was proposed
as a solution to operate at power levels above 3.5 kW. In
comparison to the interleaved boost PFC, it introduces two
MOSFETs and also replaces four slow diodes with two fast
diodes. The gating signals are 180◦ out of phase, similar to the
interleaved boost.
A detailed converter description and steady state operation
analysis are given in [13]. This converter topology shows a high
input power factor, high efficiency over the entire load range
and low input current harmonics.
B. Isolated DC–DC Converter Topologies
1) ZVS FB Converter With Capacitive Output Filter:
The ZVS full-bridge converter topology with capacitive output
filter is illustrated in Fig. 5. Current fed topologies with capacitive output filter inherently minimize diode rectifier ringing
since the transformer leakage inductance is effectively placed
in series with the supply side inductor [14]. In addition, high
efficiency can be achieved with zero voltage switching (ZVS),
3024
Fig. 5.
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 5, MAY 2015
ZVS FB converter with capacitive output filter.
Fig. 7.
Full-bridge LLC resonant converter.
cantly. However, the wide output voltage range requirements
for a battery charger are drastically different and challenging
compared to telecom applications, which operate in a narrow
output voltage range.
A detailed design procedure and resonant tank selection
for LLC resonant converters in battery charging application is
given in [17].
IV. EV/PHEV E NERGY S TORAGE S YSTEM I SSUES
Fig. 6.
Interleaved ZVS FB converter with voltage doubler.
in particular the trailing edge pulse width modulated (PWM)
full-bridge gating scheme proposed in [15] is an attractive
solution to achieve ZVS.
2) Interleaved ZVS FB Converter With Voltage Doubler: An interleaved Zero Voltage Switching Full Bridge with
Voltage Doubler, illustrated in Fig. 6, is proposed to further
reduce the ripple current and voltage stress on the output filter
capacitors, as well as component cost reduction.
Due to interleaving, each cell shares equal power and the
thermal losses are distributed uniformly among the cells and
also the input ripple is four times the switching frequency.
Moreover, the output voltage doubler rectifier significantly
reduces the number of secondary diodes and the voltage rating
of the diodes is equal to the maximum output voltage [16].
Although the proposed converter can operate in either discontinuous conduction mode (DCM), boundary conduction mode
(BCM), or continuous conduction mode (CCM), only the DCM
and BCM modes are desirable. Operation in CCM results in the
lowest RMS currents and ZVS can be achieved for all switches,
but the high di/dt results in large reverse recovery losses in
the secondary side rectifier diodes and high voltage ringing.
Moreover, to operate this converter in CCM, it requires a larger
resonant inductor which also increases the transformer turns
ratio and thus increases stress on the primary side switches.
Thus, this converter should be designed to operate in DCM,
or BCM.
3) Full-Bridge LLC Resonant Converter: Fig. 7 illustrates a Full-Bridge LLC Resonant converter. The resonant LLC
converter is widely used in the telecom industry for its high
efficiency at the resonant frequency and its ability to regulate
the output voltage during the hold-up time, where the output
voltage is constant and the input voltage might drop signifi-
It is well-known today that batteries are indeed the main
stumbling block to driving EVs. In fact, the common issues
related to lithium rechargeable cells can be summed up by one
simple topic: cell equalization. Typically, an EV battery pack
consists of a string of cells (typically 100 cells, providing a
total of about 360 Volts; 1 Li-ion cell produces approximately
3.60 V—nominal voltage). Each Li-ion cell is not exactly equal
to the others, in terms of capacity and internal resistance,
because of normal dispersion during manufacturing. However,
the most viable solution for this problem might not originate
from mere changes in battery properties or chemistries. The
aim of this section is to explain the role of power electronics
in battery cell voltage equalizers and their role in improving
battery cycle life, calendar life, power, and overall safety of
EV/PHEV battery energy storage systems.
It is imperative that most studies related to energy storage systems (ESS) for EV applications must follow a costconscious approach. For instance, taking into account that
typical lithium batteries cost about $500/kWh [5] (or $250/kWh
[6], [7], if manufactured in high volumes), a typical 16 kWh
battery, which provides about 100 km (62 miles) autonomy to a
medium-sized sedan (simulated and tested under the Federal
Test Procedure, FTP driving schedule). This amounts to a
surcharge of about $5,000 over the price of a standard vehicle,
exceeding the reasonable budget for an average consumer.
Moreover, issues related to cycle life and the calendar life
issues cannot be ignored. Depending on the intensity of usage,
an average cobalt or manganese cathode Li-Ion battery holds
about 500 cycles of 80% the capacity, before losing 20% of its
initial capacity [1]. If the battery is replaced at that point and
the cost of electricity is added, the expenses rise to $0.1/km.
Consequently, the existing scheme makes the EV option more
expensive than the traditional gasoline based vehicle. Considering newer batteries based on lithium iron-phosphate (LiFePO4 )
chemistries, these numbers are slightly better, withstanding
1000 cycles on current technology, and expecting (but still
not proven) 6000–7000 cycles for future EV applications. On
the other hand, the LiFePO4 chemistry depicts slightly lower
WILLIAMSON et al.: INDUSTRIAL ELECTRONICS FOR ELECTRIC TRANSPORTATION
3025
energy density (100 Wh/kg) [8], [9]. Although LiFePO4 seems
to be the best fit for EVs, the long term cycle life and volume
costs have to be considered seriously. As a reference, current
price per unit of LiFePO4 ranges from $1.90–$2.40/Wh, compared to $0.86/Wh, for typical manganese based Li-ion batteries [5]–[9]. Extrapolating the current unit price relationship
to high volume applications, the battery pack for a typical
medium-sized car would cost in the range of $7,000–$10,000
for a 16 kWh pack.
Another critical issue to be considered is overall safety. The
key factors that play a vital role in maintaining safety include,
usage of high quality materials and safety monitoring at the
development process. In addition, continuous monitoring of cell
current, cell voltage, temperature, and taking eventual corrective measures, also helps in critically improving the safety of
the system. However, the most viable solution for this problem
might not be originated merely from changes in the battery
chemistry. In fact, a much smarter solution relies on a power
electronic battery cell equalizer, which can improve not only
the cycle life (the quantity of charge-discharge cycles before
the end of life) of batteries, but also their calendar life (the
time, fully charged and no cycling, to end of life), power, and
safety. In the following sections, the impact of the utilization of
a battery cell equalizer will be analyzed, in terms of economical
as well as safety advantages.
V. C HARACTERISTICS OF L ITHIUM -I ON B ATTERIES FOR
EV/PHEV E NERGY S TORAGE
Lithium rechargeable battery technologies, although not mature enough to be used in EV/PHEVs, prove to be the best
solution for PHEV applications today. For instance, a 20 kWh
lithium-ion battery weighs about 160 kg (100–140 kWh/kg),
which is acceptable for PHEV applications. In contrast, current
HEV Nickel-Metal Hydride (Ni-MH) batteries weigh between
275–300 kg, for the same application. Moreover, Li-ion batteries also depict excellent power densities (400–800 W/kg)
[18]–[20], allowing more than 2C discharge rate. “C” represents the discharge of full capacity in 1 hour (at the rate
of 40–80 kW peak power, in a 20 kWh pack), and up to
10C for some chemistries [19]–[21]. However, they also suffer
from many drawbacks. One of them is the cost (projected at
about $250–$300/kWh; $600/kWh for the LiFePO4 chemistry),
which is the most expensive of all chemistries [22]. The second
drawback is that lithium is a very flammable element, whereby
its flame cannot be put off with a normal ABC extinguisher.
Finally, Li-ion batteries have a cycle life between 400 to
700 cycles, which does not satisfy EV expectations [22]. Therefore, finding a solution to these issues is extremely crucial.
To resolve safety issues, few manufacturers have modified the chemistry of the battery [23]. This is currently the
case for Lithium iron phosphate (LiFePO4 ) batteries, which
seem to have handled few issues related to EV applications,
such us reducing flammability and obtaining higher cycle life
(1000 cycles or more) [21]–[23], but the higher cost and equalization issues are still pending to be resolved. With reference
to cycle life, the battery can suffer significant degradation in its
capacity, depending on its usage. Furthermore, the internal re-
Fig. 8.
Experimental test results: (a) cell cycle life versus SOC
utilization; (b) cell total energy delivered during total lifetime versus SOC
utilization.
sistance also increases with each charge cycle. Also, according
to the chemistry and the quality of the cells, a battery typically
loses about 20% of its initial capacity after about 200 to 2000
full cycles, also known as the 100% state of charge (SOC)
cycles. The cycle life can be greatly increased by reducing
SOC, by avoiding complete discharges of the pack between
recharging or full charging. Consequently, a significant increase
is obtained in the total energy delivered, whereby the battery
lasts longer. In addition, over-charging or over-discharging the
pack also drastically reduces the battery lifetime [24]–[32].
In the context of this section, 100% SOC is the state of a
Li-ion cell after being fully charged at 4.2 V per cell, and 0%
SOC corresponds to the state of a fully discharged cell (3.0 V
per cell). The initial Capacity (C) is the capacity during the
first few cycles that go from 100 to 0% SOC, and the cycle
life is the amount of cycles after the cell loses 20% of its
initial capacity. If the battery is initially not fully charged, and
not fully discharged during the discharge period and before
charging again, then the full capacity is not used. Contrary to
the Ni-Cd batteries, in Li-ion batteries this is actually beneficial
to the cell. In fact, the cycle life is greatly increased.
In Fig. 8(a), experimental test results of cycle life of several
available cells in the market are shown, under different SOC
during the cycle. It can be appreciated how cycle life increases
as SOC utilization is reduced. In this case, SOC utilization is
considered to be centered on or around 50% SOC (half charge).
This is also confirmed by the test results of Fig. 8(b), where
the total energy delivered during the cell lifetime is also higher
3026
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 5, MAY 2015
inference that can be drawn is that the dispersion increases with
cycle life. In this case, 282 operation cycles causes the smallest
capacity cell to completely discharge, even if the demanded
capacity is 72% of the nominal capacity. The average capacity
cell, on the other hand, withstands 602 cycles of 72% nominal
capacity, before over-charging or over-discharging. Although,
this cycle life is better, it is still not enough from the point
of view of PHEV energy storage applications. Throughout this
section, several issues related to lithium batteries for EV/PHEV
applications have been exposed, particularly the unbalance in
SOC among cells. It is clear that there can be considerable
improvements in lifetime of a battery pack, if all cell capacities are suitably matched. A practical solution to obtain cell
equalization exists in the form of an electronic cell equalizer. In
the ensuing section, the most common equalizer topologies are
reviewed.
VI. I NTRODUCTION TO C LASSIC AND A DVANCED
EV/PHEV B ATTERY C ELL VOLTAGE E QUALIZERS
Fig. 9. Experimental test results: (a) cell distribution of SOC in a pack,
with initial σ = 5%, at the end of discharge; (b) cell distribution of SOC
in a pack, with initial σ = 5%, at the end of charge.
when SOC reduces. The latter is expressed in C-units (initial
capacity under 100% SOC).
The trend curves are critical for economical analysis. They
are estimated based on the tests performed by [24]–[32]. Each
point represents the value during a test performed on multiple
reference sources, and the dotted line is the calculated trend
of all those values. The plots of Fig. 8 can also be the subject
matter for further analyses. For example, in a 360 V EV battery
pack, a string of 100 cells is used. In this simulation, a 5% initial
dispersion (σ) in the capacity of the cells is considered, charging the pack at 4.1 V per cell (86% SOC), and discharging at
72% of the total capacity (up to 14% SOC). An interesting trend
can be observed. The smaller capacity cells swing initially from
about 92% SOC to 8% SOC, which is 84% of the total capacity,
instead of the average 72%. This higher SOC swing can be
translated into less cycle life (282 cycles, instead of 600), which
is a deeper degradation for the smallest capacity cells. This
effect deepens during successive cycles, producing a premature
degradation of smallest capacity cells, which brings the whole
pack into a premature “out of service.” Fig. 9(a) shows the
distribution of SOC (quantity of cells versus SOC) at the end of
discharge, in steps of 47 cycles. Fig. 9(b) shows the distribution
of SOC (quantity of cells versus SOC) at the beginning of
discharge, in steps of 47 cycles. The gradually increasing SOC
span is an indication of the reduced capacity.
From Fig. 9, it is clear that the end of life arrives faster,
because of the initial dispersion in capacity. Another critical
A battery cell voltage equalizer is essentially a power electronic controller, which takes active measures to equalize the
voltage in each cell. Furthermore, by few additional methods,
such as measuring the actual capacity and internal resistance
of each cell (followed by instantaneous SOC computation), it
is capable of equalizing the SOC of each cell. As a result,
each of the cells will have the same SOC during charging and
discharging, even in conditions of high dispersion in capacity
and internal resistance. If all the cells have the same SOC
utilization, they will degrade equally, at the average degradation
of the pack. If this condition is accomplished, then all the cells
will have the same capacity during the whole lifetime of the
battery pack, avoiding premature end of life (EOL), due to the
EOL of only one cell. If after SOC equalization, there still exists
a case of faster degradation in some cells, the equalizer will
further reduce the current demand on those cells, thus reducing
the demand and degradation. In the example presented in the
previous section, in Fig. 9, instead of 282 cycles, the pack
would last 602 cycles. For the same application, the requirement of current though the equalizer is 5 A of equalizing current
from any one cell to another, as will be explored later in this
section. In principle, there exist 3 basic groups of equalizers;
resistive, capacitive, and inductive. Their main characteristics
are explored in the ensuing sub-sections.
A. Resistive Equalizers
Resistive equalizers simply burn the excess power in higher
voltage cells, as depicted in Fig. 10. Consequently, they represent the cheapest option, and are widely utilized for laptop
batteries.
Obviously, due to inherent heating problems, resistive equalizers tend to have low equalizing currents in the range of
300–500 mA, and work only in the final stages of charging
and flotation. Due to the virtual non-existence of energy recovery, the efficiency is practically 0%. Also, as aforementioned,
because the EV battery should avoid working at high temperatures, and because in this configuration, all the equalizing
WILLIAMSON et al.: INDUSTRIAL ELECTRONICS FOR ELECTRIC TRANSPORTATION
Fig. 10.
Schematic representation of a typical resistive equalizer.
Fig. 12.
Fig. 11.
3027
Schematic representation of a typical capacitive equalizer.
current is transformed into heat, this equalizer configuration is
not recommended for high reliability battery packs [33].
B. Capacitive Equalizers
Capacitive based equalizers use switched capacitors, as
shown in Fig. 11, to transfer the energy from the higher voltage
cell to the lower voltage cell. It switches the capacitor from
cell-to-cell, allowing each cell to physically have the same
voltage. Besides, it also depicts higher current capabilities than
a resistive equalizer.
In addition, capacitive equalizers are also quite simple to
implement, without any control issues [33], [38]. At the same
time, the main drawback of capacitive equalizers is the fact
that they cannot control inrush currents, in the case of big
differences in cell voltages, leading to potentially devastating
current ripples flowing into the cells. Furthermore, they do
not allow any desired voltage difference, which are especially
critical in equalizing SOC.
C. Inductive Equalizers
Inductive or transformer based equalizers use an inductor
to transfer energy from the higher voltage cell to the lower
Schematic representation of a typical inductive equalizer.
voltage cell. In fact, this is the most popular family of highend equalizers. Due to its capability to fulfill most of the
needs for EV battery energy storage: high equalizing current,
high efficiency, and in some configurations, controllability, it is
explored in detail in forthcoming sections of this section.
1) Basic Inductive Equalizer: A basic inductive equalizer
is shown in Fig. 12. These equalizers are relatively straightforward and can transport a large amount of energy. At the same
time, they are also capable of handling more complex control
schemes, such as current limitation and voltage difference
control [34].
This allows the controller to compensate for the internal
resistance of the cells, and increase equalization current, independent of the cell voltage. On the other hand, it takes some
additional components to avoid current ripples from getting into
the cell. Typically, this configuration requires 2 switches (plus
drivers and controls) per cell. Also, due to switching losses,
the distribution of current tends to be highly concentrated in
adjacent cells. Hence, a high-voltage cell will distribute the
current largely among the adjacent cells, instead of doing it
equally in all the cells along the string. In this case, the typical
50% duty cycle switching scheme could be replaced by a more
global scheme, with a slight additional cost of more processing
power.
2) Cuk Equalizer: As the name indicates, this is an
inductive-capacitive type of equalizer, primarily based on the
Cuk converter topology. It shares almost all the positive characteristics of inductive equalizers, plus a very small cell current ripple. However, it suffers in terms of additional cost of
power capacitors and double rated switches (higher voltage and
current handling) [35]–[37]. The schematic representation of a
typical Cuk equalizer is shown in Fig. 13.
The Cuk equalizer does incur additional losses due to the
series capacitor, having slightly less efficiency than typical
inductive equalizers. Similar to inductive equalizer, the Cuk
equalizer also presents some issues while distributing equalizing current among all the cells in the string. This equalizer also
3028
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 5, MAY 2015
TABLE III
C OMPARISON OF C ELL E QUALIZER C HARACTERISTICS [33]–[38]
Fig. 13.
Schematic representation of a typical Cuk equalizer.
Fig. 14.
Schematic representation of a multi-winding transformer
equalizer.
Fig. 15.
Schematic representation of a multiple transformer equalizer.
possesses high current and complex control capability, at the
expense of additional processing power.
3) Transformer Based Equalizers: The solutions provided by transformer-based equalizers theoretically permit the
right current distribution along all cells, without any additional
losses or control issues. One such popular arrangement is
depicted in Fig. 14 [33].
Such a topology poses an additional problem of using a very
complex multi-secondary transformer. This transformer is very
difficult to mass produce, because all the secondary windings
must have exactly the same voltage and resistance. If not, the
differences will be translated into cell voltage difference, failing
to perform the equalization accurately. Hence, this option is not
a practical solution for high-count EV cell packs. Moreover,
this option also lacks the capability of handling complex control
algorithms, such as current and voltage control. An alternative
solution is presented in Fig. 15, using separate transformers for
each cell.
This solution is modified here, to use 1:1 transformers, which
are less difficult to mass produce [33]. Although this topology
represents a substantial improvement with respect to multisecondary transformers, in terms of manufacturability and cost,
only a very small dispersion can be accepted in the transformer
primary inductance. This is still very difficult to obtain in
commercial inductors, with the risk of experiencing current and
voltage imbalance.
Table III summarizes the capabilities of each type of equalizer listed above. They are classified in relation with the
main characteristics of each type of equalizer, as previously
reviewed. The ranking scheme considers the positive or negative effect over the equalizer, i.e., higher equalizer current is
positive, while higher cost is negative.
It can be appreciated, that in general, none of the equalizer
configurations are a perfect fit for a particular set of applications. For example, in very low cost, low current applications,
resistive equalizers prove to be practical. For intermediate size
batteries, where current or battery string length is limited, the
capacitive or transformer based equalizers can be envisaged.
VII. P OWER E LECTRONICS FOR EV D RIVETRAINS
This section discusses and compares three kinds of propulsion system architectures for EVs. The design constraints for
the power conditioning system for electric propulsion in fuel
cell vehicles (FCVs) are similar to Battery Electric Vehicles
(BEV) and that is adapting to dynamic load behavior. In addition, fuel cells (FC) have a unique property compared to
batteries that FC stack voltage varies with the load current.
FC deliver rated power at low voltage and transfers low power
at higher FC stack voltage. It, therefore, causes continuous
variation in FC voltage with dynamic vehicle load as discussed.
Three power propulsion system architectures are discussed
next based on source (FC or Battery) voltage, capacity, and
drive range.
A. High Voltage Variable DC Bus System (Fig. 16)
This architecture has variable dc voltage bus from FC or
battery and offers the following features.
1) Very simple system with lesser number of components.
WILLIAMSON et al.: INDUSTRIAL ELECTRONICS FOR ELECTRIC TRANSPORTATION
Fig. 16.
voltage.
3029
Fig. 18.
Fixed DC link voltage electrical system architecture.
Fig. 19.
Naturally commutated CFDAB converter.
Power system architecture with high-voltage variable DC-link
C. Low Voltage Fixed DC Bus System (Fig. 18)
Fig. 17.
voltage.
Power system architecture with low-voltage variable DC-link
2) This has isolation between the battery and the propulsion
drive but not between the FC stack and the drive.
3) It is very challenging to control three phase inverter with
input voltage varying between FC bus voltage of 150 V
to 300 V when there is not sufficient voltage to reach
high speeds due to FC characteristic. This may bring
control stability problems in the drive system.
4) Inverter controls the machine current of traction drive
and bidirectional converter controls the power flow into
and out of the battery. Regulation of power output from
the FC stack is not possible on electrical domain.
5) Owing to safety reasons, high voltage DC bus is not
preferred in small power compact vehicles.
B. Low Voltage Variable DC Bus System (Fig. 17)
This architecture has variable low voltage dc voltage bus
from FC or battery and offer s the following features.
1) More complex compared to previous system due to
multi-stage inverter instead of single-stage voltage
source inverter (VSI).
2) Both the FC stack and the battery are isolated from the
drive system.
3) Since low voltage DC bus (32 V to 68 V in case of
FC and 48 V in case of a battery) is used, the design
of a bidirectional converter becomes simpler owing to
requirement of low voltage gain. Non-isolated topologies may be sufficient to perform the required task with
desired voltage gain.
4) Six-pulse modulation technique discussed in next
section eliminates DC link electrolytic capacitor and
reduces switching losses substantially in the inverter
devices by 86.6% as compared to standard VSI.
This architecture has variable low voltage dc voltage bus
from FC or battery along with intermediate dc link and offer
s the following features.
1) Additional converter compared to previous two systems,
but design and control of power converters is very easy
as the DC bus voltage is regulated.
2) FC converter regulates the DC bus voltage at 100 V.
Bidirectional converter is designed to operate at highest
efficiency as the operating input and output voltages are
fixed. Constant DC bus voltage also eases the control
of inverter drive system. Hence, control of all the three
converters is straight forward.
VIII. P OWER E LECTRONIC C ONVERTER TOPOLOGIES
Bidirectional converter topologies are required to interface
low voltage battery and fuel cell DC bus in FCV or between
battery and inverter in low voltage EV [39]. High frequency
(HF) ripple results into waste of fuel and low overall system
efficiency. Low frequency harmonic pulsation introduces instability issues. Therefore, power electronics system design must
address these critical constraints. Numerous bidirectional converters have been proposed in literature [40]–[42] but are hard
switched. To realize higher efficiency and compact design, softswitching converters are preferred [43]–[48] pushing device
switching frequency to higher limit. Full-bridge bidirectional
DC/DC soft-switching converters are considered for this application [49]. Current-fed converters are chosen over voltagefed converters to attain low ripple in FC current. Comparison
and performance evaluation of competing alternate voltage-fed
dual active bridge (DAB) and current-fed dual active bridge
(CFDAB) and half-bridge topologies have been reported in
[50]–[52].
Voltage-fed DAB demonstrates higher circulating current,
high peak current, and reduced efficiency at light load and
varying source voltage compared to current-fed converters. In
current-fed topologies, to limit the voltage across the semiconductor devices at turn-off, various solutions are found in
3030
Fig. 20.
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 5, MAY 2015
Schematic of the proposed SRSPM and DTPM modulated inverter system without DC link.
[40], [48], [53]. Naturally clamped snubberless bidirectional
CFDAB topology is shown in Fig. 19, where modulation of
secondary switches naturally clamps the voltage across the
primary devices with zero current commutation [54] along with
the removal of external active or passive snubber circuit. The
proposed converter achieves zero-current switching (ZCS) of
the semiconductor devices on low voltage side and zero-voltage
switching (ZVS) of the high voltage side devices. Current-fed
half-bridge topology is reported in [55]. For low FC stack configuration (Figs. 17 and 18), design of single high density unit
at rated power is a challenge as it requires very low transformer
leakage inductance. Therefore, interleaving is adapted [56]. It is
challenging to control an inverter with widely varying DC bus
voltage in particular if dc bus voltage is not sufficient to reach
high speeds (Fig. 19). In literature, various control techniques
for traction drive are available like vector control, direct torque
control, and hysteresis control [57]–[59]. Standard three-phase,
two-level VSI has been used for this particular conversion.
A power electronics system is shown in Fig. 20 to implement
propulsion system shown in Fig. 17. Battery of 12 V has been
interfaced with input (32 to 68 V) through a bidirectional
converter shown in Fig. 19. The traction motor drive is fed
from a two-stage HF inverter. Because the input DC voltage is
lower than the peak value of the required output voltage, voltage
needs to be boosted. Therefore, two-stage inverter consisting of
front-end DC/DC converter followed by a standard three-phase
PWM inverter is a potential solution. It consists of a voltagefed full-bridge DC/DC converter at front-end modulated by
single reference six-pulse modulation (SRSPM) to produce
HF pulsating voltage consisting of six-pulse waveform on an
average followed by a three-phase inverter switched using 33%
modulation scheme that produces balanced three-phase voltage
by switching only one leg out of three at any given time.
Traditional requirement of electrolytic capacitor at the DC link
has been eliminated [60]–[66].
Various topologies and modulation techniques at front-end
have been adapted in literature to generate pulsating HF six
pulse waveform at the input of the inverter. [60]–[62] proposed
a DC to pulsating DC voltage using three full-bridges and three
HF transformers modulated with three references. Major issues
are complex control because of multiple references and circu-
lating current through semiconductor devices. In an alternate
topology, two full-bridge converters are interleaved by parallel
input and series output shown in Fig. 20 [63]–[65]. Dual threepulse modulation (DTPM) has been proposed, implemented,
and demonstrated to modulate two bridges using two reference voltages VA,ref and VB,ref , generated from three-phase
voltages. This produces pulsating output DC voltage at Vdc ,
which is fed to standard 3-phase inverter [65]. Switching of
both converters is modulated in such a way that, output at Vdc
has 6-pulse waveform on an average. This topology has lesser
component count but still faces circulating current problem.
If one bridge fails, then the inverter cannot produce desired
balanced output to supply to the drive. The detailed steady-state
analysis, design, and results are reported in [65].
A novel innovative SRSPM technique is proposed to modulate a full-bridge converter to produce pulsating DC voltage
at Vdc , which is fed to standard 3-phase inverter [66], which is
modulated by 33% modulation. Significant saving in switching
losses of the semiconductor devices is obtained and DC lank
capacitor is eliminated. Drop in switching losses accounts to be
around 86.6% in comparison with a standard voltage source inverter modulated by sine pulse width modulation (SPWM). The
detailed steady-state analysis, design, and results are reported in
[66]. If interleaved, in case of failure of one of the cells, inverter
can still deliver at least lower power to the motor drive due
to unique single reference control. It makes the inverter fault
tolerant. These two modulations functioning together at highfrequency presents high efficiency while achieving compact
and light weight designs, which are of important concern for
a vehicle design.
IX. S UMMARY AND C ONCLUSION
This paper thoroughly discussed the critical issues related
to the current status and future development of industrial electronics for electric transportation. The paper highlights the use
of battery energy storage system cell voltage equalizers, which
the auto industry is still looking to optimize, when used in conjunction with EV battery packs. A popular norm in the traction
battery manufacturing industry is that the key issue to commercializing EVs involves reducing the cost of the cell management
WILLIAMSON et al.: INDUSTRIAL ELECTRONICS FOR ELECTRIC TRANSPORTATION
system to less than 1% of the total battery pack cost. Key
solutions for future EV battery pack management systems, with
a special emphasis on the power electronics-intensive on-board
cell voltage management system are addressed. Key issues related to charging of future EVs/PHEVs are reported. Presently
existing standards have been summarized, whereas future DC
charging infrastructures were also discussed. Relevant power
electronic converter topologies to achieve DC as well as AC
charging for Levels 1, 2, and 3—slow, medium, and fast
charging, respectively, were discussed. Futuristic renewable
energy based charging infrastructures as well as DC or AC
charging using inductive power transfer are expected to utilize
some intuitive combination of the presented charger topologies.
Finally, various BEVs/FCVs propulsion system architectures as
well as efficient bidirectional DC/DC converter topologies have
been evaluated. Novel two-stage high-frequency inverter and
novel modulation techniques were discussed. Power electronic
system architectures presented were mainly based on the battery voltage, capacity, driving range, and applications.
R EFERENCES
[1] Surface Vehicle Recommended Practice J1772, SAE Electric Vehicle and
Plug in Hybrid Electric Vehicle Conductive Charge Coupler, SAE International, Jan. 2010.
[2] B. Singh et al., “A review of single-phase improved power quality ac–dc
converters,” IEEE Trans. Ind. Electron., vol. 50, no. 5, pp. 962–981,
Oct. 2003.
[3] C. Qiao and K. M. Smedley, “A topology survey of single-stage power
factor corrector with a boost type input-current-shaper,” IEEE Trans.
Power Electron., vol. 16, no. 3, pp. 360–368, May 2001.
[4] M. O’Loughlin, An Interleaved PFC Preregulator for High-Power
Converters. [Online]. Available: http://www.ti.com/download/trng/docs/
seminar/Topic5MO.pdf
[5] L. Balogh and R. Redl, “Power-factor correction with interleaved
boost converters in continuous-inductor-current mode,” in Proc. IEEE
Appl. Power Electron. Conf. Expo., San Diego, CA, USA, Mar. 1993,
pp. 168–174.
[6] Y. Jang and M. M. Jovanovic, “Interleaved boost converter with intrinsic
voltage-doubler characteristic for universal-line PFC front end,” IEEE
Trans. Power Electron., vol. 22, no. 4, pp. 1394–1401, Jul. 2007.
[7] P. Kong, S. Wang, F. C. Lee, and C. Wang, “Common-mode EMI study
and reduction technique for the interleaved multichannel pfc converter,”
IEEE Trans. Power Electron., vol. 23, no. 5, pp. 2576–2584, Sep. 2008.
[8] C. Wang, X. Ming, F. C. Lee, and B. Lu, “EMI study for the interleaved
multi-channel PFC,” in Proc. IEEE Power Electron. Spec. Conf., 2007,
pp. 1336– 1342.
[9] B. Lu, R. Brown, and M. Soldano, “Bridgeless PFC implementation using
one cycle control technique,” in Proc. IEEE Appl. Power Electron. Conf.
Expo., 2005, pp. 812– 817.
[10] U. Moriconi, “ A bridgeless PFC configuration based on L4981 PFC
controller,” STMicroelectronics, Geneva, Switzerland, Nov. 2012.
[Online].
[11] Y. Jang and M. M. Jovanovic, “A bridgeless PFC boost rectifier with
optimized magnetic utilization,” IEEE Trans. Power Electron., vol. 24,
no. 1, pp. 85–93, Jan. 2009.
[12] L. Huber, J. Yungtaek, and M. M. Jovanovic, “Performance evaluation of
bridgeless PFC boost rectifiers,” IEEE Trans. Power Electron., vol. 23,
no. 3, pp. 1381–1390, May 2008.
[13] F. Musavi, W. Eberle, and W. G. Dunford, “A phase-shifted gating
technique with simplified current sensing for the semi-bridgeless ac–dc
converter,” IEEE Trans. Veh. Technol., vol. 62, no. 4, pp. 1568–1576,
May 2013.
[14] I. D. Jitaru, “A 3 kW soft switching dc–dc converter,” in Proc. IEEE Appl.
Power Electron. Conf. Expo., 2000, pp. 86– 92.
[15] D. Gautam, F. Musavi, M. Edington, W. Eberle, and W. G. Dunford, “A
zero voltage switching full-bridge dc–dc converter with capacitive output
filter for a plug-in-hybrid electric vehicle battery charger,” in Proc. IEEE
Appl. Power Electron. Conf. and Expo., 2012, pp. 1381–1386.
3031
[16] D. Gautam, F. Musavi, M. Edington, W. Eberle, and W. G. Dunford,
“An isolated interleaved dc–dc converter with voltage doubler rectifier for
PHEV battery charger,” in Proc. IEEE Appl. Power Electron. Conf. Expo.,
2013, pp. 3067–3072.
[17] F. Musavi, M. Craciun, D. S. Gautam, W. Eberle, and W. G. Dunford,
“An LLC resonant dc–dc converter for wide output voltage range battery
charging applications,” IEEE Trans. Power Electron., vol. 28, no. 12,
pp. 5437–5445, Dec. 2013.
[18] Li-Ion 18650, Quantities of 50,000, AA Portable Power Corp., Richmond,
CA, USA, Jan. 2009.
[19] T. Markel and A. Simpson, “Energy storage systems considerations
for grid-charged hybrid electric vehicles,” in Proc. IEEE Veh. Power
Propulsion Conf., Chicago, IL, USA, Sep. 2005, pp. 344–349.
[20] M. Anderman, F. Kalhammer, and D. MacArthur, “Advanced batteries for
electric vehicles: An assessment of performance, cost, and availability,”
State of California Air Resources Board, Sacramento, CA, USA, Tech.
Rep., Jun. 2000.
[21] A123 Systems, Inc. [Online]. Available: http://www.a123systems.com
[22] LiFeBATT, Inc. [Online]. Available: http://www.lifebatt.com
[23] H. Webster, “Flammability assessment of bulk-packed, rechargeable
lithium-ion cells in transport category aircraft,” Office Aviation Res.
Develop., Washington, DC, USA, Sep. 2006.
[24] bq27500-System Side Impedance Track Fuel Gauge, Texas Instruments,
Dallas, TX, USA, Sep. 2007.
[25] P. Ramadass, B. Haran, R. White, and B. Popov, “Performance study of
commercial LiCoO2 and spinel-based Li-ion cells,” J. Power Sources,
vol. 111, no. 2, pp. 210–220, Apr. 2002.
[26] H. Maleki and J. Howard, “Effects of overdischarge on performance and
thermal stability of a Li-ion cell,” J. Power Sources, vol. 160, no. 2,
pp. 1395–1402, Oct. 2006.
[27] J. W. Lee, Y. K. Anguchamy, and B. N. Popov, “Simulation of
charge-discharge cycling of lithium-ion batteries under low-earth-orbit
conditions,” J. Power Sources, vol. 162, no. 2, pp. 1395–1400,
Jul. 2006.
[28] G. Ning, B. Haran, R. White, and B. Popov, “Cycle life evaluation
of pouch lithium-ion battery,” in Proc. 204th Meet. Electrochem. Soc.,
Orlando, FL, USA, Oct. 2003, pp. 1, Abstract 414.
[29] P. Liu, K. Kirby, and E. Sherman, “Failure mechanism diagnosis
of lithium-ion batteries,” in Proc. 206th Meet. Electrochem. Soc.,
Honolulu, HI, USA, Oct. 2004, pp. 1, Abstract 387.
[30] K. A. Striebel et al., “Characterization of high-power lithium-ion cellsperformance and diagnostic analysis,” Lawrence Berkeley Nat. Lab.,
Berkeley, CA, USA, Paper LBNL-54097, Nov. 2003.
[31] H. Croft, B. Staniewicz, M. C. Smart, and B. V. Ratnakumar, “Cycling
and low temperature performance of lithium-ion cells,” in Proc. IEEE
35th Intersoc. Energy Convers. Eng. Conf. Exhib., Las Vegas, NV, USA,
Jul. 2000, vol. 1, pp. 646–650.
[32] M. C. Smart et al., “Performance characteristics of lithium-ion cells for
NASA’s Mars 2001 Lander application,” IEEE Aerosp. Electron. Syst.
Mag., vol. 14, no. 11, pp. 36–42, Nov. 1999.
[33] P. A. Cassani and S. S. Williamson, “Feasibility analysis of a novel
cell equalizer topology for plug-in hybrid electric vehicle energy-storage
systems,” IEEE Trans. Veh. Technol., vol. 58, no. 8, pp. 3938–3946,
Oct. 2009.
[34] K. Nishijima, H. Sakamoto, and K. Harada, “A PWM controlled
simple and high performance battery balancing system,” in Proc.
IEEE Power Electron. Spec. Conf., Galway, Ireland, Jun. 2000, vol. 1,
pp. 517–520.
[35] P. A. Cassani and S. S. Williamson, “Design, testing, and validation of
a simplified control scheme for a novel plug-in hybrid electric vehicle battery cell equalizer,” IEEE Trans. Ind. Electron., vol. 57, no. 12,
pp. 3956–3962, Dec. 2010.
[36] Y. S. Lee and M. W. Cheng, “Intelligent control battery equalization for
series connected lithium-ion battery strings,” IEEE Trans. Ind. Electron.,
vol. 52, no. 5, pp. 1297–1307, Oct. 2005.
[37] Y. S. Lee, M. W. Cheng, S. C. Yang, and C. L. Hsu, “Individual cell
equalization for series connected lithium-ion batteries,” IEICE Trans.
Commun., vol. E89-B, no. 9, pp. 2596–2607, Sep. 2006.
[38] A. C. Baughman and M. Ferdowsi, “Double-tiered switched-capacitor
battery charge equalization technique,” IEEE Trans. Ind. Electron.,
vol. 55, no. 6, pp. 2277–2285, Jun. 2008.
[39] F. Z. Peng, H. Li, G.-J. Su, and J. S. Lawler, “A new ZVS bidirectional
dc–dc converter for fuel cell and battery application,” IEEE Trans. Power
Electron., vol. 19, no. 1, pp. 54–65, Jan. 2004.
[40] K. Wang et al., “Bi-directional DC to DC converters for fuel cell
systems,” in Proc. IEEE Power Electron. Transp. Conf., 1998,
pp. 47–51.
3032
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 5, MAY 2015
[41] M. Marchesoni and C. Vacca, “New dc–dc converter for energy storage
system interfacing in fuel cell hybrid electric vehicle,” IEEE Trans. Power
Electron., vol. 22, no. 1, pp. 301–308, Jan. 2007.
[42] R. Sharma and H. Gao, “Low cost high efficiency dc–dc converter for fuel
cell powered auxiliary power unit of a heavy vehicle,” IEEE Trans. Power
Electron., vol. 21, no. 3, pp. 587–591, May 2006.
[43] O. D. Patterson and D. M. Divan, “Pseudo-Resonant full bridge dc/dc
converter,” IEEE Trans. Power Electron., vol. 6, no. 4, pp. 671–678,
Oct. 1991.
[44] L. Zhu, “A novel soft-commutating isolated boost full-bridge ZVS-PWM
dc–dc converter for bidirectional high power applications,” IEEE Trans.
Power Electron., vol. 21, no. 2, pp. 422–429, Mar. 2006.
[45] H.-J. Chiu and L.-W. Lin, “A bidirectional dc–dc converter for fuel cell
electric vehicle driving system,” IEEE Trans. Power Electron., vol. 21, no.
4, pp. 950–958, Jul. 2006.
[46] Y. Du, S. M. Lukic, B. Jacobson, and A. Huang, “Review of high
power isolated bi-directional DC–DC converters for PHEV/EV DC charging infrastructure,” in Proc. IEEE Energy Convers. Congr. Expo., 2011,
pp. 553–560.
[47] T.-F. Wu, Y.-C. Chen, J.-G. Yang, and C.-L. Kuo, “Isolated bidirectional
full-bridge DC–DC converter with a flyback snubber,” IEEE Trans. Power
Electron., vol. 25, no. 7, pp. 1915–1922, Jul. 2010.
[48] S. J. Jang, C. Y. Won, B. K. Lee and J. Hur, “Fuel cell generation
system with a new active clamping current-fed half-bridge converter,”
IEEE Trans. Energy Convers., vol. 22, no. 2, pp. 332–340, Jun. 2007.
[49] D. Xu, C. Zhao and H. Fan, “A PWM plus phase-shift control bidirectional dc–dc converter,” IEEE Trans. Power Electron., vol. 19, no. 3,
pp. 666–675, May 2004.
[50] A. K. Rathore and U. R. Prasanna, “Comparison of soft-switching
voltage-fed and current-fed bi-directional isolated dc/dc converters for
fuel cell vehicles,” in Proc. IEEE Int. Symp. Ind. Electron., 2012,
pp. 252–257.
[51] J. S. Lai and D. J. Nelson, “Energy management power converters
in hybrid electric and fuel cell vehicles,” Proc. IEEE, vol. 95, no. 4,
pp. 766–777, Apr. 2007.
[52] P. Xuewei and A. K. Rathore, “Comparison of bi-directional voltage-fed
and current-fed dual active bridge Isolated dc/dc converters low voltage
high current applications,” in Proc. IEEE Int. Symp. Ind. Electron., 2014,
pp. 2566– 2571.
[53] J.-G. Cho, C.-Y. Jeong, and F. C. Y. Lee, “Zero-voltage and zero-currentswitching full-bridge PWM converter using secondary active clamp”,
IEEE Trans. Power Electron., vol. 13, no. 4, pp. 601–607, Jul. 1998.
[54] P. Xuewei and A. K. Rathore, “Novel bidirectional snubberless naturally
commutated soft-switching current-fed full-bridge isolated DC/DC converter for fuel cell vehicles,” IEEE Trans. Ind. Electron., vol. 61, no. 5,
pp. 2307–2315, May 2014.
[55] A. K. Rathore and U. R. Prasanna, “Analysis, design, and experimental
results of novel snubberless bidirectional naturally clamped ZCS/ZVS
current-fed half-bridge dc/dc converter for fuel cell vehicles,” IEEE Trans.
Ind. Electron., vol. 60, no. 10, pp. 4482–4491, Oct. 2013.
[56] P. Xuewei and A. K. Rathore, “Novel interleaved bidirectional snubberless soft-switching current-fed full-bridge voltage doubler for fuel-cell
vehicles,” IEEE Trans. Power Electron., vol. 28, no. 12, pp. 5535–5546,
Dec. 2013.
[57] I. Boldea, “Control issues in adjustable-speed drives,” IEEE Ind. Electron.
Mag., vol. 2, no. 3, pp. 32–50, Sep. 2008.
[58] M. Ehsani, Y. Gao, and J. M. Miller, “Hybrid electric vehicles: Architecture and motor drives,” Proc. IEEE, vol. 95, no. 4, pp. 719–728,
Apr. 2007.
[59] S. S. Williamson, A. Emadi, and K. Rajashekara, “Comprehensive efficiency modeling of electric traction motor drives for hybrid electric
vehicle propulsion applications,” IEEE Trans. Veh. Technol., vol. 56,
no. 4, pp. 1561–1572, Jul. 2007.
[60] H. Fujita, “A three-phase voltage-source solar power conditioner using
a single-phase PWM control method,” in Proc. IEEE Energy Convers.
Congr. Expo., 2009, pp. 3748–3754.
[61] S. K. Mazumder and A. K. Rathore, “Performance evaluation of a new
hybrid-modulation scheme for high-frequency-ac-link inverter: Application for PV, wind, fuel-cell and DER/storage applications,” in Proc. IEEE
Energy Convers. Congr. Expo., 2010, pp. 2529–2534.
[62] R. Huang and S. K. Mazumder, “A soft switching scheme for multiphase dc/pulsating-dc converter for three-phase high-frequnecy-link
Pulsewidth Modulation (PWM) inverter,” IEEE Trans. Power Electron.,
vol. 25, no. 7, pp. 1761–1744, Jul. 2010.
[63] Q. Lei and F. Z. Peng, “Space vector pulsewidth amplitude modulation
for a buck-boost voltage/current source inverter,” IEEE Trans. Power
Electron., vol. 29, no. 1, pp. 266–274, Jan. 2014.
[64] U. R. Prasanna and A. K. Rathore, “Dual Three-Pulse-Modulation
(DTPM) based high-frequency pulsating dc link two-stage three-phase
inverter for electric/hybrid/fuel cell vehicles applications,” IEEE J. Emerg.
Sel. Topics Power Electron., vol. 2, no. 3, pp. 477–486, Sep. 2014.
[65] A. Rahnamaee and S. K. Mazumder, “A soft-switched hybrid-modulation
scheme for a dc-link-capacitor-less three-phase pulsating dc-link inverter,” IEEE Trans. Power Electron., vol. 29, no. 8, pp. 3893–3906,
Aug. 2014.
[66] U. R. Prasanna and A. K. Rathore, “A novel single-reference six-pulsemodulation (SRSPWM) technique based interleaved high-frequency
three-phase inverter for fuel cell vehicles,” IEEE Trans. Power Electron.,
vol. 28, no. 12, pp. 5547–5556, Dec. 2013.
Sheldon S. Williamson (S’01–M’06–SM’13)
received the B.E. degree in electrical engineering with high distinction from the University of
Mumbai, India, in 1999, and the M.S. and Ph.D.
degrees (with Honors) in electrical engineering
from the Illinois Institute of Technology, Chicago,
IL, USA, in 2002 and 2006, respectively.
From June 2006 to June 2014, he held a
tenure-track Assistant Professor position, followed by a tenured Associate Professor position, in the Department of Electrical and
Computer Engineering, Concordia University, Montreal, QC, Canada.
He currently holds an Associate Professor position in the Department of
Electrical, Computer, and Software Engineering, University of OntarioInstitute of Technology (UOIT), Oshawa, ON, Canada. His research
interests include transportation electrification, electric energy storage
systems, and automotive power electronics/motor drives.
Akshay K. Rathore (M’05–SM’12) received the
M.S. degree from the Indian Institute of Technology (BHU), Varanasi, India, in 2003 and
the Ph.D. degree from the University of Victoria,
Victoria, BC, Canada, in 2008. He had two
subsequent postdoctoral research appointments with University of Wuppertal, Wuppertal,
Germany, and the University of Illinois at
Chicago, IL, USA.
Since November 2010, he has been an Assistant Professor with the Department of Electrical and Computer Engineering, National University of Singapore,
Singapore.
Dr. Rathore serves as an Associate Editor for the IEEE T RANS ACTIONS ON I NDUSTRY A PPLICATIONS , IEEE T RANSACTIONS ON
T RANSPORTATION E LECTRIFICATION, IEEE J OURNAL OF E MERGING
S ELECTED TOPICS IN P OWER E LECTRONICS, IEEE T RANSACTIONS ON
S USTAINABLE E NERGY, and IET Power Electronics. He is also a recipient of the 2013 IEEE Industry Applications Society Andrew W. Smith
Outstanding Young Member Award and the 2014 Isao Takahashi Power
Electronics Award.
Fariborz Musavi (S’10–M’11–SM’12) received
the B.Sc. degree from Iran University of Science and Technology, Tehran, Iran, the M.Sc.
degree from Concordia University, Montreal,
QC, Canada, and the Ph.D. degree in electrical
engineering with emphasis on power electronics from The University of British Columbia,
Vancouver, BC, Canada.
Since 2001, he has been with several hightech companies including EMS Technologies
Inc., Montreal, QC, Canada; DRS Pivotal Power,
Bedford, NS, Canada; Alpha Technologies, Bellingham, WA, USA, and
Delta-Q Technologies, Vancouver, BC, Canada. He is currently with CUI
Inc., Portland, OR, USA, where he is Director of Engineering and is
engaged in research and development on the simulation, analysis, and
design of point-of-load converters and intermediate bus converters for
telecom, datacom, and networking systems applications. His current
research interests include high-power high-efficiency converter topologies, high-power-factor rectifiers, electric vehicles, and sustainable and
renewable energy sources.