UNIVERSITY OF CALGARY
In-home PLC to DSL Interference Characterization and Mitigation
by
Khaled Ali
A THESIS
SUBMITTED TO THE FACULTY OF GRADUATE STUDIES
IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE
DEGREE OF DOCTOR OF PHILOSOPHY
DEPARTMENT OF ELECTRICAL AND COMPUTER ENGINEERING
CALGARY, ALBERTA
January, 2015
c Khaled Ali 2015
Abstract
It is often advocated that a solution to the problem of mutual interference between
digital subscriber line (DSL) networks and in-home power line communications (PLC)
networks is to prevent the PLC networks from utilizing the DSL spectrum. However,
this solution will render PLC networks inoperable with the introduction of wide-band
DSL technologies like G.fast. As an alternative solution, this thesis proposes to use
the common mode (CM) signal, which contains information about the electromagnetic
interference (EMI), to estimate and subtract the differential mode (DM) PLC-to-DSL
interference from the DM DSL signal.
Since the PLC-to-DSL interference environment within a residential setting has
neither been characterized via measurements nor by a model, a measurement campaign is conducted. A set of 480 measurements are collected, within two residential
test-sites, to characterize the PLC-to-DSL interference environment for two DSL modem installation scenarios. This campaign shows that the PLC-to-DSL interference
channels are frequency selective and vary significantly from room to room within the
house.
Two interference mitigation solutions are proposed in this thesis. The first solution
relies on scheduling the PLC channel access; while, the second solution pre-multiplies
the PLC symbol by the inverse of the DM cross-coupling channel before transmission.
Both solutions utilize adaptive frequency domain interference cancellers (FDICs) that
are insensitive to the non-stationarity of the PLC channel and the frequency selectivity
of the coupling channels. The performances of the proposed solutions are evaluated,
and their effectiveness in mitigating the PLC-to-DSL wide-band EMI is demonstrated
using analysis that incorporates the measured PLC-to-DSL coupling channels.
i
Acknowledgements
First and foremost, I would like to sincerely thank my supervisor, Dr. Geoffrey
Messier, for his patience, guidance, and support. Dr. Messier taught me how to
conduct high calibre research that has practical value. Dr. Messier’s commitment to
excellence and his ability to advice, guide, and encourage his students to reach their
potential is admirable. Working under his supervision made me a better researcher
and significantly improved my communication skills.
Also, I would like to thank my co-supervisor, Dr. Stephen Lai, for his advice and
support. I have been fortunate to have a co-supervisor who cared about my work and
gave me valuable feedback. I also thank the Lai and Messier families for welcoming
me into their homes to perform channel measurements.
I would like to thank my thesis committee, Dr. John Nielsen, Dr. Abu Sesay,
Dr. Kyle O’Keefe, and Dr. Michael McGuire for their valuable time and advice.
Thanks to Lincoln Zhao, Mohamed Gaafar, Michael Wasson and all my colleagues
at the FISH Lab for all their help and support. Many thanks to Mohamed Ammar
Al Masri for the various stimulating discussions and coffee runs. Thanks to Ms. Ella
Lok and the staff at the ECE department. Many thanks to Dr. Rainer Iraschko and
the staff at TRTech.
Words cannot express my gratitude to my parents, who have been very supportive
through out my life. I would have not been able to make it this far, if it were not for
their kindness, patience, forgiveness, and advice. In addition, I would like to thank
my brothers, Mohamed and Sameh, for their understanding and continuous support.
Finally, special thanks to my wife, Noha, and my father-in-law and mother-in law.
This work was financially supported by TRTech and the Natural Sciences and
Engineering Research Council (NSERC) of Canada.
ii
Dedication
To my loving parents: Mustafa and Magda.
iii
Table of Contents
Abstract . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Acknowledgements . . . . . . . . . . . . . . . . . . . . . . . . . .
Dedication . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Table of Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . .
List of Tables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
List of Figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
List of Symbols . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1
INTRODUCTION . . . . . . . . . . . . . . . . . . . . . . . .
1.1 Broadband Residential Internet Access Technology . . . . . .
1.1.1 HFC Broadband Access . . . . . . . . . . . . . . . . .
1.1.2 DSL Broadband Access . . . . . . . . . . . . . . . . . .
1.1.3 PLC Broadband Access . . . . . . . . . . . . . . . . .
1.2 PLC In-Home Networking . . . . . . . . . . . . . . . . . . . .
1.3 DSL and PLC Coexistence . . . . . . . . . . . . . . . . . . . .
1.4 Related Work on DSL and PLC Coexistence . . . . . . . . . .
1.4.1 DM PLC-to-DSL Channel Measurements . . . . . . . .
1.4.2 DSL Electromagnetic Interference Mitigation Solutions
1.4.2.1 Narrow-band EMI Mitigation Solutions . . . .
1.4.2.2 Wide-band EMI Mitigation Solutions . . . . .
1.5 Thesis Contribution . . . . . . . . . . . . . . . . . . . . . . . .
1.5.1 Cross-Coupling Channel Characterization . . . . . . .
1.5.2 Interference Cancellation . . . . . . . . . . . . . . . . .
1.6 Thesis Outline . . . . . . . . . . . . . . . . . . . . . . . . . . .
2
THE DSL and PLC TECHNOLOGIES . . . . . . . . . . . . .
2.1 DSL Technology . . . . . . . . . . . . . . . . . . . . . . . . . .
2.1.1 The Physical Network . . . . . . . . . . . . . . . . . .
2.1.1.1 Fiber to the Exchange . . . . . . . . . . . . .
2.1.1.2 Fiber to the Cabinet . . . . . . . . . . . . . .
2.1.1.3 Fiber to the Premise . . . . . . . . . . . . .
2.1.2 The DSL Families . . . . . . . . . . . . . . . . . . . . .
2.1.2.1 Basic Rate Interface . . . . . . . . . . . . . .
2.1.2.2 The High Bit-rate DSL Family . . . . . . . .
2.1.2.3 The Asymmetric DSL Family . . . . . . . . .
2.1.2.4 The Very High Speed DSL Family . . . . . .
2.1.2.5 Fast Access to Subscriber Terminal . . . . . .
2.1.2.6 DSL Families Comparison . . . . . . . . . . .
2.1.3 DSL Signalling and Frame Structure . . . . . . . . . .
2.1.3.1 Discrete Multi-Tone Modulation . . . . . . .
2.1.3.2 DSL Signalling . . . . . . . . . . . . . . . . .
2.1.3.3 DSL Frame Structure . . . . . . . . . . . . .
2.1.4 The DSL Interference Environment . . . . . . . . . . .
2.1.4.1 Intrinsic Interference . . . . . . . . . . . . . .
iv
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ii
iii
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vii
viii
x
1
1
3
3
4
5
6
8
9
10
10
11
14
14
15
16
18
18
19
20
20
20
21
21
22
22
24
25
25
26
27
33
34
35
35
2.2
2.3
3
3.1
3.2
3.3
3.4
4
4.1
4.2
5
5.1
5.2
5.3
5.4
2.1.4.2 Extrinsic Interference . . . . . . . . . . . . .
PLC Technology . . . . . . . . . . . . . . . . . . . . . . . . .
2.2.1 PLC In-home Network . . . . . . . . . . . . . . . . . .
2.2.2 PLC Signalling and Channel Access . . . . . . . . . . .
2.2.2.1 PLC Signalling and Modulation . . . . . . . .
2.2.2.2 MAC for Broadband PLC . . . . . . . . . . .
PLC Noise Environment . . . . . . . . . . . . . . . . . . . . .
CROSS-COUPLING CHANNEL MEASUREMENTS . . . . .
DSL Modem Installation Scenarios . . . . . . . . . . . . . . .
DSL and PLC Interference Environment . . . . . . . . . . . .
Measurement Methodology . . . . . . . . . . . . . . . . . . . .
3.3.1 Measurement Hardware . . . . . . . . . . . . . . . . .
3.3.2 Calibration . . . . . . . . . . . . . . . . . . . . . . . .
3.3.3 Measurement Setup . . . . . . . . . . . . . . . . . . . .
Measurement Campaign . . . . . . . . . . . . . . . . . . . . .
3.4.1 Test-Sites . . . . . . . . . . . . . . . . . . . . . . . . .
3.4.1.1 Case Study A . . . . . . . . . . . . . . . . . .
3.4.1.2 Case Study B . . . . . . . . . . . . . . . . . .
3.4.2 Results . . . . . . . . . . . . . . . . . . . . . . . . . . .
3.4.2.1 Cross-Coupling Channel Frequency Responses
3.4.2.2 Stationarity of Cross-Coupling Channels . . .
3.4.2.3 Effect of Spatial Separation . . . . . . . . . .
INTERFERENCE SYSTEM MODEL . . . . . . . . . . . . .
Current System Model . . . . . . . . . . . . . . . . . . . . . .
Effect of Mutual DSL and PLC Interference on Bit Rates . . .
4.2.1 Effect of PLC Interference on DSL Bit Rates . . . . . .
4.2.2 Effect of DSL Interference on PLC Bit Rates . . . . . .
INTERFERENCE MITIGATION SOLUTIONS . . . . . . . .
Modified System Model . . . . . . . . . . . . . . . . . . . . . .
5.1.1 Proposed System Model . . . . . . . . . . . . . . . . .
5.1.2 Variations in the DM to CM Estimated Ratio . . . . .
5.1.3 Integration of the FDIC . . . . . . . . . . . . . . . . .
Interference Mitigation Block Diagram . . . . . . . . . . . . .
Scheduling-Based Interference Mitigation Solution . . . . . . .
5.3.1 Medium Access . . . . . . . . . . . . . . . . . . . . . .
5.3.2 Cancellation Algorithm . . . . . . . . . . . . . . . . . .
5.3.2.1 C-DSL Training . . . . . . . . . . . . . . . . .
5.3.2.2 C-PLC Training . . . . . . . . . . . . . . . .
5.3.3 Performance Evaluation . . . . . . . . . . . . . . . . .
5.3.3.1 Mean Square Error Analysis . . . . . . . . . .
5.3.3.2 Improvement in Bit Rate . . . . . . . . . . .
Pre-Distortion-Based Interference Mitigation Solution . . . . .
5.4.1 PLC Symbol Pre-distortion . . . . . . . . . . . . . . .
5.4.2 Cancellation Algorithm . . . . . . . . . . . . . . . . . .
5.4.2.1 C-DSL Training . . . . . . . . . . . . . . . . .
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38
39
39
42
42
44
46
48
49
51
53
53
55
57
59
60
60
61
63
63
69
72
75
75
77
77
82
87
87
88
91
92
93
98
99
101
102
104
105
106
109
112
115
116
117
5.4.3
Performance Analysis . . . . . . . . . . . . . . .
5.4.3.1 Training Phase . . . . . . . . . . . . .
5.4.3.2 Transmission Phase . . . . . . . . . .
5.5 Comparison with Spectral Management Solutions . . .
6
CONCLUSION . . . . . . . . . . . . . . . . . . . . . .
6.1 Measurement Campaign Findings . . . . . . . . . . . .
6.2 Interference Mitigation Solutions . . . . . . . . . . . .
6.3 Recommendation for Future Research . . . . . . . . . .
6.3.1 Interference Channel Characterization . . . . .
6.3.1.1 PLC-to-DSL Cross-Coupling Channels
6.3.1.2 DM and CM DSL Direct Channels . .
6.3.2 Interference Cancellation . . . . . . . . . . . . .
Bibliography . . . . . . . . . . . . . . . . . . . . . . . . . .
vi
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119
121
123
126
133
134
135
136
136
136
137
138
140
List of Tables
2.1
Comparison of DSL Families. . . . . . . . . . . . . . . . . . . . . . .
26
3.1
Average coherence bandwidth in kHz. . . . . . . . . . . . . . . . . . .
73
5.1
5.2
5.3
5.4
Latency: Scheduling-based solution. . . . . . . . . . . . . . . . .
Training phase available bit rates. . . . . . . . . . . . . . . . . .
Transmission phase available bit rates. . . . . . . . . . . . . . .
Latency: Scheduling-based versus pre-distortion-based solutions.
vii
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101
123
124
126
List of Figures and Illustrations
2.1
2.2
2.3
2.4
2.5
2.6
2.7
2.8
2.9
2.10
2.11
2.12
Conventional DSL network. . . . . . . . . . . .
Sinc functions at various sub-carrier indexes . .
DMT transmitter block diagram. . . . . . . . .
DMT receiver block diagram. . . . . . . . . . .
DM and CM signal at the receiver. . . . . . . .
DSL super-frame structure. . . . . . . . . . . .
Near end crosstalk among twisted pairs. . . . .
Far end crosstalk among twisted pairs. . . . . .
Electrical wiring in north America [1]. . . . . .
In-home PLC network. . . . . . . . . . . . . . .
Simplified PLC coupling circuit. . . . . . . . . .
PLC frame structure for a 3-user PLC network.
3.1
3.2
3.3
3.4
3.5
3.6
3.7
3.8
3.9
3.10
3.11
3.12
3.13
3.14
3.15
3.16
3.17
3.18
Desk Modem Scenario. . . . . . . . . . . . . . . . . . . . . . . . . . .
Entry Point Scenario. . . . . . . . . . . . . . . . . . . . . . . . . . . .
Co-located DSL and PLC networks interference environment. . . . . .
North Hills 0320BF Balun. . . . . . . . . . . . . . . . . . . . . . . . .
Northern Microdesign PLC Coupler. . . . . . . . . . . . . . . . . . .
Insertion loss of the balun and the PLC coupler before calibration. . .
Modified Balun. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Modified PLC Coupler. . . . . . . . . . . . . . . . . . . . . . . . . . .
Right to Left: Open, Short, and Load (100 Ω). . . . . . . . . . . . . .
Through calibration. . . . . . . . . . . . . . . . . . . . . . . . . . . .
Setup for PLC-to-DSL coupling. . . . . . . . . . . . . . . . . . . . . .
GPIB controlled measurement setup. . . . . . . . . . . . . . . . . . .
Case Study A: test-site floor plan. . . . . . . . . . . . . . . . . . . . .
Case Study B: test-site floor plan. . . . . . . . . . . . . . . . . . . . .
Coupling in Differential Mode. . . . . . . . . . . . . . . . . . . . . . .
Coupling in Common Mode. . . . . . . . . . . . . . . . . . . . . . . .
Common mode to differential mode transfer function (C2DTF). . . .
Variation in DM cross-coupling channels from one power outlet to another. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Entry Point Scenario coupling gain probability density function. . . .
Desk Modem Scenario coupling gain probability density function. . .
Mean DM and CM PLC-to-DSL coupling. . . . . . . . . . . . . . . .
Desk Modem Scenario.: DM PLC-to-DSL coupling for Room A Plug
1 over measurement interval. . . . . . . . . . . . . . . . . . . . . . . .
Entry Point Scenario.: DM PLC-to-DSL coupling for Room A Plug 1
over measurement interval. . . . . . . . . . . . . . . . . . . . . . . . .
Desk Modem Scenario: Room A Plug 1 DM vs. CM PLC-to-DSL
coupling. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Entry Point Scenario: Room A Plug 1 DM vs. CM PLC-to-DSL coupling.
3.20
3.19
3.21
3.22
3.23
3.24
3.25
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19
28
29
32
33
34
35
36
40
41
42
45
49
50
52
54
54
55
56
56
57
57
58
59
61
62
64
65
66
67
67
68
68
69
70
70
71
3.26 Effect of relative distance between DSL and PLC modems on PLC-toDSL coupling. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
3.27 Effect of relative distance between DSL and PLC modems on crosscoupling channels coherence bandwidth. . . . . . . . . . . . . . . . .
74
4.1
4.2
4.3
4.4
4.5
4.6
4.7
4.8
4.9
Current system model. . . . . . . . . . . . . . . . .
Received DSL Signal versus PLC interference. . . .
US and DS frequencies for band plan ”998E30” [2].
Available DSL bit rates. . . . . . . . . . . . . . . .
Degradation in DSL bit rates. . . . . . . . . . . . .
Received PLC Signal versus DS DSL interference. .
Received PLC Signal versus US DSL interference. .
Available PLC bit rates. . . . . . . . . . . . . . . .
Degradation in PLC bit rates. . . . . . . . . . . . .
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76
78
79
80
81
82
84
85
85
5.1
5.2
5.3
5.4
5.5
Proposed system model. . . . . . . . . . . . . . . . . . . . . . . . . .
Interference coupling on twisted-pair. . . . . . . . . . . . . . . . . . .
Integration of the FDIC into a DMT transceiver. . . . . . . . . . . .
Interference cancelling scheme block diagram. . . . . . . . . . . . . .
Desk Modem Scenario: MSE of the proposed scheme versus the relative
distance between the DSL and PLC modems. . . . . . . . . . . . . .
Entry Point Scenario: MSE of the proposed scheme versus the relative
distance between the DSL and PLC modems. . . . . . . . . . . . . .
Desk Modem Scenario: Achieved improvement in bit rates versus the
Euclidean distance between the DSL and PLC modems. . . . . . . . .
Entry Point Scenario: Achieved improvement in bit rates versus the
Euclidean distance between the DSL and PLC modems. . . . . . . . .
Achieved total DSL bit rates for DS transmission, for various DSL
cable run length. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Desk Modem Scenario: Achieved bit rates for DSL-DS. . . . . . . . .
Entry Point Scenario: Achieved bit rates for DSL-DS. . . . . . . . . .
Desk Modem Scenario: Achieved bit rates for PLC. . . . . . . . . . .
Entry Point Scenario: Achieved bit rates for PLC. . . . . . . . . . . .
Desk Modem Scenario: Achieved bit rates for DSL-US. . . . . . . . .
Entry Point Scenario: Achieved bit rates for DSL-US. . . . . . . . . .
89
91
92
94
5.6
5.7
5.8
5.9
5.10
5.11
5.12
5.13
5.14
5.15
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109
110
112
113
125
127
129
129
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131
131
List of Symbols and Abbreviations
η c,DSL
DSL CM AWGN matrix
η c,PLC
PLC CM AWGN matrix
η d,DSL
DSL DM AWGN matrix
η d,PLC
DSL DM AWGN matrix
γ0
SNR in presence of AWGN only
γF
SINR after utilizing C-DSL
γW
SINR after utilizing Weiner filter
γ W/O
SINR before interference mitigation
φ
MSE of FDIC
φF
MSE of C-DSL
φTr
MSE of C-DSL during training period of the pre-distortion solution
φTx
MSE of C-DSL during transmission period of the pre-distortion solution
φW
MSE of Weiner filter
∆f
Sub-carrier spacing
∆fDSL
DSL sub-carrier spacing
∆fPLC
PLC sub-carrier spacing
Γ
SINR gap
x
γ(i)
SINR for frequency bin i
λ
Normalization factor
bd,DSL
v
Estimated of DM PLC interference matrix
bd,PLC
v
Estimated of DM DSL interference matrix
CDSL
DSL FDIC coefficient matrix
CW
Wiener filter coefficient matrix
CPLC
PLC FDIC coefficient matrix
HDSL
Direct DSL channel frequency response matrix
HPLC
Direct PLC channel frequency response matrix
DSL
H∆f
c,DTx−PRx
CM interference channel frequency response matrix between the DSL
receiver and a PLC transmitter, sampled at integer multiples of ∆fDSL
DSL
H∆f
c,PTx−DRx
CM interference channel frequency response matrix between a PLC
transmitter and the DSL receiver, sampled at integer multiples of
∆fDSL
DSL
H∆f
d,DTx−PRx
DM interference channel frequency response matrix between the DSL
receiver and a PLC transmitter, sampled at integer multiples of ∆fDSL
DSL
H∆f
d,PTx−DRx
DM interference channel frequency response matrix between a PLC
transmitter and the DSL receiver, sampled at integer multiples of
∆fDSL
PLC
H∆f
d,PTx−DRx
DM interference channel frequency response matrix between a PLC
transmitter and the DSL receiver, sampled at integer multiples of
∆fPLC
xi
q
Transmitted PLC signal matrix
rc,DSL
Received CM DSL signal matrix
rc,PLC
Received CM PLC signal matrix
rd,DSL
Received DM DSL signal matrix
rd,PLC
Received DM PLC signal matrix
u
Generated PLC symbol
vc,DSL
CM DSL interference matrix
vc,PLC
CM PLC interference matrix
vd,DSL
DM DSL interference matrix
vd,PLC
DM PLC interference matrix
x
Transmitted DSL signal matrix
y
Desired DSL signal matrix
z
Desired PLC signal matrix
(...)
Average of a variable
εD
DSL maximum transmission power
εP
PLC maximum transmission power
b
y
Estimate of desired DSL signal matrix
b
z
Estimate of desired PLC signal matrix
d
(...)
Estimate of a variable
xii
b(i)
Number of bits for frequency bin i
bmax
Maximum number of bits allowed per frequency bin
M
Number of DSL sub-carriers
N
Number of PLC sub-carriers
N0,c
PSD of CM AWGN
N0,d
PSD of DM AWGN
No
PSD of AWGN
R
Total number of bits
R0
Total bit rates in presence of AWGN only
RBound
Total number of bits in presence of background noise
RF
Total bit rates after utilizing C-DSL
RInterference
Total number of bits in presence of interference plus background noise
RW/O
Total bit rates before interference cancellation
RW
Total bit rates after utilizing Weiner filter
ti
Start time of Slot i
TP
PLC frame duration
TSF
DSL super-frame duration
TS
DSL frame duration
(...)∗
Complex conjugate of a variable
xiii
E[...]
Expectation of a variable
Loss %
Bit rates loss percentage
a
Normalization factor
H(i, j)
ith row and j th column of the matrix H
x(i)
ith row of the vector x
A/D
Digital to analog
ADSL
Asymmetric DSL
ADSL2
Asymmetric DSL 2
ADSL2+
Asymmetric DSL 2 plus
AM
Amplitude modulation
AMR
Automatic meter reading
AWGN
Additive white Gaussian noise
BNC
Baby N Connector
BRI
Basic rate interface
C-DSL
DSL FDIC
C-PLC
PLC FDIC
C2DTF
CM to DM transfer function
CB
Coheren Bandwidth
CM
Common mode
xiv
D/A
Digital to analog
DFT
Discrete Fourier Transform
DM
Differential mode
DMT
Discrete multi-tone
DOCSIS
Data over cable service Interface specification
DS
Downstream
DSL
Digital subscriber line
EMI
Electromagnetic interference
FDD
Frequency division duplexing
FDIC
Frequency domain interference canceller
FFT
Fast Fourier transform
FT
Fourier transform
FTTB
Fiber to the building
FTTC
Fiber to the curb
FTTCab
Fiber to the cabinet
FTTDp
Fiber to the distribution point
FTTEx
Fiber to the exchange
FTTH
Fiber to the home
FTTP
Fiber to the premise
xv
G.fast
Fast access to subscriber terminals
GPIB
General purpose interface bus
HDSL
High bit-rate DSL
HFC
Hybrid fibre-coaxial cable
HFTP
Hybrid fiber twisted pair
IDFT
Inverse discrete Fourier Transform
IFFT
Inverse fast Fourier Transform
ISP
Internet service provider
ITU-T
International telecommunication union-telecommunication standardization sector
LAN
Local area network
LPF
Low pass filter
OFDM
Orthogonal frequency division multiplexing
ON
Optical node
PAM
Pulse amplitude modulation
PDF
Probability density function
PLC
Power line communications
PLC-DC
PLC domain controller
PSD
Power spectral density
PSTN
Public service telephone network
xvi
QAM
Quadrature amplitude modulated
QAM
Quadrature amplitude modulation
RJ
Registered jack
SHDSL
Single-pair high-speed DSL
SMA
Sub-miniature version A
TB
Terminal block
TDD
Time division duplexing
US
Upstream
VDSL
Very high speed DSL
VDSL2
Very high speed DSL 2
VNA
Vector network analyzer
xTalk
Crosstalk
xvii
Chapter 1
INTRODUCTION
Recent advances in power line communications (PLC) have made it popular for inhome networking. This makes PLC an increasingly relevant source of interference
for digital subscriber line (DSL) networks within the home environment. This thesis
presents two measurement case studies that characterize the PLC-to-DSL coupling
channels, within a residential setting. In addition, this thesis proposes two interference
mitigation solutions that enhance the coexistence of in-home PLC and DSL networks.
The rest of this chapter is organized as follows. Broadband access technology for
residential Internet is introduced in Section 1.1, while home networking solutions via
PLC technology is discussed in Section 1.2. The coexistence environment between
in-home PLC networks and DSL broadband access networks are discussed in Section
1.3, while background information on topics relevant to the PLC and DSL interference
characterization and mitigation are discussed in Section 1.4. The thesis contributions
are highlighted in Section 1.5, while the outline of the thesis is presented in Section
1.6.
1.1 Broadband Residential Internet Access Technology
Broadband access technology is an integral part of the world wide communication
network. The term broadband refers to any access technology that is always-on
and can support multiple services at high data rates. Residential broadband access
has become a pillar of the world culture. Throughout the developed world, and in
some developing countries, access to the Internet is considered one of the household
essentials due to the various services that can be provided through it, along with the
1
reduction in cost of having it.
The ultimate in broadband access technology, in terms of data rates and reliability,
is optical fiber. However, due to the cost of installation and replacement of existing
access technologies, creating a world-wide communications network via optical fiber
is not feasible. Currently, hybrid optical fiber connections, where the link to any
particular subscriber runs partly on optical fiber cables and then partly on copperbased wires, are utilized to form hybrid networks that serve as the backbone of the
world wide communications network.
Residential broadband access technologies can be classified into two categories:
wired (or fixed line) and wireless broadband access technologies. Fixed line broadband access technologies utilize a physical connection between the service provider
and the customer; thus, a physical wired network has to be in place before the nodes
in the network can communicate. Wireless broadband access, on the other hand,
utilize air as the transmission medium. Wireless broadband solutions provide, along
with freedom of mobility, instantaneous wide area coverage that is especially beneficial for remote areas where the infrastructure for fixed line technologies do not
reach. However, wired broadband access technologies have several advantages over
the wireless technologies such as higher data rates and better reliability.
The most commonly used copper-based wired broadband access technologies are
digital subscriber lines (DSL) and hybrid fiber-coaxial cable (HFC). DSL utilizes the
copper twisted-pairs of the public service telephone network (PSTN) as the transmission medium, while HFC utilizes the coaxial cables used by the digital cable television
(CATV) network as the transmission medium. Both DSL and cable modems technologies incorporate hybrid fibre optic connections, where portions of the twisted-pairs or
the coaxial cables within the networks are replaced with fiber optic cables. Another
copper-based fixed line broadband access technology is power line communications
2
(PLC). PLC utilizes the power lines that are used to carry electricity to subscriber
houses as a transmission medium. The aforementioned copper-based fixed line broadband access technologies are discussed in Sections 1.1.1 to 1.1.3.
1.1.1 HFC Broadband Access
The cable television (CATV) network was created for broadband unidirectional transmission. Various optical nodes (ONs) are connected to the head unit via fiber cables,
creating HFC connections. Out of each ON, branches of shared coaxial copper cable
connect various customers. Thus a tree is formed, where the root of the tree is the
head-end [3]. Note that up to 500 customers can be serviced by a single ON. To
enable broadband access to the Internet, the CATV network is upgraded to support
bidirectional traffic.
HFC relies on dividing the bandwidth of the shared coaxial cable into non overlapping channels, where one or more of these channels are dedicated for upstream
transmission while the rest of the channels are reserved for downstream transmission
and cable TV. The most recent version of the cable modems protocol Data Over Cable Service Interface Specification (DOCSIS), which is the most widely used protocol
for HFC broadband access, provides data rates of up to 400 Mbits/s [3]. However, the
major drawback of HFC broadband access is that the cable TV network is a shared
tree network, and the bandwidth per customer is limited. The available bandwidth
per customer is dependent on the number of active users and poses a significant challenge for upstream data rates when a large number of users are active. In addition,
due to the structure of the tree, security and scalability are always a challenge.
1.1.2 DSL Broadband Access
DSL technology utilizes dedicated link for each customer, where data is transmitted
over the public service telephone network (PSTN). A PSTN utilizes twisted-pairs
3
of copper wires that were originally designed to serve as a medium for transmitting
speech signals. Since human speech is in the range of 300 Hz to 3400 Hz, higher
frequencies can be used for transmitting data over existing PSTNs [4]. The ubiquity
of the PSTN has motivated new DSL technologies that increase throughput using a
combination of advanced communications techniques and replacing the twisted-pairs
in portions of the PSTN by fibre optic cables [5]. These hybrid connections shorten
the distance the DSL signal has to travel over passive copper twisted-pairs between
the central office and the end users, and results in a wider bandwidth channel.
An advanced standardized DSL technology is the very high bit rate digital subscriber line 2 (VDSL2). VDSL2 utilizes up to 30 MHz bandwidth [2] and vectoring [6]
to achieve transmission rates up to or exceeding 100 Mbits/s. Currently, a standard
is being developed for a new DSL technology, called G.fast (fast access to subscriber
terminal) [7], that has the potential of achieving data rates of 1 Gbits/s, over a spectrum that spans 106 or 212 MHz [8]. These fast data rates being dedicated to each
customer, unlike HFC, have made DSL a popular choice among end users for residential broadband access. The DSL technology is discussed in more detail in Section
2.1.
1.1.3 PLC Broadband Access
Extensive research on utilizing PLC networks to deliver Internet to customer houses
has been performed. However, one of the main obstacles that prevented utilizing
power lines by Internet service providers (ISPs) is the need for a repeater at each
transformer, since data signal cannot pass through the transformers [3]. Installing
repeaters at each transformer, especially in North America where each transformer
serves only few houses, was proved costly. This high cost and the presence of other
telecommunication infrastructure prevented the concept of providing Internet over
power lines to materialize.
4
1.2 PLC In-Home Networking
Utilizing legacy wires, such as power lines, telephone cables, and coaxial cables, to
distribute data within the home environment has no rewiring requirement. Various
standards have been developed for the three aforementioned legacy wires to be used for
residential data distribution. Among these standards, the international telecommunication union-telecommunication standardization sector (ITU-T) home networking
standard G.hn [9] has specifications for each of the three legacy wires networks, along
with multi-domain specifications. For the multi-domain specifications, two or more
of the legacy wires networks can be utilized simultaneously to deliver data.
Among the three legacy wires, power lines are the most extensively used in houses.
This coupled with the recent advances in the area of networking within the home using
legacy wires make PLC networks increasingly common within the home. The power
lines can be utilized to form a network that transform the house into a smart home,
where appliances within the house are connected.
Creating an in-home local area network (LAN) via PLC technology has many
advantages over other in-home networking solutions such as Ethernet and wireless.
The first advantage is the low implementation cost which is due to the existence of a
pre-installed power line network in each house. Thus, no new wiring or physical installation is required, which is not the case with Ethernet LANs. A second advantage
that in-home PLC LANs has over Ethernet LANs is the presence of multiple access
nodes (in form of electrical outlets), which are spread throughout the house.
Currently, the average cost of a PLC network card is the same as the cost of a
wireless network card; however, as in-home networking via PLC gains popularity with
end users, mass production of PLC networks cards will cost about half the cost of the
wireless network card because PLC network cards do not require an RF component
[10]. Additionally, data rates over PLC networks can reach up to 200 Mbits/s, with
5
the ability of integrating multiple functions over the same network. For instance,
automatic meter reading (AMR), home automation, and triple play services (Internet,
television, and telephone services) can be integrated over an in-home PLC network
simultaneously [11].
All the aforementioned advantages make in-home PLC networks a suitable choice
for in-home networking [3] and, in cases where one medium is not sufficient, is a
cost-effective complement to other in-home technologies [12]. Further detail on the
PLC technology is presented in Section 2.2.
1.3 DSL and PLC Coexistence
In-home PLC networks operate over the same spectrum as DSL networks. This
increases the likelihood of crosstalk between PLC and DSL communications systems.
For instance, two home networks that operate at the same frequency range, one over
copper twisted-pairs (138 kHz - 30 MHz [2]) and the other over power lines (1.8
MHz - 30 MHz [9]), would interfere with each other. The DSL and PLC interference
environment is discussed in more detail in Section 3.2.
Communication standards such as ITU G.hn [9], have been developed with mechanisms that prevent any interference between various systems within the home environment. However, the interference from PLC-to-DSL is usually prevented by forcing the
PLC network to notch frequencies that affect the DSL signal. While this is a viable
solution, denying PLC networks access to the DSL spectrum will render in-home PLC
networks inoperable especially if VDSL2 or G.fast technology is employed. Further
detail on mitigating the PLC-to-DSL interference via spectral notching is discussed
in Section 1.4.2.2.1.
Other PLC-to-DSL interference reduction solutions, such as reducing the PLC
transmit power (which is known as spectral management) or increasing the distance
6
between the DSL and PLC modems (which is referred to as spatial separation), have
been proposed in the literature. As will be discussed in Section 1.4.2.2.2, spectral
management degrades the performance of the PLC network, and with the increase
in the usable DSL bandwidth, spectral management solutions will render the PLC
networks inoperable. From the measurement case studies presented in this thesis and
in [13, 14], it is shown that the interference levels between the PLC and DSL modems
do not depend on the distance separating the modems. Thus, spatial separation,
which is discussed in more detail in Section 1.4.2.2.4, does not mitigate the PLC-toDSL interference.
The goal of this thesis is to enhance the co-existence environment between DSL
and PLC networks in a residential environment. The objective is to mitigate the PLCto-DSL interference without hindering the performance of the in-home PLC network.
Ultimately, this will lead to the increase of data transmission rates in both DSL and
PLC networks.
Complementary signals are inserted onto each of the wires of the twisted-pair by
the transmitter. At the receiver, the difference between the complementary signals
is the differential mode (DM) signal, while the common mode (CM) signal is the
arithmetic mean of the complementary signals. The DM signal contains the desired
signal, while the CM signal is mainly composed of the interference. Further detail on
the DM and CM signalling in DSL systems is discussed in Section 2.1.3.2.
The main hypothesis of this thesis is that the common mode signal contains information about the differential mode interference. By utilizing the common mode
signal, the differential mode interference can be estimated and subtracted from the
differential mode signal. Removing the differential mode interference increases transmission rates in DSL networks without limiting the capacity of the in-home PLC
network. The common mode and differential mode signalling are discussed in more
7
detail in Section 2.1.3, while estimating the DM interference from the CM signal is
discussed in Chapter 5.
A thorough literature survey indicates that the PLC-to-DSL cross-coupling channel measurements in a residential setting that included both DM and CM reception
methods have never been performed [13]. Only DM PLC-to-DSL channel measurements in lab settings have been performed in the literature; further detail on work
done to date with regards to PLC-to-DSL cross-coupling channel measurements can
be found in Section 1.4.1. Characterizing the PLC-to-DSL cross-coupling channels is
central to the main hypothesis and the interference mitigation solutions proposed in
this thesis. Thus, a measurement campaign that studies the cross-coupling channels
between DSL and in-home PLC networks has to be performed.
The focus of this thesis is divided into two areas. The first focus area is PLC-toDSL cross-coupling channel characterization in a residential setting. A measurement
campaign that studies the PLC-to-DSL interference environment within residential
test sites is performed. The second focus area is interference cancellation. Two
interference mitigation solutions based on adaptive filter theory are proposed. The
effectiveness of the proposed solutions in mitigating wide-band EMI from an in-home
PLC network on to the DSL system, based on field measurements, is evaluated.
1.4 Related Work on DSL and PLC Coexistence
In this section, work related to the two focus areas of the thesis is discussed. Related
work to the PLC-to-DSL coupling channel measurements is presented in Section 1.4.1,
while work related to the PLC-to-DSL interference mitigation is given in Section 1.4.2.
8
1.4.1 DM PLC-to-DSL Channel Measurements
Measurement case studies that investigate the differential mode PLC-to-DSL coupling
channel in a laboratory environment were performed in [15, 16, 17, 18]. These studies
investigated the effects of various factors, such as the distance separating the PLC
and DSL cables and the shared length between the PLC and DSL cables, on the PLCto-DSL coupling over the VDSL2 spectrum. The main finding of these studies is the
potential for significant crosstalk between PLC and DSL systems. However, these
studies do not characterize the PLC-to-DSL coupling channel in an actual residence,
nor do they consider the CM PLC-to-DSL coupling channel.
The coupling between PLC and DSL systems over the frequency range of 138 kHz
to 17.664 MHz (band plan ”998ADE17”) was measured in [19]. The objective of this
study was to investigate the quality of service for Internet protocol television (IPTV).
Two case studies, [20] and [21], investigated the effect of PLC on the throughput of a
VDSL modem. The DM PLC-to-DSL coupling channels were neither measured in [20]
nor in [21]. Rather the effect of the PLC system on the VDSL modem was simulated
by inducing a PLC signal into a co-axial cable with variable attenuators. The main
finding of [20] and [21] is that the VDSL throughput experiences degradation when
the ratio of the received VDSL signal power to the PLC signal power is less than 20
dB. In other words, as long as the DSL cable run length is less than 300 m, the VDSL
throughput will not be hindered. Note that in [19] the DM PLC-to-DSL coupling
channels up to only 17.664 MHz was measured, while in [20] and [21] no channel
measurements were performed. In addition, the CM PLC-to-DSL coupling channel
was not studied in [19, 20, 21].
It is important to note that the studies discussed above did not measure the
PLC-to-DSL coupling channels, within a residential setting. Knowledge about the
coupling channels between in-home PLC and DSL networks is essential in determining
9
an interference mitigation solution. Since there is neither a model nor previously
conducted field measurements that characterize the interference environment between
in-home PLC networks and a DSL modem, further field measurements of the DM and
CM PLC-to-DSL coupling channels are required.
1.4.2 DSL Electromagnetic Interference Mitigation Solutions
The copper twisted-pairs, utilized as the transmission medium by the DSL networks,
were not initially designed to transmit broadband signals over a wide spectrum. As
with all copper based wires, the twisted-pairs turn into an antenna at high frequencies
[22]. This antenna radiates a portion of the transmitted signal as electromagnetic
waves which cause interference to other systems operating at same frequencies. This
radiated electromagnetic interference (EMI) is known as EMI egress. The copper
twisted-pairs also pick up the radiated electromagnetic waves from other systems,
and this causes what is known as EMI ingress.
EMI to DSL can be classified as narrow-band EMI or wide-band EMI interference. Narrow-band EMI, as the name suggests, typically affects few of the DSL
sub-channels. The most common narrow-band EMI to DSL systems are from amplitude modulation (AM) radio and amateur radio. Wide-band EMI, however, causes
interference to a large portion of the DSL spectrum, which is usually from co-located
systems that operate over the same spectrum as DSL. The most common wide-band
EMI to DSL systems is from a co-located in-home PLC network [12, 23]. Techniques
used to mitigate narrow-band and wide-band EMI to DSL systems are discussed in
Sections 1.4.2.1 and 1.4.2.2 respectively.
1.4.2.1 Narrow-band EMI Mitigation Solutions
Narrow-band EMI mitigation can be performed at the analog front of the DSL modem
or within the digital structure. No optimal solution exists, but rather the choice
10
depends on each situation [22]. Digital narrow-band EMI mitigation solutions are
usually preferred to analog solutions, since digital solutions are less expensive and
more flexible than analog ones. However, in certain scenarios an analog solution
would be necessary. For instance, if the EMI levels are high enough to saturate the
analog to digital converter, EMI mitigation has to be performed at the analog front
of the DSL modem. Note that a mixture of analog and digital narrow-band EMI
mitigation solutions are utilized in most scenarios [22].
Various techniques are utilized in the literature to mitigate the narrow-band EMI
effect on DSL signal. These techniques are either active or passive. Note that both
analog and digital narrow-band EMI mitigation solutions can utilize either active
or passive techniques. Passive techniques rely on information about the frequency of
the interference and utilize filters (usually, notch filters) to eliminate the narrow-band
EMI egress. Active techniques, on the other hand, rely on finding a reference signal
or on a priori knowledge of the EMI to mitigate the differential mode narrow-band
EMI egress.
Usually, a reference signal is obtained via satellite or antenna in wireless systems;
however, for DSL this is not required since the common mode signal can be utilized.
Studies that take advantage of the common mode signal to mitigate the effect of
narrow-band EMI were performed in [24, 25]. In [24], the authors mitigate the effect
of impulsive noise on the DSL signal by utilizing the CM signal. Similarly, in [25],
the CM signal was used to reduce the impact of EMI on the DM DSL signal.
1.4.2.2 Wide-band EMI Mitigation Solutions
Wide-band EMI mitigation solutions fall into one of the following four categories:
spectral notching, spectral management, interference cancellation, and spatial separation. In Section 1.4.2.2.1 to 1.4.2.2.4, the work done to date in each of the four
aforementioned wide-band EMI mitigation solution categories is discussed.
11
1.4.2.2.1 Spectral Notching
Home networking standards, such as ITU G.hn [9], ensure that home networks that
operate over the same spectrum as any of the DSL technologies do not interfere with
the functionality of the DSL system. This is achieved by preventing home networks
from utilizing frequencies occupied by the DSL system. Even though preventing home
networks from operating over the DSL spectrum ensures successful operation of the
DSL system, it denies home networks from benefiting from a wide range of frequencies.
In addition, with the emerging of new DSL technologies that utilize wider bandwidth,
such as G.fast (up to 212 MHz), denying PLC networks access to the spectrum that
overlaps with DSL networks will render the PLC network inoperable [26].
1.4.2.2.2 Spectral Management
As an alternative to spectral notching, spectral management is proposed in [27] and
[28] to alleviate the effects of the PLC interference by reducing the transmit power
for PLC sub-carriers that cause interference to the received DSL signal. In [27], the
authors assume a flat PLC-to-DSL cross-coupling channel and study the achieved
DSL bit rates at various PLC transmit power levels. In [28], the authors propose
reducing the transmit power for PLC sub-carriers that interfere with the downstream
frequencies of VDSL2. As will be shown in Chapter 5, the proposed solutions achieve
higher DSL bit rates than both solutions proposed in [27] and [28]. In addition,
it will be shown that, mitigating the effects of PLC interference through spectral
management negatively affects the PLC bit rates, while the proposed solution does
not hinder the PLC bit rates.
1.4.2.2.3 Cross-Coupling Channel Equalization
An interference cancellation solution was proposed in [28], which entails utilizing
adaptive filters to estimate and mitigate the effects of the DM PLC interference on
the DM DSL signal. In [28], the authors propose connecting the PLC network and
12
the DSL network, by utilizing a coupler that carries the PLC signal transmitted on
the power lines to an adaptive filter that is connected to the DSL line.
A fundamental assumption in [28] is that the coupling channel between the PLC
transmitter and DSL receiver is the same as between the DSL transmitter and PLC
receiver. As shown by the measurement campaign presented in Chapter 3, the PLCto-DSL coupling channel varies significantly from one outlet to the other. And the
coupling channel experienced at the PLC receiver outlet will be very different than
the coupling channel at the PLC transmitter outlet. In addition, the authors in [28]
assume that the retraining of the adaptive filter is only required to accommodate the
time-variation of the PLC channel, and they do not account for the outlet-to-outlet
variation experienced by different users.
Another wide-band EMI mitigation solution that utilizes an adaptive filter to
equalize the cross-coupling channel was proposed in [29]. In this study, a time domain
adaptive filter utilizes the CM signal to estimate the DM EMI over the asymmetric
DSL (ADSL) spectrum (0.025 - 1.1 MHz [30]). Because of the nature of the wide-band
interference, this filter requires a long training time [29].
Note that this time domain filter requires a long training time for a DSL network
operating over the ADSL spectrum, which is much smaller than the spectrum utilized
by VDSL, VDSL2, or G.fast technologies. Thus, utilizing a time domain adaptive
filter is not feasible for DSL technologies that operate over a spectrum that span tens
of MHz. In addition, a core assumption made in [29] is that the relationship between
the differential and common mode cross-coupling channels is a smooth function of
frequency. As will be shown in Chapter 3, the ratio of the PLC-to-DSL DM to CM
cross-coupling channels varies significantly from one sub-channel to the other.
1.4.2.2.4 Spatial Separation
The authors in [31] propose that changing the location of the PLC transmitter or
13
the DSL receiver might be utilized to mitigate the interference between PLC and DSL;
since the coupling levels vary from one power outlet to the other, the locations with
the lowest coupling levels should be chosen. However, as will be shown in Chapter
3, the spatial separation between the DSL and the PLC modems does not have a
significant impact on the coupling channels captured in the measurements presented
in this paper. The average coupling levels are relatively the same for all outlets.
1.5 Thesis Contribution
The goal of this thesis is to mitigate the mutual wide-band interference between
co-located DSL and PLC networks, within the home environment. To achieve this
goal, the focus of the thesis is on two areas: cross-coupling channel characterization
and interference cancellation. The thesis contribution to the area of cross-coupling
channel characterization is summarized in Section 1.5.1, while the thesis contribution
to the area of interference cancellation is presented in Section 1.5.2.
1.5.1 Cross-Coupling Channel Characterization
Cross-coupling channel characterization is essential to understand the various interference sources affecting the DSL networks. In this thesis, the cross-coupling channels
between DSL and in-home PLC networks are studied.
Residential field measurements of PLC-to-DSL cross-coupling channels have never
been performed. A measurement campaign, which is discussed in detail in Chapter
3, has been designed and conducted to characterize the PLC-to-DSL cross-coupling
channels. Both the DM and CM PLC-to-DSL cross-coupling channels are studied
within two residential houses. For each of the two houses, the DM and CM PLC-toDSL cross-coupling channels are measured for various rooms.
A set of 480 DM and CM cross-coupling channel measurements between the DSL
14
and PLC networks has been collected for the two DSL modem installation scenarios
discussed in Chapter 2. The main findings of the measurement campaign are:
• A strong DM PLC-to-DSL cross-coupling channel exists, with cross-coupling
channel coupling gains ranging between 40 dB and 80 dB.
• The PLC-to-DSL cross-coupling channels are frequency selective and independent of the proximity between the PLC modem and the DSL modem.
• The PLC-to-DSL cross-coupling channels vary significantly from one outlet to
the other.
• The CM to DM PLC-to-DSL interference ratio is not a smooth function of
frequency, and varies from one outlet to the other.
1.5.2 Interference Cancellation
The interference cancellation portion of this thesis is concerned with how to mitigate the mutual interference between a DSL modem and a co-located in-home PLC
network, without negatively impacting the performance of the PLC network. Two
PLC-to-DSL interference mitigation solutions are presented in Chapter 5. Both solutions utilize adaptive filters to estimate the differential mode interference on a tone
by tone basis using the common mode signal. Only adaptive time domain cancellers
that attempt to mitigate the wide-band EMI suffered by the DM DSL signal have
been researched [29]. Due to the frequency selective nature of the PLC interference,
these time domain adaptive filters cannot mitigate DM PLC interference.
The first interference mitigation solution uses an adaptive frequency domain canceller that utilizes the CM DSL signal to extract information about the DM PLC interference and relies on restricting channel access to one PLC user per DSL super-frame.
This scheduling is performed to combat the variation in cross-coupling channels from
15
one power outlet to the other. While this will increase the latency experienced by the
PLC users, it will be shown that this additional delay is acceptable for multi-media
applications.
The second interference mitigation solution eliminates the need to restrict channel
access to one PLC user per DSL super-frame by pre-multiplying the PLC symbol with
the inverse of the cross-coupling channel. This eliminates the latency caused by the
first solution but will be more expensive to implement due to the additional signal
processing required. Similar to the first solution, the second solution also utilizes an
adaptive frequency domain canceller that estimates the DM PLC-to-DSL interference
from the CM DSL signal.
As will be shown in Chapter 5, both proposed solutions successfully mitigate the
PLC-interference on the DSL network, without degrading the performance of the PLC
network. This is one of the main advantages that the proposed interference mitigation
solutions have over the interference mitigation solutions discussed in Section 1.4.2.
1.6 Thesis Outline
This thesis is organized as follows. In Chapter 2, a brief background on the DSL and
PLC technologies is provided, along with description of the PLC-to-DSL interference
environment within a residential setting.
In Chapter 3, a measurement campaign that characterizes the PLC-to-DSL crosscoupling channels within two test sites is presented. Both the differential and the
common mode cross-coupling channels were measured in two test sites.
In Chapter 4, the current system model is presented. Additionally, the effects of
the mutual wide-band interference on the achieved bit rates for both DSL and PLC
systems are discussed.
In Chapter 5, two interference cancelling solutions are presented. While both
16
solutions utilize adaptive filters to extract an estimate of the DM interference from
the usually ignored CM signal, each solution has its advantages and disadvantages.
The first solution while having the advantage of simplicity, adds restriction on the
PLC channel access per DSL super-frame. This restriction introduces latency, albeit
within the acceptable range for multimedia services. The second solution removes the
restriction on PLC channel access per DSL super-frame; thus, eliminating the added
latency introduced by the first solution. However, the second solution encompasses
pre-shaping the PLC symbols before transmission. This adds to the complexity of
the PLC transmitter, which is translated in the implementation cost.
Finally, in Chapter 6, the thesis is concluded where the main findings of the thesis
are summarized and future research direction is introduced.
17
Chapter 2
THE DSL and PLC TECHNOLOGIES
The objective of this thesis is to enhance the coexistence between DSL and PLC
networks in residential settings. DSL, despite its maturity, is still an active area of
research. Recent advancement in DSL research lead to the development of standards
that promise to provide data rates of up to 1 Gbits/s, which makes DSL the leading
technology in copper-based wired broadband access. Similarly, PLC, because of its
existence within every structure, has generated a lot of research advancement which
has made it a popular choice for in-home networks.
DSL and PLC utilize overlapping spectra; as a result, interference between both
systems occurs. In order to mitigate the interference between co-located PLC and
DSL networks, a better understanding of both networks is required. In this chapter, the DSL and PLC technologies are briefly introduced in Sections 2.1 and 2.2
respectively.
2.1 DSL Technology
In this section, we introduce DSL technology. The layout of the DSL physical network
is presented in Section 2.1.1. The various DSL families are discussed in Section 2.1.2,
and the DSL modulation, frame structure, and signal transmission techniques are
presented in Section 2.1.3. Finally, the DSL interference environment is presented in
Section 2.1.4.
18
From Core Network
FT
TE
x
Street Cabinet
(Pedestal)
TB
FT
FTTCab
Central Office
(Local Exchange)
FT
TH
Street Cabinet
(Pedestal)
Building
(remote DSLAM)
Twisted-pairs
USER
USER
USER
Fiber
Figure 2.1: Conventional DSL network.
2.1.1 The Physical Network
In a DSL network, as shown in Fig. 2.1, cables run from the core network to central
offices (COs) (or local exchanges or access nodes (ANs)). Historically, the cables between the core network and the COs were twisted-pairs. In each of these cables, there
is about 1000 twisted-pairs. From each CO, up to 100 twisted-pairs are connected
to pedestals that service a number of houses. Finally, each house is connected to its
respective pedestal by two twisted-pairs. However, only one of these twisted-pairs is
used in conventional DSL networks. Thus, a house is connected by one passive copper
twisted-pair to the CO. As a result, the DSL signal travels a very long distance which
considerably reduces the transmission bit rate.
Presently, however, sections of the twisted-pairs are being replaced by fiber optic
cables. This connection is referred to as a hybrid fiber twisted pair (HFTP) connec-
19
tion [5]. These hybrid connections shorten the distance the DSL signal has to travel
over passive copper twisted-pairs. HFTP connections have many deployment scenarios, depending on the connection from the CO to the end user. The main HFTP
deployment scenarios are discussed in Sections 2.1.1.1 to 2.1.1.3 [32, 33].
2.1.1.1 Fiber to the Exchange
In the fiber to the exchange (FTTEx) deployment, copper twisted-pairs connect the
CO to cabinets (or pedestals) that service a number of houses. Each house is connected to its respective cabinet by two twisted-pairs. However, only one of these
twisted-pairs is usually active. Thus, in an FTTEx deployment, a house is connected
by one passive copper twisted-pair to the CO. As a result, the DSL signal travels a
very long distance which considerably reduces the transmission rate.
2.1.1.2 Fiber to the Cabinet
In fiber to the cabinet (FTTCab) deployment, fiber cable is utilized to connect the
CO to a remote optical unit (ONU) within the cabinet. Thus, DSL transmission over
copper twisted pairs occurs only between the ONU and the customer premise. The
FTTCab deployment has many variants, such as fiber to the curb (FTTC) or fiber
to the distribution point (FTTDp). The main difference among these variants is the
reach of the fiber connection within the DSL network.
2.1.1.3 Fiber to the Premise
Fiber to the premise (FTTP) deployment has two variants. First is fiber to the
building (FTTB), where a fiber cable connects the CO with a multi-dwelling building.
Once in the building, a remote DSL access multiplexer (DSLAM) manages the DSL
transmission over very short runs of twisted-pairs. The second variant is fiber to the
home (FTTH), where the CO, and consequently the core network, is connected to the
user by a fiber optic cable. FTTH is considered ideal and future proof [5], since a need
20
for data rates higher than the rates achieved by this deployment cannot be envisioned.
However, installation of FTTH is both time consuming and very expensive, which is
why it is not widely deployed.
2.1.2 The DSL Families
DSL technologies have a very broad range, and as a result, they can be grouped
by modulation scheme, data rate transmission direction, or by any other common
factor between the technologies. In this section, the DSL technologies are classified
into families, where each family share similar modulation schemes and data rate
transmission directions. In Sections 2.1.2.1 to 2.1.2.5, the major DSL families are
introduced, while in Section 2.1.2.6 the spectrum, data rate, and maximum reach of
the various DSL technologies are compared.
2.1.2.1 Basic Rate Interface
The basic rate interface (BRI), which is considered as the original DSL, is based on the
integrated service digital network (ISDN) [34] technology. BRI provides the ability
to transmit both data and voice signals over the PSTN, with data rates up to 160
kbits/s over a single twisted-pair. The maximum length of a single run of the twistedpair cannot exceed 18 kft; however, with the use of repeaters, operation over longer
distances can be achieved. BRI is offered as a replacement for plain old telephone
service (POTS). However, where the demand for POTS exists, BRI and POTS can coexist. Based on the geographical region of deployment, various modulation schemes
for BRI are used. In North America, a four level pulse amplitude (4-PAM) modulation
is used [35]. BRI provides a symmetric transmission rate, i.e., the entire bandwidth is
utilized for both upstream (US) and downstream (DS) transmission. This is achieved
by the use of echo cancellation.
21
2.1.2.2 The High Bit-rate DSL Family
The high bit-rate DSL (HDSL) family of DSL technology is the most mature DSL
technology. The HDSL family has witnessed a number of technologies, all of which
have not been standardized. The common feature among theses variants is the ability
to deliver symmetric high data rates over long distances, which is suitable for business
customers. The two most common technologies in this family are the high bit rate
DSL (HDSL) and the single-pair high-speed DSL (SHDSL), which are standardized by
the international telecommunication union-telecommunication standardization sector
(ITU-T) in G.991.1 [36] and G.991.2 [37] respectively.
HDSL offers symmetric data rates of 1.544 Mbits/s over two twisted-pairs. Each
of the twisted-pairs carry a 784 kbits/s, over a single run of 12 kft. To increase the
reach of a single run, more twisted-pairs can be used, in addition to utilizing repeaters.
Similar to BRI, HDSL utilizes a 4-PAM modulation with echo cancellation. HDSL
utilizes low frequencies to increase its reach; thus, HDSL cannot coexist with either
POTS or BRI.
SHDSL, on the other hand, provides data rates that range from 192 kbits/s to 2.3
Mbits/s, in increments of 8 kbits/s, over a single twisted pair. However, there is an
option to utilize a second pair to improve the reach. SHDSL provides a multi-rate
transmission, and relies on coding gain to achieve its high data rates. The modulation utilized is a 16-PAM modulation with trellis coding (TC) and echo cancellation.
Various regional operational conditions are specified in G.991.2, with an annex dedicated to achieving data rates up to 5.696 Mbits/s. Similar to HDSL, SHDSL is not
compatible with either POTS or BRI.
2.1.2.3 The Asymmetric DSL Family
Residential customers, unlike business customers, require higher data rates in the DS
transmission direction. As a result, the asymmetric DSL (ADSL) technology emerged,
22
which is compatible with both POTS and BRI and deliver high data rates over a
long reach. The three main technologies in the ADSL family are the asymmetric
DSL (ADSL), the asymmetric DSL 2 (ADSL2), and the asymmetric DSL 2 plus
(ADSL2+), which are standardized by the ITU-T in G.992.1 [30], G.992.2 [38], and
G.992.5 [39], respectively.
ADSL offers asymmetric data rates up to 8 Mbits/s in the DS direction (from the
CO to the customer) and up to 896 kbits/s in the US direction. The maximum reach
of an ADSL line is 18 kft; however, at this reach the available DS data rate is 1.544
Mbits/s. ADSL was the first of the DSL technologies to utilize DMT modulation.
The frequency spectrum of ADSL starts from 25 kHz up to 1.1 MHz; however, as
an option, frequencies up to 80 kHz can be avoided in presence of BRI. Frequency
division duplexing (FDD) of US and DS transmission is achieved in ADSL by utilizing
the frequencies from 25 kHz to 138 kHz for US transmission, while the frequencies
from 138 kHz to 1.1 MHz are reserved for DS transmission.
ADSL2 is an extension of ADSL. The same frequency spectrum is utilized in both;
however, ADSL2 offers asymmetric rates of up to 12 Mbits/s (DS transmission) and
1 Mbits/s (US transmission). The enhancement in the data rates is achieved by
utilizing trellis coding, which was optional in ADSL, and the ability to use one-bit
constellations, such as binary PAM [4]. In addition, an all-digital mode is present in
ADSL2, where the entire bandwidth can be utilized, in absence of POTS and BRI.
ADSL2+ is an enhanced version of ADSL2, where the DS bandwidth is extended
to 2.2 MHz. As a result, the achieved DS data rates are increased to 20 Mbits/s,
especially for customers close to the CO. For various regional operational conditions,
the DS transmit PSD can be shaped to meet specific requirements.
23
2.1.2.4 The Very High Speed DSL Family
The main feature of the very high speed DSL (VDSL) family is that it provides
both symmetric and asymmetric transmission, which is suitable for both business
and residential customers respectively. While the ADSL family successfully offer high
bit rate over long distances, the demand for higher rates kept on rising. The VDSL
family of DSL technologies offers the highest data rates achieved over copper twistedpairs. However, these high data rates are only available over short distances. The
two DSL technologies within the VDSL family are the very high speed DSL (VDSL)
and the very high speed DSL 2 (VDSL2), which are standardized by the ITU-T in
G.993.1 [40] and G.993.2 [2] respectively.
VDSL offers asymmetric data rates that can reach 52 Mbits/s for DS transmission
and 1.5 Mbits/s for US transmission, for customers within a 1 kft radius from the
ONU. Symmetric data rates of 10 Mbits/s can be achieved over distances of 4.5 kft.
DMT modulation, along with FDD of US and DS transmission is used in VDSL.
VDSL uses frequency ranges from 0.138 MHz to 12 MHz, with these frequencies
divided into two DS and two US band plans. In addition to being compatible with
both POTS and BRI, VDSL is compatible with the entire ADSL family.
VDSL2 offers data rates that can reach up to 200 Mbits/s (asymmetric) and 100
Mbits/s (symmetric), over short distances. VDSL2 uses a wider bandwidth (from
25 kHz up to 30 MHz) and has a longer reach than VDSL. DMT modulation, along
with FDD of US and DS transmission is used in VDSL2. G.993.2 standard specifies
four major band plans, with eight profiles, where each of these band plans is suitable
for a specific HFTP deployment [41]. Band plan 8 (4 profiles: 8a, 8b, 8c, and 8d)
utilizes a bandwidth of 8.6 MHz with data rates up to 50 Mbits/s. Band plan 12
(2 profiles: 12a and 12b) utilize a bandwidth of 12 MHz and has a maximum data
rate of 68 Mbits/s. Both band plan 8 and 12 are suitable for FTTEx deployments,
24
with an 8 kft maximum length of twisted-pair run. Band plan 17 (1 profile: 17a)
utilizes a bandwidth of 17.7 MHz and provides a maximum data rate of 100 Mbits/s.
This band plan is suitable FTTCab deployments, where the maximum length of the
twisted-pairs is approximately 5 kft. Finally, the Band plan 30 (1 profile: 30a) utilizes
the entire VDSL2 bandwidth and offer data rates up to 200 Mbits/s. This band plan
is suitable for HFTP deployments where the twisted-pair length does not exceed 1
kft, such as FTTB deployments. Similar to VDSL, VDSL2 is fully compatible with
POTS, BRI and the entire ADSL family.
2.1.2.5 Fast Access to Subscriber Terminal
Fast access to subscriber terminal (G.fast) technology is the most advanced among
the DSL technologies. Currently, G.fast is being standardized by ITU-T in G.9700 [7].
G.fast, which has the potential of delivering aggregate data rates of up to 1 Gbits/s,
is suitable for FTTP deployment, where the maximum length of the twisted-pairs is
approximately 0.82 kft (250 m) [8].
This high data rate is achieved by utilizing a wider bandwidth. G.fast is being
studied for two band plans: from 2.2 MHz to 106 MHz and to 212 MHz. Instead
of FDD, G.fast is expected to utilize time division duplexing (TDD) of US and DS
transmission. It is expected that the US to DS ratio will be flexible, but constant
among lines served by the same distribution point. The US to DS ratio has to be
constant to mitigate the NEXT effect. Finally, it is expected that G.fast will be
compatible with all the aforementioned DSL technologies.
2.1.2.6 DSL Families Comparison
Table 2.1 compares the utilized bandwidth, the maximum data rates, the maximum
reach over twisted-pairs, and the ITU-T standards for the DSL technologies mentioned
above.
In this thesis, the focus is on profile 30a of the VDSL2 technology. Thus, any
25
Table 2.1: Comparison of DSL Families.
Family
Technology
BRA
ISDN
HDSL
Spectrum
MHz
up to 0.08
up to 0.4
SHDSL
up to 0.4
ADSL
0.025 − 1.1
ADSL2
0.025 − 1.1
ADSL2+
0.025 − 2.2
VDSL
0.138 − 12
VDSL2
0.025 − 30
G.fast
2.2 − 106 (212)
HDSL
ADSL
VDSL
G.fast
Max. Data Rates
Mbits/s
0.16
1.544
on two pairs
2.3
on a single pairs
DS: 6.8
US: 0.64
DS: 12
US: 1
DS: 20
US: 1
52 (net)
10 (net)
200 (net)
25 (net)
1000 (net)
Reach
ITU-T
kft
Standard
18
G.961
12
G.991.1
13.2
G.991.2
18
G.992.1
18
G.992.3
18
G.992.5
1.0
4.5
1.0
5.0
0.82
G.993.1
G.993.2
G.9700
referral to the DSL spectrum, indicates frequency ranges up to 30 MHz. Note that
the solutions to interference mitigation, introduced in this thesis, are not frequency
dependent. As will be shown in Chapter 5, the interference mitigation filters proposed in this proposal can be utilized by any DSL technology that is based on DMT
modulation.
2.1.3 DSL Signalling and Frame Structure
Currently, DSL systems utilize discrete multi-tone (DMT) modulation. DMT modulation, which utilizes a discrete Fourier transform (DFT) based block transceiver,
is a variant of orthogonal frequency division multiplexing (OFDM). One of the main
differences between DMT and OFDM is that the output of the DFT block in DMT
is real valued samples, while the output of the DFT block samples in OFDM are
complex.
The rest of this section is organized as follows. DMT modulation and DFT-based
26
transceivers are discussed in Section 2.1.3.1. DSL signalling and frame structure are
discussed in Section 2.1.3.2, while the DSL frame structure is discussed in Section
2.1.3.3.
2.1.3.1 Discrete Multi-Tone Modulation
In DMT, the channel bandwidth is partitioned, utilizing the inverse discrete Fourier
transform (IDFT), into orthogonal sub-channels (or tones). This division simplifies
the channel equalization needed to negate the channel dispersive effect. Moreover,
DFT-based transceivers are favoured in multi-channel modulation due to the availability of computationally efficient implementation methods for the DFT, such as the
fast Fourier transform (FFT).
Fig. 2.2 shows three sub-channels (sub-channels 10, 11, and 12), where the subchannels remain orthogonal at the sub-carrier frequency. Note that, at the sub-carrier
indexes, i.e., the center frequencies of the sub-channels, the contribution from other
sub-channels is zero. Thus, from the receiver’s point of view, sampling the received
signal at the sub-carrier frequencies is equivalent to having multiple parallel nonoverlapping sub-channels.
The number of sub-channels depends on the frequency separation between subcarriers (i.e., the sub-carrier spacing ∆f ) which varies from one system to another.
The main factor that affects the number of sub-carriers is the coherence time of the
channel. The coherence time is defined as the duration during which there is no
variation in the channel impulse response. The number of sub-carriers is usually
chosen such that the number of sub-carriers is maximized and the symbol duration
is less than the coherence time. Variation in the channel during the DMT symbol
duration degrades the orthogonality of the sub-channels.
Since the DMT symbol consists of multiple tones, it inherits the robustness of a
tone, i.e., a sinusoid signal, to dispersion. This fact is exploited to simplify the channel
27
Carrier no. 10
Carrier no. 11
Carrier no. 12
1
0.8
0.6
0.4
0.2
0
−0.2
−0.4
2
4
6
8
10
12
14
Sub−carrier index
16
18
20
Figure 2.2: Sinc functions at various sub-carrier indexes .
equalization for DMT-based systems. The insensitivity of a tone to the dispersion of
the channel can be easily seen in the frequency domain, where the Fourier transform
(FT) of an infinite duration tone is a delta function. No matter how dispersive the
channel is, distortion of the tone by the channel is limited to amplitude and phase
changes, which can be negated by a single-tap equalizer. For a signal that is composed
of multiple tones at various frequencies, the effect of the dispersive channel will be
also limited to changes in the amplitudes and the phases of each of the delta functions
corresponding to the various sinusoids.
The same concept can also be applied to discrete-time tones, where the DFT of a
discrete-time tone is also a delta function in the discrete-frequency domain. However,
during the truncated DMT symbol, the received signal is composed of the summation
of different replicas each of which is delayed differently due to the dispersive channel.
Thus, the received multiple tones are no longer orthogonal during the DMT symbol time. This, along with causing inter-symbol interference (ISI), complicates the
equalization of the channel effect.
To overcome ISI and simplify the channel equalization process, two things are
28
performed. First, a guard interval is utilized between consecutive symbols such that
the ISI occurs during this guard interval. Additionally, few samples are taken from
the end of the DMT symbol and appended to the beginning of said symbol. By doing
so, the received multiple tones are guaranteed to have suffered the same dispersion,
and thus, the DFT of these multiple tones are delta functions and the effect of the
dispersive channel will be limited to changes in the amplitudes and the phases of
each of the delta functions. This is achieved, as will be explained below, via adding
a cyclic prefix to the input sequence before applying it to the channel.
A block diagram of the DMT transmitter is shown in Fig. 2.3. The bit stream
is converted from a serial stream of bits into N parallel sets of bits via a serial to
parallel (S/P) converter. The number of parallel sets corresponds to the number of
sub-channels, and the number of bits in each of the parallel sets is based on each subchannel signal to noise ratio (SNR). Each set of bits is modulated via a quadrature
amplitude modulation (QAM) encoder, such that the output of the QAM encoder is
the complex N × 1 vector X define by
X = [X0 , X1 , ..., XN −1 ]T ,
(2.1)
where Xi is the modulated QAM symbol for sub-channel i.
Bit Stream
S/P
X1
x1
QAM X2
Encoder
x2
XN
x1, x2,....., x2N
2N-point
complex-to- real
IFFT
P/S
Add
Cyclic
Prefix
D/A and
Filter
x2N
Figure 2.3: DMT transmitter block diagram.
The IDFT of the output of the QAM encoder is obtained via a 2N -point complex
29
to real IFFT operation, which results in the 2N × 1 real vector x defined by
x = [x0 , x1 , ..., x2N −1 ]T .
(2.2)
A real output is required because, in DMT, the output of the IFFT is applied
directly to the channel after digital to analog conversion. To obtain the real vector
x from the complex vector X via the IFFT operation, a 2N × 1 vector XH with
Hermitian symmetry property has to be formed from the N × 1 complex vector X.
This can be easily performed by
T
XH = Re{XN −1 }X0 , X1 , ..., XN −3 , XN −2 , Im{XN −1 }, XN∗ −2 , XN∗ −3 , ..., X2∗ , X1∗ , ,
(2.3)
where Re{Xi } and Im{Xi } indicate the real and the imaginary parts of Xi respectively, and Xi∗ is the complex conjugate of Xi . The elements of x, i.e., the IDFT of
XH are calculated by
2N −1
1 X
XH,k ej(2π/2N )kn , ∀ n ∈ [0, 2N − 1].
xn = √
2N k=0
(2.4)
The output of the IFFT block, i.e., the time domain input samples defined by
vector x, is passed through a parallel to serial (P/S) converter, which converts the
parallel sub-symbols to a series of sub-symbols as shown in Fig. 2.3. Before the
inserting the DMT symbol into the channel, the cyclic prefix is added to the beginning
of the DMT symbol. The cyclic prefix is a copy of the last v − 1 sub-symbols of x2N ,
which is appended to the beginning of time domain input sequence as shown in Fig.
2.3. The length of v is chosen such that it equals the length of the channel impulse
response. In that manner, the ISI between consecutive symbols will be confined to
the cyclic prefix. Note that instead of appending the last v − 1 sub-symbols to the
DMT symbol before transmitting it over the channel, one could append v −1 zeros, to
confine the ISI to the cyclic prefix. However, the cyclic prefix serves another purpose.
30
As mentioned earlier, adding the cyclic prefix serves the purpose of simplifying
the channel equalization. By appending the last v − 1 samples of the time sequence
representing the DMT symbol to the beginning of the sequence, the aperiodic input
time sequence seems periodic over the length of the convolution. Thus, the linear
convolution can be represented by a circular convolution, once the cyclic prefix is
removed. Modelling the linear convolution as a circular convolution is important
because it allows the usage of DFT. This means that, after removing the cyclic prefix,
the DFT of the received sequence equals the product of the DFTs of the transmitted
sequence and the channel impulse response. Consequently, the channel effect can be
negated using a simple one-tap equalizer for each sub-channel.
Finally after adding the cycling prefix to the input sequence, the input sequence
is converted to an analog signal via digital to analog (D/A) block as shown in Fig.
2.4. Before the analog signal is applied to the channel, a transmit filter, shown in
Fig. 2.4, may be utilized to eliminate any out of band power leakage and to ensure
that the transmitted signal power spectral density (PSD) remains in a specific range.
Note that the PSD levels can be controlled digitally, but sometimes it is easier to
control it via the analog front end. [42]
Fig.2.4 shows the DMT receiver block diagram. Once the signal is received, it
passes through the receiver filter which minimizes the out of band noise, after which
it is converted from analog to digital via the analog to digital (A/D) block. The cyclic
prefix is then stripped from the received sequence y.
Through a P/S converter the received sequence is converted to 2N sub-symbols.
These sub-symbols are fed to an FFT block which performs and 2N -point real-toimaginary conversion. The FFT block takes the DFT of the received sequence and
reverses the Hermitian symmetry. Recall, the cyclic prefix is added to the input
sequence to force periodicity over the convolution interval, which results in a linear
31
y1
Y1
y2
Y2
^
X
1
^
X
2
FEQ
Filter and
A/D
Add
Cyclic
Prefix
y1, y2,....., y2N
N-point
YN
complex-to- real
IFFT
S/P
QAM
Decoder
Bit Stream
P/S
X^N
y2N
Figure 2.4: DMT receiver block diagram.
convolution being equivalent to a circular convolution. Thus, for a noiseless channel,
the received sequence after the FFT block is a multiplication of the input sequence
and the channel frequency response, which can be written in vector form as

 

X0 .H0
Y

 0  

 

 Y1   X1 .H1 

 

Y=
,
=
..

 ..  
.

 .  

 

XN −1 .HN −1
YN −1
(2.5)
where Hi is the frequency response of the ith sub-carrier defined in (2.6).
N −1
1 X
Hk = √
hn e−j(2π/2N )kn , ∀ k ∈ [0, N − 1].
N k=0
(2.6)
To extract the Xi from Yi , the received sequence is passed through a single-tap
filter per sub-channel called frequency-domain equalizer (FEQ) as shown in Fig. 2.4.
The coefficients of the FEQ is the inverse of the channel frequency response. Note
that division is not usually preferred from an implementation perspective. Thus, the
single-tap FEQ is implemented via a complex operation that involves scaling and
rotating Yi to mitigate the effect of channel impairment [22].
In presence of noise, the outputs of the FEQ are not exactly equal to X but
rather X̂, which is an estimate of X. This estimate of X is then passed through a
QAM encoder followed by a P/S converter, as shown in Fig. 2.4, to reconstruct the
32
transmitted bit stream . Note that in absence of coding, X̂ is passed through a simple
symbol by symbol decision process to recover the constellation points before utilizing
the QAM decoder. However, if trellis encoding (i.e., convolution codes) is utilized,
then a trellis decoder is required to decode the constellation points.
2.1.3.2 DSL Signalling
DSL signals are transmitted in differential mode (DM) where complementary signals are transmitted over a twisted-pair of wires. DM signalling provides resilience
to electromagnetic interference (EMI) as any external interference will couple identically on both of the pairs, and thus, any interference will be eliminated at the
receiver. However, in practical situations, external interference affects each wire in a
pair differently.
The common mode (CM) signal, which is the arithmetic mean of the signals,
can be determined at the receiver, with little extra cost. The CM signal contains
information about the external EMI; thus, the CM signal can be utilized to estimate
the DM EMI. That estimate can then be used to cancel the interference.
Balun
DM Signal
CM Signal
Figure 2.5: DM and CM signal at the receiver.
Fig. 2.5 shows a balun that is connected to a twisted-pair, and outputs two signals:
the DM and the CM signal. A balun is a resistance transformer that converts the
balanced DSL signal to a differential mode signal, at the receiver end. From the
center tap of the balun, the common mode signal, which is usually ignored since it
does not contain relevant information about the desired signal, is obtained. Ideally,
a balun will output a DM signal that is only made up of the desired signal and a
33
CM signal that only contains noise and electromagnetic interference. However, due
to imperfections in the balun, and in the twisted-pairs, the DM portion of the EMI
is found in the DM signal.
2.1.3.3 DSL Frame Structure
The DSL network utilizes a dedicated channel for each user. Data is transmitted over
the DSL channel in super-frames. Since each DSL user has its own dedicated channel,
all super-frames on the DSL channel belong to a single user.
The super-frame for VDSL2 is divided into 257 frames. The last frame within
a super-frame is the synchronization (sync) frame [2]. Sync symbols are usually
utilized for only synchronization purposes, where bits are modulated via 4-QAM
constellation; however, pilot signals can be transmitted during the sync frame for
channel state estimation purposes [6]. To facilitate parallel sub-channel estimation,
orthogonal pilot sequences are again used [43]. Note that the pilot sequences are
vendor specific [44].
DSL
Super-frame
t
TSF
Synch
Data
TS
TD
Figure 2.6: DSL super-frame structure.
A super-frame, shown in Fig. 2.6, has duration of TSF . Note that the value for TSF
varies from one standard to the other; for VDSL2 TSF =64.25 ms [2], while for G.fast
TSF < 10 ms [8]. Recall, the DSL technology utilized in this thesis is VDSL2. For
simplicity, and without loss of generality, let us assume that the first frame in each
34
super-frame is a synchronization frame and there are 256 frames per super-frame.
Thus, as shown in Fig. 2.6 a DSL super-frame constitutes a synchronization frame
followed by 255 data frames, each with a duration of TS =0.25 ms. Thus, in this thesis,
the duration of VDSL2 super-frame is assumes to be 64 ms.
2.1.4 The DSL Interference Environment
Interference in DSL can be classified as intrinsic, such as thermal noise and crosstalk,
or extrinsic, such as impulsive noise and PLC interference [45]. In this section, the
various sources of interference in DSL networks and the interference mitigating techniques utilized to combat them are discussed. The rest of this section is organized
as follows. In Section 2.1.4.1, DSL intrinsic interference sources are discussed, while
DSL extrinsic interference sources are presented in Section 2.1.4.2.
2.1.4.1 Intrinsic Interference
Crosstalk (xTalk), which occurs when a signal power leaks from one twisted-pair
to another, is a wide-band interference and is the main cause of errors in a DSL
network [22]. There are two types of crosstalk: near end crosstalk (NEXT) and far
end crosstalk (FEXT) [46].
Signal
Tx/Rx
NEXT
Tx/Rx
Figure 2.7: Near end crosstalk among twisted pairs.
NEXT occurs among twisted-pairs when an interfering signal is transmitted from
the same end of the cable as the receiver, as shown in Figure 2.7. FDD is utilized to
35
prevent NEXT among DSL lines served by the same distribution point [22]. However,
since TDD is used instead of FDD in G.fast, the US to DS ratio should be constant
among twisted-pairs served by the same distribution point. By fixing the US to DS
ratio, it is ensured that all transceivers at the customers’ premises (or at the CO) are
transmitting at the same time, which eliminates NEXT.
Signal
Tx/Rx
FEXT
Tx/Rx
Figure 2.8: Far end crosstalk among twisted pairs.
FEXT, on the other hand, occurs when an interfering signal is transmitted from
the end of the cable that is opposite to the receiver as shown in Figure 2.8. Various
mitigation techniques are utilized to mitigate the FEXT, the most prominent of which
is vectoring [6]. Vectoring is discussed in more detail Section 2.1.4.1.3.
The concept of interference mitigation is essential to any communication systems.
Various techniques over the years have been developed to estimate and cancel DSL
intrinsic interference. In Sections 2.1.4.1.1 to 2.1.4.1.3, interference mitigation techniques that are utilized in mitigating DSL intrinsic interference are highlighted.
2.1.4.1.1 Adaptive Filters
Mitigating the effects of NEXT by utilizing time domain adaptive filters was proposed in [47] [48]. For a cable containing m + 1 twisted pairs, m filters are required to
mitigate the NEXT effect on each twisted-pair. Thus, a total of (m+1)m adaptive filters are needed to eliminate NEXT in said cable. In addition, access to the disturbing
signals is required for this approach to work. The computational complexity of this
36
approach makes it impractical, especially at the customers’ premises. A variation of
this approach was proposed in [49], where the highest n NEXT sources are cancelled
using n adaptive filter. However, the same limitations on this variation still exist,
along with the task of detecting the highest n disturbers.
2.1.4.1.2 Frequency Division Duplexing
Frequency division duplexing (FDD) of US and DS transmission in DSL networks
is used to mitigate the effects of NEXT [22]. By utilizing FDD, transceivers at the
same end of the cable are transmitting and receiving at different frequency bands.
This creates a zipper pattern where the available spectrum is divided into smaller
sub-spectra. The sub-spectra are allocated in an alternating manner to US and DS
transmission; thus, eliminating NEXT.
However, as a result of utilizing FDD, the available VDSL2 bandwidth is restricted
to specific non-overlapping frequency ranges, which affects the overall transmission
rate. In addition, in presence of multiple DSL systems that employ different duplexing
schemes, utilizing FDD is rendered ineffective in mitigating the effects of NEXT [50].
2.1.4.1.3 Vectored Transmission
For a single DSL system, the utilization of FDD results in the elimination of NEXT.
Thus, the major source of intrinsic interference for DSL is FEXT. In 2001 a new
transmission technique called vectored transmission (vectoring) was proposed in [51],
which can be utilized to eliminate FEXT from a DSL network for all transceivers
within the same cable binder co-located at the DSLAM.
As long as all the twisted-pairs in the cable are connected to the same CO and are
managed by the same service provider [52], FEXT among twisted-pairs can be virtually eliminated, which results in transmission rates up to 100 Mbits/s [44]. However,
if not all the twisted-pairs within the cable binder are considered in the joint signal
processing (i.e, presence of uncontrolled lines), the resulting FEXT will negatively
37
impact the vectoring process [53].
In 2010, vectoring was standardized in [6], and is being adopted by a number of
service providers as the next-generation broadband technology [54]. While vectoring
shows promising improvement in data rates, especially over short lines, the presence
of uncontrolled lines significantly deteriorates this improvement. This is a severe
limitation of vectoring, especially when local loop unbundling (LLU). In LLU the
incumbent local exchange carriers are forced to share their infrastructure with other
operates. Since multiple operators share a cable binder, and since channel state
information is not shared among the multiple operators, mitigating the effect of FEXT
via vectoring will not be possible. In [55], it was shown that data rates dropped
from 100 Mbits/s to 70 Mbits/s in presence of a single uncontrolled line. Another
limitation of vectoring is its complexity in situations where a relatively large number
of twisted-pairs served by the CO [56].
2.1.4.2 Extrinsic Interference
Extrinsic interference can be classified as narrow-band or wide-band interference.
Radio frequency interference from AM and amateur radio, which usually affects few
tones at a time, is considered narrow-band interference. These types of narrow-band
extrinsic interference usually are geographically variable [45]. On the other hand,
wide-band extrinsic interference is usually from systems co-located with the DSL
system, and utilizes the same spectrum as DSL. One of the most prominent of these
technologies is power line communications (PLC) [12, 23]. Techniques that mitigate
the effect of EMI on DSL, both narrow-band and wide-band EMI, have been discussed
in Section 1.4.2.
Note that this thesis focuses on wide-band extrinsic interference from PLC on
DSL in a residential environment. Recall, the objective of the thesis is to propose
solutions that mitigate the interference between the DSL modem and a co-located
38
PLC network, within a residential setting.
2.2 PLC Technology
Although PLC as a technology has been proposed since the beginning of the last
century [57], it is only recently that PLC networks are seen as a viable option for
home networking. Communications over power lines utilized a narrow-band single
carrier at its early stage [58]; currently, broadband PLC utilize a wide spectrum that
reaches up to 100 MHz and has the potential of delivering data rates of up to 500
Mbit/s [3]. As discussed in Chapter 1, delivering the Internet to customers over
the power lines faced many challenges due to the cost and availability of cheaper
alternatives. However, PLC technology saw success in various areas such as vehicular
networks, smart grid applications, municipal applications, and local area networks
(LANs).
Because of the existing extensive power line infrastructure and the continuous
advancement in LAN technology, PLC is becoming an attractive solution to create
in-home networks among devices that might benefit from access to the Internet. In
this section, broadband PLC technology and its application in home networking are
introduced. The architecture of a typical in-home PLC network is presented in Section
2.2.1, while broadband PLC signalling and frame structure is discussed in Section
2.2.2.
2.2.1 PLC In-home Network
Electric power is carried to neighbourhoods over high voltage power lines in the
range of 6 to 16 kV. For residential usage, this high voltage is reduced via step-down
distribution transformers to 240 V [1]. Fig. 2.9, shows a distribution transformer
with two leads. The two leads of the transformer are connected to Line 1 and Line 2.
39
The potential between Line 1 and Line 2 is 240 V [58]. In north America, the 240 V
is split into two phases via a central tap that is connected to the neutral of the house
[59]. The potential between the neutral and each of Line 1 and Line 2 is 120 V, as
shown in Fig. 2.9. Thus, within the panel, there are two lines, the neutral, and the
earth ground. Note that the earth ground connection is not shown in Fig. 2.9.
Distribution
Transformer
Panel
Line 1
120 V
240 V
16 kV
Neutral
120 V
Line 2
Figure 2.9: Electrical wiring in north America [1].
The power lines within the house are terminated at bus bars with the panel (also
known as the circuit breaker panel). These power lines are either single-phase or
two-phase. The single-phase power lines are connected to outlets within the house
to supply light fixtures and small appliances such as TV, PLC couplers, etc, with
electricity. Large appliances such electric stoves, water heaters, etc, are connected to
the two-phase since they require the usage of the 240 V between Line 1 and Line 2.
In-home PLC networks are utilized for many purposes. An in-home PLC LAN
can have multiple functions, and support communications on various spectra. There
are two categories of PLC networks: narrow-band and broadband. Narrow-band
PLC (NB-PLC) is used for AMR and home automation; broadband PLC (BB-PLC)
is mainly used for distribution of triple play services within the house. Fig. 2.10
shows a PLC network within the home environment. The unshielded power lines run
throughout the house. PLC transceivers are connected via the mains outlets to form
40
the PLC network. To these transceivers, devices such as computers and television,
are connected.
Power Lines
Broadband
Router
PLC
Transceiver
PLC-DC
PLC
Transceiver
PLC
Transceiver
Figure 2.10: In-home PLC network.
Note that cross-phase coupling, i.e., the ability of the PLC signal to couple from
one phase to the other, is an issue for NB-PLC. The low operating frequency used
by NB-PLC prevented the signal from coupling across phases because the impedance
between phases caused the signal to attenuate significantly. A capacitive coupler
is required to enhance the coupling between phases [59]. However, since BB-PLC
utilize high frequencies, the bus bars in the panel act as a capacitor which enables
the cross-phase coupling. Additionally, BB-PLC signal couples across phases at large
appliances, which are connected to both phases. Thus, cross-phase coupling is not an
issue for BB-PLC [1].
41
As will be discussed in Section 2.2.2, the shared PLC channel access by PLC
transceivers (users) is performed in one of two ways, depending on the type of traffic.
Contention based channel access does not require a controlling entity to organize the
channel access among users. Contention free channel access, as with other broadband
access technologies with shared mediums, requires a domain controller to facilitate
access among PLC users based on each user’s requirement. Note that this is the case
when a minimum quality of service is required.
In contention free channel access, one of the PLC transceivers takes over as the
PLC domain controller (PLC-DC). The PLC-DC regulates access to the PLC channel
and is connected to the Internet via a broadband router, as shown in From Fig. 2.10.
In addition to accessing the Internet via a computer, services such as IPTV or voice
over Internet protocol (VOIP) are provided via the PLC transceivers.
2.2.2 PLC Signalling and Channel Access
In this section, the aspects of the physical and MAC layer of in-home PLC networks
relevant to this thesis are discussed. In Section 2.2.2.1, PLC signal transmission and
modulation are discussed, while the MAC standards for broadband PLC networks
are introduced in Section 2.2.2.2.
2.2.2.1 PLC Signalling and Modulation
Power Lines
PLC Coupler
DM Signal
CM Signal
Figure 2.11: Simplified PLC coupling circuit.
42
Similar to DSL, differential mode signalling is utilized in PLC. Fig. 2.11 shows
how a capacitive PLC coupling circuit used to transmit broadband signals over inhome power lines [60]. The objective of this coupling circuit is to filter out the 60 Hz
high voltage waveform and transmit a broadband signal over the power lines. The
PLC modems also have fuses before the capacitors and diodes after the transformer
for over voltage protection. The fuses and diodes are not shown in Fig. 2.11. What is
relevant to this thesis is the functionality of the transformer. Similar to the balun, the
coupler acts as an impedance transformer, where two balanced signals are converted
to two unbalanced signals. The DM signal, which is the difference between the two
balanced complementary signals, is utilized to carry the desired Signal. The CM
signal, is usually ignored, since it mainly contains EMI.
PLC utilizes DFT-based transceivers for signal modulation/demodulation [58],
which allows the partitioning of the broadband channel into narrower sub-channels.
Among the variants of DFT-based modulation techniques, DMT modulation is the
most suitable for PLC communications. This is because base-band transmission is
utilized in PLC networks. DMT modulation was discussed in detail in Section 2.1.3.1.
Note that the PLC sub-carrier spacing ∆fPLC is larger than the DSL sub-carrier
spacing, since the PLC channel state varies more rapidly than the DSL channel.
Recall, the number of sub-carriers (or sub-channels) is inversely proportion with the
coherence time of the channel. Since the DMT symbol duration has to be shorter
than the coherence time of the channel to prevent degrading the orthogonality of
the sub-channels, and since the coherence time of a PLC channel is shorter than the
coherence time of a DSL channel, the DMT symbol duration for a PLC network is
shorter than that of the DSL network. And due to the inverse relationship between
time and frequency, a shorter PLC symbol duration means a wider PLC sub-carrier
spacing.
43
2.2.2.2 MAC for Broadband PLC
As discussed in Section 2.1.3.3, the DSL network utilizes a dedicated channel for each
user. The PLC channel, on the other hand, is shared among all the users in the PLC
network. As will be discussed in Chapter 3, the cross-coupling channel varies significantly from one PLC user to the other (i.e., from one power outlet to the other). Thus,
to propose solutions that overcome the outlet-to-outlet variations in the PLC-to-DSL
cross-coupling channels, it is essential to understand the channel access mechanism
utilized by in-home broadband PLC networks, for multimedia applications.
Various standard bodies have developed standards for the MAC layer of a broadband PLC network. Among the most wide-spread standards are HomePlugAv [61],
IEEE P1901 [62], and G.hn [9], which are standardized by the HomePlug Consortium,
the IEEE, and the ITU-T standard bodies, respectively. In all the aforementioned
three standards, PLC nodes are divided into two classes, PLC domain controllers
(PLC-DCs) and PLC users. In [62], the PLC-DC is called the local administrator,
while in [61] and [9], the domain controller is referred to as the connection manager
and the domain manager, respectively. PLC users are referred to slave stations or
PLC nodes.
PLC-DCs have various functions such as transmitting beacons to provide info on
the contention periods and contention free periods, broadcasting information about
channel, and assigning time-slots to PLC users. Usually, the first node that joins the
network takes over as a PLC-DC [9]; however, as will be discussed in Chapter 5, the
PLC user closest to the DSL modem is the PLC-DC in this thesis.
All the above standards employ one of two channel access mechanisms, either
carrier sense multiple access with collision avoidance (CSMA/CA) or time division
multiple access TDMA [63],[64]. CSMA/CA is reserved for best effort traffic, where
channel access is contention-based. Best effort traffic, such as web-based or email
44
applications, is a classification of traffic that does not have a quality of service (QoS)
requirement, such as packet loss, latency, etc. Sensitive traffic, such as VOIP and
video conferencing, is traffic that requires its packets to be delivered on time with a
certain error threshold. For this type of sensitive traffic, TDMA is used as an access
mechanism where the PLC-DC assigns time-slots to the various PLC users according
to their QoS requirements.
In this thesis, it is assumed that the in-home PLC LAN is utilized for QoS traffic;
thus, the PLC network employs TDMA as a channel access mechanism. The TDMA
channel access for a three-user PLC network (User A, User B, and User C) is depicted
in Fig. 2.12. The vertical axis indicates the names of the users and the combined
channel, while the horizontal axis indicates the time, which is divided into time-slots.
The user’s packets only represent when traffic becomes ready for transmission at each
node. Thus, Fig. 2.12 shows how the time-slots are allocated to the various PLC
users.
PLC
t1
User A
A
User B
Slot 1
-
t6
t5
t4
A
B
User C
Channel
t3
t2
A
B
C
C
Slot 2
B
Slot 3
A
B
A
A
B
B
C
Slot 4
C
Slot 5
A
t8
t7
Slot 6
C
C
Slot 7
B
t
TP
Figure 2.12: PLC frame structure for a 3-user PLC network.
Assuming that all three PLC users have the same QoS requirement, the PLC-DC
allocates the next available time-slot to the PLC user with the earliest frame arrival
45
time, (i.e., the PLC user whose frame was queued for transmission before all other
PLC frames in the network). In Fig. 2.12, the first frame arrival is between t1 and
t2 from User B followed by User A then User C, where ti is the start time of Slot
i. Thus, the PLC-DC assigns the next available time slot, Slot 2, to User B; after
which, the PLC-DC allocates Slot 3 and Slot 4 for User A and User C, respectively.
Note that Slot 1 is not used because the first frame arrival occurs after the beginning
of Slot 1.
Note that it is assumed that the PLC transceivers follow the recommendations
of the ITU-T G.hn home networking standard. While the interference mitigation
solutions discussed in Chapter 5 propose some modification to MAC layer of the PLC
network, these modifications comply with the recommendations of the G.hn standard.
2.3 PLC Noise Environment
PLC utilizes unshielded power-lines which are more sensitive to EMI than the twistedpairs utilized by DSL. The power lines within the home pick up EMI from various
noise sources such as AM and amateur radio, appliances within the house, etc. In the
literature, there are three main classes of noise present on the PLC channel. However,
in this thesis, a fourth class is introduced.
The first class of noise is the coloured background noise, which is the sum of
various noise sources and is considered time-invariant since the noise level does not
change for various consecutive AC cycles [58]. Typically, the coloured background
dBm
noise has a PSD of -145
[65].
Hz
The second class of noise is the impulsive noise which is characterized with shorter
durations and higher amplitudes than the background noise. Impulsive noise within
PLC system is either synchronous or asynchronous to the mains frequency, or in form
of isolated impulses. Synchronous impulsive noise occurs at the frequency of the AC
46
mains and is usually due to silicon controlled rectifiers and from electronic circuits.
Asynchronous impulsive noise occurs at frequencies much higher than the AC mains,
and a typical source of asynchronous impulsive noise is switching regulators. In the
dBm
dB [65].
literature, the PSD of impulsive noise is in the range of -105
Hz
The third class of noise to PLC systems is the narrow-band interference from
AM and amateur. As was the case with narrow-band DSL interference, this type
of interference affects few sub-channels at a time and is usually mitigated via notch
filters. Finally, the fourth class of noise to PLC systems is wide-band interference
from co-located DSL systems, which is introduced in this thesis in Chapter 4. This
class of noise typically affects a number of sub-channels. For PLC, the dominant
wide-band interference source occurs due to the US-DSL transmission. Further detail
on the effect of DSL on the performance of a co-located PLC system is presented in
Section 4.2.2.
47
Chapter 3
CROSS-COUPLING CHANNEL
MEASUREMENTS
The main goal of this thesis is to mitigate the interference between co-located PLC
and DSL networks, within the home environment. The interference mitigation solutions proposed in this thesis utilize adaptive filters to extract an estimate of the DM
interference from the CM signal. As will be discussed in Chapter 5, both the DM and
CM PLC-to-DSL cross-coupling channel frequency response matrices are used in the
training of the adaptive filters, and in the performance evaluation of the proposed
interference mitigations solutions.
Field measurements of the interference environment between a DSL modem and
a co-located in-home PLC network have never been performed. In addition, a model
that describes the PLC-to-DSL cross-coupling channels within a residential house does
not exist. Thus, a field measurement campaign is required to study the interference
environment between a DSL modem and an in-home PLC network.
The measurement campaign, introduced in this chapter, characterizes both the
DM and CM PLC-to-DSL cross-coupling channels in two test-sites. A vector network
analyzer (VNA) based measurement system is developed to measure the complex
PLC-to-DSL cross-coupling channels. Measurements were performed in residential
houses to study the effect of the co-located DSL and PLC systems on each other.
Among other findings, analysis of the measurement data reveals that the PLC-to-DSL
cross-coupling channels are frequency selective and the spatial separation between the
PLC and DSL modems have no significant impact on the interference levels.
The rest of this chapter is organized as follows. In Section 3.1, the DSL modem
48
installation scenarios are discussed. The PLC-to-DSL interference environment is
discussed in Section 3.2; the methodology used to measure the DM and CM PLCto-DSL cross-coupling channels is discussed in Section 3.3. Finally, the measurement
campaign, which is composed of two case studies, is presented in Section 3.4.
3.1 DSL Modem Installation Scenarios
The DSL modem installation scenarios used in the measurement campaign are presented in this section. Within a house, the DSL copper twisted-pair is terminated
by a modem using one of two DSL modem installation scenarios: the Desk Modem
Scenario and the Entry Point Scenario. The PLC-to-DSL cross-coupling channels,
presented in Section 3.4, are measured for both the Desk Modem Scenario and the
Entry point Scenario. Additionally, the performance of the interference mitigations
solutions, proposed in Chapter 5, is evaluated for both scenarios.
TB
LPF
Telephone Lines
LPF
DSL
Transceiver
Broadband
Router
Figure 3.1: Desk Modem Scenario.
In the Desk Modem Scenario, shown in Fig. 3.1, the DSL modem is located on
a desk within the house. The twisted-pair carrying the DSL signal to and from the
49
house is connected to a terminal block (TB). The TB is connected to the house’s
interior telephone wiring, over which the broadband DSL signal is carried through
the house to a DSL modem. Low pass filters (LPFs) are installed on each telephone
to prevent audio distortion from the DSL signal.
TB
LPF
DSL
Transceiver
Telephone Lines
Broadband
Router
Figure 3.2: Entry Point Scenario.
The Entry Point Scenario, on the other hand, considers the situation where the
DSL modem is installed where the telephone cable enters the house, as shown in
Fig. 3.2. The signal from the DSL modem is distributed throughout the house either
via Wi-Fi, Ethernet, or over the house interior coaxial TV cable to one or more set
top boxes. In addition, an in-line LPF is installed after the TB for the household
telephone cable so that the DSL signal is blocked from travelling within the house.
This makes installing LPFs on each telephone unnecessary.
Both scenarios have their own advantages and disadvantages. The Desk Modem
Scenario carries the DSL signal deep into the house, which is beneficial if the Internet
is carried throughout the house via a WLAN. The Entry Point Scenario, on the other
hand, minimizes the EMI suffered by the DSL signal, because the LPF that prevents
the DSL signal from travelling within the house over the house’s interior telephone
50
wiring also blocks the EMI that coupled on the house’s interior telephone wiring from
affecting the DSL signal.
Thus, to reduce the cross-coupling between DSL and in-home PLC networks, the
Entry Point scenario is preferred. In addition, installing the DSL transceiver within
the house requires the installation of extra LPFs for each telephone outlet to prevent
interference to the voice band. However, since most customers utilize wireless routers
to carry the Internet throughout the house, terminating the modem within the house,
i.e., the Desk Modem Scenario, allows for a better coverage.
3.2 DSL and PLC Interference Environment
Power lines are not designed to communicate data at high frequencies; rather, the
main purpose of power lines is to supply alternating high voltages at very low frequency. Since power lines are not shielded, portions of the signals transmitted over
the power lines will radiate.
Fig. 3.3 shows the interference environment between co-located DSL and PLC
networks, within the home environment. In this figure, a DSL modem and an inhome PLC-network that operate over the same spectrum co-exist. Note that the
DSL modem is installed via the Entry Point Scenario discussed in Section 3.1 and it
supplies the house with Internet, while the PLC network forms a LAN that carries
the Internet within the house.
The power lines within the house form a tree with various branches that connect
the various rooms of the house. Any signal transmitted over the power lines travel
through all the branches of the power line tree; as the signal travels down the power
line tree, portion of it is radiated. Due to the radiation from the various branches
of the power line tree, the house is transformed into a large antenna. This radiation
is picked up by the copper twisted-pair. As will be confirmed by the measurements
51
TB
LPF
Power Lines
DSL
Transceiver
PLC
Transceiver
Broadband
Router
Telephone Lines
PLC-DC
PLC
Transceiver
PLC
Transceiver
Figure 3.3: Co-located DSL and PLC networks interference environment.
presented in Section 3.4, this PLC radiation causes significant interference to the DSL
network.
Similarly, due to imperfections in the twisted-pairs, a portion of the DSL signal
is radiated. This radiation is picked up by the power lines; however, since the DSL
signal travels from and to the house, the level of DSL-to-PLC interference varies
significantly between upstream and downstream DSL transmission. Further detail on
the levels of DSL interference on a co-located PLC network is discussed in Section
4.2.2.
52
3.3 Measurement Methodology
In this section, the methodology behind the channel measurement campaign is described. Recall, the main purpose of the measurements is to characterize the DM and
CM PLC-to-DSL cross-coupling channels. Since both DSL and PLC systems utilize
multi-carrier FFT based modulation, as discussed in Chapter 2, measuring the crosscoupling channels in the frequency domain is appropriate. The hardware utilized in
the measurement campaign is described in Section 3.3.1, while the calibration process
is discussed in Section 3.3.2. Finally, the measurement setup is introduced in Section
3.3.3.
3.3.1 Measurement Hardware
To study the cross-coupling channels between PLC and DSL networks within the
home environment, an Agilent E5071B vector network analyzer (VNA) with an operating range of 300 kHz to 8.5 GHz is utilized. One port of the VNA is connected to
the twisted-pairs within the house, while the second port of the VNA is connected to
the power lines. The scatter matrix between the two ports of the VNA is obtained,
which is essentially the cross-coupling channels between the PLC and DSL networks.
However, since a VNA has coaxial cable outputs (which are unbalanced with
respect to the ground), and both DSL and PLC utilize DM signalling (which is
composed of two complementary signals that are balanced with respect to the ground),
the unbalanced coaxial ports of the VNA have to be matched to the balanced DSL
and PLC networks [14].
To connect the balanced DSL network to the VNA, a balun is used. A balun is
essentially a resistance transformer that converts a balanced input to an unbalanced
output and vice versa. Fig. 3.4 shows the North Hills 0320BF Balun utilized in the
measurement campaign. The 0320BF Balun operates over the range of 10 kHz to 30
53
Figure 3.4: North Hills 0320BF Balun.
MHz. This balun has two 50 Ω unbalanced coaxial ports (labelled J1 and J2 in Fig.
3.4) and two balanced ports with a combined 100 Ω resistance (labelled 1 and 2 in
Fig. 3.4). To use the balun in differential mode, it is connected to the VNA via port
J1; while for common mode readings, the balun is connected to the VNA via port J2.
Note that the twisted-pairs are connected to the Balun via ports 1 and 2, shown in
3.4.
Figure 3.5: Northern Microdesign PLC Coupler.
Fig. 3.5 shows the Northern Microdesign PLC coupler that operates over the
range from 1.8MHz to 39MHz, which is utilized to connect the VNA to the power lines
54
within the house. The PLC coupler has two sub-miniature version A (SMA) ports
and a three-pin plug to connect to the the power lines via the mains outlets within
the house. Both SMA ports are connected to the differential output of the coupler.
Thus, to inject a DM signal onto the power lines, the PLC coupler is connected to
the VNA via one of the SMA ports. Note that an SMA to BNC (baby N connector)
adaptor is required to connect the coupler to the VNA coaxial ports.
3.3.2 Calibration
−0.5
Balun
PLC Coupler
−1
Gain in dB
−1.5
−2
−2.5
−3
−3.5
−4
0
5
10
15
20
Frequency in MHz
25
30
Figure 3.6: Insertion loss of the balun and the PLC coupler before calibration.
Fig. 3.6 shows the insertion loss due to the balun and the PLC coupler. Thus,
before collecting the measurements, the VNA is calibrated to remove the effects of
the balun and the PLC coupler. However, before calibration both the Balun and the
PLC coupler have to be modified. A registered jack (RJ) is connected to ports 1
and 2 of the balun to facilitate the calibration procedure and promote a more secure
connection to the twisted-pairs as shown in Fig.3.7. A female 3-pin socket is soldered
to a BNC adaptor to enable the calibration of the PLC coupler as shown in Fig. 3.8.
Calibration of the VNA, Balun, and PLC coupler is required to eliminate any
55
Figure 3.7: Modified Balun.
Figure 3.8: Modified PLC Coupler.
effect on the collected measurements. Each port of the VNA is calibrated for four
scenarios: open, short, load, and through (between ports). A standard calibration kit
provided by the manufacturer is usually utilized in the calibration process; however,
since standard calibration kits are designed for the 50 Ω VNA ports, and since both
the balun and PLC coupler require a 100 Ω calibration kit, a customized calibration
kit is utilized in the calibration process. Three RJ 45 jacks are utilized for the open,
short, and load calibration, as shown in Fig. 3.9. The through calibration, shown in
Fig. 3.10, is performed via a pair of wires that are connected to a BNC adaptor at
one end and to an RJ 45 jack at the other.
56
Figure 3.9: Right to Left: Open, Short, and Load (100 Ω).
Figure 3.10: Through calibration.
3.3.3 Measurement Setup
After calibration, the scatter matrix between port 1 and port 2 of the VNA is used
to determine the PLC-to-DSL cross-coupling channels. One port of the VNA is
connected to the power lines of the house through the PLC coupler, while the second
port is connected to the telephone wires in the house through the balun, as shown
in Fig. 3.11. The VNA is set to output 0 dBm sinusoidal signals and to collect 1500
samples, between frequencies 300 kHz and 30 MHz. To measure the DM PLC-to-DSL
channel frequency response matrix the balun is connected in the DM mode to the
VNA, as shown in Figure 3.11(a). To measure the CM PLC-to-DSL cross-coupling
channel, the balun is connected in the CM mode to the VNA, as shown in Figure
3.11(b).
57
VNA
PLC adapter
Power
Network
Balun
Port 2
Coupling
Port 1
Telephone
Network
Port 1
VNA
Port 2
PLC adapter
Power
Network
Balun
Coupling
(a) Setup for measuring DM coupling.
Telephone
Network
(b) Setup for measuring CM coupling.
Figure 3.11: Setup for PLC-to-DSL coupling.
To study the stationarity of the PCL-to-DSL cross-coupling channels, measurements of the the channels have to be collected over time. An Agilent E5810A general
purpose interface bus (GPIB) controller is used to control an Agilent E5071B vector
network analyzer (VNA), via MATLAB as shown in Fig. 3.12. Note that the setup
shown in Fig. 3.12 is required when consecutive measurements of the channel are
needed. If the variation of the channel over time is not a concern, the GPIB is not
necessary, and thus, the setup shown in Fig 3.11 is sufficient.
In the literature, the background noise on the DM DSL line is usually assumed
dBm
to be -140
[4]. However, a similar assumption for the CM background noise
Hz
does not exist. Both the DM and the CM background noise levels are required to
study the performance of the adaptive filters utilized by the interference mitigation
solutions proposed in this thesis. The background noise present on the CM DSL line
was measured with a spectrum analyzer which showed the background noise on the
dBm
CM DSL channel is approximately -120
.
Hz
58
PLC adapter
Power
Network
Port 2
Balun
Telephone
Network
VNA
GPIB
Controller
MATLAB
Balun
Balun
Differential Mode
Setup
Common Mode
Setup
Coupling
Port 1
Figure 3.12: GPIB controlled measurement setup.
3.4 Measurement Campaign
The measurement campaign, introduced in this section, is composed of two case
studies: Case Study A and Case Study B. The measurements were conducted in two
residential test-sites (one site per case study).
Since no field measurements have been conducted in a residential house before
Case Study A, the case study was designed with two objectives in mind. The first
objective is identifying the level of interference between a DSL modem and a colocated PLC network in the home environment. Identifying and determining the
relationship between the DM and CM PLC-to-DSL cross-coupling channels within a
residential setting is the second objective of Case study A. Note that only the Desk
Modem Scenario was considered in Case Study A because the test-site’s wiring did
not permit measuring the cross-coupling channels for the Entry Point Scenario.
From Case Study A, the existence of strong frequency selective cross-coupling
channels between the DSL and PLC networks is identified. In addition, it was noticed
that the cross-coupling channels were not dependent on the spatial separation between
the PLC and DSL modems. However, Case Study A considered only the Desk Modem
59
Scenario. In addition, the variation of the cross-coupling channels over time was not
studied. Case Study B was a more extensive case study, where a set of 480 DM
and CM cross-coupling channel measurements between the DSL and PLC networks
is collected, within a residential house. Also, both DSL modem installation scenarios
discussed in Section 3.1 are considered in Case Study B.
The rest of this section is organized as follows. In Section 3.4.1, the test-sites
for both case studies are discussed, while the measurements results are presented in
Section 3.4.2.
3.4.1 Test-Sites
The test-sites for the two case studies of the measurement campaign are discussed in
this section. In Section 3.4.1.1, the test-site of Case Study A is presented, while the
test-site of Case Study B is discussed in Section 3.4.1.2
3.4.1.1 Case Study A
In this case study, only the Desk Modem Scenario discussed in Section 3.1, is considered. By not terminating the copper twisted-pair at the entrance of the house,
the distance over which the twisted-pair DSL line coexists with the PLC power lines
increases. This, in-turn, increases the coupling between the two systems.
Both DM and CM cross-coupling channels were measured in four rooms, all on
the same level, within a 1000 ft2 residential house. The floor plan of the house is
shown in Figure 3.13. The circle indicates the location of the telephone outlet (i.e.,
the location of the DSL modem within the house), while the triangles indicates the
various locations of the power outlets for which the cross-coupling channels were
studied. Note that in Case Study A, the variation of the PLC-to-DSL cross-coupling
channels over time is not considered, and thus, the setup shown in Fig 3.11 was
utilized.
60
40 ft
Room B
25 ft
Room D
Room C
Room A
Telephone Outlet
Power Outlet
Figure 3.13: Case Study A: test-site floor plan.
3.4.1.2 Case Study B
In Case Study B, a new measurement case study is conducted. The PLC-to-DSL
cross-coupling channels within a residential house are measured. Note that the setup
shown in Fig. 3.12 was used in this case study. A set of 480 measurements is collected
to characterize the DM and CM PLC-to-DSL cross-coupling channels, for both DSL
modem installation scenarios discussed in Section 3.1 (i.e., the Desk Modem Scenario
and the Entry Point Scenario).
The floor plan of the upper and lower floors of the residential single storey bungalow, which is used as a test-site, are shown in Fig. 3.14(a) and Fig. 3.14(b), respectively. The bungalow was built in 1930 and renovated to modern wiring standards in
1992, when the telephony network was professionally installed. Note that the distance
between the two floors is approximately 9 ft. Within each room, there are multiple
power outlets. Each power outlet where a PLC-to-DSL coupling measurement was
collected is labelled. Thus, Room A Plug 1 refers to the PLC-to-DSL cross-coupling
channels for Plug 1 in Room A. Note that the location of the DSL modem for the
61
Telephone
Plug
Room F
Power
Outlet
48 ft
Room E
Cable T.V.
1
1
Room G
Room I
Room1
H
1
Room J
DN
92 ft
(a) Upper floor.
3
2
Telephone
Plug
2
Power
Outlet
Room D
42 ft
Cable T.V.
1
Room A
1
Room C
3
Room B
2
1
UP
72 ft
(b) Lower floor.
Figure 3.14: Case Study B: test-site floor plan.
Desk Modem Scenario and the Entry Point Scenario within the measurement house
are in Room J in Fig. 3.14(a) and Room C in Fig. 3.14(b), respectively.
For each of the measured power outlets, 10 scatter matrix measurements are collected 0.5 seconds apart. This is performed for both the Desk Modem Scenario and
the Entry Point Scenario to study the changes in the CM and DM PLC-to-DSL crosscoupling channels over time. A single set of measurements, which consists of 1500
measurement points, is completed in 0.5 seconds. Both the smoothing (averaging over
frequency) and averaging (averaging over time) functions of the VNA were turned off
62
during the measurements, and the VNA was not synchronized with the AC cycle.
Given that the AC cycle has a duration of 1/60 seconds, over the 10 measurement
sets, each measurement point has a different location in the AC cycle. Thus, any variation detected in the PLC-to-DSL cross-coupling channel measurements is attributed
to the change in the loads of the power line network.
3.4.2 Results
This section presents the results of the measurement campaign. The frequency responses of the measured DM and CM PLC-to-DSL cross-coupling channels are presented in Section 3.4.2.1. The stationarity of the cross-coupling channels is discussed
in Section 3.4.2.2, while the effect of the spatial separation on the coupling gain and
coherent bandwidth of the cross-coupling channels is studied in Section 3.4.2.3.
3.4.2.1 Cross-Coupling Channel Frequency Responses
To determine the frequency response matrix for a given channel, the scatter matrix
between the ports of the VNA is used. In Section 3.4.2.1.1, the cross-coupling channels
measured in Case Study A are presented, while the cross-coupling channels measured
in Case Study B are discussed in Section 3.4.2.1.2.
3.4.2.1.1 Case Study A
Fig. 3.15 shows the DM PLC-to-DSL cross-coupling channels, for Rooms A, B, C,
and D. From Fig. 3.15 the following is noticed. First, the measurements indicate a
very high degree of DM coupling between the PLC and DSL channels. In addition,
it also indicates that the DM PLC-to-DSL cross-coupling channels are frequency
selective. The figure also shows that the relative distance between the telephone
outlet and the power outlet has no significant impact on the PLC-to-DSL coupling.
Fig. 3.16 shows the CM PLC-to-DSL cross-coupling channels, respectively, for
Rooms A, B, C, and D. Similar to the findings of the DM PLC-to-DSL cross-coupling
63
−30
DM coupling Channel Gain in dB
−40
−50
−60
−70
−80
Room A
Room B
Room C
Room D
−90
−100
0
5
10
15
20
Frequency in MHz
25
30
Figure 3.15: Coupling in Differential Mode.
channel measurements, the CM PLC-to-DSL cross-coupling channel measurements
show the expected high degree of coupling between the PLC and DSL channels in
the common mode. Additionally, Fig. 3.16 indicates that the CM PLC-to-DSL crosscoupling channel is frequency selective and the relative distance between the telephone
outlet and the power outlet has no significant impact on the PLC-to-DSL coupling.
Finally, by comparing Figs. 3.15 and 3.16, it is noted that the DM coupling is lower
than the CM coupling, which is expected since DM transmission is used to mitigate
the effects of EMI.
The CM to DM transfer function (C2DTF), which is defined as the ratio of the
DM to CM cross-coupling channels, is calculated by
G(i) =
gd (i)
gc (i)
(3.1)
where, G(i) is the C2DTF ratio for frequency bin i, and gd (i) and gc (i) are the DM
and CM channel gain vectors for frequency bin i respectively. Fig. 3.17 shows that
the C2DTF is quite different for each room. This is due to the tree topology of the
power line network within the house. The various branches of the power line network
cause either constructive or destructive interference to the DSL network. This tree
64
−20
CM coupling Channel Gain in dB
−30
−40
−50
−60
−70
Room A
Room B
Room C
Room D
−80
−90
0
5
10
15
20
Frequency in MHz
25
30
Figure 3.16: Coupling in Common Mode.
also causes the relationship between the DM and CM cross-coupling channels to vary
significantly from one power outlet to another. In essence, the topology of the network
changes from room to room which causes the variation in the C2DTF.
3.4.2.1.2 Case Study B
Fig. 3.18 shows the outlet-to-outlet variation in the DM PLC-to-DSL cross-coupling
channels, for the Desk Modem Scenario. Only 5 cross-coupling channels are shown in
Fig. 3.18 for the sake of clarity; however, variations among the DM and the CM PLCto-DSL cross-coupling channels for all labelled outlet in Fig. 3.14 have been studied.
It was observed that both the DM and CM PLC-to-DSL cross-coupling channels are
frequency selective and vary significantly from one power outlet to another, for both
the Desk Modem Scenario and the Entry Point Scenario. Note that, as discussed
in Section 3.4.2.1.1, the frequency selectivity of the cross-coupling channel, and its
variation from one outlet to the other, is due to the tree structure of the power lines
within the house.
Similar to Room A Plug 1, the CM and DM PLC-to-DSL cross-coupling channels for the rest of the labelled power outlets in Fig. 3.14 were measured, for both
65
30
20
C2DTF Gain in dB
10
0
−10
−20
−30
Room A
Room B
Room C
Room D
−40
−50
0
5
10
15
20
Frequency in MHz
25
30
Figure 3.17: Common mode to differential mode transfer function (C2DTF).
Desk Modem Scenario and Entry Point Scenario. From these measurements, it was
observed that the PLC-to-DSL cross-coupling channels are frequency selective, and
vary significantly from one power outlet to another. In addition, the relationship
between the DM and the CM cross-coupling channels is not the same for all power
outlets. Note that these findings agree with the finding of Case Study A.
To study the distribution of the coupling gains of the DM and CM PLC-to-DSL
cross-coupling channels, the empirical probability density functions (PDFs) of the
DM and CM PLC-to-DSL coupling gains, for both the Desk Modem Scenario and
the Entry Point Scenario are generated. To generate the PDF for each PLC-toDSL cross-coupling channel, a histogram of the gains at all frequency points, for
the measured channels, is generated and normalized such that area under the curve
tracing the heights of the histogram bins is equal to one.
66
−30
Channel Gain in dB
−40
−50
−60
−70
−80
−90
−100
0
5
10
15
Frequency in MHz
20
25
30
Figure 3.18: Variation in DM cross-coupling channels from one power outlet to another.
0.05
Emperical Probability Density Function
0.045
0.04
DM
Mean DM
CM
Mean CM
0.035
0.03
0.025
0.02
0.015
0.01
0.005
0
−120 −110 −100 −90
−80 −70 −60 −50
Coupling Gain in dB
−40
−30
−20
−10
Figure 3.20: Entry Point Scenario coupling gain probability density function.
For each scenario, the means of the DM and CM PLC-to-DSL cross-coupling
channels, represented by vertical lines in Figs. 3.19 and 3.20, are calculated. Note
that, on average, the DM coupling for the Entry Point Scenario is approximately
20 dB lower than the DM coupling for the Desk Modem Scenario, which indicates
the significant effect of the in-line LPF on the reduction of the DM interference.
67
0.05
Emperical Probability Density Function
0.045
0.04
DM
Mean DM
CM
Mean CM
0.035
0.03
0.025
0.02
0.015
0.01
0.005
0
−120 −110 −100 −90
−80 −70 −60 −50
Coupling Gain in dB
−40
−30
−20
−10
Figure 3.19: Desk Modem Scenario coupling gain probability density function.
However, as will be shown in Chapter 4, this reduction is not sufficient to prevent the
PLC system from degrading the performance of the DSL system.
−30
−40
Coupling Gain in dB
−50
−60
−70
−80
Desk Modem: Mean DM
Desk Modem: Mean CM
Entry Point: Mean DM
Entry Point: Mean CM
−90
−100
0
5
10
15
Frequency in MHz
20
25
30
Figure 3.21: Mean DM and CM PLC-to-DSL coupling.
The means of the DM and CM PLC-to-DSL cross-coupling channels for all rooms,
for both the Desk Modem Scenario and the Entry Point Scenario, are shown in Fig.
3.21. The mean for each scenario represent the average of the gains for a particular
frequency across all the measurement locations shown in Fig. 3.14, i.e., these plots
68
show the cross-coupling channels averaged across all outlets used to inject the PLC
signal. From Fig. 3.21, it is noted that both the DM and CM PLC-to-DSL crosscoupling channels are frequency selective. Furthermore, it is noted that the in-line
LPF present in the Entry Point Scenario reduces the amount of DM coupling relative
to the Desk Modem Scenario. However, the CM coupling for both scenarios is not
affected by the in-line LPF. In addition, it is evident that the CM coupling is higher
than the DM coupling, for both scenarios. This is expected since the DM reception
mode partially cancels EMI.
3.4.2.2 Stationarity of Cross-Coupling Channels
Figure 3.22: Desk Modem Scenario.: DM PLC-to-DSL coupling for Room A Plug 1
over measurement interval.
Figs. 3.22 and 3.23 show the changes in the DM PLC-to-DSL coupling for Room
A Plug 1 over time for the Desk Modem Scenario and the Entry Point Scenario
respectively. Recall, for each power outlet, the PLC-to-DSL cross-coupling channels
are measured for 10 consecutive times, with a separation interval of 0.5 seconds, to
study the changes in the channel over time.
69
Figure 3.23: Entry Point Scenario.: DM PLC-to-DSL coupling for Room A Plug 1
over measurement interval.
−20
−30
Channel gain in dB
−40
−50
−60
−70
−80
−90
−100
−110
DM Coupling
CM Coupling
0
5
10
15
20
Frequency in MHz
25
30
Figure 3.24: Desk Modem Scenario: Room A Plug 1 DM vs. CM PLC-to-DSL
coupling.
70
−10
−20
Channel gain in dB
−30
−40
−50
−60
−70
−80
−90
DM Coupling
CM Coupling
−100
−110
0
5
10
15
20
Frequency in MHz
25
30
Figure 3.25: Entry Point Scenario: Room A Plug 1 DM vs. CM PLC-to-DSL coupling.
For each scenario, the average channel gains of the DM and CM PLC-to-DSL
cross-coupling channels, for Room A Plug 1, are calculated, and the variations in the
channel gains over time is determined by calculating the standard deviation of the
channel gains at each frequency point. The average DM and CM PLC-to-DSL crosscoupling channels for Room A Plug 1, for the Desk Modem Scenario and the Entry
Point Scenario, are shown in Figs. 3.24 and 3.25 respectively. Also, the variations in
the channel gains over time are indicated via error bars in Figs. 3.24 and 3.25.
From Figs. 3.22, 3.23, 3.24, and 3.25 it is noted that the channel gain variations
of the PLC-to-DSL cross-coupling channels over time are insignificant. However,
over frequency, the variation of the PLC-to-DSL cross-coupling channel gains is quiet
significant. Also, it is evident that the CM coupling is higher than the DM coupling,
for both scenarios. In addition, it is noted that the DM and CM cross-coupling
channels are frequency selective, for both scenarios.
Additionally, for each room, the standard deviation for each frequency bin over
the 10 measurements is determined; after which, the standard deviation values are
71
averaged across all the rooms and then across all the frequency bins. The average
standard deviations for the DM and CM PLC-to-DSL cross-coupling channels for the
Desk Modem Scenario are 1.06 and 0.98 dB respectively, while for the Entry Point
Scenario the average standard deviations for the DM and CM PLC-to-DSL coupling
are 1.87 and 1.24 dB respectively. These numbers indicate that the variation in the
channel coupling gains over time is insignificant, and thus, the PLC-to-DSL crosscoupling channel is considered stationary.
The measurements were performed in a furnished house, albeit, most of the appliances were not actively drawing power during the measurements. Only the fridge,
deep freezer, and coffee maker were on during the measurement campaign. Thus,
the test-site was not very electrically active. As will be discussed in Chapter 5, the
proposed interference mitigation solutions presented in this thesis are insensitive to
changes in the cross-coupling channels due to the non-stationarity of the PLC channel. For a given outlet, any variations in the PLC interference is reflected in both
the DM and CM cross-coupling channels simultaneously. Additionally, the proposed
interference mitigation solutions utilize adaptive filters that estimate the ratio of the
DM to CM PLC-to-DSL cross-coupling channels. Since variations in the PLC interference simultaneously affect both the DM and CM PLC-to-DSL cross-coupling
channels, the ratio of the DM to CM PLC-to-DSL cross-coupling channels for a given
room remains constant over time.
3.4.2.3 Effect of Spatial Separation
Fig. 3.26 shows the average DM and CM PLC-to-DSL cross-coupling channel gains
versus the relative distance between the DSL and PLC modems for all the measurement locations. Each point represents the cross-coupling channel gain averaged
across the measurement frequency range. Note that the Euclidean distance between
the three dimensional positions of the DSL modem and the PLC modem is used in
72
−30
Average Coupling Gain in dB
−40
−50
−60
DM: Desk Modem
DM: Entry Point
CM: Desk Modem
CM: Entry Point
−70
−80
−90
0
10
20
30
40
50
60
70
Euclidean Distance in ft
80
90
100
Figure 3.26: Effect of relative distance between DSL and PLC modems on PLC–
to-DSL coupling.
Fig. 3.26. Information on the actual run length of the cables was unavailable. From
this figure, it is noted that the relative distance between the DSL modem and the
PLC modem does not have a significant impact on the coupling, since the coupling
levels do not decrease as the relative distance between the modems increase. This
is likely because the unshielded power lines radiate interference approximately uniformly throughout the home due to the length of the power lines and the relatively
small size of the home.
Finally, the 90% coherence bandwidth (CB) of both the DM and CM PLC-to-DSL
cross-coupling channels versus the Euclidean distance between the PLC and DSL
modem is shown in Fig. 3.27. The 90% coherence bandwidth (CB) was determined
by calculating the autocorrelation of each channel and determining the width of the
bandwidth at which the magnitude of autocorrelation drops to 90% of its maximum
value.
Table 3.1: Average coherence bandwidth in kHz.
Scenario
DM
CM
Desk Modem 157.1765 144.8123
Entry Point 282.4361 178.7780
73
300
DM: Desk Modem
CM: Desk Modem
DM: Entry Point
CM: Entry Point
Coherence Bandwidth in kHz.
250
200
150
100
50
0
10
20
30
40
50
60
Euclidean Distance in ft.
70
80
90
Figure 3.27: Effect of relative distance between DSL and PLC modems on cross-coupling channels coherence bandwidth.
It is noted that, on average, the CB for the DM and CM PLC-to-DSL crosscoupling channels for Desk Modem Scenario is lower than the CB for Entry Point
Scenario, as shown in Table 3.1. This due to the fact that by allowing the DSL signal
to travel within the house, more branches of the PLC tree contribute (constructively
or destructively) to the cross-coupling channels.
74
Chapter 4
INTERFERENCE SYSTEM MODEL
In Chapter 3, the cross-coupling channels between DSL and PLC networks within
a residential setting have been measured. Two case studies have been performed,
and among the findings was a strong DM cross-coupling channel between DSL and
PLC networks. As a result of this coupling, the mutual interference between two
co-located DSL and PLC networks, operating over the same frequency band, will
inevitably affect the performance of both systems. Thus, an interference mitigation
solution is required.
In this chapter, we quantify the effect of the mutual DSL and PLC interference
on the data rates of existing DSL and PLC systems. The current system model,
which describes the interference environment between PLC and DSL systems as it
exists today, is discussed in Section 4.1. The data rates achieved by a DSL modem
in presence of an in-home PLC network and the data rates achieved by an in-home
PLC network in presence of a DSL modem are studied in Section 4.2.
4.1 Current System Model
Within a residential house, the DSL and PLC systems utilize the twisted-pairs and the
power lines, respectively, as transmission mediums. The twisted-pairs are composed
of two twisted copper wires, designed to minimize both EMI egress and ingress. On
the other hand, power lines are made of two or three wires. The power lines are
neither twisted nor shielded, which result in significant electromagnetic radiation.
Fig. 4.1 shows the current system model for the PLC-to-DSL interference environment within a residential setting. In each house, there is a single DSL transceiver
75
DSL system
DSL Transceiver
Twisted-pair
DMT
Transceiver
Balun
Wide-band EMI
PLC-D.C.
Power Lines
PLC
Transceiver
PLC
Transceiver
PLC
Coupler
PLC
Transceiver
DMT
Transceiver
PLC Transceiver
PLC system
Figure 4.1: Current system model.
while there are multiple PLC transceivers. The measurement campaign presented
in Chapter 3 indicates that the cross-coupling channels between the DSL and PLC
systems vary significantly from one PLC transceiver to the other.
The DSL transceiver is connected to the copper twisted-pairs, over which the DSL
signal is carried to and from the house. The DSL transceiver is composed of a balun
that outputs a DM signal, which is then passed to a DMT transceiver, as shown in
Fig. 4.1. Similarly, the PLC transceivers form a network through the power lines
which run through the house. Each PLC transceiver is composed of a PLC coupler
that feeds its output to a DMT transceiver, as shown in Fig. 4.1. Note that the
76
structure of the DMT transceiver was discussed in Section 2.1.3.
4.2 Effect of Mutual DSL and PLC Interference on Bit Rates
The measured cross-coupling channels presented in Chapter 3 are utilized to visualize
the effect of the PLC interference on the received DSL signal in Section 4.2.1; while,
the degradation of the PLC system performance due to the DSL interference is studied in Section 4.2.2. Note that the measured cross-coupling channels are reciprocal;
however, as will be shown in the following sections, the effect of the PLC and the
DSL networks on one another is not.
4.2.1 Effect of PLC Interference on DSL Bit Rates
To determine the effect of the PLC interference on the received DSL signal, the measured PLC-to-DSL cross-coupling channels are used to calculate the power spectral
density (PSD) of the PLC interference. The PSD of the PLC interference is then
compared with the PSD of a typical received DSL signal. Note that the direct DSL
channel, used to determine the PSD of the received DSL signal, is obtained from
the standard two-port model defined in [29]. Similarly, the average of the measured
DM PLC-to-DSL cross-coupling channels, presented in Section 3.4 and shown in Fig.
3.21, is utilized to determine the PSD of the PLC interference.
The PSDs of the received DSL signal for various DSL cable run lengths versus the
PSD of the PLC interference, for both the Desk Modem Scenario and the Entry Point
Scenario, are shown in Fig. 4.2. The maximum transmit power of both the DSL and
PLC systems, as specified in their respective ITU-T standards, is multiplied by the
DSL channel and the mean DM PLC-to-DSL cross-coupling channel to determine the
dBm
PSD of the signals. For VDSL2, the maximum transmit power is -50
[2]; while
Hz
dBm
for in-home PLC networks, the maximum allowed transmit power is -60
[9].
Hz
77
−60
−80
dB m
Hz
−120
PSD in
−100
−140
−160
−180
−200
−220
0
DSL: Recieved at 2kft
DSL: Recieved at 4kft
PLC: Desk Modem Scen.
PLC: Entry Point Scen.
5
10
15
20
Frequency in MHz
25
30
Figure 4.2: Received DSL Signal versus PLC interference.
Fig. 4.2 illustrates two points. First is the extremely high level of PLC interference
on the DSL line for both the Desk Modem Scenario and the Entry Point Scenario.
Although, in the Entry Point Scenario, the in-line LPF reduces the PLC interference
levels, this reduction is not sufficient to prevent the DSL system from experiencing
low signal to interference plus noise ratio (SINR) over the VDSL2 spectrum.
The second observation is, while the PLC interference levels are unaffected by the
distance between the central office (or distribution point) and the house, the received
DSL signal power decreases as the distance increases. Thus, the PLC interference
does not only impact VDSL2 services, but also it poses a significant risk to DSL
services that utilize narrower spectrum, such as ADSL2+ (up to 2.2 MHz [39]) and
VDSL (up to 12 MHz [40]), as the DSL cable run increases.
Bit loading, where the number of bits allocated to each sub-channel is dependent
on the sub-channel’s SINR, is used in both DSL and PLC [22]. The number of bits
b(i) that can be loaded on frequency bin i is calculated by
γ(i)
,
b(i) = min bmax , log2 1 +
Γ
(4.1)
where γ(i) is the SINR of frequency bin i, Γ is the SNR gap, and bmax is the maximum
78
number of bits that can be allocated to a frequency bin. After determining the number
of bits that can be loaded on each frequency bin, the total bit rate R is then calculated
by
R = λ∆f
X
b(i),
(4.2)
i
where λ is the normalizing factor and ∆f is the sub-carrier spacing [27]. Note that the
value of bmax , λ, and ∆f varies from one system to the other. In presence of AWGN
dBm
, the signal to the interference plus noise power is used
with PSD of No = -140
Hz
to determine the available bits that can be loaded to each of the DSL sub-channels
via (4.1). After which, from (4.2), the total number of bits is determined. Note that
for VDSL2, bmax =15 bits , Γ=9.45 dB, λ=0.79, and ∆fDSL =8.6 kHz [27].
Recall, DSL utilizes FDD, where the sub-channels are grouped and the groups
are allocated to US and DS transmission in an alternating fashion. The number of
sub-channels per transmission direction varies from one band plan to the other. Note
that band plan ”998E30” [2], shown in Fig. 4.3, is utilized in this thesis.
Downstream
Upstream
Figure 4.3: US and DS frequencies for band plan ”998E30” [2].
Fig. 4.4 shows the available DSL bit rates vs the length of the DSL cable run
lengths. Six plots are shown in Fig. 4.4, where Upstream and Downstream indicate
79
the available bit rates for US and DS transmission direction respectively, based on the
bandwidth allocation shown in Fig. 4.3. Bound DSL denote the maximum available
bit rates in presence of AWGN only, i.e., in absence of PLC interference. Similarly,
Desk Modem and Entry Point indicate the available bit rates in presence of both
AWGN and PLC interference for both the Desk Modem Scenario and the Entry
Point Scenario, respectively.
250
Upstream: Bound DSL
Downstream: Bound DSL
Upstream: Desk Modem
Downstream: Desk Modem
Upstream: Entry Point
Downstream: Entry Point
Available Bit rates in Mbit/s
200
150
100
50
0
500
1000
1500
2000
DSL cable run length in ft
2500
Figure 4.4: Available DSL bit rates.
From Fig. 4.4, it is noted that the PLC interference effect on the DS transmission
is higher than the PLC interference effect on the US transmission. This is expected
since in DS transmission the DSL signal is attenuated as it travels from the central
office to the house over the twisted pairs, and thus, it arrives at low power levels at
the house.
The data rate loss percentage is calculated to determine the loss percentage in
each case. The loss percentage Loss % is calculated by
Loss % =
RBound − RInterference
× 100,
RBound
(4.3)
where RBound is the total number of bits in presence of background noise only and
80
RInterference is the total number of bits in presence of interference plus background
noise. Fig. 4.5 shows the degradation in DSL bit rates in terms of data rate loss
percentage. From this figure, it is note that DSL suffers up to 83% and 52% loss in
available data rates for DS transmission for the Desk Modem Scenario and the Entry
Point Scenario respectively. While, on the other hand, the highest loss in available
data rates for US transmission are 38% and 3% for the Desk Modem Scenario and
the Entry Point Scenario respectively. Note that for US transmission, the receiver is
at the central office; thus, the PLC interference is attenuated along with the desired
signal as it travels over the twisted pair.
90
80
Upstream: Desk Modem
Downstream: Desk Modem
Upstream: Entry Point
Downstream: Entry Point
Bitrate Loss Percentage
70
60
50
40
30
20
10
0
500
1000
1500
2000
DSL cable run length in ft
2500
Figure 4.5: Degradation in DSL bit rates.
Another observation from Fig. 4.5 is that the impact of the PLC interference on
the DS DSL available data rates depends significantly on the DSL cable run length.
For instance, the reduction in DS DSL bit rates for the Entry Point Scenario is 8% for
a DSL cable run length of 200 ft, which increases to 52% for a DSL cable run length
of 1600 ft, and then decreases to 31% for a DSL cable run length of 2600 ft. This is
explained as follows. For short DSL cable run length, the SINR is very large; thus,
the limiting factor on the number of bits that can be loaded onto each sub-channel
81
is bmax in (4.1). However, as the signal power decreases due to the attenuation from
travelling down the twisted-pairs, the PLC effect becomes more prominent, which
limits the available bit rates. As the DSL cable run length increases, the signal
power decreases significantly, especially at high frequency. This makes the AWGN
the limiting factor on the number of bits that can be loaded to each sub-channel.
Finally, it is noted that the impact of the PLC interference on the US DSL data
rates does not vary significantly with the DSL cable run length. This is because both
the DSL signal and the PLC interference originate from the house.
4.2.2 Effect of DSL Interference on PLC Bit Rates
To study the impact of the DSL network on a co-located in-home PLC network, the
same approach used in Section 4.2.1 is utilized. However, unlike the PLC signal that
is confined to the house, the DSL signal travels to and from the house (i.e., DS and
US traffic). Thus, the DSL interference impact on the received PLC signal has to be
studied for both traffic directions. Note that the US and DS frequencies are based on
the band plan ”998E30” [2], shown in Fig. 4.3.
−50
−100
PSD in
dBm
Hz
−150
−200
−250
PLC Desired Signal
DSL: Desk Modem Received at 2kft
DSL: Desk Modem Received at 4kft
DSL: Entry Point Received at 2kft
DSL: Entry Point Received at 4kft
−300
0
5
10
15
20
Frequency in MHz
25
30
Figure 4.6: Received PLC Signal versus DS DSL interference.
82
Fig. 4.6 shows the PSD of the received PLC signal versus the interference caused
by DS traffic for various DSL cable run lengths, for both the Desk Modem Scenario
and the Entry Point Scenario. Note that the effective DM DSL-to-PLC cross-coupling
channels and the direct PLC channel are utilized in the analysis presented in this
section. The effective DM DSL-to-PLC cross-coupling channels are obtained by multiplying the direct DSL channel defined in the two-port model in [29] by the mean
DM DSL-to-PLC cross-coupling channels shown Fig. 3.21. Note that the DM crosscoupling channels between the DSL and PLC systems are reciprocal. The direct PLC
channel is obtained from the model defined in [66].
From Fig. 4.6 it is noted that the DS DSL interference is below the DM noise
dBm
) for Entry point scenario; thus, it is negligible. However,
floor (i.e., below -140
Hz
for Desk Modem scenario, the DS DSL interference for short DSL cable run length
might affect some of the PLC sub-channels. This degradation will be more noticeable
if the PLC transmit power is reduced to mitigate the PLC-to-DSL interference, which
is the solution proposed by [27, 28]. As will be shown in Chapter 5, the solutions
proposed in this thesis do not have this negative impact on the PLC network.
Fig. 4.7 shows the PSD of the received PLC signal versus the interference cause by
US traffic for both the Desk Modem Scenario and the Entry Point Scenario. Note that
US DSL traffic is not affected by the DSL cable run length; thus, the mean DM DSLto-PLC cross-coupling channel is multiplied by the DSL maximum transmit power to
simulate the effect of the US DSL interference on the received PLC signal. From Fig.
4.7, it is noted that interference from the US DSL traffic on the received PLC Signal
is significantly higher than the DS DSL interference shown inf Fig. 4.6.
Similar to what was performed in Section 4.2.1, the ratio of the PLC signal power
to the DSL interference power is translated into achievable bit rates to quantify the
effect of the DSL interference on the PLC bit rates. From (4.1), the SINR is used
83
PLC Desired Signal
DSL: Desk Modem
DSL: Entry Point
−60
−80
PSD in
dBm
Hz
−100
−120
−140
−160
−180
0
5
10
15
20
Frequency in MHz
25
30
Figure 4.7: Received PLC Signal versus US DSL interference.
to determine the available bits that can be loaded to each of the PLC sub-channels.
Note that it is assumed that the AWGN on the PLC DM channel has a PSD of No
dBm
. The total number of bits that can be transmitted is then determined
= -140
Hz
from (4.2).
For PLC, bmax =12 bits , λ=0.75, and ∆fPLC =24.4 kHz. Recall, PLC utilizes
TDMA, where the channel time is divided into slots and only a single PLC user
can transmit at a time. However, since DSL utilizes FDD, the effect of the DSL
interference for both US and DS transmission, for band plan ”998E30” [2] shown in
Fig. 4.3, is considered.
Fig. 4.8 shows the available DSL PLC rates versus the length of the DSL cable.
Five plots are shown in Fig. 4.8. Upstream and Downstream indicate the available
PLC bit rates in presence of US and DS DSL interference respectively. Bound PLC,
Desk Modem, and Entry Point denote the available bit rates in presence of AWGN
only, in presence of both AWGN and DSL interference for Desk Modem Scenario, and
in presence of both AWGN and DSL interference for Entry Point Scenario, respectively.
84
250
Available Bit rates in Mbit/s
240
230
Bound PLC
Upstream: Desk Modem
Downstream: Desk Modem
Upstream: Entry Point
Downstream: Entry Point
220
210
200
190
500
1000
1500
2000
DSL cable run length in ft
2500
Figure 4.8: Available PLC bit rates.
From Fig. 4.8, it is noted that the DS DSL interference effect on the PLC signal
is lower than the US DSL interference effect on the PLC signal. This is due to the
attenuation of the received DSL signal as it travels down the length of the copper
twisted-pairs, and thus, it arrives at low power levels at the house.
25
Bitrate Loss Percentage
20
15
Upstream: Desk Modem
Downstream: Desk Modem
Upstream: Entry Point
Downstream: Entry Point
10
5
0
500
1000
1500
2000
DSL cable run length in ft
2500
Figure 4.9: Degradation in PLC bit rates.
85
This is confirmed by Fig. 4.9, where the DS DSL interference effect on the PLC
signal results in a loss in bitrates below 5%, except for short DSL cable run lengths
(less than 600 ft). On the other hand, US DSL interference causes a 20% loss in the
available PLC data rates for Desk Modem scenario for all DSL cable run lengths.
Note that the loss percentage is calculated by (4.3).
From the analysis conducted in this section and Section 4.2.1, it is concluded
that there is significant potential for PLC networks to degrade DSL performance in a
home environment, regardless of whether or not an in-line LPF is used for the home
telephone wiring. On the other hand, the interference from the DSL network on the
PLC network is not as significant; albeit, potential for PLC network performance
degradation due to a co-located DSL network does exit. This motivates the need for
the interference cancellation solutions presented in this Chapter 5.
86
Chapter 5
INTERFERENCE MITIGATION SOLUTIONS
Both DSL and PLC systems are subject to various sources of interference such as
radio frequency interference from amateur radios, impulsive noise, etc.; however, only
mutual interference between the DSL and the PLC systems, in presence of thermal
noise, are considered in this thesis. As discussed in Chapter 3, field measurements
indicate the presence of significant cross-coupling levels between DSL and PLC. This
level of coupling negatively impacts the performance of both DSL and PLC systems,
as was confirmed in Chapter 4. In this chapter, we introduce our proposed interference
mitigation schemes.
The rest of this chapter is organized as follows. The modified system model for
the interference environment between DSL and in-home PLC networks is presented
in Section 5.1, and in Section 5.2, the interference mitigation block diagram is introduced. Based on this block diagram, two interference mitigations solutions are
presented. A scheduling-based interference mitigation solution is presented in Section 5.3; while, a pre-distortion-based interference mitigation solution is discussed
in Section 5.4. Finally, in Section 5.5, the performance of the proposed interference
mitigation solutions are compared with the performance of the spectral management
solutions proposed in [27] and [28].
5.1 Modified System Model
In Chapter 4, the current system model of the interference environment between a
DSL modem and an in-home PLC networks have been discussed. In this section, a
modified system model is introduced. Recall, within a residential house, the DSL
87
modem (transceiver) is connected to the distribution point, outside the house, via a
twisted-pair. PLC transceivers, on the other hand, are connected to each other via
the power lines within the house. Traditionally, both networks are not connected;
however mutual wide-band EMI interference affects their performances as was shown
in Section 4.2.
The rest of this Section is organized as follows. The system model, which is the
basis of the proposed interference mitigation solutions, is presented in Section 5.1.1.
Time variations in the PLC interference and its effect on the estimate of the DM to
the CM cross-coupling channels ratio is discussed in Section 5.1.2. Finally, integration
of the FDIC within the structure of the DMT tranceiver is shown in Section 5.1.3.
5.1.1 Proposed System Model
Both DSL and PLC utilize DM transmission, where complementary signals are inserted via the transmitter over the transmission medium, and the difference between
the signals is obtained at the receiver as the desired signal. The CM signal, on the
other hand, is the arithmetic mean of the complementary signals. If EMI couples
identically on each of the wires of either the twisted-pairs or the power lines, the DM
signal will only contain the transmitted signal, while the CM signal will be composed
of only EMI. Since the CM signal contains no useful information about the signal, it
is usually ignored at the receiver. In practical situations, EMI does not couple identically on the wires, and thus, the DM signal is composed of the transmitted signal
and interference, while the dominant component of the CM signal is the interference.
Since the CM signal contains information about the EMI, we propose utilizing this
information to mitigate the effect of the EMI on the DM signal.
Fig. 5.1 shows the system model for our proposed scheme. In each house, there is
a single DSL transceiver while there are multiple PLC transceivers. One of the PLC
transceivers is assigned as a PLC-DC. This thesis proposes that the PLC transceiver
88
DSL system
DSL Transceiver
Balun
CM
Signal
Twisted-pair
PLC-D.C.
DM
∑
Signal
Power Lines
FDIC
PLC
Transceiver
PLC
Transceiver
DM ∑
PLC Signal
Coupler
PLC
Transceiver
CM
Signal
PLC system
DMT
Transceiver
DMT
Transceiver
FDIC
PLC Transceiver
Figure 5.1: Proposed system model.
closest to the DSL modem takes over as the PLC-DC. Note that the PLC-DC is connected to both the twisted-pair carrying the DSL signal and to the DSL transceiver
FDIC. Further detail on the functionality of these connections are presented in Sections 5.3 and 5.4.
Frequency domain interference cancellers (FDICs) are utilized by the interference
mitigation solutions presented in Sections 5.3 and 5.4. As shown in Fig. 5.1, there
is one FDIC per transceiver. The main role of the FDIC is to extract an estimate
89
of the DM interference from the CM signal. Each canceller is composed of multiple
single-tap filters. The number of single-tap filters per FDIC equals the number of subcarriers. Thus, the FDIC utilized by the DSL system has more single-tap filters than
the FDICs utilized by the PLC system because DSL systems utilize more sub-carriers
than a PLC system for a given bandwidth.
The tap coefficients of each FDIC are an estimate of ratio of the DM cross-coupling
channel to the CM cross-coupling channel. This ratio is utilized to estimate the DM
interference affecting the desired signal from the CM signal. This interference signal
estimate is the product of the CM signal and the ratio of the DM cross-coupling
channel to the CM cross-coupling channel. Thus, the FDIC installed in the DSL
transceiver is utilized to estimate and mitigate the PLC interference on the DSL
system; while, the FDIC installed in a PLC transceiver estimates and mitigates the
DSL interference on the PLC system. Note that the FDICs have to be trained to
obtain the ratio of the DM interference to the CM interference. The procedure used to
train the tap-coefficients of the FDICs depends on the utilized interference mitigation
solution. In this thesis, FDIC training is discussed in Sections 5.3 and 5.4.
The modified DSL transceiver is composed of a balun, an FDIC, and a DMT
transceiver; while, the modified PLC transceivers are composed of a PLC coupler, an
FDIC, and a DMT transceiver, as shown in Fig. 5.1. The balun outputs two signals:
DM signal and CM signal. The CM signal is passed through the FDIC to obtain
an estimate of the DM interference. This estimate is then subtracted from the DM
signal, before it goes through the DMT transceiver to retrieve the desired DSL signal.
Similarly, the PLC coupler converts the complementary signals transmitted over the
power lines into a DM signal and a CM signal. From the CM signal, an estimate
of the DM interference is obtained via the FDIC. This estimate is subtracted from
the DM signal; then, the resultant is then demodulated by the DMT transceiver to
90
obtain the desired PLC signal.
5.1.2 Variations in the DM to CM Estimated Ratio
After training the FDIC, the weights of its tap-coefficients remain valid until the PLC
user generating the interference changes. The measurements discussed in Chapter 3
indicate that the PLC-to-DSL cross-coupling channels are stationary when few appliances were drawing power. In the event that more appliances are actively drawing
power from the power lines the PLC-to-DSL cross-coupling channels might vary significantly within a DSL super-frame. However, despite the non-stationarity of the
PLC-to-DSL cross-coupling channels, the FDIC will still operate properly because
the FDIC tap-coefficients are an estimate of the ratio of the DM interference to the
CM interference.
k.s
s
Balun
j.s
Interference
DM
Interference
s.(k-j)
CM
Interference
s.(k+j)/2
Figure 5.2: Interference coupling on twisted-pair.
Fig. 5.2 illustrates how the radiated interference is coupled on a DSL twistedpair, where s is the interference radiated from the power lines, and k and j are the
imbalance factors on the twisted-pair wires. Any variation in the PLC cross-coupling
channel would manifest as a variation in s, and any variation in s will change both
DM and CM interference by the same amount. As a result, the variation would cancel
in the ratio of DM to CM interference which, in Fig. 5.2, would be s(k − j) divided
by s(k + j)/2.
Note that the PLC channel is a known for its non-stationarity. In addition to the
PLC interference, impulsive noise on the power lines cause by in-home appliances will
couple through the cross-coupling channels. As discussed in Section 2.3, the PSD of
91
dBm
the impulsive noise is -105
, while the transmit power of the PLC signal is -50
Hz
dBm
. Thus, the dominant interferer to DSL in the home environment is the PLC
Hz
signal. In this thesis, the PLC noise sources are not considered in the analysis of the
PLC-to-DSL interference mitigation solutions.
5.1.3 Integration of the FDIC
DSL
Symbols
DSL Signal
Direct
DSL
Channel
+ ∑
- +
∑
-
AWGN
∑
Filter &
Remove
CP
S/P
FFT
FEQ
+
PLC
Symbols
DM
PLC-to-DSL
Channel
QAM
Decoder
P/S
Bit
Stream
∑
-
AWGN
CM
PLC-to-DSL
Channel
∑
Filter &
Remove
CP
S/P
FDIC
FFT
PLC Interference
Figure 5.3: Integration of the FDIC into a DMT transceiver.
To accommodate the FDIC, a modified version of the DMT transceiver discussed
in Section 2.1.3 is shown in Fig. 5.3. As shown in this figure, the FDIC can be easily
integrated into the current DSL modems. The CM signal is passed through a receiver
filter, where the out-of-band noise is minimized and the signal is converted to digital
via a digital to analog converter. After that, the cyclic prefix is removed and the
serial samples are converted to a set of parallel sub-symbols, which are fed to an FFT
block. Up to this point, both the DM and CM signal paths have the same blocks.
At the CM signal path, the output of the FFT is fed to the FDIC, whose output
is subtracted from the output of the FFT block utilized by the DM signal. The result
of the subtraction, i.e., the estimate of the desired DSL signal, is passed through an
FEQ. The output of the FEQ is an estimate of the transmitted DSL symbols, which
92
are then fed to a QAM decoder to determine the transmitted bits. By comparing
Figs. 2.4 and 5.3, it is noted that most of the modification occurs in the CM signal
path.
5.2 Interference Mitigation Block Diagram
For the rest of this chapter, a boldface lower case letter will indicate a vector, while a
boldface upper case letter will denote a matrix. When referring to the DSL system,
all vectors are M ×1 column vectors and all matrices are M ×M square diagonal
matrices, where M is the number of DSL sub-carriers. Similarly, all vectors and
matrices that belong to the PLC system are N ×1 column vectors and N ×N square
diagonal matrices respectively, where N is the number of PLC sub-carriers.
When needed, a superscript is added to variables to indicate whether variables
belongs to the DSL or the PLC system. The notation x(i) indicates the element in the
ith row of the column vector x. Similarly, the notation H(i, j) refers to the element
in the ith row and the j th column of the square diagonal matrix H. To indicate the
complex conjugate of u, the notation u∗ will be used. Finally, the notation eb and e
will be utilized to indicate the estimate and the average of e, respectively.
DSL and PLC utilize bit loading on orthogonal sub-carriers, based on the subcarrier’s signal to interference plus noise ratio (SINR). The frequency separation
between PLC sub-carriers ∆fPLC is usually greater than the frequency separation
between DSL sub-carriers ∆fDSL [9]. Thus, over a given spectrum, the number of
PLC sub-carriers is smaller than the number of DSL sub-carriers. This means the
cross-coupling channel is sampled at slightly different frequencies by the DSL and
PLC systems.
The cross-coupling channels between the DSL and the PLC systems are shown
in Fig. 5.4. The DM and CM cross-coupling channels between a PLC transmitter
93
ηd,DSL
x
HDSL
rd,DSL
y
∑
vd,DSL
vc,DSL
^
y
∑
v^ d,DSL
rc,DSL
∑
ηc,DSL
CDSL
DSL system
Δf
PLC
Hc,DTx-PRx
Δf
DSL
Hd,PTx-DRx
H
H
Interference
Channels
ΔfPLC
d,DTx-PRx
ΔfDSL
c,PTx-DRx
PLC system
vd,PLC
q
HPLC
z
vc,PLC
ηc,PLC
rd,PLC
∑
ηd,PLC
rc,PLC
∑
^
z
∑
v^ d,PLC
CPLC
Figure 5.4: Interference cancelling scheme block diagram.
and the DSL receiver, sampled at integer multiples of ∆fDSL , are denoted by the
∆fDSL
DSL
channel frequency response matrices H∆f
d,PTx−DRx and Hc,PTx−DRx . A signal transmit-
ted through these channels correspond to the electromagnetic waves radiated by a
PLC transceiver, which then couples onto the twisted-pair connected to a balun that
outputs a DM and a CM signal, as shown Fig. 5.1. Note that since the DM and CM
PLC-to-DSL cross-coupling channels are sampled at integer multiples of ∆fDSL , both
∆fDSL
DSL
H∆f
d,PTx−DRx and Hc,PTx−DRx are M ×M square diagonal matrices.
94
Similarly, the DM cross-coupling channel between the PLC transmitter and the
DSL receiver when sampled at integer multiples of ∆fPLC is defined by the channel
PLC
frequency response matrix H∆f
d,PTx−DRx , which is an N ×N square diagonal matrices.
∆fPLC
DSL
Note that while both H∆f
d,PTx−DRx and Hd,PTx−DRx define the same channel, these
two matrices are not equal due do the difference in the sampling frequencies (i.e.,
∆fDSL 6= ∆fPLC ).
∆fPLC
PLC
The channel frequency response matrices H∆f
d,DTx−PRx and Hc,DTx−PRx define the
DM and CM cross-coupling channels between the DSL transmitter and the PLC
transceiver, when sampled at integer multiples of ∆fPLC . Note that since the DM
and CM DSL-to-PLC cross-coupling channels are sampled at integer multiples of
∆fPLC
PLC
∆fPLC , both H∆f
d,DTx−PRx and Hc,DTx−PRx are N ×N square diagonal matrices.
Fig. 5.4 also shows the block diagram for the DSL and the PLC systems. The
desired DSL signal at the customer’s premise is y, which is given by
y = HDSL x,
(5.1)
where x is the transmitted DSL symbol and HDSL is the channel frequency response
matrix for the direct DSL channel. Note y is received at the house via the twisted-pair
shown in Fig. 5.1.
Similarly, the desired PLC signal is z and is defined by
z = HPLC q,
(5.2)
where HPLC is the channel frequency response matrix for the direct PLC channel and
q is the transmitted PLC symbol. Note y is transmitted within the house from the
PLC transmitter to the PLC receiver via the power lines shown in Fig. 5.1.
The PLC interference is composed of the transmitted PLC symbol q radiated
within the house and picked by the twisted-pair. At the DSL receiver, the DM PLC
interference vd,DSL can be seen as the signal q transmitted through the differential
95
DSL
PLC-to-DSL cross-coupling channel H∆f
d,PTx−DRx . Similarly, the CM PLC interference
vc,DSL is the signal q transmitted through the common mode PLC-to-DSL crossDSL
coupling channel H∆f
c,PTx−DRx . Thus, the DM PLC interference vd,DSL and the CM
PLC interference vc,DSL are defined by (5.3a) and (5.3b) respectively. The DM and
CM PLC interference corresponds to the interference component of the DM and CM
DSL signals at the output of the balun shown in Fig. 5.1.
DSL
vd,DSL =H∆f
d,PTx−DRx q
(5.3a)
DSL
vc,DSL =H∆f
c,PTx−DRx q
(5.3b)
Moreover, the DSL interference is composed of the received DSL symbol y radiated within the house and picked up by the power lines that form the in-home PLC
network. At the PLC receiver, the DM PLC interference vd,PLC (shown in (5.4a))
is the signal y transmitted through the differential DSL-to-PLC cross-coupling chanPLC
nel H∆f
d,DTx−PRx , while the CM PLC interference vc,DSL (shown in (5.4a)) is the sig-
nal y transmitted through the common mode DSL-to-PLC cross-coupling channel
PLC
H∆f
c,DTx−PRx . Both vd,DSL and vc,DSL are the interference component of the DM and
CM PLC signals at the output of the PLC coupler shown in Fig. 5.1.
PLC
vd,PLC =H∆f
d,DTx−PRx y
(5.4a)
PLC
vc,PLC =H∆f
c,DTx−PRx y
(5.4b)
The received DSL DM signal rd,DSL , shown at the output of the balun in Fig. 5.1,
is composed of the desired DSL signal y, the DM PLC interference vd,DSL , and the
DM additive white Gaussian noise η d,DSL , as shown in (5.5a). The received DSL CM
signal rc,DSL , also shown at the output of the balun in Fig. 5.1, is composed of only
96
the CM PLC interference vc,DSL and the CM additive white Gaussian noise η c,DSL ,
as shown in as shown in (5.5b).
rd,DSL =y + vd,DSL + η d,DSL
(5.5a)
rc,DSL =vc,DSL + η c,DSL
(5.5b)
Similarly, the received DM PLC signal rd,PLC , which is given in (5.6a), is the
summation of the desired PLC signal z, the DM DSL interference vd,PLC , and the
DM additive white Gaussian noise η d,PLC . However, the received CM PLC signal
rc,PLC , shown in (5.6b), is composed of only the CM DSL interference vc,PLC and the
CM additive white Gaussian noise η c,PLC . Both rd,PLC and rc,PLC are the DM and
CM PLC signals at the output of the PLC coupler in Fig. 5.1.
rd,PLC =z + vd,DSL + η d,DSL
(5.6a)
rc,PLC =vc,DSL + η c,DSL
(5.6b)
The tap-coefficients matrices of the adaptive frequency domain interference cancellers C-DSL and C-PLC are shown in Fig. 5.4. Note that C-DSL and C-PLC
correspond to the FDICs utilized by the DSL and PLC systems, shown in Fig. 5.1,
respectively. The tap-coefficients of the canceller C-DSL are defined by the M ×M
square diagonal matrix CDSL , while the tap-coefficients of the canceller C-PLC are
defined by the N ×N square diagonal matrix CPLC . The estimate of the DM PLC
bd,DSL is extracted from rc,DSL via CDSL as shown in (5.7a), while the
interference v
bd,DSL is extracted from rc,PLC via CPLC as
estimate of the DM DSL interference v
shown in (5.7b).
97
bd,DSL =CDSL rc,DSL = CDSL vc,DSL + η c,DSL
v
(5.7a)
(5.7b)
bd,PLC =CPLC rc,PLC = CPLC vc,PLC + η c,PLC
v
The canceller C-DSL is utilized to mitigate the effects of vd,DSL on rd,DSL . The
b in Fig. 5.4, which is the difference between the received DSL signal rd,DSL
signal y
bd,DSL , is defined by (5.8a). Similarly, the canceller
and the output of the canceller v
C-PLC is utilized to mitigate the effects of vd,PLC on rd,PLC . The signal b
z in Fig.
5.4 is the difference between the received PLC signal rd,PLC and the output of the
bd,PLC , as shown in (5.8b).
canceller v
b =rd,DSL − CDSL rc,DSL
y
(5.8a)
b
z =rd,PLC − CPLC rc,PLC
(5.8b)
The manner in which the tap-coefficients of the FDICs are trained, in this thesis,
vary according to the mitigation solution in which they are utilized. In Sections
5.3 and 5.4, two interference mitigation solutions are presented. For each solution,
the FDICs are trained and their abilities to negate the effect of the cross-coupling
channels are evaluated.
5.3 Scheduling-Based Interference Mitigation Solution
In this section, the first interference mitigation solution, which is based on scheduling
the PLC users’ access to the PLC channel along with utilizing an adaptive FDIC, is
introduced. The medium access techniques used by both the DSL and PLC networks,
as well as the proposed modification to the PLC medium access, is discussed in
Section 5.3.1. The proposed cancellation algorithm utilized to mitigate the PLC-to98
DSL interference is presented in Section 5.3.2; while, its performance in mitigating
the PLC-to-DSL interference is analyzed in Section sub:PerformanalysisI.
Note that this interference cancellation scheme does not require modification to
either G.993.2 [2] or G.hn [9], which are the ITU standards for VDSL2 and home networks respectively. Although, the proposed cancellation scheme would require some
changes in the silicon design of existing DSL and PLC customer premise equipment,
these changes are limited the receiver section of the DSL modem and the PLC domain
controller. The necessary modifications to the DSL and PLC transceivers have been
shown in Fig. 5.3.
5.3.1 Medium Access
The frame structure of both the DSL and the PLC networks are discussed in this
section. This section also describes how the PLC-DC schedules the transmissions of
PLC users in order to facilitate interference cancellation.
Data is transmitted over the DSL channel in super-frames. For VDSL2, a superframe has a duration of TSF = 64.25 ms [2]. The super-frame is divided into 257
frames, the last of which is the synchronization frame [2]. Since each DSL user
utilizes a dedicated physical link, all super-frames on the DSL channel belong to a
single user.
The PLC channel, on the other hand, is a shared medium divided into time-slots
of duration TP [9]. Only a single PLC user can transmit per time-slot. One of the
PLC users takes over as a PLC-DC and allocates time-slots for various users, based
on the quality of service requirements of each user.
In order for the FDIC to successfully mitigate the effects of the PLC interference,
the PLC network needs to restrict channel access so only one PLC user is allowed
to transmit per DSL super-frame. It was shown in Chapter 3 that the PLC-to-DSL
cross-coupling channels vary significantly from one outlet to the next. Thus, the
99
trained FDIC tap-weights are only viable for a specific PLC user. As a result, the
FDIC has to be retrained to compensate for the change in the cross-coupling channel
each time a different user starts to transmit. As will be discussed in the next section,
the FDIC can only be trained during the super-frame sync symbols. Thus, it is
essential that only the PLC user for which the FDIC was trained transmits during a
given super-frame.
To prevent more than one PLC user from transmitting during a given super-frame,
the PLC-DC is connected to the DSL line, as shown in Fig. 5.1. The PLC-DC does
not decode the full super-frame but does synchronize itself to it. It then schedules
the PLC users so that only one user transmits per DSL super-frame.
The retransmission and acknowledgement protocol of ITU-T G.hn [9] standard
specifies two acknowledgement techniques for uni-cast transmission, namely: immediate acknowledgement (Imm-ACK) and delayed acknowledgement (delayed-ACK).
When Imm-ACK is specified by the transmitter, the receiver is required to transmit
an Imm-ACK frame after the successful reception of a given frame; when delayedACK is required, the receiver delays the acknowledgement frame until it is assigned
a transmission opportunity (TXOP). In this thesis, it is assumed that delay-ACK
procedure is employed by the PLC network, which would allow a DSL super-frame
duration to hold both the current PLC data frame and the ACK frame for a previous
transmission. Thus, an entire super-frame duration is not dedicated to a relatively
short ACK frame.
Given a delayed-ACK procedure, the latency experienced by a PLC user in our
proposed solution is dependent on the number of PLC users sharing the PLC network
and on the duration of the DSL super-frame. Assuming a round-robin style of access,
the latency experienced by a user is TLAT,Scheduling = (n−1)TSF , where n is the number
of users and TSF =64.25 ms [2]. Using this simple model, it is possible to satisfy the
100
latency requirements for the multimedia applications in [67], which is between 100
and 300 ms, with up to 5 active PLC transceivers, as shown in Table 5.1. Note D is
the number active PLC users.
Table 5.1: Latency: Scheduling-based solution.
D (active users) TLAT,Scheduling (ms)
2
64.25
3
128.5
4
192.75
5
257
It is possible that this PLC scheduling scheme may reduce PLC throughput if the
transmission from a particular user is much shorter than a DSL super-frame. However,
this loss in throughput will be more than made up by allowing PLC networks full
use of the DSL operating spectrum. Furthermore, while VDSL2 has a super-frame
duration of 64.25 ms [2], it is expected that the super-frame duration for G.Fast will
be less than 10 ms [8].
5.3.2 Cancellation Algorithm
The interference cancelling scheme is composed of two phases: training phase and
cancellation phase. The objective of the training phase is to determine a ratio between
the differential mode interference and the common mode interference. This ratio
allows the FDIC to use the CM signal to estimate the DM interference component in
order to subtract it off of the desired signal.
As discussed in Section 5.2, the C-DSL and C-PLC are utilized to estimate a ratio
between the DM and CM cross-coupling channels. The measurements in Chapter
3 show that the relationship between DM and CM cross-coupling channels is very
frequency selective. However, since the FDICs processes each of the sub-channels
101
independently, this frequency selectivity does not affect the interference cancellation
operation.
To trigger the training process of the C-DSL, interference from the PLC network
has to be present. Similarly, the training of C-PLC only occurs if and only if the DSL
interference is significant. Various techniques could be used to trigger the training
process. For instance, the interference could be identified by applying a threshold
to the CM signal, which if surpassed the training of the FDICs begins. A similar
concept is proposed in [24], where once the CM interference is detected, the FDIC is
trained and utilized to remove the DM interference. In this scheme, however, since
the PLC-DC is connected to both the DSL and PLC transceivers, it is responsible
for triggering the training process.
Note that C-PLC does not require re-training unless a change in the physical
channel occurs, while C-DLC has to be re-trained each time a new PLC user occupies
the channel. This is due to the fact that there is only one DSL transceiver and its
effect on each of the PLC transceiver does not change unless the location of the PLC
transceiver changes or a change in the DSL channel occurs. On the other hand, as
shown in Chapter 3 the effect of the PLC transceivers vary significantly from one
power outlet to the other [14, 13].
5.3.2.1 C-DSL Training
C-DSL estimates the ratio of the DM interference to the CM interference by dividing
the DM signal by the CM signal and then averaging the result. In order for training to
occur, it is necessary to subtract the desired signal component from the DM received
signal. This is possible only when the desired signal is known. Thus, training of the
C-DSL occurs only during the sync frame of each DSL super-frame when y can be
correctly estimated and subtracted from the received DM signal rd,DSL .
As the C-DSL trains, interference will be present in the DSL super-frame sync
102
symbols and the synchronization of the DSL super-frame will degrade. However,
Section 5.3.3 will show that the C-DSL trains extremely quickly, with acceptable
performance achieved after only a handful of symbols. As a result, it is assumed the
C-DSL converges quickly enough for the sync symbols to benefit from interference
cancellation so that DSL modem synchronization will not be adversely affected.
After removing the DSL desired signal component, the received DM signal rd,DSL
defined (5.5a) is reduced to
rd,DSL = vd,DSL + η d,DSL .
(5.9)
The C-DSL tap-coefficient for sub-channel i during the k th symbol index, CDSL,k (i, i),
is calculated by
CDSL,k (i, i) =
rd,DSL (i)
vd,DSL (i) + ηd,DSL (i)
=
rc,DSL (i)
vc,DSL (i) + ηc,DSL (i)
(5.10)
Note that since the C-DSL tap-coefficient matrix is a square diagonal matrix, CDSL (i, j) =
0 when j 6= i.
To minimize the effect of the background noise on the estimate of the ratio of
the DM interference to the CM interference, we average CDSL,k (i, i) over K symbols.
This average is utilized in the cancellation phase of the interference cancelling scheme.
Thus, for sub-channel i, the averaged C-DSL tap-coefficient C DSL,K (i, i), defined by
(5.11), is the estimate of the ratio of the DM interference to the CM interference.
K
1 X
C DSL,K (i, i) =
CDSL,k (i, i)
K k=1
(5.11)
After training the C-DSL, an estimate of the DM interference is extracted from
bd,DSL = CDSL,K rc,DSL ). This estimate of
the CM signal through the C-DSL (i.e., v
the DM interference is subtracted from the corresponding DM signal rd,DSL (i) to
extract the DSL signal y(i). Finally, the DSL signal is passed through a frequency
domain equalizer (FEQ) to calculate the estimate of the transmitted DSL symbol
103
x
b(i). Note that the FEQ coefficients are the inverse of the the elements of the DSL
channel frequency response matrix. The effectiveness of this cancellation scheme
will be demonstrated in Section 5.3.3 using measured PLC-to-DSL cross-coupling
channels.
5.3.2.2 C-PLC Training
In Chapter 4, it was shown that the DSL interference effect on the PLC network is
not as substantial as the effect of the PLC interference on the performance of the DSL
network; however, as was shown in Fig. 4.9, interference due to US DSL transmission
causes up to 20% loss in the achieved PLC data rates. Thus, an adaptive frequency
domain interference canceller to mitigate the effects of the DSL-to-PLC interference
is required to mitigate the effect of the DSL interference on the PLC system.
Each time a PLC transceiver joins the PLC network, all PLC transceivers are
forced into listening mode to estimate the interference caused by the DSL transceiver.
Note that it was found in [13] that the cross-coupling channel between the PLC
and DSL systems are stationary; thus, only the PLC transceiver joining the PLC
network requires training. However, the PLC transceivers that previously trained
their cancellers, if needed, can retrain their respective cancellers during the listening
mode. Note that, unlike C-DSL, there is no restriction on when the C-PLC should
be trained.
DSL
The ratio of the DM to the CM cross-coupling channels (i.e., H∆f
d,DTx−PRx to
DSL
H∆f
c,DTx−PRx in Fig.
5.4) is estimated by each PLC transceiver.
Since all PLC
transceivers are listening to the PLC channel, rd,PLC is reduced to vd,PLC and η d,PLC ,
while rd,PLC is equal to vc,PLC and η c,PLC . Note that vd,PLC and Vc,PLC are equal to
∆fDSL
DSL
H∆f
d,DTx−PRx y and Hc,DTx−PRx y respectively, sampled at integer multiples of ∆fDSL .
The training of C-PLC is similar to the training of C-DSL, which has been discussed in Section 5.3.2.1. In essence, to estimate the ratio of the DM to CM DSL-
104
to-PLC cross-coupling channels for the ith sub-carrier, while in listening mode, each
PLC transceiver divides rd,PLC (i) by rc,PLC (i) to obtain the tap-coefficient CPLC (i, i, p)
during the pth symbol interval, as shown in (5.12a). Note that since the background
noise has an effect on C PLC , the tap-coefficient for each sub-carrier is averaged over
P consecutive symbols to minimize the effect of the background noise on the estimate
CPLC (i, i, p), as shown in (5.12b).
vd,PLC (i) + ηd,PLC (i)
rd,PLC (i)
=
rc,PLC (i)
vc,PLC (i) + ηc,PLC (i)
P
1 X
CDSL (i, i, p)
C PLC,P (i, i) =
P p=1
CPLC (i, i, p) =
(5.12a)
(5.12b)
5.3.3 Performance Evaluation
In this section, we evaluate the efficacy of the interference cancelling scheme presented
in Section 5.3.2 in mitigating the effect of the DM PLC interference on the DSL signal.
Recall, our proposed interference cancelling scheme estimates the DM interference
from the CM signal. The DSL and both the DM and CM cross-coupling channel
frequency response matrices are required to evaluate the performance of the proposed
cancelling scheme. The DSL channel frequency response matrix HDSL is obtained
from the standard two-port model defined in [29]. However, the measured DM and
CM PLC-to-DSL cross-coupling channels (i.e., matrices Hd,PTx−DRx and Hc,PTx−DRx
respectively), which were presented in Chapter 3, are utilized to determine the PLC
interference.
The maximum transmission power for DSL symbols and PLC symbols are εD =
−60 dBm
[2] and εP = −50 dBm
[9] respectively. Note that we assume that the backHz
Hz
ground noises on the DM and CM channels are AWGN and not correlated. The PSDs
of η d,DSL and η c,DSL are N0,d and N0,c , respectively. In the literature, N0,d is usually
assumed to be -140
dBm
.
Hz
As shown in Chapter 3, the measured background noise on
105
the DSL line in the common mode setup, N0,c , has a PSD of approximately -120
dBm
.
Hz
The interference cancelling scheme is evaluated in three steps. In Section 5.3.3.1,
the mean square error (MSE) of the proposed scheme when the C-DSL utilized is
determined and compared with the MSE of the proposed scheme when an optimum
Wiener filter is utilized. After determining the MSE, the SINR after utilizing the
interference cancelling scheme is used to calculate the achieved improvement in available bit rates, which is then compared with the bit rates in absence of the cancellation scheme in Section 5.3.3.2. Finally, in Section 5.5, the performance of the
scheduling-based interference mitigation solution is compared with the performance
of the spectral management mitigation solutions proposed in [27] and [28].
5.3.3.1 Mean Square Error Analysis
When determining MSE for both the Desk Modem Scenario and the Entry Point Scenario, the measured DM and CM PLC-to-DSL cross-coupling channels are assumed
to be the noise free actual channels. Using these channels, our proposed cancelling
scheme estimates the DM interference from the CM interference in presence of simulated background noise, represented by η d,DSL and η c,DSL in Fig. 5.4.
As discussed in Section 5.3.2, the proposed interference cancelling scheme utilizes
the C-DSL to estimate the ratio of the DM interference to the CM interference. From
(5.3), the DM PLC interference for the ith sub-channel is defined as
vd,DSL (i) = Hd,PTx−DRx (i, i)q(i),
(5.13)
while the CM interference for the ith sub-channel is calculated by
vc,DSL (i) = Hc,PTx−DRx (i, i)q(i).
(5.14)
The coefficient of the canceller C(i, i) (5.11), which is an estimate of the ratio of the
DM interference to the CM interference, is multiplied by the corresponding CM signal
106
to yield an estimate of the DM PLC interference vbd,DSL (i). Thus, the estimate of the
DM PLC interference for the ith sub-channel is calculated by
vbd,DSL (i) = C(i, i) (vc,DSL (i) + ηc,DSL (i)) .
(5.15)
Note that the MSE has the same units as the square of the estimated quantity and
is therefore expressed in
dBm
.
Hz
After training the canceller, the estimation error for sub-channel i, which is the
difference between vd,DSL (i) and its estimate vbd,DSL , is given by
e(i) = vd,DSL (i) − vbd,DSL =vd,DSL (i) − C(i, i) (vc,DSL (i) + ηc,DSL (i)) .
(5.16)
Thus, the MSE for sub-channel i, φ(i) = E[e(i)e∗ (i)], is defined by (5.17).
φ(i) =E[vd,DSL (i)u∗d,DSL (i)] − E[u∗d,DSL (i)C(i, i) (vc,DSL (i) + ηc,DSL (i))]
− E[vd,DSL (i)(C(i, i) (vc,DSL (i) + ηc,DSL (i)))∗ ]
(5.17)
+ E[C(i, i)C
∗
∗
(i, i)vc,DSL (i)vc,DSL
(i)]
∗
+ E[C(i, i)C ∗ (i, i)ηc,DSL (i)ηc,DSL
(i)]
It is assumed that all the signals are complex wide sense stationary with zero
mean, and the PLC symbol transmit power is εP . Thus, E[vd,DSL (i)u∗d,DSL (i)] is
equal to E[qq ∗ ]|Hd,PTx−DRx (i, i)|2 , where E[qq ∗ ] = εP . Similarly, E[vc,DSL (i)u∗c,DSL (i)]
is equal to εP |Hc,PTx−DRx (i, i)|2 . Furthermore, the term E[vd,DSL (i)u∗c,DSL (i)] is simpli∗
fied to E[qq∗]Hd,PTx−DRx (i, i)Hc,PTx−DRx
(i, i), where as the term E[u∗d,DSL (i)vc,DSL (i)]
∗
is simplified to E[qq∗]Hd,PTx−DRx
(i, i)Hc,PTx−DRx (i, i), respectively. Finally, the term
∗
E[ηc,DSL (i)ηc,DSL
(i)] is equal to the PSD of the CM background noise N0,c . Thus,
107
(5.17) simplifies to
∗
φ(i) =εP |Hd,PTx−DRx (i, i)|2 − εP C(i, i)Hc,PTx−DRx (i, i)Hd,PTx−DRx
(i, i)
∗
− εP C ∗ (i, i)Hc,PTx−DRx
(i, i)Hd,PTx−DRx (i, i) + εP |C(i, i)|2 |Hc,PTx−DRx (i, i)|2
+ |C(i, i)|2 N0,c .
(5.18)
The canceller tap-coefficient C(i, i) affects the value of the MSE. Thus, the weights
C DSL,K (i, i) calculated by (5.11) are substituted in (5.18) to calculate the MSE achieved
by the C-DSL, which is denoted φF,K (i). Similarly, to calculate the MSE achieved by
an optimum Wiener filter φW (i), the weights CW (i, i) are substituted in (5.18). Note
that CW (i, i) (5.19) is determined from the Wiener-Hopf equation [68].
CW (i, i) =
∗
εP Hd,PTx−DRx (i, i)Hc,PTx−DRx
(i, i)
2
εP |Hc,PTx−DRx (i, i)| + N0,c
(5.19)
The average MSE achieved by the C-DSL, φF,K , is calculated for various values
of K and compared to the MSE achieved by an optimum Wiener filter φW , to study
the effect of background noise on the performance of the C-DSL. The average MSE
is calculated by averaging the achieved MSE for each sub-channel according to φ =
PM
1
i=1 φ(i). For both measurement scenarios, φF,K is determined for K=1, 5, and
M
10 symbols.
Figs. 5.5 and 5.6 show the average achieved MSE versus the relative distance
between the DSL and the PLC modems for the Desk Modem Scenario and the Entry
Point Scenario respectively. Note that it was established in Chapter 3 that relative
distance between the DSL and the PLC modems does not have a significant impact
on the cross-coupling channel gains. However, for ease of comparison and to maintain
consistency, the euclidean distance is used as a reference point.
Note that, as mention at the beginning of Section 5.3.3, the maximum transmission
power for DSL symbols and PLC symbols are εD = −60 dBm
and εP = −50 dBm
Hz
Hz
respectively. Additionally, η d,DSL has a PSD of N0,d =-140 dBm/Hz, while η c,DSL has
108
a PSD of N0,c =-120 dBm/Hz[13].
−124
φF,1
−126
φF,5
φF,10
−128
M SE in
dB m
Hz
φW
−130
−132
−134
−136
−138
0
3.7
18
2
2
41.2
4
6
8 47 48.9
10 5
Eucledean Distance in ft (not to scale)
50.7
12
Figure 5.5: Desk Modem Scenario: MSE of the proposed scheme versus the relative
distance between the DSL and PLC modems.
From Figs. 5.5 and 5.6, we can first observe that φW is at the same level of
N0,d , which highlights the potential of the proposed interference cancelling scheme to
eliminate the DM PLC interference and offer performance comparable to a noise only
environment. In addition, the number of training symbols required for the C-DSL
to achieve an MSE comparable to φW is quite small. This is due to the fact that
the CM signal provides a very low noise estimate of the EMI corrupting the DM
signal. Finally, the achieved MSE for the Entry Point Scenario, shown in Fig. 5.6,
is lower than the achieved MSE for the Desk Modem Scenario, shown in Fig. 5.5.
This is expected since in the Entry Point Scenario the in-line LPF reduces the DM
PLC interference levels while the CM PLC interference levels remains unaffected, as
shown in Fig 3.20.
5.3.3.2 Improvement in Bit Rate
The proposed interference cancelling scheme, presented in Section 5.3.2, estimates the
DM interference signal and subtract this estimate from the DM signal. The power of
109
−136
φF,1
φF,5
−137
φF,10
−138
M SE in
dB m
Hz
φW
−139
−140
−141
−142
00 13.7
31.1
62.3
4 31.3 34.4
6 45.4 48
10
Eucledean Distance in ft (not to scale)
15.9
2 16.4
80.9
12
Figure 5.6: Entry Point Scenario: MSE of the proposed scheme versus the relative
distance between the DSL and PLC modems.
the remaining interference is equal to the MSE of the C-DSL. In this section, we use
the MSE of the C-DSL and the optimum Wiener filter, discussed in Section 5.3.3.1,
to calculate the SINR when the proposed interference cancelling scheme is utilized.
The SINR achieved after utilizing the C-DSL and the optimum Wiener filter is used
to calculate the available bit rates. These bit rates are compared with the bit rates
in absence of the proposed interference cancelling scheme.
The SNR in presence of only background noise for sub-channel i, γ 0 , is the ratio of
the signal power to the background noise power (5.20). If a co-located PLC network
exists, the SINR for sub-channel i, γ(i), is the ratio of the desired DSL signal power
to the DM PLC interference power and the background noise (5.21).
εD |HDSL (i, i)|2
N0,d
(5.20)
εD |HDSL (i, i)|2
N0,d + εP |Hd,PTx−DRx (i, i)|2
(5.21)
γ0 (i) =
γ(i) =
When the interference cancelling scheme is utilized, the DM PLC interference
power is reduced to the MSE of the utilized canceller. Let the SINR when the C-DSL
and Wiener filter are utilized be γ F,K and γ W respectively. Thus, for frequency bin
110
i, γF,K (i) and γW (i) are calculated by (5.22a) and (5.22b) respectively.
εD |HDSL (i, i)|2
N0,d + φF,K (i)
εD |HDSL (i, i)|2
γW (i) =
N0,d + φW (i)
γF,K (i) =
(5.22a)
(5.22b)
Recall, bit loading, where the number of bits allocated to each sub-channel is
dependent on the sub-channel’s SINR, is used in DSL [22]. The number of bits b(i)
that can be loaded on frequency bin i, in absence of any interference cancelling scheme
is calculated by (5.23), where the SNR gap Γ for DSL is 9.45 dB and the maximum
number of bits that can be allocated to a DSL frequency bin bmax is 15 bits [27].
γ(i)
b(i) = min bmax , log2 1 +
Γ
(5.23)
To calculate the number of bits for frequency bin i when CDSL,K and CW are
utilized, γ(i) in (5.23) is replaced by γF,K (i) and γW (i) respectively. Finally, in
presence of AWGN noise only, the number of bits for frequency bin i is determine by
replacing γ(i) in (5.23) with γ0 .
After determining the number of bits that can be loaded on each frequency bin,
the total bit rate R is then calculated by (5.24), where, for VDSL2, the normalizing
factor λ is 0.79 and the sub-carrier spacing ∆f is 8.6 kHz [27].
R = λ∆f
X
b(i)
(5.24)
i
To study the effectiveness of our proposed interference cancelling scheme, we calculate the total bit rate when the C-DSL and Wiener filter are utilized (i.e., RF,K and
RW respectively) and compare the results with the total bit rates in presence of only
AWGN R0 and in absence of any interference cancellation scheme RW/O , for both the
Desk Modem Scenario and the Entry Point Scenario.
111
160
140
DSL Bit Rate in Mb/s
120
RF,1
100
RF,5
RF,10
80
RW
R0
60
RW/ O
40
20
0
0
3.7
18
2
25 28.3
6 28.8 41.2
8 47.2 48.9
10 50.1 50.7
12
Euclidean Distance in ft (not to scale)
20.6 22.1
4
Figure 5.7: Desk Modem Scenario: Achieved improvement in bit rates versus the
Euclidean distance between the DSL and PLC modems.
Figs. 5.7 and 5.8 show the total bit rate, achieved by CDSL,K and CW , in absence
of any interference cancellation, and by complete removal of the PLC interference
versus the relative distance between the DSL and the PLC modem, for the Desk
Modem Scenario and the Entry Point Scenario respectively. It is evident that the
C-DSL successfully reduces the DM PLC interference levels. As K increases the
C-DSL approaches the optimum performance by a Wiener filter. Note that despite
the optimality of the Wiener filter, it does not achieve the maximum bit rates of
R0 because the weights of the Wiener filter are dependent on the cross-correlation
between the DM and the CM interference, as shown in (5.19).
5.4 Pre-Distortion-Based Interference Mitigation Solution
Utilizing the scheduling-based interference mitigation solution, proposed in Section
5.3, restricts channel access to one PLC user per DSL super-frame. Restricting channel access to one PLC user per DSL super-frame has two drawbacks. First drawback is
underutilization of the PLC channel, which affects the overall PLC network throughput. This occurs in the event that a PLC user that is granted access to the channel
112
160
140
DSL Bit Rate in Mb/s
120
100
RF,1
80
RF,5
60
RF,10
RW
40
R0
20
0
0
0
RW/ O
13.7 15.9
2 1
46
4 31.3 346
8 49
10 66.2
Euclidean Distance in ft (not to scale)
80.9
12
Figure 5.8: Entry Point Scenario: Achieved improvement in bit rates versus the
Euclidean distance between the DSL and PLC modems.
does not have enough information to transmit for the entire duration of the DSL
super-frame. The second drawback is the latency experience by the PLC users due to
the scheduling restriction of one PLC user per DSL super frame. This latency affects
the QoS of interactive multimedia applications.
The pre-distortion-based interference mitigation solution, proposed in this section,
is suitable for interactive multimedia applications and is not significantly affected
by the number of active PLC users. Since the PLC-to-DSL cross-coupling channel
is reciprocal, the PLC transceivers can estimate the cross-coupling channel using
the DSL super-frame training symbols when directed by the PLC-DC. The PLC
transceivers then pre-multiply their symbols by the inverse of the DM cross-coupling
channel. The PLC transceivers transmit two types of symbols: training symbols and
data symbols.
Training symbols are known to all PLC transceivers and to the DSL transceiver.
Additionally, the same sequence of training symbols are always transmitted during
the training of the FDIC used by the DSL system, i.e., C-DSL. This means that all
interfering signals, during the training of the FDIC, arrive at the DSL receiver at a
113
constant gain offset relative to the desired DSL signal, regardless of source or subcarrier. Note that the constant gain offset is removed from the desired DSL signal,
during the training of C-DSL. This allows multiple PLC users to access the PLC
channel during a single a DSL super-frame, which was not the case for the schedulingbased interference mitigation solution. Note that the proposed modification to the
PLC frame structure complies with the requirement of G.hn [9].
Recall, the DM and CM cross-coupling channels between a PLC transmitter and
the DSL receiver, sampled at integer multiples of ∆fDSL , are denoted by the channel
∆fDSL
DSL
frequency response matrices H∆f
d,PTx−DRx and Hc,PTx−DRx . Similarly, the DM cross-
coupling channel between the PLC transmitter and the DSL receiver when sampled
at integer multiples of ∆fPLC is defined by the channel frequency response matrix
PLC
H∆f
d,PTx−DRx .
Additionally, recall that the received DSL DM signal rd,DSL is the summation of
DSL
the desired DSL signal y = HDSL x, the DM PLC interference vd,DSL = H∆f
d,PTx−DRx q,
and the DM additive white Gaussian noise η d,DSL . Note that HDSL is the direct
DSL channel, while x and q are the transmitted DSL and PLC symbols, respectively.
Similarly, the received DSL CM signal rc,DSL is the summation of only the CM PLC
DSL
interference vc,DSL = H∆f
c,PTx−DRx q and the CM additive white Gaussian noise η c,DSL .
Finally, recall that the received DM PLC signal rd,PLC is the summation of the
PLC
desired PLC signal z = HPLC q, the DM DSL interference vd,PLC = H∆f
d,DTx−PRx y,
and the DM additive white Gaussian noise η d,PLC . Also, the received CM PLC signal
PLC
rc,PLC is the summation of only the CM DSL interference vc,PLC = H∆f
c,DTx−PRx y and
the CM additive white Gaussian noise η c,PLC .
The rest of this section is organized as follows. The technique with which the
PLC symbol is pre-distorted is presented in Section 5.4.1. In Section 5.4.2, the predistortion cancellation algorithm and the training of C-DSL are discussed. Finally,
114
in Section 5.4.3, the performance of pre-distortion solution is analyzed and compared
with the performance of the scheduling-based solution.
5.4.1 PLC Symbol Pre-distortion
Before transmission, the generated PLC symbol u is pre-multiplied by the inverse
PLC
of the DM PLC-to-DSL cross-coupling channel H∆f
d,PTx−DRx and the normalization
factor a to form the transmitted PLC symbol q. Thus, the transmitted PLC symbol
q, which causes interference to the DSL system as shown in Fig. 5.4, is defined by
q=
au
DSL
H∆f
d,PTx−DRx
.
(5.25)
Recall, the main obstacle to allow more than one PLC user to access the PLC
channel during a single DSL super-frame is the outlet-to-outlet variations of the PLCto-DSL cross-coupling channels. This obstacle is overcome in two steps. The first step
is adding a training prefix (TP) at the beginning of each PLC frame, during which a
training sequence that is known to all PLC transceivers and the DSL transceiver is
transmitted. Thus, each PLC frame is composed of known training symbols followed
by data symbols. The second step is pre-multiplying the PLC symbol by the inverse
of the DM cross-coupling channel before transmission.
This means that during the PLC frame TP transmission, no matter which PLC
user is transmitting, the effect of the PLC interference on the DSL receiver is at
a constant level across all sub-carriers. This constant level can be estimated and
removed from the DSL DM signal. Additionally, during the PLC frame TP transmission, the C-DSL is trained to estimate the DM to CM cross-coupling channel ratio of
each individual PLC user. This ratio is required to mitigate the effects of the PLC
interference during the transmission of the PLC data symbols. Further detail on the
training of the C-DSL is presented in Section 5.4.2.1.
The normalization factor a, on the other hand, is utilized to confine the PLC
115
signal power to the values stated in [9]. In addition, the normalization factor controls
the interference level of the PLC symbol on the DSL system, which prevents the PLC
interference from masking the DSL desired signal.
Note that the effective direct PLC channel has to be estimated by each PLC
transceiver. Recall, each time a PLC transceiver joins the PLC network, all PLC
transceivers are forced into listening mode to estimate the interference caused by
the DSL transceiver. In the event that an C-PLC is utilized to mitigated the effect of the DSL-to-PLC interference, the C-PLC is trained as discussed in Section
5.3.2.2. After which, the PLC transceiver estimates the effective PLC channel (i.e.,
∆fPLC
HPLC /Hd,PTx−DRx
).
In absence of the pre-multiplication imposed by the proposed pre-distortion-based
interference mitigation solution, a PLC receiver would have to estimate the direct
PLC channel HPLC . However, since in our proposed cancellation scheme the PLC
transmitter pre-multiples the generated PLC symbols u by the inverse of the DM
PLC
cross-coupling channel H∆f
d,PTx−DRx before transmission, the PLC receiver has to esti-
∆fPLC
mate HPLC /Hd,PTx−DRx
and not only HPLC . Direct PLC channel estimation is beyond
the scope of this thesis; however, the standard channel estimation approach utilized
by G.hn to estimate the direct PLC channel could be utilized to estimate the product
of the pre-distortion and the actual PLC channel.
5.4.2 Cancellation Algorithm
In this section, we introduce the pre-distortion based interference cancellation scheme,
where the training of the canceller C-DSL is discussed in 5.4.2.1. Recall, the block
diagram on which this scheme is based was introduced in Section 5.2.
The C-DSL has to be re-trained each time a new PLC user occupies the channel
because of the outlet-to-outlet variation in the PLC-to-DSL cross-coupling channels.
Note that the training of the C-DSL occurs during the transmission of the PLC
116
frame TP. Thus, impact of the PLC interference on the DSL DM signal during the
TP duration is reduced to a constant interference level, which can be removed from
the DSL signal because the training sequence is known to the DSL transceiver.
Once the C-DSL is trained, it is utilized to mitigate the effect of the PLC interference during the data symbol transmission portion of the PLC frame. As will be
discussed in detail in Section 5.4.2.1, the TP added to each PLC frame along with the
pre-distortion of the PLC symbols before transmission allow multiple PLC users to
access the PLC channel during a DSL super-frame. Thus, the latency introduced by
the solution proposed in Section 5.3 is eliminated, which allows the PLC network to
support interactive multimedia traffic, with very low latency tolerance. Additionally,
by allowing more than one PLC user to access the channel during the DSL superframe, the unused time-slots issue is overcome. Note that during the PLC frame TP
transmission, the C-DSL is trained with no significant impact on the DSL system, as
will be shown in Section 5.4.3.
5.4.2.1 C-DSL Training
A predefined training sequence is transmitted by the PLC transceiver during the
training portion of the PLC frame. The same sequence is used by all PLC transceivers,
and it is known to the DSL transceiver. The PLC-DC initiates the training of the
C-DSL each time a new PLC user access the channel. Recall, once the C-DSL is
trained, the tap-coefficient of C-DSL remains valid until a new PLC user access the
PLC channel. Each PLC frame is preceded with a training period, where the TP is
transmitted. During this training period, the transmitting PLC transceiver transmits
q (5.25), where the generated PLC symbol u is a pre-defined training sequence.
The DSL canceller C-DSL tap-coefficient matrix CDSL is an M ×M square diagonal
matrix. During training, the received CM DSL signal rc,DSL is utilized to estimate the
ratio of the DM to CM PLC-to-DSL cross-coupling channels. Recall, the transmitted
117
PLC symbol q is an N × 1 column vector, which is converted to an analog signal via
an A/D before transmission; the received interference signals vd,DSL (5.3a) and vc,DSL
(5.3b), on the other hand, are sampled at integer multiples of ∆fDSL (i.e., vd,DSL and
vc,DSL are an M × 1 column vector).
During the training period, the CM PLC interference for the ith sub-carrier vc,DSL (i)
is defined in (5.26), where q ∆fDSL (i) is the transmitted PLC symbols sampled at the
DSL sampling frequency. During the training period, u is the pre-defined training
sequence, which is known to all PLC transceivers and to the DSL transceiver.
∆fDSL
vc,DSL (i) =Hc,DTx−PRx
(i)q ∆fDSL (i)
∆fDSL
=Hc,DTx−PRx
(i)
au(i)
!∆fDSL
(5.26)
∆fPLC
Hd,PTx−DRx
(i)
To extract an estimate of the DM to the CM PLC cross-coupling channel ratio, the
C-DSL is trained by dividing the known training sequence by the received CM signal.
Thus, for the lth symbol interval, the C-DSL tap-coefficient for the ith sub-carrier
CDSL (i, i, l) is calculated by (5.27), where vc,DSL (i) is defined in (5.26). Note that the
PLC-DC is connected to the C-DSL to communicate the value of the normalizing
factor a as shown in Fig. 5.4.
CDSL (i, i, l) =
au(i)
au(i)
=
rc,DSL (i)
vc,DSL (i) + ηc,DSL (i)
(5.27)
To minimize the effect of the AWGN noise on the estimate of CDSL , the value
of CDSL (i, i, l) is averaged over L successive training symbols. Thus, for the ith subcarrier, the averaged tap-coefficient C DSL,L (i, i), defined by (5.28), is the estimate of
the the DM interference to the CM interference.
L
1X
C DSL,L (i, i) =
CDSL (i, i, l)
L l=1
(5.28)
Note that while training C-DSL, the pre-defined training sequence is subtracted
from rd,DSL . During the training of C-DSL, for the ith sub-carrier, the DM PLC inter118
ference vd,DSL (i) is approximately equal to au, as shown in (5.29). Thus, subtracting
au(i) from rd,DSL (i) mitigates the PLC DM interference during the training phase of
C-DSL, as will be confirmed by the analysis presented in Section 5.4.3.
∆fDSL
vd,DSL (i) =Hd,DTx−PRx
(i)q ∆fDSL (i)
au(i)
∆fDSL ≈
H
(i) d,DTx−PRx
(5.29)
DSL
∆f
∆fPLC Hd,PTx−DRx
(i)
The reason than vd,DSL (i) is not exactly equal au(i) is the difference in sampling
frequencies utilized by the DSL and PLC systems. Since each system samples the
cross-coupling channel at different frequencies, the resulting samples are not equal
∆fDSL
∆fDSL
∆fPLC
(i.e.,Hd,DTx−PRx (i) is only equal Hd,PTx−DRx (i)
when the samples are taken at
the same frequency). However, as will be shown in Section 5.4.3, this difference does
not impact the performance of the DSL system, either during the training phase or
during the transmission phase of the C-DSL canceller.
5.4.3 Performance Analysis
To evaluate the performance of the pre-distortion-based interference mitigation solution, the MSEs during the the training phase and transmission phases of C-DSL
are determined. During the training phase of C-DSL, the known training sequence is
subtracted from the received DSL signal. Thus, the MSE of the pre-distortion-based
solution during the training phase of C-DSL does not depend on its tap-coefficients.
However, during transmission phase, the tap-coefficients of C-DSL affects the MSE of
the solution because the error during transmission phase is not simply the difference
between the known training sequence and the interference due to the transmission of
the TP. Rather, the error is the difference between the output of the C-DSL canceller
and the DM PLC interference due to the transmission of the data symbols portion of
the PLC frame, as shown in Fig. 5.4.
119
In this section, we evaluate the performance of C-DSL during both the training
phase and transmission phase in Sections 5.4.3.1 and 5.4.3.2 respectively. To evaluate
the performance of C-DSL the mean square error (MSE) is analyzed for both training
and transmission phase. After which, the available bit rates based on the SINR per
sub-carrier is calculated before and after utilizing the proposed canceller for both the
training and transmission phase.
It was shown in Fig. 5.2, the PLC interference s couples on each of the DSL
twisted-pair differently, according to the imbalance factor of each of the twistedpairs. Let us assume s = a.u, where a is the normalization factor, and u is the
generated PLC symbol. For the twisted-pair imbalance factors k and j, the DM to
− j)
a.u(k
. Thus, variations in a will appear in both the
CM interference ratio is
+ j)
0.5
a.u(k
DM and the CM PLC interference signals, and since C-DSL estimates a ratio, these
variations should cancel out.
Therefore, if needed to maintain the performance of the PLC system, during
the transmission phase each PLC transceiver can choose a normalization factor that
maintains the power spectral density of the received PLC signal. However, during
the training phase, all PLC transceiver have to utilize the same known value for the
normalization factor a. By ensuring that, no matter which PLC user is active, the
PLC interference level at the DSL receiver is constant during the training phase of
C-DSL, the PLC interference can be easily removed while training C-DSL.
The DSL direct channel frequency response matrix, along with the DM PLC-toDSL cross-coupling channel response matrices sampled at both ∆fDSL and ∆fPLC , and
the CM PLC-to-DSL cross-coupling channel response matrices sampled at ∆fDSL are
required in the evaluation process. Recall, the DSL direct channel is obtained from
the standard two port model discussed in [29]; while, the cross-coupling channels are
obtained from the measurements presented in Chapter 3. Note that we utilize the
120
measurements for the Entry Point Scenario in the following analysis.
Recall, DSL and PLC utilize frequency separations of ∆fDSL = 8.6 kHz and
∆fPLC =24.4 kHz respectively. However, for simplicity, we assume that ∆fPLC is three
times ∆fDSL (i.e., ∆fDSL = 8.6 kHz and ∆fPLC = 25.8 kHz). Thus, the DSL and PLC
system utilize M and N orthogonal sub-carriers respectively, where N = M/3. The
maximum transmission power for DSL symbols and PLC symbols are εD = −60 dBm
Hz
[2] and εP = a dBm
respectively. Note that we assume that both η d,DSL and η d,PLC
Hz
have a PSD of N0,d =-140 dBm/Hz, while η c,DSL and η c,PLC have a PSD of N0,c =-120
dBm/Hz [13].
Based on each sub-carrier’s SINR, a number of bits are allocated to each subcarrier [22]. The number of bits b(i) that can be transmitted over the ith sub-carrier
is calculated by (5.23).
5.4.3.1 Training Phase
During the training phase, the known training sequence au is subtracted from the
received DSL signal rd,DSL = vd,DSL + η d,DSL . Therefore, the error for the ith subcarrier , eTr (i) = vd,DSL (i) − au(i). Note that if DSL and PLC system utilize the same
frequency separation with aligned sub-carrier locations, vd,DSL (i) will be equal to
au(i). However, since the PLC-to-DSL cross-coupling channel is sampled at different
frequencies by the PLC and DSL systems, vd,DSL (i) 6= au(i), which results in errors.
Thus, the MSE during the training phase of the C-DSL for sub-carrier i, φTr (i) =
E[eTr (i)e∗Tr (i)], is defined by (5.30).
φTr (i) =|vd,DSL (i)|2 + a2 u2 (i)
(5.30)
− au(i)(vd,DSL (i) +
∗
vd,DSL
(i))
The SNR in presence of only background noise is the maximum bound achieved
by complete removal of the interference. Thus, for the ith sub-carrier, the SNR bound
121
γBound (i) is the ratio of the signal power to the background noise power (5.20). If a colocated PLC network exists, the SINR for the ith sub-carrier, γW/O (i), is the ratio of
the desired DSL signal power to the DM PLC interference power and the background
noise (5.21). Utilizing the MSE calculated by (5.30), we calculate the SINR after
interference mitigation during the training phase. This is performed by substituting
the interference power with the calculated MSE. Thus, for the ith sub-carrier, the
SINR after interference cancellation is calculated by (5.22).
To calculate the available bit rates for the DSL system during the training phase
of the C-DSL, we calculate the number of bits that can be loaded to each of the DSL
sub-carriers, based on each sub-carrier’s SINR, from (5.23). The number of bits for
the SNR bound, SINR without interference cancellation, and the SINR during the
training phase are calculated from the SNR (or SINR) for each case in (5.23). After
determining the number of bits that can be loaded on each sub-carrier, for each case,
the total bit rate R is then calculated by (5.24). where, for VDSL2, the factor λ is
0.79 and the sub-carrier spacing ∆fDSL is 8.6 kHz [27].
Table 5.2 shows the available bit rates during the training phase of C-DSL. Four
cases are considered. The first and the second are the total bit rates in presence of
only background noise RBound,Tr and without interference cancellation RW/O,Tr . The
other two cases are when the interference cancellation scheme is utilized; however, we
study the total bit rates if both DSL and PLC utilize the same frequency separation
RM=N,Tr (i.e.,∆fPLC = ∆fDSL ) and the total bit rates when the frequency separation
between the PLC sub-carriers is 3 times the frequency separation utilized by the DSL
system RM=N/3,Tr (i.e.,∆fPLC = 3∆fDSL ). From Table 5.2, it is evident that during
the training phase, subtracting au from vd,DSL achieves optimal bit rates if both PLC
and DSL utilized the same frequency separation, and near-optimal bit rates when the
frequency separation utilized by PLC is 3 times the frequency separation utilized by
122
Table 5.2: Training phase available bit rates.
Case
Bits in Mb/S
RBound,Tr
160.79
RW/O,Tr
49.80
RM=N,Tr
160.79
RM=N/3,Tr
157.65
DSL.
5.4.3.2 Transmission Phase
During transmission phase, the canceller tap-coefficient CDSL (i, i) affects the value
of the MSE. Thus, the number of symbols utilized in training C-DSL will affect the
available bit rates. The error during transmission phase is not simply the difference
between the known training sequence and the DM PLC interference estimate vd,DSL ,
bd,DSL
but rather the error is the difference between the output of the C-DSL canceller v
and vd,DSL , as shown in Fig. 5.4.
The transmission phase error for the ith sub-carrier , eTx (i) = vd,DSL (i)−b
vd,DSL (i) .
Thus, The MSE during the transmission phase for sub-carrier i, φTx (i) = E[eTx (i)e∗Tx (i)],
is defined by (5.31).
φTx (i) =|vd,DSL (i)|2 + |b
vd,DSL (i)|2
(5.31)
−
∗
vbd,DSL
(i)vd,DSL (i)
−
∗
vbd,DSL (i)vd,DSL
(i)
Similar to the approach used to evaluate the C-DSL performance during training,
we utilize the MSE to calculate the SINR. The SINR is then utilized to calculate the
available bit rates for the various cases presented in Table 5.2. However, since the the
number of symbols used in training affects the MSE, and consequently the achieved
SINR and available bit rates, we calculate the available bit rates for various numbers
of training symbols. Recall, CDSL (i, i, l) is averaged over l symbols to mitigate the
123
Table 5.3: Transmission phase available bit rates.
l (symbols) RM=N,Tx (Mb/s) RM=N/3,Tx (Mb/s)
1
151.37
148.22
5
153.22
150.13
10
153.65
150.44
25
153.84
150.79
effect of the AWGN noise on the estimate, as shown by (5.28).
Table 5.3 shows the available bit rates for two cases, where both the DSL and PLC
systems utilize the same frequency separation and when the frequency separation
between the PLC sub-carriers is 3 times the frequency separation utilized by the
DSL system during the transmission phase of C-DSL (i.e., RM=N,Tx and RM=N/3,Tx
respectively), for DSL cable run length of 2000 ft and for various values of l. The
total available bit rates for the bound case RBound,Tx and in absence of interference
cancellation RW/O,Tx are 160.79 and 60.3 M b/s respectively. From Table 5.3, it is
clear that the canceller C-DSL successfully mitigates the effect of the PLC interference
on the DSL system, which is reflected in the available bit rates achieved by the
canceller.
Fig. 5.9 shows the total bit rates achieved by the C-DSL, for N=M and N=M/3,
and by the same C-DSL if it was trained according to the scheme discussed in Section
5.3 versus the total bit rates in presence of only background noise (denoted by Bound)
and without interference cancellation (indicated by W/O), for various DSL cable run
lengths. Note that the results in Fig. 5.9 are the achieved DSL DSL bit rates for band
plan ”998E30” [2], after training each of the C-DSLs for 5 symbols. It is evident that
C-DSL achieve slightly better performance that the canceller presented in Section
5.3, both for N=M and N=M/3.
It was shown in Section 5.3.1 that the scheduling based interference mitigation
solution is suitable for multimedia applications that require the latency to be in the
124
240
Bound
C−DSL in Section 5.3
W/O
C−DSL N=M
C−DSL N=M/3
220
200
DSL Bit Rate in Mb/s
180
160
140
120
100
80
60
40
200
400
600
800
1000 1200 1400 1600 1800 2000 2200
DSL loop length in ft
Figure 5.9: Achieved total DSL bit rates for DS transmission, for various DSL cable
run length.
range of 100 to 300 ms, such as IPTV and high definition video streaming. However,
interactive multimedia applications, such as VoIP and Internet video conferencing,
have very low latency tolerance that lies is the range of 10 to 30 ms [67]. The purpose
of the pre-distortion-based interference mitigation solution is to mitigate the PLCto-DSL interference without introducing significant latency to the PLC system.
For a saturated PLC system, i.e., all users have packets to transmit, the latency
is calculated by
TLAT = (n − 1)τ,
(5.32)
where n is the number of active PLC users and τ is the duration the PLC channel is
restricted to a single user. In Section 5.3.2, once a PLC user takes over the channel,
other PLC users cannot utilize it for the duration of the DSL super-frame TSF . Thus,
for the scheduling-based solution, and in presence of a VDSL2 system, Tτ = TSF =
64.25 ms is substituted in (5.32) to calculate the latency of the scheduling-based
algorithm TLAT,Scheduling . On the other hand, if C-DSL is trained according to the
algorithm discussed in Section 5.4.2, which does not impose any additional restriction
125
on the PLC channel access, Tτ = TPLC =100 µs, where TPLC is the duration of the PLC
time-slot duration. Thus, to calculate the latency due to the pre-distortion algorithm
TLAT,Pre−distortion , Tτ = 100 ms is substituted in (5.32).
Table 5.4: Latency: Scheduling-based versus pre-distortion-based solutions.
D (active users) TLAT,Scheduling (ms) TLAT,Pre−distortion (ms)
2
64.25
0.1
3
128.5
0.2
4
192.75
0.3
5
257
0.4
Table 5.4 shows the latency experience by the PLC network due to both the
scheduling-based and the pre-distortion-based interference mitigation solutions. For
each algorithm, the latency is calculated by (5.32) based on the number of active PLC
users. It is evident that the latency introduced by the pre-distortion based solution
is significantly lower than the latency caused by the scheduling based solution. Additionally, only the pre-distortion based solution is suitable for interactive multimedia
applications with very low latency tolerance.
5.5 Comparison with Spectral Management Solutions
In this section, we compare the achieved bit rates of the scheduling-based interference
mitigation solution presented in Section 5.3 with the achieved bit rates of the spectral
management (SM) solutions proposed in [27] and [28], over the VDSL2 spectrum.
Note that both scheduling-based and pre-distortion-based solutions achieve similar
bit rates; however, in the pre-distortion-based solution, the simplicity of the scheme
is sacrificed to reduce the latency experienced by the PLC users.
In [27], the PLC transmit power for all sub-carriers within the VDSL2 spectrum is
reduced; however, only the transmit power for the PLC sub-carriers that overlap with
126
the DSL downstream (DS) frequencies is reduced in [28]. Henceforth, the solutions
proposed in [27] and [28], will be referred to as SMFR (SM flat reduction) and SMSR
(SM selective reduction) respectively.
The factor used to compare the performance of the three PLC-to-DSL interference
mitigation solutions is the available bit rates. We utilize band plan ”998E30” [2]
and study the bit rates achieved by each solution for three cases, namely: DSL
downstream (DSL-DS), DSL upstream (DSL-US), and PLC. In each case, we calculate
the available bit rate based on the SINR as discussed in Section 5.3.3.2.
In Figs. 5.10 to 5.15, the available bit rate in presence of only AWGN noise
is indicated by “Bound”, while the bit rate without any interference cancellation
is denoted by “W/O”. The achieved bit rates by the C-DSL when trained for 5
symbols CDSL,5 are indicated by “F,5”. If the performance of both SMFR and SMSR
is identical (i.e., in the case of DSL-DS), “SM” is used to indicated the achieved bit
rates by both SMFR and SMSR. Note that the notation “@ -50 dBm/Hz”, “@ -60
dBm/Hz”, and “@ -75 dBm/Hz” is utilized to donate that the achieved bit rates
when the PLC transmit power εP is set to -50, -60, and -70 dBm/Hz, respectively.
240
Bound
F,5 @ −50 dBm/Hz
F,5 @ −75 dBm/Hz
W/O @ −50 dBm/Hz
SM @ −60 dBm/Hz
SM @ −75 dBm/Hz
220
200
DSL Bit Rate in Mb/s
180
160
140
120
100
80
60
40
20
0
500
1000
1500
DSL loop length in ft
2000
Figure 5.10: Desk Modem Scenario: Achieved bit rates for DSL-DS.
127
Fig. 5.10 shows the DSL-DS bit rates achieved by the PLC-to-DSL interference
mitigation solutions versus the DSL loop length for the Desk Modem Scenario. The
percentage improvement of a bit rate R relative to no interference cancellation is
calculated according to R − RW/O /RW/O . For a short DSL loop length (i.e., DSL
loop length less than 500 ft in Fig. 5.10), the percentage of improvement achieved
by the proposed scheduling-based interference mitigation solution ranges from 50%
to 80%, while for long DSL loops (i.e., DSL loop length greater than 500 ft in Fig.
5.10), the percentage of improvement achieved by our proposed scheme ranges from
100% to 300%. The variation in the improvement percentage is because for short DSL
loops the limiting factor for the number of bits per frequency bin is bmax , while for
longer loops, the PLC-to-DSL interference plays a role in reducing the SINR, which
consequently affects the total bit rates.
On the other hand, the improvement achieved by both SMFR and SMSR by
reducing εP to -60 dBm/Hz is significantly lower than the improvement achieved by
our scheme. While both SMFR and SMSR achieve comparable results to our scheme
when εP to -75 dBm/Hz, this reduction in PLC transmit power significantly hinders
the achievable PLC bit rates, as will be shown in Figs. 5.12 and 5.13.
Similarly, for the Entry Point Scenario, the scheduling-based interference mitigation solution achieves higher SINR than both SMFR and SMSR, which is reflected
in the available bit rates shown in Fig. 5.11, especially for εP less than -75 dBm/Hz.
However, for εP equal to -75 dBm/Hz, both SMFR and SMSR achieve bit rate improvement that is identical to the scheduling-based interference mitigation solution.
Note that the percentage of bit rate improvement in the Entry Point Scenario is lower
than that of the Desk Modem Scenario, because the PLC-to-DSL interference levels
are lower due to the in-line LPF.
To study the effect of reducing εP on the PLC bit rates, we simulated a PLC
128
240
220
200
DSL Bit Rate in Mb/s
180
160
140
120
100
80
60
40
20
0
Bound
F,5 @ −50 dBm/Hz
F,5 @ −75 dBm/Hz
W/O @ −50 dBm/Hz
SM @ −60 dBm/Hz
SM @ −75 dBm/Hz
500
1000
1500
DSL loop length in ft
2000
Figure 5.11: Entry Point Scenario: Achieved bit rates for DSL-DS.
channel based on the model presented in [66]. A medium PLC channel [69] was
generated and the simulation included both AWGN on the PLC channel at No =
-140 dBm/Hz and the interference from the DSL signal that couples across to the
PLC receiver through HD,d channel shown in Fig. 5.1. For the PLC simulation, we
assumed Γ = 9.45 dB, λ = 0.75, ∆f = 24.4 kHz, and bmax = 12 bits [27].
200
180
160
PLC Bit Rate in Mb/s
140
120
100
80
60
FDIC @ −50 dBm/Hz
SMFR @ −60 dBm/Hz
SMSR @ −60 dBm/Hz
SMFR @ −75 dBm/Hz
SMSR @ −75 dBm/Hz
40
20
0
0
500
1000
1500
DSL loop length in ft
2000
Figure 5.12: Desk Modem Scenario: Achieved bit rates for PLC.
In Figs. 5.12 and 5.13, “FDIC” represents the PLC bit rates available when the
129
240
220
200
PLC Bit Rate in Mb/s
180
160
140
120
100
80
FDIC @ −50 dBm/Hz
SMFR @ −60 dBm/Hz
SMSR @ −60 dBm/Hz
SMFR @ −75 dBm/Hz
SMSR @ −75 dBm/Hz
60
40
20
0
500
1000
1500
DSL loop length in ft
2000
Figure 5.13: Entry Point Scenario: Achieved bit rates for PLC.
scheduling-based interference mitigation solution is utilized, which are the available
PLC bit rates without reducing the PLC transmit power. The achieved DSL-DS bit
rates by both SMFR and SMSR was comparable to the DSL-DS bit rates achieved by
the scheduling-based interference mitigation solution when εP is set to -75 dBm/Hz.
However, as shown in Fig. 5.12, reducing εP to -75 dBm/Hz results in a 65%
to 85% reduction in the PLC bit rates for SMFR and a 45% to 55% reduction in
the PLC bit rates for SMSR, for the Desk Modem Scenario. Our proposed solution,
on the other hand, does not hinder the PLC bit rates because our scheme relies on
estimating the DM PLC interference and subtracting it from the DSL signal rather
than reducing the PLC transmit power.
Similarly, for the Entry Point Scenario, shown in Fig. 5.13, reducing εP to -75
dBm/Hz results in a reduction in the PLC bit rates for both SMFR and SMSR.
However, the reduction is not as large as for the Desk Modem Scenario, because the
interference from the DSL signal via the HD,d channel is reduced by the in-line LPF.
Figs. 5.14 and 5.15 show the DSL-US bit rates achieved by the PLC-to-DSL
interference mitigation solutions versus the DSL loop length for both the Desk Modem
130
130
Bound
F,5 @ −50 dBm/Hz
SMSR @ −50 dBm/Hz
120
110
DSL Bit Rate in Mb/s
100
90
80
70
60
50
40
30
0
500
1000
1500
DSL loop length in ft
2000
Figure 5.14: Desk Modem Scenario: Achieved bit rates for DSL-US.
Scenario and The Entry Point Scenario respectively. SMSR achieves higher PLC bit
rates than SMFR, because SMSR only reduces the PLC transmit power for subcarriers that overlap with the DS frequencies of the VDSL2 spectrum. However, not
reducing the transmit power of sub-carriers that overlap with the US frequencies of
the VDSL2 spectrum negatively affects the DSL-US bit rates.
130
Bound
F,5 @ −50 dBm/Hz
SMSR @ −50 dBm/Hz
120
DSL Bit Rate in Mb/s
110
100
90
80
70
60
50
40
0
500
1000
1500
DSL loop length in ft
2000
Figure 5.15: Entry Point Scenario: Achieved bit rates for DSL-US.
For the Desk Modem Scenario, shown in Fig. 5.14, the DSL-US bit rates when
131
SMSR is utilized are significantly lower than DSL-US bit rates achieved by our proposed solution, especially for short DSL loops. For short DSL loops, bmax is usually
the limiting factor, if the PLC-to-DSL interference is removed. However, in presence
of PLC interference, which is the case when SMSR is utilized, the numbers of bits
loaded to each sub-carrier is dependent on the SINR. For the Entry point Scenario,
shown in Fig. 5.15, where the PLC interference is reduced by the in-line LPF, the
DSL-US bit rates achieved by SMSR is lower than the rates achieved by our proposed
solution. However, the gap between the achieved bit rates between our solution and
SMSR is not as pronounced as it was for the Desk Modem Scenario.
132
Chapter 6
CONCLUSION
Recent advances in PLC have made it popular for in-home networking. This makes
PLC an increasingly relevant source of interference for DSL networks within the home
environment. Current solutions proved to be favouring the performance of the DSL
network over the PLC achievable bit rates. In fact, with the increase of the usable
DSL bandwidth, these current solutions will render the PLC network inoperable.
The objective of this thesis is to provide solutions that mitigate the interference
between co-located DSL and in-home PLC networks, without degrading the performance of the PLC networks. The main hypothesis of this is that the common mode
signal, which can be determined at the receiver and is usually ignored, contains information about the external EMI. Via an adaptive frequency domain canceller, the
common signal can be utilized to estimate differential mode interference EMI.
To the best of my knowledge, neither field measurements of PLC-to-DSL crosscoupling channels have been performed nor a model that characterizes the interference
environment between DSL and in-home PLC networks exists. Thus, field measurements that characterize the interference environment in a residential setting are required. As discussed in Chapter 3, a measurement campaign that has been designed
to characterize the PLC-to-DSL cross-coupling channels was conducted in two residential test-sites. The findings of the measurement campaign are summarized in
Section 6.1. Both the DM and CM PLC-to-DSL cross-coupling channels are studied
within two residential houses as test sites. For each of the two test sites, the DM and
CM PLC-to-DSL cross-coupling channels are measured for various rooms.
Mitigating the PLC-to-DSL interference via utilizing adaptive filters have never
133
been performed in the literature. Only time domain adaptive cancellers have been
proposed to mitigate the near end crosstalk in ADSL networks. However, in spite of
the relative narrow-bandwidth utilized by the ADSL technology, these time domain
filters required a long training time. In addition, due to the frequency selective nature
of the PLC interference, these time domain adaptive filters cannot mitigate the DM
PLC interference.
Two PLC-to-DSL interference mitigation solutions were presented in Chapter 5.
Both solutions utilize frequency domain adaptive cancellers to combat the frequency
selectivity of the PLC-to-DSL cross-coupling channel. Via these cancellers, the proposed solutions estimate the differential mode interference from the usually ignored
common mode signal, on a tone by tone basis. The functionality of the proposed
solutions is summarized in Section 6.2. Finally, in Section 6.3, recommendations for
future research is proposed.
6.1 Measurement Campaign Findings
The main findings of the measurement campaign are:
• The interference from an in-home PLC network to a co-located DSL modem
is significant.
• Effect of the PLC interference on the DSL bit rates varies according to the
distance between the DSL modem and the CO.
• For short DSL cable run length, interference from DSL negatively impacts the
performance of the PLC network.
• Utilizing an LPF to prevent the travel of the DSL signal over the house internal
telephone wires reduces but does not eliminate the degradation of the DSL
system performance due to the PLC interference.
134
• Spatial separation between the DSL modem and the PLC modems has no
significant impact on the interference.
• The in-home cross-coupling channels, both DM and CM, are frequency selective.
• The ratio of the DM to CM PLC-to-DSL cross-coupling channels is not a
smooth function of frequency.
• Both cross-coupling channels significantly vary from one power outlet to the
other.
• In a non-electrically active test site, the cross-coupling channels are stationary.
6.2 Interference Mitigation Solutions
Both interference solutions utilize FDICs. The FDICs are composed of parallel singletap filters, where the number of taps equals the number of sub-channels. Each tap of
the FDIC is an estimate of the ratio of the DM to the CM interference for a given subchannel. Since each sub-channel is processed independently, the frequency selectivity
of the cross-coupling channels is overcome. Additionally, since the FDIC estimates a
ratio, variations in the PLC channel due to its non-stationarity does not affect the
performance of the FDIC.
The first interference mitigation solution is a scheduling-based solution, where
access to the PLC channel is restricted to one PLC user per DSL super-frame. This
restriction overcomes the variation in cross-coupling channels due to changes in the
location of the PLC transceiver. For less than 6 active users in the PLC network, the
added delay due to this restriction does not deteriorate the QoS for non-interactive
multimedia applications. However, as number of active users in the PLC network
increases, this restriction severely degrades the QoS for the multimedia applications.
135
The second interference mitigation solution is a pre-distortion based solution,
where the PLC symbols are multiplied by the inverse of the DM PLC-to-DSL crosscoupling channel and a training period is added to the beginning of each PLC frame.
This allows the FDIC to train during the PLC frame training period instead of restricting channel access to one PLC user per DSL super-frame. The pre-distortion
based solution does not add any latency to PLC frame transmissions, and it meets
the necessary QoS requirements for interactive multimedia applications. However,
it is more complex to implement than the scheduling-based solution because of the
required signal processing.
6.3 Recommendation for Future Research
Future work motivated by the findings of this thesis is discussed in this section. For
consistency, the future work is categorized according to the two focus points of the
thesis. First, field measurements for cross-coupling channels over the G.fast spectrum
are discussed in Section 6.3.1. The proposed approach to solve the interference source
in presence of interference from multiple sources and developing a replacement system
for the current DSL transmission scheme is presented in Section 6.3.2.
6.3.1 Interference Channel Characterization
Two measurement campaigns are required to characterize the DM and CM PLCto-DSL cross-coupling channels and the DSL DM and CM direct channels over the
G.fast spectrum are discussed in Sections 6.3.1.1 and 6.3.1.2, respectively. Note that
G.fast has a usable bandwidth that goes up to 212 MHz.
6.3.1.1 PLC-to-DSL Cross-Coupling Channels
A measurement campaign that studies the PLC-to-DSL coupling environment over
the VDSL2 spectrum has been performed in [13, 14]. While this study provides insight
136
into the relation between the CM interference and the DM interference, results from
only two test-sites is not sufficient to extract a model for this relation. In addition,
only interference over the VDSL2 spectrum, i.e., up to 30 MHz, was considered in the
measurement campaign. The goal of this future campaign is to define a model for the
relationship between the CM and DM PLC-to-DSL coupling channels that considers
frequencies up to 212 MHz.
To achieve this goal, more measurements in residential houses are need. Note that
these measurements will be performed using the setup discussed in Chapter 3; however, modification to the balun and the PLC coupler is necessary to accommodate the
wider spectrum of G.fast. Various factors will affect the number of houses in which
the measurements will be conducted. First is the variance of the measurements. If
it is determined that the C2DTF does not significantly vary from house to house,
fewer houses will be utilized to extract a model for the C2DTF of the PLC-to-DSL
cross-coupling channels. Another factor that will influence the number of houses
used in this measurement campaign is the availability of the houses. Collaboration
with a Canadian telecommunications company is underway to determine the possibility of locating various houses for measurements. Once these houses are identified,
measurements to study the PLC-to-DSL coupling channels will be performed.
6.3.1.2 DM and CM DSL Direct Channels
To characterize the interference in DSL networks over the 212 MHz spectrum, the
DSL direct channels are needed. The DM DSL channels over the G.fast spectrum do
not conform to the standard two-port model [8]. Despite various research efforts and
proposed models, such as the model proposed in [70], there is much to learn about
the behaviour of the DSL channel in the 30 to 212 MHz frequencies [8]. In addition,
the current DSL channel models do not model the CM signal.
This measurement campaign will be performed in two phases. The first phase will
137
constitute measurements in the lab environment, since these high frequencies are only
suitable for short distances. The setup discussed in Chapter 3 is required to perform
these measurements with the exception of utilizing a balun with a 212 MHz range.
Once lab measurements have been conducted, field measurements will be performed
to substantiate the lab measurements. It is suggested that these filed measurements
should be performed in collaboration with a local exchange carrier because access to
both ends of the twisted-pairs is required. Note that since utilizing a VNA to perform
these field measurements is impractical, software radios will be utilized.
6.3.2 Interference Cancellation
Currently, for VDSL2 and similar DSL technologies, NEXT is eliminated via FDD,
while FEXT is mitigated via vectoring. The FDIC has the potential to eliminate
intrinsic interference, without the complexity of vectoring and the restriction of FDD.
In addition, given that TDD is proposed as the duplexing scheme, for G.fast DSL
technology, NEXT will return as the dominant crosstalk. However, the performance
of the FDIC in mitigating multiple sources of interference has to be studied in further
detail.
Preliminary results indicate that the FDIC has the ability to reduce interference
from multiple sources; however, the achieved improvement in SINR in presence of
multiple sources is not as high as the achieved improvement in SINR in presences of
a single source of interference. This is because the FDIC estimates the equivalent interference levels based on the average (or maximum) transmission power. Thus, once
actual transmission occurs in presence of multiple sources, the ratio of the summed
differential mode interference to the summed common mode interference does not
remain constant, while this ratio remains constant for a single interference source.
A potential solution to this problem is to utilize FDICs at the DSLAM (an FDIC
for each of the twisted-pair). At the DSLAM, all transceivers are co-located, and thus,
138
information about the DSL signals on each of the twisted-pairs can be shared among
the transceivers. This information can be utilized by the FDIC in the calculation of
the C2DTF. For instance, the actual transmitted symbol for each of the disturbers,
rather than the average (or maximum) transmission power, can be utilized by the
FDIC to calculate the C2DTF. This solution has to be examined in further details
to ensure its functionality and feasibility. In addition, the effect of uncontrolled lines
on this solution has to be studied as well.
139
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