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TRY THIS VERSATILE CROSSOVER
JANUARY 2007
US $7.00 • Canada $10.00
The Audio Technology
Authority
BUILD a Coax Horn
for better sound
How Do Cathode
Followers Stack Up?
A Switching System
for guitarists
Sub Amp Design
project begins
www.audioXpress.com
Expert Tips on Grounding
& System Interfacing
Cover-107.indd 1
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Industry News
H
IFICRITIC is a new bimont h l y s tere o aud io
magazine entirely funded
by subscribers. Delivering
definitive, well-researched reporting,
HIFICRITIC is headed by Editor Paul
Messenger and Technical Editor Martin Colloms. Covering a generous mix
of features and equipment and music
reviews, the first issue was published
on December 1. For more information,
visit www.hificritic.com.
Acoupower has dropped the price
for the 15 Bully Subwoofer, to $299 in
quantities of two, including US shipping. The 12 Bully Driver is now being
developed, and will soon be available.
The Acoupower website now has a page
devoted to amplifier recommendations
for the company’s subwoofer drivers. See
www.acoupower.com/amplifier.php for more.
Design Build Listen recently released an assortment of solid brass knobs,
available in 30mm or 50mm diameters
and designed to complement the company ’s ezChassis pre-punched cabinets. For more information, visit www.
designbuildlisten.com.
DH Labs has moved to their new
headquarters. The new address is DH
Labs, Inc., 9638 NW 153rd Terrace,
Alachua, FL 35615.
The TM400, f rom Lectrosonics,
simplifies the measurement process by
eliminating long cable runs between the
calibrated microphone and the test equipment. Now, however, some are using the
system for recording ambient sounds at
concerts. Due to its Digital Hybrid Wireless technology, the TM400 offers higher
dynamic range than wireless systems with
comparators. To learn more about Lectrosonics, go to www.lectrosonics.com.
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audioXpress 1/07
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After significantly simplifying the
creation of electronic circuits for hobbyists, SchmartBoard has been nominated
for a 2006 World Technology Award in
the category of IT Hardware. Honoring
individuals from 20 technology-related
sectors viewed by their peers as being
the most innovative and doing the work
of the greatest likely long-term significance, the World Technology Awards
are presented by the World Technology
Network, in association with the New
York Stock Exchange, Dow Chemical,
TIME Magazine, and CNN.
The IDS-25, from Haig Audio, based
on a design by Roger Russell, is a singleline array, full range system, requiring
minimal power. The original design was
one that Russell originated during his
career at McIntosh Labs, and was built
under two of his patents. For more information, visit www.IDS25.com.
New product literature is now available for illbruck acoustic, inc.’s SONEXvalueline Panels. The Panels, effective at absorbing excess sound at middle
frequencies where unwanted noise and
reverberation can interfere with communication, have noise reduction coefficients from 0.75 to 1.05. Visit www.
illbruck-acoustic.com/vlit to obtain the
product literature.
Aperion Audio has released the Intimus 533-T Tower Speaker, featuring
beautiful midrange in a compact package. Also from the company is the 634VAC, the first speaker to use VoiceRight
technology, which compensates for the
effects of reflected sound off large screen
TVs and cabinets. To see the company’s
products and more, visit the newly redesigned website at aperionaudio.com.
Digi-Key Corporation received Pelco’s “Supplier of the Year, New Product Support” award, and the President’s
Corporate Award for 2005, from Murata Manufacturing Company, based on
sales expansion and Digi-Key’s overall
contribution to Murata. The company has also signed a global distribution
agreement with semiconductor brand
SMSC, providing application specific solutions. Matrix Orbital, a manufacturer
of LCD solutions, transducer manufacturer CR Magnetics, antenna solutions
developer Antenova, Fox Electronics,
Future Designs, Inc., Quatech, Amphenol Connex, Delta Products Corporation, Ember, and Conec have all
inked global agreements with Digi-Key
as well.
Model 465, from TDL Technology,
is a six-channel volume controller, featuring an all analog circuit, master vol-
ume control, individual channel volume
controls, and IR remote control. Also
from TDL, the Model 444A Stereo
Headphone Amplifier features a noninverting output, low noise, wide bandwidth, and works with virtually all types
of headphones including low sensitivity
and low impedance models. For more
information, visit www.tdl-tech.com.
Mouser Electronics, Inc. has signed
a global agreement with SchmartBoard,
manufacturer of electronic prototyping system products. Mouser has also
penned an agreement with SMSC to
distribute the company’s embedded I/O
controllers, USB transceivers, Ethernet
products, and more. The company released its third catalog of 2006, featuring
1,808 pages of the newest products and
latest technologies. To find more information, go to www.mouser.com.
Consumer electronics manufacturer
ARCHOS, Inc. introduced the Generation 4 line of portable media players,
which hold up to 700 hours of TV content. Products include the ARCHOS
www.audioXpress .com
11/21/2006 3:16:33 PM
404 and 404 Camcorder, the 160GB ARCHOS 504, and
the ARCHOS 604 and 604 Wi-Fi. Featuring full DVD
resolution, the products can play all standard video formats,
and offer TV recording with the DVR Station accessory. For
more information, please visit www.archos.com.
New from OPPO Digital, the DV-970-HD Universal
DVD Player with HDMI, with DivX certification, can playback various media, and converts video to high definition
resolutions from standard definition for HD compatibility.
To learn more, visit www.divx.com.
Klipsch introduced the new iGroove HG, an MP3 shelf
system now available in high-gloss black, as well as the KL7800-THX in-wall LCR and KS-7800-THX in-wall surround, versions of the
THX Ultra2 custom
home theater system.
In other company
news, klipsch.com was
re-launched to better
serve goal-oriented visitors, while the brand is partnered with
LivingHomes, LLC, a developer of prefabricated homes,
which will feature Klipsch speakers. Visit klipsch.com for more.
The MPX1000 HD Multipoint Extender, from Avocent,
provides connectivity for moving high-definition content
from one source to multiple destinations. Extending video
and audio over standard 10/1000 Ethernet wiring up to
3000´ and wirelessly through walls up to 150´, the MPX1000
HD has interchangeable modules for input of analog VGA
signals or digital HDMI/DVI signals. Also from Avocent,
the ECMS2000U is a digital workstation extender, providing hardware-based extension for digital/analog video, USB
keyboard and mouse, USB media, and audio signals. Consisting of a computer node and user node interconnected
in a point-to-point manner at Gigabit Ethernet rates using
IP protocols over a single UTP cable, the ECMS2000U
removes the requirement for conditioned power in the edit
suite and greatly reduces ambient noise. For more information, please go to www.avocent.com. aX
CONTRIBUTORS
Ed Simon (“A Combination Horn You Can Build,” p. 10) received his B.S.E.E.
at Carnegie-Mellon University, and has installed over 500 sound systems at venues
including Jacob’s Field, Cleveland, Ohio; Museum of Modern Art Restaurants, New
York; The Forum, Los Angeles; and Fisher Cats Stadium, Manchester, N.H.
Christopher Paul (“The Cathode Follower and Its Weaker Siblings,” p.
22) has written a number of tube-circuit articles for audioXpress.
Gary Galo (“Grounding and System Interfacing,” p. 26), Audio Engineer at
The Crane School of Music, SUNY Potsdam, has authored over 230 articles and
reviews on audio technology, music, and recordings.
Rudy Godmaire (“A Flexible Subwoofer Amp Pt. 1,” p. 34), a sales consultant for Bell Canada, has been interested in DIY audio since 1998.
Steve Stokes (“A Unique Crossover Design With Waveform Fidelity,” p.
42) is a former member of the AES and co-inventor of a Dipole Speaker System for
Surround Sound. This is his first article for audioXpress.
Dennis Hoffman (“Low-Level Analog Switching,” p. 51) is an associate
engineer in the Controls and Power Electronics Department of The Stanford Linear
Accelerator Center. This is his first article for audioXpress.
Dennis Colin (“Book Review: The Art of Linear Electronics,” p. 62) has
demonstrated the audibility of phase distortion at Boston Audio Society, and
has designed the “Omni–Focus” speaker (bipolar coincidental with phase–linear
first–order crossover), ARP 2600 analog music synthesizer, 1kW biamp and PWM
supply at A/D/S, and Class D amps.
audioXpress January 2007
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11/21/2006 3:16:37 PM
speakers
A Combination Horn You Can Build
In designing this coax horn speaker, the author gives much
consideration to wide coverage in the unit’s application.
I
magine going to a concert and
finding out that your seat had a
great sight line, but the sound system covered only one person in
the audience. Would you join the line of
people asking for their money back?
Many loudspeakers intended for
home use have a very narrow sweet spot.
Is there an implicit assumption that people who listen to music have no friends?
Maybe these speaker designers have
never been moved enough by the music
that they wanted to get up and dance?
Do these people really sit in a chair and
just listen to music?
The reason to design loudspeakers
with a narrow coverage angle is to reduce the effect of the room’s acoustic
character on the reproduced sound. A
difficulty is keeping the coverage angle
uniform over the entire audio frequency
range, due to the large variation in wavelengths.
rooms. Some folks even prefer directional control in otherwise good rooms.
This is one of those areas of audio open
to enlightened debate.
Because I like to listen to music while
working at my desk, in the shop, or pretty much everywhere, sometimes other
people listen with me, which requires a
different set of conditions. My preference for music reproduction is for the
room acoustics to enhance the sound.
To me this requires a room with less
absorption and rising reverb at low frequencies, no hard focused echoes, and a
smooth short reverb tail.
For those who think the room should
add nothing to what is coming out of
the reproducer loudspeakers, I suggest
you visit an anechoic chamber. Many
recording studios approach that level of
absorption; this is one of those audio
points on which opinions may differ,
and the other guys are just wrong.
CONTROLLING DIFFICULT ROOMS
BOOKSHELF EFFECTS
The simplest method to produce pattern
control is the sound column, in which
multiple drivers all reproduce the same
range. Due to the length of the column, the resulting interference and reinforcement pattern reduces the long
axis coverage angle. A 20Hz tone has a
wavelength about 56´. This would require your sound column to be 28´ tall
to limit the dispersion on a single axis to
about 45°. At 20kHz the same column
would need to be just under ⅜˝. One advantage is that by confining the energy
to a smaller area, more energy is delivered there, thus the on-axis sensitivity is
higher.
There are other pattern control devices besides a sound column, such as a
horn system or a phased array. Controlling the coverage angle or dispersion is
certainly a valid approach for difficult
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To achieve a rising reverb time at low
frequencies requires solid walls, floor,
and ceiling. If the surfaces are flimsy,
some low-frequency energy will flow
right out of the room, be lost moving
the wall materials, and to a minor extent reradiate back in, sometimes even
at a different frequency! This is different
than noise control where the goal is to
keep sound from annoying others.
The methods for isolation differ from
enhancing reverberation. For isolation
it is possible to use diaphragmatic absorbers, add mass, or (my favorite) loose
particle-filled floors. Imagine a normal
hollow floor filled with perlited gypsum
(kitty liter); as the low frequencies move
the particles, they rub against each other
and thus absorb the energy.
Normal room furnishings such as
carpet, drapes, and furniture absorb the
By Ed Simon
PHOTO 1: The completed coax horn speaker.
midrange energy and more so the highs.
It is possible to have too much absorption if there is good low-end containment; the unbalanced combination will
produce a muddy-sounding room. The
treatment is either less absorption or
special bass absorbers. The other end
is not enough high-frequency absorption. You can improve this with rugs,
furniture, or, for tweaks, foam or other
products from advertisers in this very
magazine.
Obtaining a uniform sound field—if
that is your goal—requires the basics:
no parallel surfaces that are untreated
and objects in the room that refract or
scatter the sound field. Think of the
sound field as a balloon. As you add air,
it becomes bigger, just as a sound wave
would propagate. If you press it against
a flat wall, it will give you a single reflection. Press it against a wall of furniture,
and it will show you the multiple small
surface imprints which will model the
smaller and smoother reverb.
www.audioXpress .com
11/21/2006 3:05:22 PM
A while back a friend with a TV studio asked me about the acoustics of his
control room. I did a quick survey: a
large room with enough volume to have
a true reverberant field for most of the
frequency ranges of concern, two speakers on the front wall, equipment racks to
the side, carpet on the floor, mediumquality acoustic tile for the ceiling and
drywall on the back wall. The reverb
time was good for a room this size; almost all of the sound hitting the mix
position was well behaved except for a
little too much echo from the back wall.
The solution was “some bookshelves on
the back wall.” Someone had suggested
that it was more fitting to install some
specialty panels on the back wall, add
diffusers to the ceiling, and cover many
of the walls with closed cell foam.
Shortly thereafter I had the opportunity to meet with many of the major
manufacturers of audio test gear, so I
scheduled a measurement session in his
space with five or six of the equipment
manufacturers to demonstrate their
gear. After the measurements (T.E.F.,
T.D.S., S.T.I., and so on), my friend
asked the group for their suggestions.
One of the invited engineers said in his
impeccable English (with just enough of
his native Danish showing), “Oh, some
bookshelves on the back wall are all you
need.” The rest of the group agreed. The
friend was sure I put them up to it, but
he installed three bookshelf units and
saved a small fortune.
In most small rooms, bookshelves or
other large furniture will provide enough
short diffuse reverb to complement the
music. If the length-to-width-to-height
ratio and sizes in the listening room
are wrong—causing buildup of specific
frequencies—lots of padding will help,
but not fix the problem. If the walls or
ceiling is not substantial, low frequencies
will just flow right out of the room. The
simplest fix is to avoid bad rooms for
your listening area.
SPEAKER REQUIREMENTS
Now what you need is a loudspeaker
that sounds good off-axis as well as dead
center. It would also be nice if it was efficient (or sensitive), went smoothly low
and high, had great transient response
and low distortion, and was small, cheap,
and easy to build.
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A symphony orchestra plays a Forte at
I want the tweeter to be about ear
95dBa (slow weighting) in rehearsal and height to allow the sound not to be
somehow manages to get this to 105dBa blocked by furniture. I could build a tall
(slow weighting) in a performance. The narrow version of a two- or three-way
players will tell you a Forte is a Forte no sealed box or perhaps bass reflex speaker
matter when they play it. So my meter to meet these parameters, but I suspect
must be wrong.
using a horn-type loudspeaker will give
A rock concert contract rider fre- me the increased sensitivity the smaller
quently asks for 102dB at 100´. A sym- amplifiers I prefer require. A well-dephony requires about 30dB of headroom signed horn can also decrease the distorwith a Class AB amplifier to prevent my tion of the driver.
hearing clipping in the sound system.
To get a match to the high-frequency
Rock music needs only about 20.
horn requires a low-frequency horn of
So if I want to play music at concert enormous size or attenuating the highlevel, my loudspeakers must be capable frequency driver. One of the early highof 135dB peak level for a performance efficiency loudspeaker designs placed
but only 125dB for a more relaxed lis- the loudspeaker in a corner as part of
tening session. Allowing for two speak- the horn design using the three planes
ers, room reflections, the 10dB advan- to extend the horn size. This should give
tage of a Class A amplifier, and listening great bass response.
position, 115-120dB peaks from each
Unfortunately, bending the midrange
loudspeaker should be fine. This is quite around corners is not a good idea. So
a bit more than is available from many for a first try I will use a direct radiathome loudspeakers.
ing midbass to midrange, a horn on the
If you put a
500W Class A
amplifier on an
88dB per watt
loudspeaker, the
power compression will probably leave you 4
or 5dB short. Try
a 2000W amp,
which will get you
there very quickly.
It might also not
sound as good at
lower volumes.
Most folks understand it is easier
to build a goodsounding (or more
precisely, a not
bad-sounding)
small amp than a
PHOTO 2: JBL driver and crossover.
large one.
Engineering is knowing how to cal- mid to highs, and a horn off the back of
culate and adjust each parameter of the the low-frequency driver to get the very
design to get the desired overall result. lows. That way I can use a fairly stanArt knows which trades to make and dard two-way driver system.
still achieve pleasing results.
For a matching three-way horn sysI am willing to give up size for a loud- tem, I probably would want the midspeaker, but not floor space. In a listen- range horn to be at least 64˝ in length.
ing room bookshelves should be on the I could go a bit shorter and buy a comopposite wall from the loudspeakers. A mercial horn. There are three strikes to
floor-standing loudspeaker is a reason- that approach: one, it would make the
able first try for a design.
speaker bigger than can be unobtrusive;
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11/21/2006 3:05:31 PM
two, there would be midrange to high
crossover issues; and three, it’s more fun
to build it all.
A CLASSIC DESIGN
To keep the crossover region smooth and
coverage uniform, the high-frequency
driver should be close to the midrange
source. One of the classic designs is the
coaxial loudspeaker, in which the tweeter
is mounted centered inside the woofer.
The problem with many coaxes is that
the tweeter blocks the higher midrange
frequencies.
One design that avoids this is a
through-the-magnet horn design, in
which the magnet structure for the
woofer is hollow and shaped to form a
horn section for the tweeter’s output to
pass through the woofer. In addition, the
woofer cone forms the rest of the horn.
This usually results in a wide dispersion
driver system. There are, of course, times
when you would want a different horn
for better pattern control, but that is not
the goal here.
This idea has been around since at
least 1930. Advantages are that there is
complete symmetry of coverage at all
angles around the loudspeaker because
there is no offset between drivers. This
allows the room to add its sound without being colored by a single wall, the
ceiling, or the floor. The tweeter does
not block the midrange, and it is also
easier to get a good crossover match.
The disadvantage usually cited is that
the voice coils of the woofer and tweeter
are not in the same plane. One of the
terms that is often used and not well
defined is acoustic center, so I will avoid
using that term. (I get lots of folks telling me the definition, all different, but
Rudy Bozak’s is the earliest I know of.)
In a dynamic loudspeaker a voice
coil is suspended inside a magnet structure. (Except those in which the magnet moves and the coil is fixed!) A current applied to the voice coil causes it
to move. The coil is firmly attached to a
piston, which moves the air and you hear
the sound. The assumption is then made
that for time coincidence of two drivers covering the same frequency region
(think crossover overlap), the voice coils
should be even (in the same plane) with
each other.
One big problem with this idea is
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Simon2701-2.indd 14
when you attach the voice coil to a long light a woofer cone as possible. This will
very low mass pipe and then connect allow faster propagation at the crossover
the pipe to the cone. The movement of frequency (smoother midrange), and a
the cone is not changed by the length of low mass cone is more efficient in a lowthe pipe! The time at which the sound frequency horn.
comes from the cone is not changed.
The maximum interface with the air in WOOFER SELECTION
a typical cone occurs near the forward One of the limitations of cone loudedges of the cone! With a horn this oc- speakers is the “mass break frequency.”
curs at the outer edges.
Many loudspeakers show wonderfully
I have performed experiments to dem- flat frequency response curves on-axis.
onstrate this. If you build a loudspeaker When you look off-axis you see they
out of two drivers, the coverage narrows start rolling off at a much lower freas though the center of propagation of quency. That’s because as the frequency
the wave front is closer to the edge of increases the cone is large enough to
the cone, not the center. It doesn’t make begin to control directivity all by itself.
it all the way to the edge at higher freA 15˝ rigid piston would be almost
quencies because
of cone breakup or
something yet to
be determined.
If the tweeter is
smaller and lighter
than the woofer,
there will be some
time delay to get
from the voice coil
to the edge of the
horn, but it will
be different from
the same motion
PHOTO 3: The speaker glued and clamped with every piece except
propagating to the
the last side.
edge of a more
massive cone. I would need to measure 6dB lower in output at an angle 45°
the result to properly design a time delay from dead center when producing a freand align these wave fronts to ensure a quency of only 450Hz. If the piston were
smooth crossover region. This is one of to move the same distance at 450Hz as
the reasons some folks prefer a single it did at 225Hz, the SPL on-axis would
driver system. This is an area where phi- need to rise by about 5dB. The energy
losophy must meet finite element analy- must go somewhere, and if not off-axis,
sis or actual measurements to yield truth. then on. So most models of direct radiThe tweeter requires a horn to get ating loudspeakers have the design pathe desired control, loading for efficien- rameters adjusted to yield a flat on-axis
cy, and wide dispersion. The through- response.
the-magnet coaxial loudspeakers under
Most real loudspeakers have a limit
consideration use the woofer cone as that as you move them faster and fastthe horn, so there is not a large horn to er the mass of the cone will cause the
block the midrange. The concern raised motor, the cone, or connection to the
then is Doppler shift distortion caused motor to run out of capacity. Rememby the interaction between horn walls ber the equation is ½ × Mass × Velocity
(the cone) moving and the waveform squared! It is the V squared component
(high frequencies) being shaped. Be- that rises rapidly.
cause the cone will be loaded by a very
The point at which the loudspeaker
large horn at low frequencies and will cone can no longer act as a rigid piston
not move much, and with Doppler dis- is called the mass break point. Of course,
tortion not being very distracting, this is there is some flopping around as to the
not an area to really worry about.
exact point.
It would also seem that you want as
The larger the loudspeaker, the lower
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11/21/2006 3:05:38 PM
the mass break frequency. A larger woofer requires a larger tweeter to reach the
frequency where the crossover must be
placed. The larger tweeter does not go as
high. The question then becomes what
frequency range is desired? Or the other
version is how loud?
You could use a 5˝ woofer, which
would allow a very good high-end
tweeter, but you would not have much
low-end energy. There is not enough
piston area to move the air at low frequencies. Even if you could get a long
excursion 5˝ woofer, you need to worry
about power dissipation, breakup under
horn load, and just plain small piston
area.
As you use a loudspeaker, some of the
energy causes the voice coil to heat up.
In a well-designed loudspeaker, the voice
coil can double in impedance before it is
damaged. The problem is that when the
impedance doubles, the current draw
for the same amp voltage is cut by one
half. Thus the speaker has 3dB less output than it should. The heat also more
slowly changes the magnet.
This loss of output is called power
compression. As the loudspeaker cools,
it gets back most of what it lost, but in
some loudspeakers the magnet slowly
weakens from use.
A good compromise for this design is
an 8˝ woofer. You can get a big enough
voice coil to not only handle the power
but also keep its cool. It has enough piston area to give 115dB output at the low
end. To get a light and stiff cone requires
some sort of reinforced material.
Obviously I am not the first person to
try this design philosophy. The idea of
an 8˝ woofer with a through-the-magnet compression driver without any horn
blocking the cone has existed for at least
60 years. It meets the criterion of a wide
coverage angle and is efficient in the
right enclosure. The transient response
is somewhat inherent in the design of a
light woofer with a compression driver
if it is a good crossover choice. Picking
one that is low cost and low distortion
should allow me to reach my design
goals.
Although the trend today is for small
bookshelf speakers or perhaps towers, I
can buy ready-made drivers from at least
six manufacturers. Looking at websites
for reasonable engineering data helps
narrow the choices.
ENCLOSURE DESIGN
For a first cut at design I need to choose
a moderate-cost loudspeaker that meets
these requirements. Keep in mind this
includes a crossover with decent-quality
capacitors and inductors, a compression
driver, a horn, and a woofer. I picked the
JBL Professional (not JBL Consumer)
Control 328C (Photo 2), which comes
in either a 70V or 8Ω version. Be sure to
get the 8Ω version.
This loudspeaker is designed to be a
wide-coverage ceiling unit rated at 93 to
98dB/W at 1m. Power handling is 1kW
peak. It really won’t do a peak of 128dB,
but then I won’t be using a 1kW amp. It
has a Kevlar-reinforced woofer cone, a
real compression driver, and comes with
a crossover, which even uses plastic film
capacitors that are glued to the PC card.
It is produced in reasonable quantity,
and as such is less costly.
The 328C comes attached to a ported
baffle that also doubles as part of the
audioXpress January 2007
Simon2701-2.indd 15
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11/21/2006 3:05:45 PM
horn. I will recycle the baffle, keeping the amp’s output. The curves they pubin mind the new enclosure must act as a lished were predicted by home-grown
horn extension. These units even come software and never confirmed!
with serial numbers—mine were 10403
Part of the design process is to be sure
and 10405. It is unusual for a ceiling you can actually make what you design.
speaker to have serial numbers! If you I had some 1˝ particleboard left over
prefer you can try a different speaker, but from making counter tops, so I made a
because this is not a common design you 3ft3 enclosure to try out the loudspeaker
will need to look around a bit. List price driver. My program showed that with
for the JBL is $320. Try not to pay that.
a 3˝ diameter port 2˝ long the low-freThe hard way to design an enclo- quency driver/box combo should be dead
sure (or a listening room) is with finite flat to 30Hz and then smoothly roll off.
element analysis (FEA). With this sys- My measurements showed a 40Hz
tem the air surrounding the loudspeaker rolloff with a slight bump. This means
system is divided into small blocks (fi- that either I can’t measure very well, the
nite elements), and each block is given loudspeaker parameters are wrong, my
a model value of resistance, capacitance, prediction software is not perfect, or any
and inductance. A stimulus is applied of several other causes.
(the speaker cone moves) and the comThere were also big bumps in the mid
puter then calculates how each block in- frequencies. One dip was caused by the
teracts with its neighbors. The best way wavefront coming off the back of the
to figure out how big each block should cone hitting the enclosure back wall,
be is to try a size and then do the same then bouncing forward to cancel some
problem again with a smaller size block. of the outgoing wave. Two inches of
Quit making the blocks smaller when fiberglass on the back was not enough.
you can no longer see the difference in Six inches fixed that. There were also
bounces caused by locating the speaker
outcome.
An easier way
is to use one of a
number of classical equations that
you can figure out
with a pocket calculator. Be sure to
measure the result
to see whether the
equation you used
worked. Mine
never do, which is
why I use an FEA
program.
It is really imPHOTO 4: The box just before final glue-up.
portant to measure
after you build to see how accurate your on my workbench. Moving the microdesign was. It may seem silly, but after phone and loudspeaker showed which
you measure enough, eventually some- bumps they were.
thing you never saw pops out at you.
Now I could try some horn designs,
I once did a job using a specific loud- having a feel for the limit of my despeaker, whose published f requency sign method. I was able to almost preresponse curves did not match what I dict the low-frequency response and its
measured. I measured the amps provid- smoothness. The program was able to
ing in excess of 100V to the tweeters, show several design options based on
yet there was inadequate output and no my design goals of size, speaker position,
tweeters were blowing up! An examina- ease of construction, and low-frequency
tion of the loudspeaker showed that the response.
crossover passed no signal above 8kHz
I saw that using a good corner load
to the tweeter. The actual power making with a reasonable length horn would get
it into the tweeter was less than 3% of me close to what I wanted. I had a mid16
audioXpress 1/07
Simon2701-2.indd 16
www.audioXpress .com
11/21/2006 3:05:48 PM
bass dip I didn’t like, so I tried placing a filter before the horn
mouth to limit the horn to the lowest frequencies. Because
the programs’ limit seemed to be 20%, or about 2dB, going
much more refined without testing seemed pointless.
CONSTRUCTION CONCERNS
Being cheap, I wanted to use as little wood as possible in the
design. Wood comes in either 48 × 96 (sometimes 49 × 97)˝
sheets or in 60 × 60˝ sheets. Most of the lower-cost wood
products are stocked in 48 × 96˝ sheets. I figured setting the
cabinets about 60˝ high at ear level would allow use of either
size of wood.
I do not like harmonically related dimensions for acoustic
enclosures. Ratios such as 1:√2:√3 are good. Using one sheet
of wood would require getting eight sides out of the 48˝
width. That would make the small side 4.75˝ and the big side
7˝. An 8˝ woofer would not fit in the box.
With two sheets you could get 9.5˝ × 13.25˝, including
overlap at the sides. That would give an internal volume of
over 4ft3. Allowing half that for the woofer chamber and the
other half for the horn would allow you to have a low-frequency speaker contribution down to about 38Hz, or so the
prediction program tells me. That is almost low enough. But
I was almost wrong before, so caution is in order.
Spring for one more sheet. You now have enough to allow
for mistakes. If the box internal dimensions are 12.75˝ ×
18.25˝ × 60˝, there should be adequate volume without any
overlapping resonances. You have a good size. It should be
able to go down to the 20Hz range and still fit nicely in the
corner of the room.
Being lazy, I designed the box with a straight horn. At the
low frequencies this seems to have very little effect on the
horn. A simple low-pass filter was predicted, so I used the
simplest filter I thought I could get away with. This amounts
to a single piece of wood. I placed a small thin piece of fiberglass in the filter passage just in case there were any side wall
problems.
For ease of construction you can make the design of 1˝
MDF, HDF, particleboard, or plywood. The preparation of
the material requires only straight cuts. To make the speaker
“furniture,” either paint or veneer it. If you use ¾˝ material,
you might choose a slightly fancier edge, either a dado or a
miter. Be sure to double-up the front baffle if you use thinner
material to allow for recessing the driver into the baffle.
No matter how you decide to build it, this is a two-person job. The finished speaker should be heavy. If you use ¾˝
material, you can keep the same external dimensions—it will
not make much difference. There is enough extra material to
make the extra baffle backer pieces.
The first step after obtaining the parts is to disassemble
the loudspeaker. Carefully unplug the woofer and tweeter
connectors. JBL has cleverly used different size connectors
for each terminal. It requires real imagination to reassemble
them incorrectly. I know.
Next remove the screws that come with the baffle. Remove
the crossover from the baffle. Keep the rubber crossover
mounts, which will help decouple the crossover from the enclosure. Be careful not to get dust into the drivers. The dust
audioXpress January 2007
Simon2701-2.indd 17
17
11/21/2006 3:05:53 PM
cap on the woofer is really just a grille
for the tweeter and will allow small stuff
in, so be careful.
The cutting pattern is shown in Fig.
1. If you use ¾˝ material, don’t forget the
baffle backers, which should be 12.75˝ ×
about 14˝.
When using any power tool try to end
FIGURE 1: The cutting pattern.
the job with the same number of fingers
you started with. Long, straight cuts may
seem simple, but saws kick, people place
their digits in the strangest places, and
rip cuts with power tools account for
most of the lost fingers in small shops.
Of course, you will wear eye protection.
As someone interested in audio, you will
also use hearing protection. And because
breathing seems to be a hard habit to
break, you will also don a dust mask.
You can set up the wood on four sawhorses or some other support system.
If you have a tablesaw, you can cut to
rough size and finish on it. The pieces
18
audioXpress 1/07
Simon2701-2.indd 18
will fit better if you cut all of the same
dimensions at the same time without resetting the fence. If you don’t have a tablesaw, just be careful, clamp a straightedge to the board, and follow it with the
saw to get a finish cut. If you don’t have
a straightedge, cut one from the third
board. Then use it to cut the rest.
I prefer to start
with the shorter
cut across the
material on each
piece. Then I do
the longer straight
cuts. That way my
material is balanced better on
the sawhorses and
I am not trying to
move many heavy
pieces.
You c an use
square cuts at the
edges of the two
boards that form
the “Z ” of the
horn—a small
gap will have no
effect. However,
be sure they are
securely glued,
or a buzz could
occur if the pieces
rub. If you prefer,
you can bevel the
edges about 17° to
form a tighter fit.
You can cut the
bottom of the “Z”
last. Trim it flush
with the back or
recess it a bit if
you want to place
a grille on the horn mouth.
After you’ve cut all the pieces, rout the
driver cutout. If you use two pieces of ¾˝
wood, you can probably get away with cutting two circles. The one in the front piece
needs to clear the entire driver basket. The
rear hole should be smaller to allow you to
screw the driver into the baffle.
The driver hole should be 6½˝ down
from the top of the baffle board and 1˝
off center. I just don’t like symmetry in
resonant locations.
I used a template to rout the hole.
Using collars and a template allows me
to use a big collar to rout all the way
through the baffle. I can then use a
smaller collar to rout the recess or I can
use a bit with a guide bearing to follow
the outline of the hole and cut the recess. I did one speaker each way.
On the first try I did not recess the
driver. After listening to it, I used a
panel cutting bit with a bearing guide to
simulate the horn flare that came with
the speaker. Try to get the speaker lip
about 1/8˝ inside the baffle board with
a smooth curve to the surface. The goal
is to copy as close as possible the baffle
that came with the driver.
If you do not own a router, just cut
a round hole and surface-mount the
driver. There will be a small difference
in the final sound. If you use ¾˝ wood,
mount it to the backer and clear through
the baffle. You might want to round the
edge with a file, a surform, sandpaper, a
dremel tool, or scrape it with a knife.
If these directions seem to lack more
detail, it is because you can use just about
any method to make the opening: a reciprocating saw, a router, or even a handsaw (HDF cuts very easily). Just make a
round hole about where it should be. It
is better if you make a super precise recess mount that copies the original, but
it is not that big a deal.
ASSEMBLY
The horn seems to form a classic “Z”
shaped folded horn, but it doesn’t. The
top board of the “Z” is parallel to the
top of the speaker. It is spaced to form
a chamber with exactly 2 × 12.75˝ of
cross-section area. It is long enough
(14˝) to form a resonant chamber. This
is the low-pass filter to keep low mids
out of the horn. It seems way too simple,
but I tried it both ways and the filter is a
big improvement.
The middle board of the “Z” fits
tightly into the bottom front corner
of the box. It is the same size as the
front baffle. It should end about 2˝ from
the top and back. The exact placement
should be close. It is more important for
the filter board to mate cleanly and stay
parallel to the top.
The board at the bottom of the “Z” is
part of the horn. Its exact placement is
not critical. It should touch the middle
“Z” board about 6˝ from the bottom and
go to the back bottom. You can test-fit it
and trim to length.
www.audioXpress .com
11/21/2006 3:05:54 PM
Be sure to place all the pieces together
before gluing to be sure they fit. If it is
a small piece that is being disagreeable,
you should have enough scrap to make
another piece. If it is one of the bigger pieces, try trimming all the affected
pieces. A ¼˝ change in any dimension is
not radioactive.
If you are as sloppy at cuts as I can
be, you will really appreciate the modern urethane glues, which foam up as
they set. This seals all those annoying
gaps you get when your saw cuts are not
perfect. I prefer Elmer’s ultimate highperformance glue. The cap closes tighter
than the other well-known brand so the
glue stays good longer. The downside
is you should wear gloves when using
this glue and be sure to put a drop cloth
under the workpiece. Of course, ordinary
wood glue or the improved yellow stuff
is also a good choice.
You can just glue and clamp the pieces, or, as in the more classic home-built
method, use glue and screws every 6 to
8˝. You don’t need to clamp it if you use
screws, but it can’t hurt. You didn’t hear
this from me, but nails and glue will
work also.
I started with the side down and fitted the bottom piece, then the front
baffle, followed by the top. The long
diagonal of the horn fits into the front
bottom corner. It does not need to be
tapered; a small leak here is unimportant. At this time, mark two lines about
6 and 7˝ up from the bottom end, and
then put it in place.
The top of the horn “Z” is next. Be
sure you have a uniform 2˝ channel from
the top. The bottom of the “Z” should
fit between the two lines you made on
the long horn diagonal. Check it for fit
and trim the length. Finally, fit the back.
Check the other side fit. Glue everything except the second side.
FINISHING TOUCHES
As you can see in Photo 3, the speaker
is glued and clamped with every piece
except the last side. This allows placing
the crossover and fiberglass into the box
when it is easy.
Photo 4 shows the box just before
final glue-up. The crossover and fiberglass are installed. A very small thin
peeling of fiberglass is placed in the passage that forms the low-pass filter. This
should be about ½˝ thick and about 10˝
long.
Five screws hold in the crossover. I
kept the factory mounting grommets
and just used drywall screws to mount it.
Be careful so that it will come out later
if needed.
You can put a cup for the terminals
anywhere in the back of the loudspeaker.
The crossover has a well-marked removable connector for the wire. It is
designed for 12-gauge or smaller wire.
I had no trouble using 10-gauge wires.
The connector has four terminals to
make it possible to daisychain the speakers in ceiling use.
If you want to be a tweak, rebraid your
speaker wire and use all the terminals to
reduce the resistance of the connection.
If you use solid wire, you can use either
just one cup or a jumper. I simply ran a
piece of West Penn Wire /CDT 25210
10 Ga. wire from the speaker connector
out the back of the horn.
When the innards are done you can
glue on the remaining side. Use screws,
clamps, or both. After the glue sets you
can finish the box. Carpet or fabric is
good if you want the ’80s band look. I
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181 x 120January
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19
11/21/2006 3:05:56 PM
suggest painting it the same color as the
room where the loudspeaker will live. I
recommend veneer and a grille frame.
The crossover has quick connects for
the driver and a plug for the speaker
wire. The driver mounts with drywall
screws. I did not use a gasket because
my routed edge was reasonably smooth.
There is not a great pressure differential
as in a sealed box speaker, which is why
the seal is not very important. If you
prefer, run a light bead of silicone sealant
to seal the driver to the box.
It will take two humans to place the
speaker, which should be at a 45° angle,
tight to the corner of the room, but ½˝
from the walls. If the box is out too far
you will get bumps in the bass response.
You may wish to jiggle the placement to
get the low-frequency response to suit
your taste.
The first loudspeaker I tried had the
mounted driver protruding from the
baffle by a small bit. I also did not put
in the horn’s low-pass filter. When I first
turned it on, the results were not pleasing. Using an old Ashly PQ-66 parametric notch filter, I swept the speaker.
As you probably know, when there is
a bump up in the frequency response of
a speaker, it sounds bad. When it’s a dip,
you must listen for what is missing. The
sweep immediately showed that there
was not enough fiberglass in the box
(resonance around 400Hz), there was a
harshness around 3200Hz (impedance
dip), and I needed the bass horn’s filter.
I assembled the second speaker with
a routed recess approximating the original baffle, used way more fiberglass,
and added the filter. This speaker was
much better, so it was time to modify the
first one. I broke in the speakers for
three days.
TESTING
I tried three different amps with the
loudspeakers. One was a typical massproduced 60W class AB amp that produced quite pleasing results. I also used
the solid-state single-ended power amplifier from the April 2006 issue of audioXpress. It was more detailed than the
first amp. I used my improved “butterfly”
amplifier for most of the listening.
The loudspeaker proved quite disappointing. I know this is where the author is supposed to list some recordings
20
audioXpress 1/07
Simon2701-2.indd 20
he/she used to evaluate the loudspeaker
The plots (Fig. 2) show the speakers’
and pretend that most folks know the performance on-axis and as far off-axis
recordings intimately and can share the as I could get in the room. I used white
experience through common knowledge. noise with an FFT analysis plotted on a
This loudspeaker had the audacity log scale. This is unsmoothed data with
to make many of what had been satis- a 10Hz resolution. The low end is flat to
factory recordings of musical perfor- the limits of the measurement. I placed
mances sound like pale studio imitations the speakers on the narrow 20´ wall of
of music. I could identify that some of my office; the length of 45´ allows a
the choruses on the recordings were re- low-frequency resonance of 12 to 13Hz.
ally just sampled and replayed, not done
The limit on perceived low-frequency
live as I had previously thought. There performance was, to my surprise, afwas one song in which I used to think fected by the amp. I previously did not
the singer was tapping her foot to keep worry about low-frequency performance
time; it was actually the bass drum in the in amps, which used to be one of those
background keeping the singer on track. it-does-not-matter issues. Now for the
One of the CDs of a popular singer now first time I got almost unbelievable lows.
sounded so bad it could pass for a spoof I don’t yet have a feel for what I can and
of a performance.
cannot hear, but I will play with what
I am slowly learning that I prefer live I send to the speaker to see what are
recordings. Studio recordings are ad- my limits of audibility. The engineering
justed to the producers’/engineers’ tastes, process allows you to design a system,
while they are spending large amounts build it, test it, and see what you have
of time in a small padded room. I have learned.
better speakers, amps, and a more realThe problem with such low-frequency
istic listening environment. Somewhere response is twofold. First, some amps rebetween 4% and
20% of the recordings I listened
to with this loudspeaker could be
mistaken for an
actual live performance, depending on the type of
music and the recording methods.
Worse yet, a recording that really
did not impress
me now sounded
FIGURE 2: Speaker measurements.
wonderful. The
drums that could be heard before now ally do not have the frequency response
match the piano and balanced the piece to get the full response out of the loudto where it was downright nice (Edward speaker. Second, many recordings are
Simon 1995 KOKOPELI Records).
“engineered” on loudspeakers that do
The low-frequency response of this not go this low. If the bass is boosted to
system is far better than any loudspeak- sound good on the studio monitors, it is
er I have ever heard. The folks I have overpowering on this loudspeaker.
played it for all find that the low freBecause a loudspeaker is also supquencies are quite different than what posed to produce more than just bass,
they heard. The first thought is that I must admit that the design has some
there must be some obscene boost in low weaknesses. There is an impedance dip
frequencies. My measurements show the about 1.6kHz, which shows up as a
low frequencies are flat to the limit of slight harshness in the midrange that is
my test setup. I expected it flat only to amplifier-dependent. The upper mid25Hz. This time it is possible my model range seems a bit suppressed. The highs
erred by predicting too high a rolloff.
roll off around 18kHz.
www.audioXpress .com
11/21/2006 3:05:57 PM
I put two loudspeakers on a single
amp with a monaural source to judge
imaging and driver match. There was
one range of mid to highs and one mid
bass region where the image moved left.
This indicates the loudspeaker or the
room is not perfect. The movement was
about one-quarter of the soundstage.
This is not that bad for unmatched
speakers.
I do not hear the crossover between
the woofer and tweeter. I can look at my
measured curve to see where it is, but
the timbre difference you sometimes get
with horn-loaded compression drivers
does not jump out at me. Neither driver
strains to reproduce its range at normal
listening levels.
LET’S PARTY!
One demonstration I like to do is play
30kHz through a small loudspeaker that
has a sharp output resonance and ask
people whether they can hear it. Everyone says no. I then turn it off. Everyone
then agrees that it has gone off. I switch
it on and off a few times so that the subjects realize that they are hearing it, but
do not perceive it as a tone, just a pres-
ence. It is simply a matter of level as to
what you can sense.
This loudspeaker does not go that
high. I can still hear an amazing amount
of detail through the speaker, so I suspect much of the openness is not due
to frequency response on the high end
but rather good transient response in the
midrange. But this is something I will
need to play with to be able to quantify
it better.
This is not the be-all, end-all loudspeaker. It covers a whole room, plays
loud even with small amps, and goes
way low. It’s OK going high. It is reasonably smooth, but a few details may
be missing. Of course, this speaker driver
is designed for distributed ceiling use. It
does, however, make several of my older
high-end loudspeakers sound broken by
comparison.
The loudspeaker is great for three
particular applications. One is for aerobics—loud music, great bass, fits out of
the way in a corner, and is modest in
cost for a professional loudspeaker. The
second application is home theater. You
will have guests jump out of their seats
the first time they hear what is really on
some of the soundtracks. The third great
use is for music at a party. With its good
low-end uniform coverage, I recommend
mixed drinks or microbrews—this is not
a loudspeaker for cheap beer.
There are still things to play with.
First, I will try substituting air-core inductors for the two iron-core ones. This
is not a big issue, because of the low level
of power used with the loudspeaker. Second, I will try it bi-amplified.
I will use this loudspeaker with a digital processor such as a BSS Soundweb
or Ashly Protea. My CD will typically
send an AES data stream to the processor, which will be programmed to act as
source selector, volume control, equalizer,
and, if needed, noise gate. With only the
D/A converter and clock in what is now
my “preamp,” there are far fewer parts to
get in the way of the sound.
Oh, yes, I don’t mind using tone controls. . . how do you think the recordings
got made?
If you think that perhaps this design
could be adapted into just a monstrously
good subwoofer/speaker stand for existing speakers, you are right. But then that
is a different loudspeaker. aX
audioXpress January 2007
Simon2701-2.indd 21
21
11/21/2006 3:06:00 PM
tubes
The Cathode Follower and Its Weaker Siblings
This author’s results confirm that cathode followers are even more capable
of driving capacitive loads than equivalent common-cathode amplifiers.
I
n a recent article (“Rehabilitating Cathode Followers,” aX 5/06),
I compared the abilities of the
Cathode Follower (CF) and the
Common Cathode Amplifier (CCA)
to drive capacitive loads. I showed that
CFs and CCAs made from identical
triodes biased to identical plate voltages
and currents and driving identical loads
“disconnected” from their loads at the
same peak currents and voltages—one
on the load voltage downswing, and the
other on the upswing. The article also
discussed that the higher output impedance of the CCA left it at a disadvantage when it came to driving capacitive loads—that is, attenuation occurred
more readily at higher frequencies for
the CCA.
In a subsequent exchange of private
letters, it was pointed out that the CCA
could address this deficit by paralleling
resistances across a capacitive load (thus
extending bandwidth) and then increasing drive to reestablish the original level.
However, it also became clear that these
higher drive levels would increase distortion at the mid and low frequencies and
limit the peak voltage across the load.
It seemed that the CF still retained the
advantage.
If the CF is the preferred driver for
capacitive loads, it makes sense to more
fully explore and understand its characteristics, which you should consider in
comparison to those of CF-like circuits
that nevertheless operate distinctly differently from a CF. Over the years, I
have noticed that some people attribute
CF-like characteristics to many CF-like
circuits, apparently mainly because their
outputs are cathodes. Recently, I found
a manufacturer selling a preamp with an
SRPP output stage that claimed to have
a CF-like output impedance. The manufacturer subsequently confirmed by mea22
audioXpress 1/07
Paul-2727-4.indd 22
surement, to his surprise, that the SRPP
does not have the output impedance of
a CF, and changed the specification accordingly. A similar assertion regarding
an SRPP with a “low output impedance” was made even more recently in
the pages of audioXpress (p. 10, 10/06.)
I’d like to explore this and other differ-
By Christopher Paul
ences between the CF and its variants in
the rest of this article.
THE CIRCUITS UNDER
INVESTIGATION
There’s value both in performing circuit
derivations and in building and measuring the circuit on the bench. There’s no
FIGURE 1: Circuit for testing a CF, and SRPP or an MF.
www.audioXpress .com
11/21/2006 3:07:31 PM
denying bench results, but a validated
derivation is priceless when it comes to
getting a feel for how a circuit works
and for applying and optimizing it. So
I’ll start with a general-purpose schematic (Fig. 1) that illustrates the CF and
its variants and how to test them.
I’ll present measurements and calculations I made from them in Tables 1 and
2 to determine certain circuit parameters
of interest. I’ll also provide calculations
(Table 3) of the same parameters using
the equations I derived in the sidebar.
If you’re not interested in the derivations, you can skip the sidebar, although
I think you’ll gain some insight if you
review it. I’ll compare and discuss the
results of the measurements and the derived equations later.
In Fig. 1, 1100Ω resistors R2-R19
are strung together with 402Ω R1. You
can conveniently connect each point
between a connected pair of resistors
to Tap_? (one side of C1), affording a
means to test different cathode follower
(CF) variants. A connection of Tap_? to
Tap_V2 produces a cathode follower. A
connection to Tap_1 yields the top of an
SRPP (although in this case, C1 is not
really needed). Connections to Taps 219 yield the top of a Mu Follower (MF),
while a connection to Tap_0 provides
a convenient means of measuring the
rp (plate resistance) of the 6DJ8, as I’ll
show.
For the SRPP and MF, think of V3
and the resistors between it and the connection to Tap_? as the controlled source
and the resistance seen at the plate of
the bottom triode in those two-triode
configurations. With the CF, V3 is uniformly 0 (V2 is the active source), and
the resistors from R1-R19 are simply
cathode resistors. Note that the DC bias
of the triode is fixed for all variants at
a 5mA plate current and a 100V plate
voltage with the 6DJ8 I used.
Only one of the very low impedance
(under 10Ω) sources V1-V4 supplies a
signal at any time; consider the remainder to be short circuits. These 1kHz
signal sources are used to support the
measurement of three important parameters of the circuits I’m examining.
GAIN, OUTPUT
IMPEDANCE, AND POWER-SUPPLY
REJECTION RATIO (PSRR)
These three parameters represent excellent figures of merit for a circuit. Consider the equations (derived in the sidebar) which determine them in the Fig.
1 circuit. Capacitors are assumed to be
AC shorts and the 1M resistors are AC
open circuits. RT is defined as the sum
of R1 + R2 +. . . R19. For SRPPs and
MFs, Rk is the value of that portion of
RT between Tap_0 and the selected connection (Tap_0, Tap_1,. . . Tap_19) to
Tap_?. For CFs, Rk = RT. rp and gm are
the plate resistance and transconductance, respectively, of the 6DJ8.
Measure output impedance (Z0) by
first activating V4 and setting R22 to a
finite, non-zero value (I used 1100Ω in
my measurements, but its value is unimportant in the derivation). Other sources
are off and Ep is shorted to +200V. De-
TABLE 1: MEASUREMENTS AND THE CALCULATIONS OF GAIN, ZO, AND PSRR FROM THEM.
Rk Ω
0
402
1502
2602
3702
4802
5902
7002
8102
9202
10302
11402
12502
13602
14702
15802
16902
18002
19102
20202
RT = 20202
Ek, mV RMS
IR22, mA RMS Z0 = Ek/iR22Ω
(Z0 test)
(Z0 test)
(Z0 test)
398.0
364.0
287.0
240.0
204.6
180.9
161.6
145.7
132.3
124.7
114.4
106.3
100.8
95.0
89.0
84.8
79.7
73.8
69.9
68.1
0.093
0.124
0.194
0.236
0.268
0.290
0.307
0.322
0.334
0.341
0.350
0.358
0.363
0.368
0.373
0.377
0.382
0.387
0.391
0.392
4292
2943
1483
1018
762
624
526
453
396
366
326
297
278
258
238
225
209
191
179
174
Ek/V3 for MF
Ep/Ek
& SRPP: Ek/V2
for CF (gain test) (PSRR test)
0.217
0.483
0.727
0.812
0.858
0.886
0.906
0.922
0.934
0.942
0.946
0.950
0.952
0.958
0.962
0.966
0.966
0.968
0.970
0.972
V2 = V3 =
500mV RMS
Comments
1.3
test for rp
1.9
SRPP
3.5
MF
5.1
MF
6.7
MF
8.3
MF
9.9
MF
11.5
MF
13.1
MF
14.6
MF
16.3
MF
17.8
MF
19.4
MF
21.0
MF
22.6
MF
24.1
MF
25.8
MF
27.5
MF
29.1
MF
30.8
CF and gm test
Ep = 1V RMS
TABLE 2: CALCULATIONS OF RP AND GM FROM MEASUREMENTS USING
AUDIO TRANSFORMERS
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RTΩ
RkΩ
Ek, mV RMS
(Z0 test)
iR22, mA RMS
(Z0 test)
Z0 = Ek/iR22 Ω
(Z0 test)
rp Ω
20202
20202
0
20202
398
68.1
0.093
0.392
4292
174
5450
gm 1/Ω
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ELECTRA-PRINT AUDIO COMPANY
0.00553
4117 Roxanne Dr., Las Vegas, NV 89108
702-396-4909 Fax 702-396-4910
electaudio@cox.net www.electra-print.com
audioXpress January 2007
Paul-2727-4.indd 23
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11/21/2006 3:07:32 PM
termine Z 0 by the ratio of the cathode voltage to the cathode current fed
through R22 and C2, as was derived in
the sidebar:
PSRR is determined by the ratio of Ep/
Ek (which is always greater than 1 and
improves as it increases in value). I also
derived PSRR in the sidebar:
Z0 (measured) = Ek/IR22
Z0 (derived) = 1/(1/rp + (1 +gm×Rk)/RT)
PSRR (measured) = Ep/Ek
PSRR (derived) = rp/Z0
The Power-Supply Rejection Ratio
(PSRR) specifies how well power-supply noise is rejected at the circuit output.
Measure it by first activating V1. Other
sources are off and R22 is removed.
Measure gain by first activating source
V2 for a CF or V3 for an SRPP or an
MF. Other sources are set to 0, Ep is
shorted to +200V, and R22 is removed.
Gain is determined by the ratio of Ek to
the level of the active source (V2 or V3)
and was derived in the sidebar:
GainM,S (measured) = Ek/V2
GainM,S (derived) = ((1 + gm× Rk)/RT) × Z0
for the MF and SRPP and
GainCF (measured) = Ek/V3
GainCF (derived) = gm × Z0
for the CF.
TABLE 3: CALCULATIONS OF GAIN, ZO, AND PSRR FROM THE SIDEBAR
DERIVATIONS, USING TABLE 2 VALUES OF RP AND GM.
RkΩ
Z0Ω
gain
PSRR
Comments
0
402
1502
2602
3702
4802
5902
7002
8102
9202
10302
11402
12502
13602
14702
15802
16902
18002
19102
20202
RT = 20202
4292
2916
1553
1059
803
647
541
466
408
364
328
298
274
253
235
220
206
194
183
174
0.212
0.465
0.715
0.806
0.853
0.881
0.901
0.915
0.925
0.933
0.940
0.945
0.950
0.954
0.957
0.960
0.962
0.964
0.966
0.968
1.3
1.9
3.5
5.1
6.8
8.4
10.1
11.7
13.3
15.0
16.6
18.3
19.9
21.5
23.2
24.8
26.5
28.1
29.7
31.4
SRPP
MF
MF
MF
MF
MF
MF
MF
MF
MF
MF
MF
MF
MF
MF
MF
MF
MF
CF
FIGURE 3: Measured and derived values of gain vs. Rk/RT.
FIGURE 2: Measured and derived values of output impedance vs. Rk/RT.
24
audioXpress 1/07
Paul-2727-4.indd 24
FIGURE 4: Measured and derived values of PSRR vs. Rk/RT.
www.audioXpress .com
11/21/2006 3:07:35 PM
MEASUREMENTS AND
CALCULATIONS
A summary of the measurements I made
is given in Table 1, which also contains
the values of the parameters Gain, Zo, and
PSRR calculated from those measurements. You can also calculate those same
parameters (Table 3) from their derivations if you first calculate gm and rp (Table
2) from certain measurements listed in
Table 1. (You could read gm and rp from
a chart of plate curves, but it is better to
use the values of the tube under test to
properly compare measured and derived
results.) Comparing calculations from the
measurements and the derivations validates both, and can be done graphically
(Figs. 2, 3, and 4), or more tediously and
precisely from Tables 1 and 3.
ANALYSIS
A glance at the three graphs tells the story.
The gold standard for performance is the
CF with the highest gain and PSRR and
the lowest Z0. The SRPP top is the worst,
quite a bit poorer than the CF. The MF
top occupies a range between the two.
Why is the SRPP so poor? Two words:
positive feedback (from the cathode to the
grid). The grid is so close to the cathode
electrically that very little AC voltage develops between the two terminals to drive
DERIVATIONS OF
GAIN, Z0, AND PSRR
Refer to Fig. 1 . Capacitors are assumed to be AC shorts and 1M resistors are AC opens. RT is defined as the
sum of R1 + R2 +. . . R19. Rk is the
value of that portion of RT between
Tap_0 and the connection to Tap_?
for SRPPs and MFs, or simply RT for
CFs. rp and gm are the plate resistance
and transconductance, respectively, of
the 6DJ8. All derivations start from
the knowledge that the sum of all currents flowing out of a node is zero
(else there would be net accumulation
of charge).
Determine the equation for output
impedance Z0. Here, Ep is shorted to
+200V, and V1, V2, and V3 are set to
0. IR22 is the current through R22. The
sum of the currents out of node Ek are:
–IR22 + Ek/rp + Ek/RT – gm×Ek×((RT – Rk)/RT – 1)
the transconductance, so Z0 suffers. The
desired source for the grid, V3, is not that
much closer electrically than the plate,
so the PSRR suffers. And the triode acts
more like a simple resistive load than a
controlled source, so the gain suffers.
The MF is an attempt to win back
some of the CF’s stellar AC performance
by reducing the amount of positive feedback from the cathode to the grid. It does
this by maximizing the ratio of Rk to RT.
The ratio can’t get all that close to 1 because of the “plate resistance” (the part of
RT which is not Rk) of the “triode” (whose
other part is V3). Practical factors limiting
the ratio include 1) the voltage drop across
Rk, which must be made up by the plate
supply if the MF bias is to be maintained,
and 2) heater-cathode voltage ratings if
the entire MF is made from one dual
triode. Fortunately, even modest voltage
drops across Rk can make a big difference.
Consider the case where another 6DJ8
and R1 form the bottom part of an MF by
replacing V3 and the portion of RT which
is not Rk. The resistance seen at the plate
would be R1 × (1 + gm × rp) + rp = 18K
(see Table 2 for values of gm and rp). This
would be similar to an MF where Tap_?
was connected to Tap_3, increasing Rk/RT
to about .13 from the .02 of the SRPP.
The drop across R2 + R3 adds only 11V,
Ek×(1/rp + 1/RT + gm×Rk/RT) = IR22
Ek / IR22 = 1/(1/rp + 1/RT + gm×Rk/RT)
but the MF’s Z0 improves over that of the
SRPP by a factor of about 3, the PSRR
improves by a factor of about 2.5, and the
Gain almost doubles (see Tables 1 or 3
or Figs. 2–4). It’s easy to achieve even
greater improvement by further modest
increases in the value of Rk. You can see
this yourself by substituting larger values
for Rk into the derived equations.
But why not use the CF in place of
the MF top? Because the MF provides a
much higher impedance load to the plate
of the “bottom” triode than would a plate
resistor that would need to be used in
its place along with a CF. And common
cathode triode gain stages love high impedance plate loads for maximizing gain
and PSRR and minimizing distortion.
Also, the excellent PSRR of the CF is
rendered useless by the poor PSRR of
the resistively plate-loaded stage driving
it. To make full use of the CF PSRR, you
need to drive it from a low PSRR circuit
such as cascade feedback pair with a good
amount of open loop gain.
CONCLUSION
I believe this kind of result should make
you think twice about using an SRPP.
But if you still decide to use one, now you
can calculate exactly what you’re losing by
doing so!
aX
the MF and SRPP. Here V1, V2, and
V4 are set to 0, and R22 is open (not
used). Ep is shorted to +200V. The sum
of the currents out of node Ek are:
Z0 = Voltage across the output/Current into output = Ek/IR22 = 1/(1/rp + (1 +gm×Rk)/RT))
Ek/rp + (Ek – V3)/RT – gm×((Ek×(RT - Rk)/RT +
V3×Rk/RT) - Ek)
Z0 = 1/(1/rp + (1 +gm×Rk)/RT))
Ek×(1/rp + 1/RT + gm×Rk/Rp) = V3×(1 + gm×
Rk)/RT
Determine the equation for the
PSRR. Here V2, V3, and V4 are set
to 0V and R22 is open (not used). The
sum of the currents out of node Ek are:
(Ek – Ep)/rp + Ek/RT – gm×Ek×((RT-Rk)/RT – 1)
Ek×(1/rp + 1/RT + gm×Rk/RT) – Ep/rp = 0;
GainM,S = Ek/V3 = ((1 + gm× Rk)/RT) × Z0
Determine the equation for gain for
the CF. Tap_? is connected to Tap_V3,
V1, V3, and V4 are set to 0, and R22 is
not used. Ep is shorted to +200V. The
sum of the currents out of node Ek are:
Ek/Z0 – Ep/rp = 0
Ek/rp + Ek/RT – gm×(V2 – Ek)
PSRR = Ep/Ek = rp/Z0
Ek/rp + Ek/RT + gm×Ek = gm×V2
Determine the equation for gain for
GainCF = Ek/V2 = gm/Z0 = gm/(gm + 1/rp + 1/RT)
audioXpress January 2007
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solid state
Grounding and System Interfacing
It's time to clear up some misconceptions regarding grounding and audio gear.
By Gary Galo
T
he annual conventions of the
Audio Engineering Society
are packed with technical papers, tutorials, seminars, and
exhibits. Press coverage of the convention
has become increasingly difficult due
to the sheer size of the event. This year,
rather than attempting an overview of
the convention’s proceedings, I decided
to highlight one session that stood out
as being particularly outstanding. On
October 7, 2005, at the Jacob Javits Convention Center in New York City, Bill
Whitlock, President of Jensen Transformers, Inc., gave a three-hour tutorial
seminar titled Audio System Grounding
and Shielding: An Overview. (Whitlock
became President of Jensen following
the untimely death of company founder
Deane Jensen in 1989.)
Whitlock sought to dispel many of
the myths surrounding grounding and
system interfacing, noting that the subject
abounds in black art and myths. Basic
rules of physics are routinely ignored, and
even many manufacturers “don’t know
ground loops from Froot Loops” (the
last comment was typical of the touches
of humor that Whitlock brought to his
presentation, though there was serious
intent behind each of them). If a system
contains two or more pieces of grounded
equipment, a ground loop may be formed
if the chassis of the connected equipment are at different potentials. This will
typically happen if the various pieces of
equipment are powered by different AC
branch circuits.
The most common approach to solving hum problems due to ground loops
is by lifting safety grounds with “ground
lifters” sold in any hardware store. Whitlock warned against this approach, noting that safety grounding keeps AC line
voltages between equipment safe even
if equipment fails. His view is that you
26
audioXpress 1/07
Galo-2713-2.indd 26
should never use ground lifters even if a
manufacturer’s instructions tell you that
it’s OK to do so.
Whitlock noted that the ground adapters sold in hardware stores, while superficially appearing as ground lifters, are actually intended to provide a safety ground
when grounded (3-pin) power cords are
used with 2-prong receptacles. They are
designed to add a ground, not remove one,
by putting the outlet plate screw through
the ground tab on the adapter! If ungrounded equipment suffers an internal
failure of insulation or components at
the AC line input, the equipment chassis
can turn live with 120V AC if the safety
ground is not connected (Fig. 1; I thank
Bill Whitlock for granting permission to
use slides from his Power Point presentation for these illustrations).
Whitlock also recommends GFCI
(Ground Fault Circuit Interrupter) outlets
for safety. These outlets sense differences
between line current and neutral current
(fault currents). Any difference between
line and neutral current may be current
flowing through a human, which can be a
deadly problem. GFCI outlets trip at 4 to
7mA of current, protecting the individual
in contact with the “live” equipment.
He also dispelled some of the myths
FIGURE 1: Internal
equipment failure
can render a chassis
live with 120V AC. The
safety ground ensures that the chassis
surrounding earth grounds, noting that
earth grounds are for lightning protection.
The safety ground in a modern electrical
system is tied to neutral at the entrance
panel, and earth ground plays no role
in protecting people from electrocution.
Earth grounds are invariably not at 0V,
and two earth grounds will rarely be at
the same potential. Earth ground rods are
useless for protection against fault currents, and are equally useless for reducing
noise.
HUM CAUSES
Myths abound regarding the causes of
hum in audio systems, and hum is rarely caused by bad audio cable shielding
or AC distribution. You can trace the
vast majority of hum problems to faulty
system interfacing. Pro sound engineers
routinely engage in dangerous practices in the field—adding ground lifters to
AC power cords—in order to quickly
eliminate hum problems. Defeating safety grounds is dangerous and illegal, and
makes the person who did it legally liable.
Whitlock emphasized that such practices
have resulted in people being killed and
companies forced out of business because
of lawsuits.
The only safe way to solve hum prob-
remains at 0V potential, tripping
the breaker in the panel for that
branch circuit, and protecting the
user from potentially lethal voltages. Earth ground is for lightning
protection, and plays no role in
fault protection. (Illustration courtesy of Bill Whitlock)
www.audioXpress .com
11/21/2006 3:14:02 PM
lems is through proper system interfacing.
Whitlock provided a great deal of useful
background on the causes of hum in unbalanced interfaces, and gave an excellent
overview of the requirements for balanced
interfaces.
Balanced lines are not balanced because
they carry audio signals of equal level but
opposite polarity (though they normally
do). They are balanced because of careful
impedance matching of the two signal
carriers with respect to ground, especially
the source (driving) impedances. If the
impedances of balanced lines are not carefully matched, by definition they are not
truly balanced. Common-mode rejection
will be compromised, which reduces the
system’s ability to reject noise coupled to
the signal carriers.
In unbalanced interfaces, leakage currents flowing in signal cables are the
root cause of most hum problems (Fig.
2). Leakage currents are normally carried by the grounded conductor—the
shield—which does not have a zero ohm
impedance. Ohm’s law tells us that the
resistance of the shield generates noise
voltage over the length of the cable. This
FIGURE 2: When two grounded chassis are at different AC potentials, circulating interference
current will be carried by the interconnecting shield. Noise voltage will be generated over the
length of the shield due to the shield’s resistance. This noise voltage is added to the signal arriving at the receiver due to common-impedance coupling. (Illustration courtesy of Bill Whitlock)
FIGURE 3: Ground isolators eliminate the electrical connection
between the driver and the receiver. No noise current flows in
the cable shielding. With properly designed Faraday-shielded
input transformers, noise coupling is effectively eliminated. (Illustration courtesy of Bill Whitlock)
audioXpress January 2007
Galo-2713-2.indd 27
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11/21/2006 3:14:04 PM
noise, in turn, is added to the signal at
the receiver.
The correct way to eliminate hum
problems in audio interfacing is with
transformer isolation ( Fig. 3 ). Transformers transfer the signal voltage from
the primary to the secondary windings
without any electrical connection between them. So, there are no noise currents flowing between the connected
devices. With a properly designed transformer, hum is eliminated while safety
grounds remain intact.
Reducing ground noise in a transformer-coupled interface depends on the type
of transformer used. Output transformers typically have closely spaced primary
and secondary windings. This increases
the capacitance between the windings
allowing efficient high-frequency noise
coupling between the windings, which
is undesirable. Input transformers typically employ a Faraday shield between
the windings, which virtually eliminates
capacitive coupling. This, in turn, greatly
improves noise rejection of a transformer-coupled interface, including RF and
ultrasonic interference.
Many transformer-based “black boxes”
sold to eliminate noise caused by ground
loop problems employ output transformers. Their low output Z allows these devices to be placed anywhere in the signal
path, because their outputs are not sensitive to the capacitive loading caused
by long cable runs. Input transformers,
on the other hand, offer a 30dB improvement in noise rejection over output
types. But, their relatively high output
Z requires that they be placed within 2
or 3´ of the receiver in order to prevent
degradation of their high-frequency response by the output cabling.
Whitlock noted that normally an unbalanced interface using a single isolation
transformer close to the load will solve
the noise problem. Only in unusual circumstances is a balanced interface necessary. Converting the entire interface to
balanced is necessarily more complicated
because it requires an unbalanced to balanced conversion at the source, and a
balanced to unbalanced conversion at the
load. In other words, two transformers
rather than one.
One pet peeve of mine is the notion
that professional audio systems should
be designed to operate at matched impedances of 600Ω. Naturally, I was delighted when Whitlock dispelled this
nonsense. He correctly noted that 600Ω
impedances are holdovers from early
telephone practice, but are not applicable
to modern audio systems that are driven
by signal voltage.
Modern audio circuitry is designed
with low output impedances and high
input impedances. This way, audio circuits are not subject to unnecessarily low
Z loads. He also correctly noted that signal level, impedance, and line balance are
three independent parameters. Impedance
has nothing to do with level, and only its
matching in the two signal conductors
has anything to do with whether the line
is balanced or unbalanced.
FIGURE 4: Conventional instrumentation-style active differential amplifier. The bias return resistors R1 and R2 lower the common-mode
input impedance when compared to transformers. Raising the values
of R1 and R2 would compromise the noise performance. (Illustration
courtesy of Bill Whitlock)
28
audioXpress 1/07
Galo-2713-2.indd 28
INGENIUS ICS
The near-infinite common-mode input
impedance of a transformer-coupled line
receiver gives it a huge noise-rejection
advantage in real-world systems, but
transformers have fallen out of favor in
some pro audio circles. Transformer-less,
actively-balanced circuitry has become
increasingly popular over the past 20
years. Figure 4 shows a typical op-amp
based, instrumentation-type of active
differential input. This type of circuit has
a disadvantage over transformers because
the bias return resistors for A1 and A2
also lower the common-mode input impedance. You could raise the values of R1
and R2, but this would compromise the
noise performance.
Bill Whitlock has designed an elegant
solution to this problem (Fig. 5). The
common-mode voltage is extracted at the
junction of R3 and R4. A4 buffers the
common-mode voltage and bootstraps
R1 and R2 via capacitor C. This patented technique is being used in the InGenius® line integrated circuit differential
line receivers manufactured by THAT
Corporation (Photo 1). Common-mode
input impedance for this circuit is 10MΩ
at 60Hz and 3.2MΩ at 20kHz, without
the noise penalty that raising the resistor
values would produce.
Whitlock noted a number of features
of the InGenius chips:
• Thin-film silicon-chromium resistors are
used, which yield better stability over
time and temperature compared to nickel-chromium or tantalum-nitride types.
FIGURE 5: THAT Corporation’s InGenius line of differential line receiver ICs uses a bootstrapping technique patented by Bill Whitlock.
The circuit raises the common-mode input impedance into the megohm region without compromising noise performance. InGenius is a
registered trademark of THAT Corporation. (Illustration courtesy of
Bill Whitlock)
www.audioXpress .com
11/21/2006 3:14:05 PM
• 90dB common-mode rejection ratio and
gain accuracy are achieved by careful resistor matching, typically 0.005%. Laser
trimming is used in the manufacturing,
which keeps cost reasonable.
• Chips are manufactured using a 40V
complementary bipolar Dielectric Isolation (DI) process, which allows NPN
and PNP transistors with performance
as good as discrete circuits. Isolation between transistors is high, with no substrate connection. Stray capacitances are
low, yielding high bandwidth and slew
rates.
• Front ends use a folded-cascode, PNP
design, yielding superior noise performance, high gain with simple stability
compensation, and greater input voltage
range.
Whitlock also highlighted key performance specifications for the InGenius
chips:
• High CMRR maintained with real-world
sources:
• 90dB at 60Hz, 85dB at 20kHz with zero
imbalance source
• 90dB at 60Hz, 85dB at 20kHz with IEC
±10Ω imbalances
• 70dB at 60Hz, 65dB at 20kHz with
600Ω unbalanced source!
• THD 0.0005% typical at 1kHz and
+10dBu input
• Slew rate 12V/µs typical with 2kΩ +
300pF load
• Small signal bandwidth 27MHz typical
• Gain error ±0.05dB maximum
• Maximum output +21.5dBu typical with
±15V rails
• Output short-circuit current ±25mA
typical
• 0dB, -3dB, -6dB
gain versions =
THAT 1200, 1203,
1206
PHOTO 1: THAT Corporation’s InGenius line of integrated circuit differential line receivers. The 1200, 1203, and 1206 have gains of 0dB,
-3dB, and -6dB, respectively.
He also summarized the performance advantages:
• Conventional ac-
•
•
•
•
•
•
tive receivers are far cheaper, smaller,
and lighter than a quality transformer,
but
Transformers consistently outperform
them for reasons that need to be widely
understood and appreciated.
The main transformer advantage stems
from its inherently very high commonmode impedances.
The InGenius IC exhibits the very high
CM impedances previously associated
only with transformers.
Excellent noise rejection even with unbalanced sources!
Its bootstrap feature lends itself to novel
and very effective RF interference suppression.
Its high-quality internal op amps give it
great sound.
For further information on the InGenius chips, visit THAT Corporation’s
website at www.thatcorp.com.
Whitlock highlighted his seminar
with a lengthy Power Point presentation of over 150 slides, which he offered to e-mail to any interested parties.
He will also send his 40-page seminar
handout Understanding, Finding and
Eliminating Ground Loops in Audio and
Video Systems on request. The preceding
highlights only scratch the surface of
audioXpress January 2007
Galo-2713-2.indd 29
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11/21/2006 3:14:08 PM
his presentation, so these documents are
required reading for anyone interested
in this subject. Both are in PDF format,
and encompass virtually everything discussed at the AES seminar. E-mail him
at whitlock@jensen-transformers.com, and
kindly mention this article.
ISO-MAX PRODUCTS
Jensen manufactures a multitude of transformers for every possible application,
including microphone preamplification,
moving coil phono cartridges, as well as
balanced and unbalanced line-level usage
(www.jensen-transformers.com). They also
make a variety of transformer-based devices for audio interfacing. Perhaps the
most useful for consumer applications is
the CI-2RR Dual Ground Audio Isolator
(Photo 2). You can download the .PDF
datasheet for the CI-2RR at www.jensentransformers.com/datashts/ci2rr.pdf.
This stereo device consists of a pair of
their JT-11P-1HPC Line Input Transformers. You can download the .PDF
datasheet for this transformer at www.
jensen-transformers.com/datashts/11p1hpc.pdf.
Install the CI-2RR close to the load.
In an audio/video application, a stereo
RCA cable would be run from the video
system’s main audio output to the CI2RR. The output of the CI-2RR is then
fed to the preamp, integrated amplifier, or
AV amplifier line input. Jensen notes that
to avoid excessive high-frequency losses,
the output cable lengths should be no
more than 3´.
There may be cases where it is desirable to connect an audio/video system for
both record and playback (Fig. 6). One
possible application might be to use the
audio recording capability of a Digital
Video Recorder, DVD recorder, or VHS
Hi-Fi recorder to capture an FM broadcast for delayed listening. For this application, use a pair of CI-2RR isolators,
one in the record signal path, and one for
playback. To keep output cabling as short
as possible, place the CI-2RR used for
the record signal path close to the video
system’s audio record inputs. Place the
playback isolator close to the audio pre-
PHOTO 2: Jensen’s Iso-Max CI-2RR transformer-coupled stereo audio ground isolator is ideal for
interfacing audio and video systems. The transformers are among the world’s finest, with very
transparent sonics.
FIGURE 6: A typical audio/video application for recording and playback. A pair of Jensen
CI-2RR stereo isolators are used in both the recording and playback signal path, providing
ground isolation in both directions.
30
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11/21/2006 3:14:10 PM
amp, amplifier, or A/V receiver.
In his review of the ISO-Max (Oct.
’06 aX), Charles Hansen notes the highfrequency phase shift of the CI-2RR—
about 35° at 20kHz—appears to be considerably higher than specified by Jensen.
I made this measurement myself and got
the same results. The graph in Jensen’s
datasheets for the CI-2RR and the JT11P-1HPC shows essentially 0° deviation
from linear phase at 20kHz.
It seems strange to me that Jensen
would use the same graph for the transformer and the CI-2RR, because the CI2RR has a series R/C damping network
at its output, consisting of 13k in series
with 680pF. The R/C network should
introduce phase shift of its own, beyond
that of the transformer. I disconnected
the R/C network in one channel of one of
the CI-2RR devices supplied by Jensen.
This reduced the phase shift to around
10° at 20kHz.
The datasheet for the JT-11P-1HPC
shows a schematic of a typical application
for this transformer, with a series R/C
network of 13k and 620pF (the capacitor is a slightly different value than the
680pF used in the CI-2RR). You would
assume that the phase graph on this datasheet would be for the transformer as a
standalone device, and not with the series
R/C network, but Jensen doesn’t really
specify. None of the test circuits on page
2 of the datasheet show the series R/C
network.
However, conventional phase measurements may not tell the whole story. Whitlock offered the following comments on
the phase measurement:
“Regarding phase measurements
on the Jensen samples, let me emphasize that ‘phase shift’ is not necessarily
phase distortion! A totally benign time
delay is represented in the phase domain as a phase shift that increases linearly with frequency. . . this is not distortion. Phase distortion occurs when
there is a nonlinear relationship between frequency and phase—in other
words, a time delay that changes with
frequency. The widespread confusion
about this subject is why Deane Jensen
published a 1986 AES paper, HighFrequency Phase Specifications—Useful
or Misleading, and why Jensen always
specifies Deviation from Linear Phase,
or DLP, for our parts—not phase shift!
DLP is the true measure of phase distortion. The secondary R-C networks
specified for Jensen input transformers are chosen to provide two-pole
Bessel high-frequency rolloff, which
guarantees minimal DLP as evidenced
by zero-overshoot and zero-ringing
in square-wave response. This timedomain behavior is the hallmark of
Jensen designs and a very important
contributor to the sonic transparency
of our products.”
You can purchase the AES paper mentioned by Whitlock as a PDF download
f rom the AES website www.aes.org/
publications/preprints/search.cfm. Enter 2398
in the box labeled “Preprint/Paper Number.” Cost is $5 for AES members and
$20 for non-members. This paper is well
worth reading.
Jensen uses standard PC-mount RCA
jacks for the CI-2RR. The parts in the
output R/C network appear to be Panasonic P-series polypropylene capacitors
and 1% metal film resistors that look like
audioXpress January 2007
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audioXpress 1/07
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SUBJECTIVE TESTS
I evaluated the CI-2RR by inserting it
in the signal path between my outboard
D/A converter and my preamplifier. I use
D.H. Labs’ Air Matrix cables fitted with
their Ultimate RCA Connector for all
of my system interconnects, so I used an
additional 1/2-meter pair of these cables
to connect the CI-2RR to my preamp
inputs. My procedure was to listen to
a reference CD recording without the
CI-2RR, then insert it in the signal path
and listen to the same selection again.
I compensated for the insertion loss of
the CI-2RR when the device was in the
signal path.
Sonic evaluation of the CI-2RR proved
difficult, because the device is incredibly
transparent. Sonic anomalies I normally
associate with transformers simply didn’t
appear when I auditioned the CI-2RR.
The important virtues of my audio system
were left unscathed, including soundstage
size and localization, ability to resolve
subtle inner details, low-level resolution
including hall ambience, and the overall
clarity and purity of the sonic presentation.
There were times when I thought that
the CI-2RR softened the treble slightly,
but I could not verify this consistently.
I did find that the CI-2RR seemed to
make difficult CDs more listenable by
softening the edge on these discs. The
CI-2RR also maintained the weight and
impact in the bass region. The only time
I thought that the bass may have been degraded was on discs that have substantial
energy below 20Hz, and this was more a
matter of “feel” than audibility. But, the
effect was very subtle.
Overall, the CI-2RR acquitted itself
impressively in my listening tests, and it
is no wonder that so many manufacturers
of high-end consumer and professional
audio equipment choose Jensen transformers for their designs. As examples,
The Jeff Rowland Design group uses Jensen transformers for moving coil inputs,
as well as for balanced line inputs and
output on their multi-thousand dollar audiophile preamplifiers. The John Hardie
Company, maker of some of the world’s
finest microphone preamplifiers, uses Jensen transformers for both microphone
inputs and balanced line outputs.
The CI-2RR retails for $177.95, but
transformers of this quality are expensive
to manufacture. Like all Jensen products,
the CI-2RR is made in the USA, and is
heartily recommended in any application
requiring isolation of unbalanced lines.
Jensen manufactures a sizable assortment
of Iso-Max products for a wide variety
of balanced and unbalanced applications,
and you can purchase all of their transformers separately. Check their website
for details. Iso-Max products are also
available from Old Colony Sound Lab
(www.audioxpress.com).
VIDEO ISOLATION
Many audio/video installers have encountered hum when interfacing video and
audio systems. The hum usually appears
when the cable TV coax is connected to
the video system. Remove the CATV line
and the hum disappears.
I first ran into this problem around 20
years ago, before CATV isolators were
readily available. My solution was to connect the 300Ω ends of a pair of 75Ω to
300Ω balun transformers back-to-back.
Garden-variety balun transformers typically have an insertion loss of around 3dB,
so the total loss of this solution was around
6dB. But, it did cure the hum problem.
Anyone opting for this “cob job” approach should beware of certain devices
being passed off as balun transformers. I
have seen cheap ones that were actually
nothing more than feed-through adapters—no transformer, no impedance matching, and no ground isolation. Some devices
labeled “matching transformers” use an
autotransformer. They are not baluns, and
do not provide ground isolation. When in
doubt, use an ohmmeter to check for continuity between the 300Ω and 75Ω sides,
both signal and ground—resistance should
be infinite.
A more elegant solution to CATV
ground isolation is Xantech’s model 63400
Ground Breaker. This 75Ω to 75Ω isolation transformer costs less than $10 from
Hometech.com: www.hometech.com/video/
attn.html or Smarthome.com: www.smarthome.com/81285.html. Parts Express sells a
similar device, the Dayton VIT-1, part no.
180-075, for also less than $10. Subjectively, I find the picture a bit sharper with
the Xantech Ground Breaker.
The Jensen VRD-1FF CATV isolator is unusual in that it is the only Jensen
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11/21/2006 3:14:12 PM
product that does not use a transformer
(Photo 3). The VRD-1FF is a capacitive
isolator. According to my measurements,
the device consists simply of two series capacitors, one in the signal path and one in
the ground path. Both capacitors measure
2nF. At the power line frequency of 60Hz
the device has extremely high isolation
impedance, 1.3MΩ, but the impedance is
very low in the RF region.
Jensen claims a bandwidth of 1MHz
to 1.3GHz ±3dB. The VSWR (Voltage
Standing Wave Ratio) is specified as 1.08:1
from 50MHz to 866MHz. One advantage
of the capacitive isolator is extremely low
insertion loss in the operating range. Jensen specifies the 50MHz insertion loss at
0.01dB typical and 0.1dB maximum. The
device is compatible with analog, digital,
and HD systems, as well as modems. You
can download the VRD-1FF datasheet at
www.jensen-transformers.com/datashts/vrd1ff.
pdf.
According to the datasheet, the capacitors in the VRD-1FF are rated at 200V
DC. However, Jensen notes that the VRD1FF should not be used in situations where
the DC or peak AC voltage between input
and output exceeds 34V. Check the voltage ter conductor.
potential between your CATV line and the
Subjectively, the VRD-1FF yields the
RF input to your video system. If it exceeds clearest, sharpest picture of all of the iso34V DC or peak AC, you have a grounding lation devices I have tried. The capaciproblem that must be corrected before using tive isolation approach is superior to the
cheaper transformer alternative, and is well
this or any other isolation device!
Retailing at $60, the VRD-1FF is not worth the extra expense. I recommend
cheap, but it boasts the high-quality con- using the VRD-1FF along with one or
struction and performance I have come more CI-2RR isolators to provide comto expect of Jensen products. Because the plete effective ground isolation between
VRD-1FF has female F-connectors on video and audio systems. Note that you
each end, Jensen also supplies the short can’t use the VRD-1FF between a DSS
jumper cable shown in Photo 3. No cor- dish and receiver, because the capacitors
ners have been cut here either—the jumper can’t pass the DC power that the receiver
uses Canare LV61S cable and Canare FP- supplies to the dish. aX
C4 F-connectors.
The LV61S is a
flexible, high-quality RG-59 replacement, and the Canare F-connectors
are the finest made.
These connectors
incorporate a goldPHOTO 3: The Iso-Max VRD-1FF is the only product made by Jensen that
plated, crimped
doesn’t use a transformer. This CATV isolator is capacitive, resulting in
center pin, rather
extremely low losses over a wide bandwidth. Picture quality is visibly
superior to transformer-coupled CATV isolators. The supplied pigtail
than relying on the
cable uses high-quality Canare F-connectors and Canare LV61S cable.
cable’s flimsy cen-
audioXpress January 2007
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solid state
A Flexible Subwoofer Amplifier, Pt.1
This four-part series explains how to build a full-featured subwoofer
amp that will enhance your audiophile experience. Part 1 focuses on
By Rudy Godmaire
theoretical aspects.
T
his journey started four years
ago when I needed a new
amplifier to power my passive
subwoofer. At first, I intended to integrate an amplifier module to
the three-channel home theater amplifier I was building at that time. Though
the idea of a 3.1 channel amplifier was
seductive, my chassis was lacking space
to accommodate this fourth channel.
Building a separate unit then became
the only option.
At the same time, I was reading about
op amps and active filters. I was eager to
learn about their various uses in audio
applications. As I was learning quickly,
I soon realized that my amplifier could
greatly benefit from a custom low-pass
active filter and an all-pass active filter.
This led me to design the full-featured
subwoofer amplifier I describe here.
Most of the theoretical background
behind my design comes from the study
of three books. The first one is called
The Analysis and Design of Linear Circuits1 and provided me with all the fundamentals I needed to know about op
amps and simple active filters. Though
seemingly academic, most of the book
is quite easy to understand due to the
pedagogical approach.
The second one, National Semiconductor Audio/Radio Handbook2, contains a
wealth of interesting information. Most
notably, chapter 5 titled “Floobydust” is
of special interest for the DIYer. This is
where I picked up the schematic of my
third-order Butterworth low-pass active
filter that I will describe later. The title
of the third book speaks for itself: Op
34
audioXpress 1/07
Godmaire2689-4.indd 34
PHOTO 1: The subwoofer amplifier.
Amp Applications3. I believe that this reference book, edited by Walter G. Jung,
is a must have for any DIYer using op
amps in his projects.
DESIGN OBJECTIVES
Adjustments: Here is a key word you
must consider if you want to achieve
harmonious integration between the
subwoofer and the main loudspeakers. Among the parameters to consider,
cutoff frequency, phase alignment, and
sound level are certainly the most important. These features are all part of the
design illustrated in the block diagram
(Fig. 1).
I explain each of these blocks from
a theoretical perspective in Part 1. In
the following parts, I will discuss the
practical aspects of the project, namely,
the making of the PCBs (Part 2) and
the construction of the subwoofer amplifier (Part 3). The fourth and last part
will address the performance of the subwoofer amplifier, including some measurements and listening results. As a
“bonus feature,” I will also explain how
to build two preamplifiers derived from
the same circuitry.
NON-INVERTING BUFFERS &
INVERTING SUMMER
This stage of the circuitry deals with
the input signals. As shown in Fig. 2,
you can connect three sources to the
amplifier. In my case, I wanted to use
my subwoofer amplifier for both a home
theater system and a stereo system. Be-
FIGURE 1: Block
diagram of the
subwoofer amplifier.
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11/21/2006 3:12:58 PM
cause I neither intended nor had the
room to install two subwoofers for stereo
operation, I needed to be able to convert
the stereo signal to a monophonic one.
Hence the two non-inverting buffers
and the inverting summer. Should you
need only a single input for whatever
reason, then this part of the circuitry
could be greatly simplified by keeping
only one buffer and skipping the rest of
the components.
It is well known that switching both
signal and ground helps preserve the
integrity of the signal coming from the
source. When all the grounds of the
RCA connectors are tied together, the
common ground of the selected source
and of the amplifier is now shared by all
other sources connected to the amplifier.
The network therefore produced may
introduce distortion. This issue seems to
be especially sensitive for preamplifiers
in which you generally connect multiple
sources.
For this reason, I chose a four-pole
three-position rotary switch (S1). This
enables me to switch the signals and
grounds of all my sources. The first position receives the stereo signals coming
from my preamplifier. The next position
is dedicated to my DVD player. Note
that in this case, only two poles are connected. I left the last position free for
future use.
Following the source selector S1 are
the input capacitors C1/C2, which contribute to the isolation of the amplifier by blocking unwanted DC voltages
that might come from the source. These
capacitors form a first-order high-pass
filter with the input resistors R1/R2. The
cutoff frequency of this filter is determined by:
fcHP = 1/2πRC
It is possible that your application requires more gain. In this case, you could
replace the non-inverting buffers with
non-inverting AC amplifiers as shown
in Fig. 3. In this diagram, C0 should remain at 0.47µF and R2 at 100k in order
to maintain the 4Hz high-pass filter.
Select the value of R1 to obtain the ap-
propriate gain. Finally, adjust C1 so that
R2C0 = R1C1.
The signals emerging from the noninverting buffers converge toward the
inverting summer. The gain of the inverting summer is given by the following
equation:
FIGURE 2: Input buffer
and inverting summer
stages.
(1)
This filter provides a low cutoff frequency of 4Hz so that maximum gain is
obtained at frequencies around 20Hz.
The two input buffers, whose main
function is to isolate the amplifier
from the source, provide unity gain as
it should be by definition. IC1 as well
as IC2 and IC3 are OPA-2604 dual op
amps4. I selected these devices throughout my project because they benefit from
an excellent reputation in audio applications while being reasonably priced.
audioXpress January 2007
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11/21/2006 3:13:01 PM
FIGURE 3: Non-inverting AC amplifier.
FIGURE 4: General fourth-order Butterworth low-pass active filter.
K = K1 + K2
where K1 = - R7/R5 and K2 = - R7/R6
(2)
Note that the gain of the inverting summer will vary depending on the source
selected. When the stereo inputs are
selected the gain is –2, while it is –1 for
the DVD player.
THIRD-ORDER BUTTERWORTH
LOW-PASS ACTIVE FILTER
Designing this stage has been the most
challenging activity of the project. My
first thought was to build a fourth-order
Butterworth low-pass active filter with a
cascade of two op amps as in Fig. 4. The
topology had the advantage of a very
steep attenuation slope of –24dB per
octave. Replacing the four resistors R
with a four-ganged potentiometer would
enable the adjustment of the cutoff frequency. However, I was not enthusiastic
about the idea of using multiple cermet
potentiometers in the signal path because I thought that it would degrade
the signal too much.
In fact, I had another idea in mind. I
wanted to produce a variable low-pass
active filter that would use precision resistors instead of cermet potentiometers.
A switching system would enable the
simultaneous connection of the resistors
and provide a few positions to change
the cutoff frequency. Easier said than
done; this idea quickly became very
complex to implement on a PCB due to
the cascaded op amps.
National Semiconductor Audio/Radio
Handbook provided me with an elegant
solution. The “Floobydust” chapter contains an example of a third-order Butterworth low-pass active filter that is
based on a single op-amp design. Figure
5 shows the general presentation of this
filter with its equations. This was the
36
audioXpress 1/07
Godmaire2689-4.indd 36
key element I needed to design an audiophile-grade switching network.
A few sketches and days later, I found
a way to implement a four-pole threeposition rotary switch that would enable
the selection of three different cutoff
frequencies. The final circuit is depicted
in Fig. 6.
I have selected three cutoff frequencies based on my particular needs:
• 48Hz, a custom value that was
selected empirically;
• 80Hz, the THX standard;
• 100Hz, the
Dolby ProLogic standard.
The first frequency
is specially tuned
for stereo music in
conjunction with
my own system.
I determined this
cutoff f requency
following a process of trial and error,
which you would think would be painful.
But the experimentation has been made
easy thanks to the use of exchangeable
“chips” of resistors, which I will describe
in Part 2.
The last two frequencies are dedicated to home theater and should fit
your needs, too. But you may also want
to consider a cutoff frequency of 120Hz,
the upper limit of the Dolby Digital and
DTS standards. These formats offer a
low-frequency effect channel (LFE) that
FIGURE 5: General third-order Butterworth
low-pass active filter.
FIGURE 6: Final design of the third-order Butterworth low-pass active filter.
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11/21/2006 3:13:05 PM
reproduces low bass sounds from 3 to
120Hz.
The values of the components of Fig.
6 were established with the formulas
appearing within Fig. 5. For your convenience, Table 1 provides pre-calculated
values for different cutoff frequencies. I
recommend that you stick with the values suggested for C A , CB, and CC (C3, 4 ,
C5, and C6 in Fig. 6) because they meet
very closely the theoretical values calculated with these formulas. However, you
can calculate other custom values of R
and 2R using the following formula:
R = 2.4553/2πfcLPC1 = 710 497/fcLP
where fcLP is the cutoff frequency (3)
VARIABLE 180° ALL-PASS FILTER
Phase alignment is an important aspect
of a successful implementation of a subwoofer. You can do this mechanically or
electronically. In my opinion the best
placement for a single subwoofer is right
in the middle of the main loudspeakers,
which enables physical alignment of the
cones.
However, in most home installations
this choice is not an option. The subwoofer will lie generally somewhere else
in the room, which makes mechanical
phase alignment impossible to achieve.
This is where the variable all-pass filter
finds its use.
Figure 7 shows the general presentation of a first-order all-pass active filter.
Theoretically, this circuit enables a phase
shifting that goes from –180° at DC
to 0° at high frequencies, all without
changing the amplitude of the signal. In
other words, an all-pass filter provides
unity gain at all frequencies. The cutoff frequency fcAP is determined by the
high-pass function of C1R1 and is calculated using equation 1. At fcAP, the phase
shift of this first-order all-pass filter will
be –90°.
As mentioned in a technical brief
available on the website of Maxim/Dallas Semiconductors5, the phase shift realized by this circuit at any given frequency can be found by:
(4)
where ω is the frequency in rad/s, or 2πf,
when f is in Hertz.
You can draw some conclusions from
this equation. Changing the value of R,
C, or ω (2πf ) will have an impact on
the phase shift. You can definitely take
advantage of this. Replacing R1 of Fig.
7 with a potentiometer will provide you
with a very simple solution to control
and adjust the phase shift.
Another conclusion is that when the
frequency f of the musical signal changes, the phase shift changes as well. This
tells you that perfect alignment will be
reached only for a specific frequency. I
analyze the effect of this in Part 4.
For now, it is important to know that
FIGURE 7: General first-order all-pass
active filter.
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TABLE 1 Pre-calculated values of R and 2R as functions of fcLP
Theoretical values
Fixed real values
S2
fcLP
(Hz)
R
(Ω)
2R
(Ω)
CA
(µF)
CB
(µF)
CC
(µF)
25
25k3
56k6
0.55
0.47
0.044
Switch
Position
Ratings
120W into 4Ω
30
23k6
47k2
0.55
0.47
0.044
35
20k2
40k4
0.55
0.47
0.044
40
17k7
35k4
0.55
0.47
0.044
-3dB frequency range
5Hz to 40kHz
45
15k7
31k4
0.55
0.47
0.044
THD at 2W into 4Ω
0.08%
THD at 90% Pmax. into 4Ω
0.09%
Intermodulation distortion at 90% Pmax.
0.09%
Signal to noise ratio
107 dBA
48
14k5
29k0
0.55
0.47
0.044
50
14k1
28k2
0.55
0.47
0.044
55
12k9
25k8
0.55
0.47
0.044
60
11k8
23k6
0.55
0.47
0.044
Output power, RMS
1
100W into 8Ω
70
10k1
20k2
0.55
0.47
0.044
Input voltage for Pmax. into 4Ω
1V
80
8k9
17k8
0.55
0.47
0.044
2
Input impedance
22kΩ
100
7k1
14k2
0.55
0.47
0.044
3
www.amplimo.nl
www.plitron.com
120
5k9
11k8
0.55
0.47
0.044
as a practical matter, a first-order allpass active filter will generally provide
an overall shifting of about 120°. This
is why I decided to cascade two firstorder all-pass filters in order to get at
least 180°. Figure 8 shows the circuit
I developed for my amplifier. By carefully selecting the values of the components, I was able to produce a circuit
that provides a means to modify the
phase—within the subwoofer’s frequency range—by over 180°. Because you
have two cascaded first-order all-pass
filters, fcAP will now be at –180°.
As mentioned earlier, I replaced R1 of
Fig. 7 with a stereo potentiometer (P 1 /
P 2 ) to enable easy adjustment of the
phase. I placed the additional resistors
R20 / R23 in series to prevent the signal
from being shunted to ground when P 1 /
P 2 is positioned at 0Ω.
Finally, the all-pass filter is followed
by a 10k log potentiometer (P3) that
acts as a volume control. I have decided
to introduce the volume control at this
stage of the circuitry because it enables
the use of a mono potentiometer. You
could also install it right at the beginning of the first stage in place of R 1 / R 2
in Fig. 2. Should you prefer this configuration, you simply select a 100k stereo
potentiometer.
100W POWER AMPLIFIER
This was the easiest part of the project.
Instead of building an amplifier from
scratch, I decided to purchase a readyto-use power amplifier module, the
FIGURE 8: Dual first-order all-pass
active filter.
38
TABLE 2 Some ratings of the Amplimo A120
power amplifier module
audioXpress 1/07
Godmaire2689-4.indd 38
Amplimo A120, a nice solid-state amp.
These modules are available either from
Amplimo in the Netherlands or Plitron
in Toronto. Table 2 gathers some information available on their websites.
You might consider other solutions
here. For instance, you could build an
amplifier based on the well-known
LM3875 power amplifier manufactured
by National Semiconductor. Mike Gustafson published an interesting article6
(GA 2-3-4/00) in which he compared
a tube-based subwoofer amplifier to a
LM3875 based solid-state amplifier.
More recently, I have seen a very powerful subwoofer amplifier built around the
same LM3886 in the French magazine
Led 7 . The circuit described in the article
contained eight of these ICs mounted in
a push-pull configuration and produced
280W into 8Ω! The choice is up to you
whether you prefer to build a unit or to
use a power amplifier module such as
those manufactured by Amplimo.
Figure 9 provides a schematic view
of the A120 power module and its surrounding network. The amplifier is designed to operate with a relay, which
performs several protection functions.
One of these is to provide a one-second
delay after the application of mains voltage so that the subwoofer is protected
from the switch-on noise. The A120
will function without the relay, but neither the subwoofer nor the module will
be protected under fault conditions.
There is also an automatic volume
control feature that prevents the amplifier from clipping. Connecting the LIM
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pin to ground will disable this automatic
volume control, and then the amplifier
will clip on overload. You can install a
switch between the LIM pin and the
ground to activate or deactivate the feature. I chose to permanently enable this
feature by leaving the LIM pin unconnected.
Note that the purpose of the LED
feature is to indicate when the automatic
volume control is in operation. It does
not indicate when the amplifier is on or
off, as you might imagine. I have decided not to connect this LED. Following
the relay is a double-pole double-throw
switch (S3), which acts as a phase inverter and combines with the phase shifter
to provide a 360° adjustment capability.
UNREGULATED POWER SUPPLY
As you can see in Fig. 10, this circuit is
pretty straightforward. Nevertheless, I
will give some explanations. I selected
Z1 in a time-delay version due to the
high inrush current at power on. The
160 VA power transformer provides two
secondary taps of 33V at 2.42A. This
model is recommended by Amplimo
and Plitron to complement the Amplimo A120. Should you use another type
of power amplifier, then you will need to
choose the right power transformer to fit
your needs.
The full-wave bridge rectifier uses
Schottky diodes (D1 to D4) paralleled
with small value polypropylene capacitors (C10 to C13). I selected these diodes
for their low noise characteristics. You
could use another type depending on
your personal preferences. Make sure,
however, they can pass at least 3A.
Following the full-wave rectifier, you
find two power resistors (R26/R27), whose
main purpose is to limit the inrush cur-
FIGURE 9: Amplimo A120, relay, phase inverter switch, and binding posts.
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REGULATED POWER SUPPLY
FIGURE 10: Unregulated power supply circuit.
rent at power on. They also form a lowpass filter with capacitors C14–17 / C18–21.
The cutoff frequency of 17Hz contributes to the reduction of the 120Hz ripple
voltage. Note that the 4700µF capacitors
need not have 80V tolerance; 63V would
be perfect. I used 4700µF/80V simply
because I had these on hand. It is also
important to mention that C16 / C20 and
C 17 / C 21 are bypass capacitors that are
meant to reduce ripple voltages, and also
note the ESR of the large electrolytic
capacitors C14, 15 /C18, 19.
The red LED D5 indicates whether
the amplifier is on or off. The zener diodes ZD5, 6 simply decrease the voltage
to around 2V, within the operating range
of the LED.
This power-supply circuit generates
DC voltages of ±48V that directly power
the Amplimo A120. These voltages are
too high for the regulated power-supply
circuit that follows (Fig. 11). This is why
each rail finds two 16V/5W zener diodes
(ZD1, 2 / ZD3, 4) in series to bring these
voltages down to ±16V. Beware that
these zener diodes will become quite hot
when the amplifier is in operation.
On my first prototype, I had them
installed on the regulated power-supply
board. Their leads conducted considerable heat to the input capacitors C22 /
C23, which was really not a good thing.
This is why I moved the zener diodes
to the unregulated power-supply circuit,
where the components are quite rugged.
The wires between ZD2 / ZD4 and the
regulated power-supply board are long
enough to alleviate this heat transfer
problem.
FIGURE 11: Regulated power supply.
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This is the last circuit of the subwoofer
amplifier. Its purpose is to provide a very
clean voltage source to the op amps. To
fulfill this task, I selected two adjustable
voltage regulators manufactured by Linear Technology: the LT1085CT (IC 4 )
and the LT1033CT (IC5 ) . Both regulators will deliver up to 3A output current.
Figure 11 depicts the circuit of my regulated power-supply board.
As mentioned in the datasheets8, the
input capacitors C 22 / C 23 are required
because the regulators are more than 4˝
from the filter capacitors C14–17 / C18–21
located on the unregulated power-supply board.
The output voltages are controlled by
resistors R28 / R29 and potentiometers
P4 / P5. The following simplified formulas enable their calculation within 1% of
accuracy:
VOUT+ = 1.25V (1 + P4 / R28 ) = V3+
(5)
VOUT- = 1.25V (1 + P5 / R29 ) = V3(6)
C 24 / C 25 are optional components.
Bypassing the adjustment pin with a capacitor reduces the output ripple, noise,
and impedance. The output capacitors
C26 / C27 are required to ensure frequency compensation and stability.
Note, however, that in the case of the
LT1033CT (IC5), this capacitor will necessitate the use of an external protection
diode D6. Without this diode, the voltage regulator might be damaged by the
discharge of C27 should the input voltage become shorted. The LT1085CT’s
(IC4) internal protection is sufficient
to avoid such damage as long as C26 is
smaller than 1000µF.
You will notice in Fig. 11 an unusual orientation for R28 / R29. Following
the recommendations of Linear Technology, I soldered these resistors very
close to the output pin of the regulators. The drawing also reveals that the
ground leads of P4 / P5 are connected
to the ground leads of C26 / C27 instead
of being connected directly to the star
ground point. These subtleties are recommended by Linear Technology to
ensure the best load regulation.
One last word about the power-supply
circuits: All op amps have been bypassed
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with 10µF/16V electrolytic capacitors
and 0.01µF/63V polypropylene capacitors in order to shunt to ground any
remaining ripple voltages. If you wish
to learn more about this topic, I encourage you to read an interesting article9 by
Kevin Ross about bypass capacitors.
CONCLUSION
This first article in this series has explained all the theoretical aspects of my
project. Part 2 will focus on making the
PCBs and will support the circuits I
have just described. aX
REFERENCES
1. Roland E. Thomas and Albert J.
Rosa, The Analysis and Design of Linear
Circuits, Second Edition, Prentice Hall,
1998.
2. Martin Gilres, editor and contributor, Audio/Radio Handbook, National
Semiconductor Corporation, 1980.
3. Walter G. Jung, editor and contributor, Op Amp Applications, Analog
Devices, 2002.
4. OPA2604 Datasheet, available at
www.burr-brown.com.
5. Maxim Dallas Semiconductors,
Tech Brief 3: Digitally Control Phase
Shift, Application Note 559, available at
www.maxim-ic.com, 1996.
6. Mike Gustafson, “A Basic 50W
Stereo System Part 3: A Solid-State
Amplifier,” 4/00 Glass Audio.
7. Bernard Duval, “Un bloc mono de
forte puissance,” Led No. 168, Nov./Dec.
2001.
8. LT1033 and LT1085 Datasheets,
available at www.linear-tech.com.
9. Kevin Ross, “Basic Circuits-Bypass Capacitors,” available at www.
seattlerobotics.org/encoder/jun97/basics.
html.
Rudy Godmaire works as a sales consultant for Bell Canada in Quebec City.
DIY audio has become a passion ever since
his first attempt in 1998. He deeply enjoys
sharing his passion with other DIYers and
especially with friends Francois and Pierre.
Above all, listening to music and attending classical concerts and opera with his
beloved wife Elaine remains among the
most fulfilling activities he cherishes. Readers may visit his personal website at www.
sympatico.ca/r.godmaire.
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speakers
A Unique Crossover Design
with Waveform Fidelity
This author uncovers a heretofore virtually unknown crossover design
that offers both linear phase and flat frequency response.
A
s you know, a single loudspeaker driver is simply incapable of reproducing the entire 20 to 20kHz frequency
range. The physical qualities needed
at one frequency extreme are the exact
opposites of those needed for success at
the other extreme. So you are left with
basically two choices: either use a single
(so-called) full-range driver and accept
some loss of response at both the highand low-frequency extremes along with
a loss of dispersion in the higher frequencies, or choose to “divide and conquer” with crossover filters and multiple
drivers, which will allow you to cover a
much wider frequency range with better
overall dispersion.
However, almost all crossover networks introduce their own problems either in the form of frequency and/or
phase aberrations, or have rolloff slopes
that are much too shallow to be very
useful. Fortunately, there is one type of
crossover that you can use to produce a
multi-way system with flat frequency response, linear phase response, and useful
crossover slopes.
First look at the full-range driver option. This choice is appealing not only
because of its simplicity, but because a
good full-range driver has the ability to
sound smooth and very cohesive. This
cohesiveness is likely due to the very fact
that the full-range speaker system has
no crossover. With no crossover network
to disturb its phase characteristics, the
driver’s phase curve, while it isn’t flat, is
continuous, smoothly changing throughout the driver’s bandwidth from a phase
lead at the low frequency end, moving
through 0° somewhere in the midband,
and finally ending with a phase lag in
the higher frequencies.
As the late Richard Heyser, one of
the deepest thinkers on the subject of
loudspeaker phase, stated in a December
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1974 article on Speaker Phase Measurements for Audio Magazine, “Even
if you do not subscribe to the philosophy that all sound should recombine
as though from one source, you should
note the behavior of the transitions between (loudspeaker) drivers. The transition in phase should be uniform without
severe discontinuities (breaks or abrupt
changes).”
Among conventional crossovers, there
are many choices, and each type has its
problems. While first-order crossovers
have linear frequency and phase response, they are difficult to implement in
a system because their shallow -6dB/octave slopes require the tweeter to withstand significant power below the crossover frequency, and they need smooth
response from both the woofer and the
tweeter well beyond the crossover frequency to maintain a flat summed response. Depending on the phasing, 2ndorder filters result in frequency response
with either a sharp null at the crossover
frequency or a +3dB rise. A 3rd-order filter can give you flat frequency response,
but not linear phase. Likewise with both
2nd and 4th-order Linkwitz-Riley crossovers.
The unique and little-known crossover presented here is alternately known
as a Kido-Yamanaka, or “Filler Driver”
crossover. It exhibits both flat frequency
and phase response giving the speaker
system the potential for real waveform
fidelity, limited only by the characteristics of the individual loudspeaker drivers
used in the system. It is a three-way network that uses 2nd-order slopes for the
woofer and the tweeter plus first-order
slopes for the midrange. The outputs of
the Yamanaka crossover’s low, mid, and
high frequency bands can recombine
electrically to form a perfect replica of
the input signal and can pass perfect
square waves (Fig. 1).
By Steve Stokes
Some would argue that you don’t listen to square waves, and speaker phase
response is irrelevant. I’m certainly not
interested in listening to square waves
either, but would you consider buying
an amp that was incapable of reproducing a decent-looking square wave? Why
should you give your speaker systems
a free pass when it comes to waveform
accuracy? Both frequency response deviations and phase shift are linear distortions of the input signal, and all else
being equal, a speaker system that minimizes these errors is more accurate than
the one that does not.
With the Yamanaka crossover and
good drivers, other parameters are not
sacrificed and the result is simply better
sound. I hope you will be encouraged by
the knowledge of this crossover design
to take up the noble goal of full speaker
waveform fidelity.
FIGURE 1: A square wave signal (D) input to
the Yamanaka crossover network is split
into three passbands: low (A), band (B),
and high (C). Because this type of crossover
has perfectly linear phase and frequency
response, the three bands have the ability
to recombine into an exact replica of the
original square wave input signal (D).
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ENTER THE YAMANAKA
I first became aware of the Kido-Yamanaka linear phase crossover network
in 1975 while managing a stereo store
when one of the store’s buyers brought
back a brochure from the Consumer
Electronics Show about a new line of
speakers that had been introduced by
Technics. The flagship model was the
SB-7000, a large three-way system with
staggered driver spacing that claimed to
have a linear system phase response. The
brochure showed good acoustic square
waves reproduced by the speaker system,
along with oscilloscope photos of actual
music waveforms into and out of the
speaker: a trumpet blast, drum beats, and
piano chords. It was practically impossible to tell the difference between the
input signals and the speaker’s output.
I requested a pair of SB-7000s for
audition, evaluation, and a little reverse
engineering. When they arrived I found
that the Technics SB-7000s—though far
from perfect—had very good resolution
and a solid cohesive sound much like a
crossover-less full-range system. I was
intrigued enough that I disassembled
one of the speakers, removed its crossover, snuck it into the business office,
and photocopied the circuit board so I
could trace out the crossover circuit. The
PCB held well over 16 components—six
for the three filter sections, ten zobel elements to flatten the driver impedances
so the filters would work correctly into
resistive loads—plus an assortment of
other parts for level adjustments. From
the complexity of the crossover, I could
tell that there was much more to this
“Linear Phase” thing than just marketing hype, and the speaker left a lasting
impression on me as I later went on to
design speakers professionally.
print No. 1059 “Design of Linear Phase
Multi-Way Loudspeaker System” by
Ishii and Takahashi
2. US Patent No. 4,015,089, Ishii et
al. “Design Method for a Linear Phase
Multi-way Loudspeaker System,” US
Patent and Trademark Office website
www.uspto.gov/. (The US Patent Office
website is a fantastic free resource on
anything that’s ever been patented and
is probably the best free knowledge-base
available anywhere.)
3. Engineering Brief, Eric Baekgaard, “A Novel Approach to Linear
Phase Loudspeakers,” May 1977 issue of
the AES Journal.
Bunkichi Yamanaka realized that subtractive crossovers could theoretically
produce all manner of perfect crossover
functions, but they require active implementation and multi-amplifier systems
that were considered too expensive to
have the mass-market appeal desired
by a corporate giant like MatsushitaPanasonic. As mentioned before, a conventional 2nd-order crossover with the
high- and low-pass sections summed in
phase results in a response with a deep
notch, or null, at the crossover frequency.
Yamanaka used the subtractive crossover
method to restore the response lost to
this cancellation.
Yamanaka’s associates, Ishii and Takahashi, replaced the active subtractor with
a simple passive circuit and added it to a
conventional 2nd-order crossover, eliminating the requirement for multiple amplifiers (Fig. 2). The complex math in
Fig. 3 shows the bandpass function that
is required to restore flat response after
the low-pass and high-pass functions are
subtracted from the input signal (Vin).
Simply stated, a Yamanaka dividing
network is a conventional two-way 2ndorder crossover network connected with
both drivers in electrical and acoustic
phase, which causes a cancellation in
the overlapping range of the woofer and
tweeter outputs. To this is added a midrange driver, also connected in phase, fed
by a bandpass filter centered on the same
frequency, which fills in the void (Fig.
4). When the combined acoustic outputs
of all three drivers are positioned so that
they are time adjusted and each of their
signals blend together arriving at the
YAMANAKA EVOLUTION
AND OPERATION
The Kido-Yamanaka crossover was first
described by Bunkichi Yamanaka of
Matsushita Corp. (Panasonic) in 1967.
The following references give a full
mathematical treatment of the Yamanaka crossover network and design method. (Note that this crossover is patented
and its commercial use is restricted.)
The first reference is by far the most
thorough and informative.
1. Audio Engineering Society Pre-
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listener’s ear at exactly the same time,
the resulting acoustic waveform will be
flat in frequency and phase response,
limited only by the imperfections of the
individual drivers.
Eric Baekgaard of Bang and Olufsen
Co. published an Engineering Brief in
the May 1977 issue of the AES Journal showing an identical circuit, with
the only difference being a scaling of
the filter Qs from .5 in the Yamanaka
design to .707. Because the level at the
crossover point is given by the formula
20Log(Q), in the Yamanaka version the
electrical crossover between the woofer
and tweeter is at 20Log(.5) = -6.02dB,
and -3.01dB in the Q = .707 Baekgaard
version.
Baekgaard coined the very appropriate term “Filler Driver” for the midrange
speaker used with this type of crossover
network. He also showed that an alternate version was possible using 3rd-order
filters for the low and high passbands.
However, Baekgaard’s 3rd-order version
requires the midrange level to be +6dB
greater than the woofer and tweeter lev-
els in order to achieve flat response. The
circuit and formulas for Baekgaard’s 3rdorder filler driver crossover are included
at the end of this article.
DESIGNING A YAMANAKA LINEAR
PHASE SPEAKER SYSTEM
The design of a perfect linear phase
loudspeaker system requires three things:
1. A crossover with perfect frequency
and phase response
2. Perfect individual loudspeaker drivers, each with
A. a constant and purely resistive impedance
B. equal sensitivity
C. flat frequency response
D. flat phase response
3. Each driver must be positioned
in space so that all frequencies in each
driver’s passband arrive simultaneously
at the listening position.
Fortunately, the Yamanaka crossover
meets criterion No. 1. Criterion No. 2
need only be approximated, and No. 3
can be approximated over a relatively
FIGURE 2: Sample Yamanaka three-way crossover network based on a
Design Center Frequency (see text) of 2kHz, 6Ω impedance equalized
speakers (nominal 8Ω), and filter Qs of 0.500.
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wide range of frequencies with proper driver positioning. While the phase
shift inherent in all loudspeaker drivers prevents you from ever achieving
perfection, Fig. 5 shows that the actual
phase response of one system designed
using the Yamanaka method was able to
achieve linear phase within ±10° from
400 to 5400Hz and ±25° from 225 to
15kHz. This is better performance than
even most full-range drivers are able to
attain.
SELECTING THE DRIVERS AND DESIGN CENTER FREQUENCY
The system design begins with the simultaneous selection of the drivers and a
design center/crossover frequency, which
I will call the Design Center Frequency
(DCF, Fig. 6). The woofer and tweeter
FIGURE 3: Equation for the bandpass filter
required to restore the missing response in a
conventional two-way 2nd-order crossover.
FIGURE 4: The top group of curves shows the electrical response
of the Yamanaka three-way crossover’s individual frequency
bands: lowpass, bandpass, and highpass. Note that 2kHz is both
the crossover frequency between the woofer and tweeter and also
the center frequency of the midrange bandpass filter. The lower
group of curves shows the output of the speakers driven through
the crossover. The woofer and tweeter operating together causes
an acoustic cancellation at 2kHz which is filled in by the midrange
speaker driven through its bandpass filter. With all three drivers
operating through the crossover, the system has the ability to sum
to flat frequency response and linear phase within the limitations of
the loudspeaker drivers.
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11/21/2006 3:04:17 PM
are chosen as you would if designing a -6dB (.25X system input power) at the
conventional two-way system, and their DCF, -14dB (.04X system input power)
crossover frequency is also the DCF, at .5DCF, and -24dB (.004X system
chosen to be in the center of the mid- input power) at .25DCF, so the tweeter
range driver’s linear operating range. The is not going to be taxed terribly hard if
DCF is therefore the center frequency the crossover is properly terminated and
of the midrange bandpass filter, and is allowed to attenuate as intended. The
the only frequency used in any of the DCF can be shifted higher (or lower) as
calculations of filter values. Those for- long as the woofer’s dispersion extends
mulas are included in Fig. 2.
to match the higher DCF. The example
The chosen DCF must simultane- (Fig. 2) has the actual crossover compoously meet several requirements:
nent values that were used in the Tech1. The DCF must be high enough nics SB-7000 speaker system for a DCF
above the resonant frequency of the of 2kHz, Q of .500, and drivers that had
tweeter to prevent excessive acoustic their impedance curves flattened with
phase shift in the tweeter from making impedance equalization networks to apsystem alignment difficult or impossible. proximate 6Ω resistors.
An octave is adequate.
It is optimal that each of the speak2. The DCF must be high enough to ers have smooth, extended response as a
prevent the likelihood of tweeter dam- starting point, without any serious peaks
age from low frequencies not sufficiently or dips in the range to be left unattenuattenuated by the crossover network.
ated; however, this design is no more, or
3. The DCF must be high enough less, critical than any other design in this
that the tweeter’s unfiltered response is regard. The midrange driver should be a
not excessively rolled off in the desired good-quality wide range unit in its own
passband and transition to the stopband. sub-enclosure large enough to keep its
4. The DCF must be in the central resonance low, as the midrange will be
range of the midrange driver’s linear
operating range.
5. The DCF must
be low enough that
the woofer’s unfiltered response is not
excessively rolled off
in the desired passband and transition
FIGURE 5: Actual measured acoustic phase response of three-way
to the stopband.
speaker system using Yamanaka linear phase crossover network.
That may seem
like a tough list at
first, but the criteria
are not really that
hard to satisfy with
good quality drivers, and the results
are well worth the
effort and expense.
The most str ingent requirements
are for the tweeter,
where the example’s
DCF of 2kHz requires a tweeter
with a resonance of
about 1kHz. With
filter Qs of .500 the
tweeter crossover is
Note that individual chart divisions are 10° and that the phase response is actually ±10° the critical middle range from 400-5400Hz.
FIGURE 6: The system is designed on a Design Center Frequency
which is chosen to be in a range that is overlapped by the unfiltered
response of both the woofer and the tweeter and is also in the approximate center of the midrange speaker’s unfiltered response in
the enclosure or subenclosure in which it will be used. In the chart
the relative levels of each band have been separated vertically to
make them easier to visualize.
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responsible for an important part of the
system output and you want its response
to be reasonably flat. Its phase response
will probably be reasonably close to 0°
through that range and the system phase
will be, too.
SOLVING THE CONSTANT
IMPEDANCE PROBLEM—
ZOBEL, ET AL.
Crossover networks work properly only
when they are properly loaded with an
impedance that matches the one for
which they were designed. So unless
the crossover sees the proper impedance
load, at least in the important frequency
ranges where the crossover is required to
attenuate, the crossover will not perform
as required.
A loudspeaker’s true impedance is
a combination of resistance and reactance that varies widely with frequency, matching its nominal value at only
one or two spot frequencies. A dynamic
speaker in free air, in a closed box, or a
closed back tweeter, will display a single
large peak in the lower frequency end of
its impedance curve due to its resonant
frequency, and an increasingly rising impedance in its higher frequency range
Third-Order Filler Driver Crossover Network
E
rik Baekgaard of Bang and Olufsen showed that a 3rd-order filler driver
system is also possible. The same methods and requirements go into its
design with one very important difference. In order for the 3rd-order filler system to sum to flat frequency and phase response, the output of the midrange
driver must be +6dB higher in level than the woofer and the tweeter. If this
condition is not met, the frequency response will droop in the midrange. Figure 9 is the schematic for the 3rd-order version of the crossover and formulas
for the passive components.
FIGURE 9: Linear Phase “Filler Driver” crossover using 3rd-order filters
for lowpass and highpass filtering. This alternate circuit was described
by Eric Baekgaard of Bang and Olufsen and requires that the midrange
driver be operated at a level +6dB higher than the woofer and tweeter
for flat response. Otherwise the design process for this crossover is the
same as for the Yamanaka 2nd-order network described in the text.
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due to the inductance of its voice coil.
Wherever the impedance is higher than
the crossover design value, the crossover
will attenuate less than desired.
The driver impedance variations that
concern us most here are:
1. the impedance rise in the woofer’s
impedance curve caused by its voice coil
inductance, LE
2. the midrange impedance peak caused
by its closed box resonant frequency
3. the midrange impedance rise due to
voice coil inductance
4. the tweeter’s impedance peak
caused by its resonant frequency.
The impedance rise due to resonant
frequency of the woofer and the rise due
to the inductance of the tweeter’s voice
coil do not affect the crossover’s operation and can be safely ignored. If the
tweeter resonance (4.) is highly damped,
you might ignore it, too.
When fixed resistive pads, or continuously variable L-pads, are used to balance the sensitivity of the midrange and
tweeter drivers to equal out and match
their volume levels to the sensitivity of
the woofer, the effect of the impedance
variations of the midrange and tweeter
on the crossover network will be lessened to some extent. It is worthwhile to
make this level adjustment first and then
measure the impedance curve and see
whether zobel networks are still needed
to produce a nearly constant resistive
load for the crossover network allowing
it to attenuate as required. In my passive implementation of the Yamanaka
system, I use Radian compression drivers for the midrange and tweeter, and
installed fixed resistor L-pads to bring
down their levels. Because the compression drivers are so much more efficient
than the woofers, the resistive L-pads
are completely effective in isolating the
crossover from the actual impedance
variations of the drivers.
If the woofer or midrange voice coil
inductance is unknown, you can determine it by measuring the impedance at
a high frequency between 10kHz and
20kHz with a sine wave generator and
an AC volt-ohmmeter. The formula in
Fig. 7 gives the value for the voice coil
inductance, LE, in henries, where FM
= the frequency at which the measurement of impedance is taken, M is the
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measured Magnitude of the impedance
at that frequency in ohms, and RE is
the speaker’s voice coil DC resistance in
ohms.
With knowledge of the inductance
value of the woofer and midrange voice
coils and Thiele/Small parameters of
the midrange driver measured in the
enclosure in which it will be used, the
formulas and circuit in Fig. 8 adapted
from Robert M. Bullock III’s January
1985 Speaker Builder article on passive
crossovers will give reasonably accurate
values for your zobel impedance equalizing components. The value RE should
be equal to the voice coil’s DC resistance
value. (See back issues of Speaker Builder
for more information on measuring a
speaker’s Thiele/Small parameters and
on impedance compensation networks.)
The most effective way to avoid the
problem of driver impedance variations
and sensitivity differences that complicate the design of passive crossover networks is to make the crossover an active
one instead. This allows the use of filters
containing precision resistors and capacitors working into close tolerance, resistive loads, without the need for large,
expensive, power-robbing inductors.
FIGURE 7: You can calculate LE, the inductance of a driver’s voice coil, from this
formula by measuring the value of its impedance in ohms at any high frequency in
the 10kHz to 20kHz range. In the formula,
M is the magnitude of the impedance in
ohms, RE is the voice coil DC resistance in
ohms, and FM is the frequency at which the
impedance value was measured.
In the next installment I will present
an active vacuum tube version of the Yamanaka crossover circuit, and conclude
with an article describing proper setup
and alignment of both passive and active
systems using the Yamanaka crossover
network. aX
Steve Stokes caught the hi-fi bug while
in college when he encountered a friend’s
system of four JBL LE-15 woofers, LE375 midrange, and LE-85 treble compression drivers driven by a 200W amplifier
and fed by professional Ampex recording
gear.* After college he took a part-time job
selling stereo components, soon became the
store manager, and decided to stay in the
audio f ield. He soon left sales and began
designing home speaker systems as the Chief
Engineer of Lancer Electronics (orginally
part of Soundcraftsmen) for the following
20+ years. He also served as the Director of
Technical Information for Trusonic Corp.
when it was relocated to Fountain Valley,
Calif. in 1978. He is a former member of
the AES and the co-inventor of US Patent
No. 5,212,732 for a Dipole Speaker System
for Surround Sound. He enjoys designing
and building vacuum tube electronics and
digital SLR photography.
*As an interesting side note, the friend’s
15ft3 speaker cabinets were built for him
by Warner Bros. recording engineer Donn
Landee (Doobie Bros., Van Morrison, Van
Halen, etc.) and Dennis Dragon, who was
an occasional tour and studio drummer for
the Beach Boys and the Byrds, the recording
engineer for his brother, Daryl Dragon of the
Captain and Tenille, and the son of famous
Hollywood Symphony Conductor Carmen
Dragon.
FIGURE 8: Circuit and formulas for driver impedance equalization, after Robert M. Bullock III,
Speaker Builder 1/85. You can use this circuit to flatten the driver’s impedance curve to closer
resemble a resistive load and allow for proper crossover operation. Component RE models the
voice coil’s DC resistance and CE nulls the voice coil inductance LE. Components RM, CM, and LM
null the single resonant peak like that of a speaker in a closed back tweeter.
audioXpress January 2007
Stokes-2697-1.indd 47
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XPRESS MAIL
CORRECTIONS
In my article “The Venue Loudspeaker”
(Nov. aX), there are two errors: on p. 17,
top: “3. 1.1kHz dip (L3, R3, C3)” should
be (L3, R2, C3); and on p. 18, right:
“0.34g/15µS = 23kHz” should be 0.349.
Dennis Colin,
Gilmanton I.W., NH
The graphs in Figure 3 and Figure 4 of
my article “Build a Flat Panel Speaker”
(aX 11/06) are switched. The graph with
the title “Radiator ‘A’ ” should be with the
Figure 3 caption: Traditional loudspeaker
in free space and with boundary. The
graph with the title “Radiator ‘B’” should
be with the Figure 4 caption: DML in
free space and with boundary.
Daisuke Koya
dkoya@mac.com
48
audioXpress 1/07
XPressMail-107-3.indd 48
Figure 1.
I’ve long enjoyed Joseph Norwood Still’s QUICK CHECK
excellent and practical tube amp articles, I enjoyed reading Charles Hansen’s arincluding “Two SE Power Amps. . .” (aX ticle on a phase-meter tester (aX 11/06).
Nov. ’06). However, the power supply In fact, I have enjoyed all the articles he
(Fig. 2, p. 31) has an error: the junction has written.
of C1/C2 will charge to the negative
I thought readers might appreciate
peak of half the transformer secondary this circuit (Fig. 2) used by Hewlettvoltage, destroying C2. He describes it Packard for checking the accuracy of the
as a FW bridge voltage doubler; T1 is HP3575A Phase-Gain Meter. It prolisted as 200V AC output. If this is for vides a very accurate 90° phase shift at
each half, then there’d be 565V DC (less 10kHz. Of course, matched silver mica
diode drops) across
C1, if the center top
weren’t grounded.
Also, the heater
transformer is listed
as 12.6V AC, but
the schematic shows
6.3V heaters—the
tubes would light
FIGURE 2:
up like a Christmas
Phase calculator.
tree, for two seconds!
Regardless, I commend you for prov- capacitors and high-quality resistors will
ing that you can have your cake (SE ensure a high degree of accuracy.
amp) and eat it too (low distortion)!
Thanks for the excellent articles.
Jack Walton
Short Hills, NJ
Dennis Colin
dcolin@worldpath.net INVERTER DESIGN
My thanks to Robert Bennett for his
Joseph Norwood Still responds:
response (aX 10/06) to my questions reThank you for your comments. Although garding his article “An Improved Splitthe heater voltage in the parts list states it Load Phase Inverter” (aX 7/06). His
is 12.6V, it is actually 6.3V—per Hammond comments were well thought out and
P.N. P-T 16656A. So if you order P.N.P.-T clearly presented.
166 S6, you will have a 6.3V transformer
While the actual drive “connection”
and not a 12.6V transformer and all is well.
into his inverter stage from the precedThe C.T. of high voltage transformer T1 ing pentode stage can be debated, we
is shown as grounded. This is incorrect. The both agree on the outcome of the design:
center-top should be shown as ungrounded The effect of the bootstrap connection
(see Fig. 1). I should have caught these does increase the gain of the preceding
errors when I reviewed the page. Unfortu- pentode stage rather dramatically, as I
nately, I didn’t! Again thank you, Mr. Colin, originally suggested. Furthermore, as Mr.
for doing my job for me.
Bennett has now shown, this gain inwww.audioXpress .com
11/21/2006 3:17:38 PM
crease can be “modified” by the bootstrap
connection to help reduce the splitter’s
ill effects if the cathode channel should
become momentarily shorted. Taken as a
whole then, it does, in fact, represent an
“improved” phase splitter design.
Robert’s reminder to consider the circuit as a whole is right on target. My
original “red flag” centered on his statement that the pentode’s gain is unaffected
by the bootstrap connection, when in fact
it is. However, I did not take my analysis
further to consider the circuit in total
under shorted cathode channel operation
as he did. I will explain why in a moment,
but for now, I thank Mr. Bennett for his
reminder and analysis of his circuit under
that condition.
When the total circuit is considered,
then a relative value can be placed on
the improvement the bootstrap connection makes in splitter operation under
adverse conditions. Using his own figures
(with which I generally agree), Robert’s
complete circuit without the bootstrap
connection has a total gain of about 207
from input to either splitter output. If the
cathode channel becomes shorted in this
configuration, the total gain from input
to splitter plate output rises over 16 times
to nearly 3500, because the splitter stage
is now providing active gain rather than
a loss. With the bootstrap in place, the
total gain starts at nearly 1100.
When the cathode channel is shorted in
this condition, the total gain to the splitter
plate output rises to nearly the same 3500
as before, but in this case, it is only an
increase of about three times as opposed
to that of over 16 times before. Hence the
improvement, all as Mr. Bennett suggests.
While the improvement is significant, it
is still well short of eliminating the effect
as suggested. But the question is, can the
improvement really help?
It’s understood that this discussion
deals with a phase splitter that is directly
driving a class AB1 push-pull output
stage, which is the most likely configuration in which an overload of the following (output) stage can cause a shorted
cathode channel condition at the splitter
stage in the first place. So assuming an
overload signal of sufficient length and
the splitter stage itself does not overload,
let’s see what happens in this configuration when the output stage does.
On the first positive going cycle pre-
sented to the phase splitter that’s capable
of overloading the output stage, the cathode channel will drive its output tube to
a zero bias condition, and basically clamp
the splitter’s cathode signal at that point
to prevent any further increase during
this cycle—although the setup for R/C
“blocking” in this channel is already beginning. As the overload signal continues
to increase beyond this point at the splitter’s input, the splitter’s plate channel will
continue to drive its output tube further
negative, but now multiplied (in Mr.
Bennett’s design) by a factor of three because of the splitter’s cathode clamp.
But so what? This tube was already
cut off when the cathode channel’s output tube reached its zero bias point by
definition. Then the overload cycle to the
splitter goes negative, and problems start
to develop real quickly. This time, the
cathode channel output tube’s operating
point will start to drift negatively because
the blocking effect previously set up in
this channel has started to kick in, cutting the tube off earlier.
The channel will continue to drive this
output tube further negative, but again at
this point, so what? The tube was already
cut off when the plate channel’s output
tube reached its zero bias point, so what
does a little more cutoff matter? The
plate channel has driven its output tube
to a zero bias condition during this cycle,
but this time without the aid of any extra
amplification by the splitter stage because
its cathode was not clamped during this
cycle—although the setup for blocking
in this channel is now beginning as well.
Therefore, on the second consecutive
positive overload cycle into the splitter,
the plate channel output tube’s operating
point also starts to drift negatively, with
the process repeating itself on each cycle
until the overload is removed.
So once again, blocking is the main
culprit in this configuration at overload
(it usually is when R/C coupling is used
into the output stage), as realistic shorted
cathode channel operation at the splitter
causes it to misbehave in a sort of half
wave benign way per say, and is the reason I did not consider it in my original
letter. However, the bootstrap connection in Mr. Bennett’s splitter does help
to control overload in the splitter itself
when its cathode does become clamped.
This minimizes needless excessive cur-
audioXpress January 2007
XPressMail-107-3.indd 49
49
11/21/2006 3:17:44 PM
rent flow through the stage during positive going overload cycles presented to
it, and therefore works to contain (but
hardly eliminate) the overload blocking
effect in the cathode channel, were it not
in place. I would add that using cathode
bias in the output stage and minimizing
any global feedback will help to soften
the overall blocking effect if this configuration is used.
Ultimately for me, the beauty of Mr.
Bennett’s design is the increased gain it
provides under normal operating conditions, without the necessity of an additional gain stage to achieve it. As for
the phase splitter itself, I choose to all
but eliminate its potential misbehaviors
through its location and design, in relation to the associated circuits it interacts
with. You can then decide for yourself
how to best apply the merits of Mr. Bennett’s design to your own projects. I thank
Mr. Bennett for putting his idea forth,
the time to prepare it into article form,
and for his thoughtful response to me.
Robert Bennett responds:
Mr. Gillespie again raises some interesting
points in his latest letter. With the bootstrap
circuit, there is a tradeoff between the gain
and the symmetry of output impedances.
By making the pentode’s load resistor very
small in comparison to its anode resistance,
the output impedances approach the same
value, but the gain is less. The values given
in my article are a compromise, but lean
more toward high gain. It might be possible
to get both features by running the pentode
at very low screen voltage, but I suspect
that attempts to get much higher gain from
the circuit would result in hum and distortion problems.
The second point of Mr. Gillespie’s is
one that I had not considered. I think he
is correct that cutoff of the output valves
reduces the effects of an overload. It is an
interesting thought that a Class AB1 amplifier could have a smoother overload or less
distortion than a Class A one.
Once again, I would like to thank Mr. Gillespie for his very interesting analysis and
comments. aX
David Gillespie
Atlanta, GA
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Many months ago I took a few DVDs to Radio
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Needless to say, I was overjoyed to find a Sony
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Chesky Contest Winners
Grand Prize winner of the five CDs:
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Congratulations to our winners and
thank you to all who participated!
Visit www.audioXpress.com
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audioXpress 1/07
XPressMail-107-3.indd 50
www.audioXpress .com
11/27/2006 2:31:20 PM
solid state
Low-Level Analog Switching
Various support gadgets can distort your electric
guitar sound. Here’s how you can keep that
sound clean.
U
sing effects between an electric guitar and amp can be a
superb and inspiring way to
produce new sounds. However, the interconnections and bypass
switching systems that control the effects can also blur and distort the original sound. In an attempt to minimize
the destruction of tone, I have experimented with ways to retain as much of
the guitar’s fundamental character as
possible. There are few things worse
than bad tone.
You can accomplish switching lowlevel analog signals such as the voltage and current generated by an electric
guitar pickup with an electronic relay if
you consider specific details. By using
self-latching low thermal emf relays, you
can best preserve the analog components
inside the electronic device. Control signals sent from the outside world must be
optically coupled.
SELF-LATCHING RELAYS
As a first approximation, an electric guitar pickup generates a full-scale signal
of 2V peak-to-peak, or 0.707V RMS.
This is usually terminated into a 1Meg
impedance at the guitar preamp, and the
power is 500nW.
Signals transmitted between effects devices in
front of the guitar
amp also run at
this level.
A Pa n a s o n i c
Electric Works
DS2E-S-DC12V
12V signal relay
uses a holding current to pull in the
contacts. Its coil
By Dennis Hoffman
resistance is 720Ω, and this is a power
of 0.2W. Relay power is 400,000 times
greater than the power in a guitar signal. The magnetic field is theoretically
constant, but any power-supply ripple or
noise components can electromagnetically couple into the analog signal. There
is a very good chance that part of the
control circuit power will be fed into the
guitar signal. This is what led me to use
self-latching relays that have zero holding current.
Self-latching relays have a small permanent magnet attached to the end of
the coil’s armature. A brief pulse will
cause the coil to move to one of its two
positions. There will be an iron post or
another magnet that holds the armature
in position. Reversing the polarity of the
control pulse will send the armature to
the other of its two positions.
Again, it will use the permanent magnet to latch itself in position after you
remove the control pulse. The armature
pushes against the contacts in one position and releases them in the other.
These self-latching relays can also have
two separate control coils that generate
opposite magnetic control fields to pull
the armature one way or the other. Be-
PHOTO 1: Complete switching
system with battery powered
effects. Photo by Jeanne
Hoffman.
audioXpress January 2007
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cause there is zero holding current, there
is virtually no leakage.
To be specific, the solid-state circuit
that drives the control coil will be in its
cutoff condition, yet the transistor will
have a small leakage current typically
in the nanoamps. Also, by running the
control circuit from a battery, you can
further eliminate the usual electronic
power-supply noise. At this point control current is infinitesimally small.
As an additional note, any effect powered by AC runs the risk of picking up
pollution from the power line. Some
devices may run on AC with a wall wart
PHOTO 2: Switching system
inputs and outputs. Photo by
Jeanne Hoffman.
that outputs a low voltage DC to the effect, so you can substitute a rechargeable
sealed lead acid battery with a high amp
hour rating. I use a 12V/7AH battery
with the Moog MF-104Z Analog Delay,
and for me there is no question that the
battery is the superior power source.
LOW THERMAL EMF CONTACTS
Any time current passes through two
dissimilar metals, the connection generates a voltage. The voltage depends on
the type of metal and the temperature of
the junction. This is called the thermoelectric voltage, or thermal electromotive
force. Either inside
the relay or when
the relay is connected to a wire
(or printed circuit
board), there will
be a change from
one metal composition to another.
A relay designed
to keep this voltage generation to
a minimum will
have a thermal
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OPTICALLY COUPLED
CONTROL SIGNALS
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FIND THE ENTIRE PRODUCT SELECTION ON-LINE AT
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52
audioXpress 1/07
Hoffman-2719-2.indd 52
emf of less than 10µV. A normal switch
or relay can have a generated thermal
emf several times greater.
While this thermal emf is mostly a
DC offset error, it is dynamic because it
will change with temperature and relay
contact force. The error is a voltage artifact produced by the circuit, and this
voltage is not a part of the information
in the signal. The offset becomes one
more flaw in the system.
Perhaps I am being overly analytical,
and the imperfection is rather small in
the larger scheme of things. Think about
how many connections there are in a
switching system and add all those errors
together. You can minimize this problem
by using a low thermal emf relay, such as
the Panasonic Electric Works SX series.
The only limitation is the signal must be
less than 10V and 10mA.
A positive thermal emf offset will raise
the signal above its normal 0V reference.
You can reduce an A/D converter’s input
gain to keep it from being overdriven in
the positive direction because of the offset. Due to the reduced gain, the negative excursion of the signal will never get
to full-scale negative.
As a result, one or more least significant bits of accuracy are removed. You
only get those low-order bits and your
maximum signal-to-noise ratio if the
signal can travel from 0V to full scale.
A 20-bit digital converter will resolve a
674nV signal (.707V/2 exp 20), but that
accuracy is now lost.
When you connect a copper wire control
cable carrying electricity to a system, this
can be an antenna for unwanted noise. A
simple way to solve this problem is to
use light to connect signals from one
system to another. The use of optocouplers—an idea that has been around a
long time—is a cheap and excellent solution. Current from the control circuit
forward-biases a light emitting diode,
and the photons from the LED turn on
a photo transistor. Light does not pick
up EMI.
And the optocouplers can hold
off high voltages. The possibility of
a ground loop also disappears. These
details apply to the control systems of
channel switching amplifiers as well as
www.audioXpress .com
11/21/2006 3:11:38 PM
FIGURE 1:
Self-latching
relay and remote
control circuits.
effects switching systems.
The remote control unit will have a
battery to provide the pulses to temporarily turn on the optocouplers, which
then turn on a transistor that momentarily energizes the self-latching relay’s
coil. Under normal operation there is
no current flowing in either the remote
control unit or the switching system.
Only during switching pulses does current flow (Fig. 1).
The battery return and chassis of the
remote control unit are connected together. This common point should be
connected to the chassis of the switching
unit, which may or may not have its signal ground connected to its own chassis
at one single point, but this is a connection independent of the remote control
unit. Do not connect the remote control
unit ground directly to the switching
unit’s signal ground.
Inside the switching unit all of the
control circuit relays and transistors are
wired to the battery return. A single wire
runs from the control circuit battery
return to the signal ground, which helps
to isolate control circuit leakage current
from signal current. Finally, the typical
¼˝ phone jack at the output is connected
to the chassis, while the other jacks are
isolated with shoulder washers.
Continues on p. 59
audioXpress January 2007
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11/21/2006 3:11:46 PM
From p. 53
TABLE 1: LOW LEVEL ANALOG SWITCHING
CIRCUIT PARTS LIST.
Part #
Part
Vendor
Vendor Number
BT1-3
SW1, 2
9V battery
momentary switch (pushbutton)
switch cap
amp footswitch plug
plug sockets
plug cable clamp
amp control signals receptacle
receptacle pins
¼˝ phone jack long bushing
⅜˝ shoulder washer
ASX22012 relay
1k ¼W 5% resistor
12k ¼W 5% resistor
510R ¼W 5% resistor
240R ¼W 5% resistor
1N4003 diode
TLP372 optoisolator
2N2907/PN2907 PNP
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Mouser
Mouser
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Digi-Key
Digi-Key
N145-ND
450-1094-ND
450-1054-ND
A1304-ND
A25137-ND
A1332-ND
A1305-ND
A25136-ND
502-L-12A
534-3069
255-1589-5-ND
1.OKQBK-ND
12KQBK-ND
510QBK-ND
240QBK-ND
1N4003/4GICT-ND
TLP372-ND
PN2907-ND
Carling heavy duty stomp
switch (expensive, long lead time)
Mouser
691-216-PM-OFF
P1
J1
J2-5
K1
R1, R3
R2, R4
R6, R7
R5
D1, D2
U1, U2
Q1, Q2
Optional
SW1, SW2
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Note that an effects device that
radiates an electric
disturbance into a
guitar and amp system by itself will
still emit this interference when connected to a switching system. Sometimes using the bypass on the device
itself will help. For
me, The Analogman
Tube Screamer is a
remarkable distortion pedal, yet the
noise it radiates
when not in use is
r ubbish. Conse quently, you must
switch this device
using its own true
bypass switch. In
a live performance
situation many of
these things might
not be a problem,
but they can become
evident when playing solo or recording.
SUBJECTIVE TONE
One last consideration is my subjective opinion of the effect on the sound.
When playing an electric guitar, you
should listen to the sound directly into
the amp. Become comfortable with
your direct tone. With switching units
and effects connected, a distortion or
blurring of the signal may sound like a
damper has been placed on the string,
which can’t vibrate to its full excursion—it sounds flat and two-dimensional. The bass might sound muddy, or
fat and bloated. Often the treble will be
hard and brittle. The rhythmic flow of
the boogie is just not there.
Ultimately, if you and I can keep playing and listen to the music, then all is
well. But if the equipment is distracting
and keeps calling attention to itself, we
will know something needs tweaking.
All of the errors in a system obscure
and distort the details of the music.
While one error in itself may be small,
the addition of many errors can present
serious disturbances. 500nW of guitar
signal is a very small quantity. aX
Installer
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audioXpress January 2007
Hoffman-2719-2.indd 59
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11/21/2006 3:10:53 PM
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CLASSIFIED
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11/21/2006 3:14:44 PM
Book Review
The Art of Linear Electronics
by John Linsley Hood
Audio Amateur Press., 339 pages, $39.95
Reviewed by Dennis Colin
T
his book is highly recommended for all who are interested in highly detailed (but
clearly explained) descriptions of analog components, circuits,
and systems. Particular emphasis is focused on audio systems, but RF circuits
are also covered.
All that is required is a basic understanding of math relationships used in
analog circuits; i.e., Ohm’s law, reactive
impedance concepts, dB (log/exponential) versus linear, and so on. But this is
not a design “cookbook.” While a wealth
of practical circuits are shown (power
supplies, audio preamps and power
amps, both tube and solid-state; radio
receivers, audio and RF oscillators, and
more), the emphasis is on understanding
the role of analog components (linear
and nonlinear, contrary to the title) in
providing the desired function.
Chapter 1 is titled “Electronic Component Symbols and Circuit Drawing,” while Chapter 2 discusses “Passive
Components.” After this introductory
material, Chapters 3-5 explain, in great
and well-illustrated detail, the internal
workings and applications of tube and
transistor devices and circuits.
Following are chapters 6-8 on DC
and low-f requency (LF) amplifiers,
feedback, and passive and active filters.
INSIGHTFUL CHAPTER
Chapter 9, “Audio Amplifiers,” begins
with the basic requirements of power,
bandwidth, response flatness, influence
of acoustics and music type, and distortion audibility and its subjective effects.
Circuits described begin with classic
antique radio valve (British for “tube”)
amps, the (D.T.N.) Williamson amp,
triode/pentode/UL and Class A or AB
configurations, and so on.
Then follows a discussion of the evolution of solid-state power amps, with a comprehensive description of how (especially
the earlier) transistor amps can produce
62
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Colin-book-2758-2.indd 62
“listener fatigue” due to low-level nonlinearities (not just crossover distortion).
While the failure of rigorous comparisons between very good power amps
to show conclusive sonic differences
is mentioned, the author states, “For
myself, I think that there are still some
small remaining differences in sound
quality between different power amp
circuit designs. . .”
I personally believe this to be the most
balanced, observant, and common-sense
attitude to the highly-charged “golden
ear versus meter reader” debate.
This chapter (of greatest interest to aX
readers) concludes with descriptions of
preamps, EQ/tone controls, and low-noise
circuitry for phono preamps (five circuits
shown). The 34 comprehensive pages of
this chapter are, I would say, of greater
value to the serious audio enthusiast than
some complete books on amp design that
I’ve read! I consider myself a very proficient audio designer, yet I’ve learned some
valuable insights from this material.
The remaining chapters (10-17) cover
non-audio material, except for Chapter 15 (Power Supplies), 16 (Noise and
Hum), and 17 (Test Equipment). However, the material is presented with the
same clarity, comprehensiveness, and
illustration as in the audio material. The
topics are oscillators (audio and RF),
radio receivers, tuned circuits, waveform
generators, and noise sources. Also included are two appendices, “Component Manufacturing Conventions” and
“Circuit Impedance and Phase Angle
Calculations.”
If you are getting the impression that
The Art of Linear Electronics is an absolutely first-rate book that superbly fills
the present “digital age” vacuum of needed analog circuit coverage, you would be
correct! aX
(Available from Old Colony Sound Lab,
PO Box 876, Peterborough, NH 03458, 888924-9465, custserv@audioXpress.com)
www.audioXpress .com
11/21/2006 3:01:33 PM