Low-Voltage Fluxgate Magnetic Current Sensor
Interface Circuit with Digital Output for Portable
Applications
M. Ferri∗ , A. Surano∗ , A. Rossini∗ , P. Malcovati∗ , E. Dallago∗ , A. Baschirotto†
∗ Department
of Electrical Engineering, University of Pavia, Italy
{massimo.ferri, antonio.surano, andrea.rossini, piero.malcovati, enrico.dallago}@unipv.it
† Department of Physics, University of Milano Bicocca, Italy
andrea.baschirotto@unimib.it
Abstract—In this paper we present a complete low-voltage, lowpower and high linearity CMOS interface circuit for fluxgate
magnetic sensors for current measurement applications. The
integrated circuit provides the correct excitation signal to the
fluxgate sensors and reads-out the sensor signals from the sensing
coils. The proposed circuit allows us to deal with sensors featuring
different values of the excitation coil resistance and to process
the sensing coil signals with a power consuption lower than
1 mW. The interface circuit consists of three different modules,
namely a timing block, an excitation block and a read-out chain.
The interface circuit, has been implemented with two different
excitation circuits, operating at 5 V and 3.3 V, respectively,
without any high-voltage process options. The read-out chain
performs a synchronous demodulation of the even harmonics,
in order to extract the value of the external magnetic field.
Furthermore, it is possible to switch-on a 13 bit ADC, to provide
at the output the demodulated signal as a digital word.
I. Introduction
Magnetic sensors represent an essential block in several
industrial and research applications, such as environmental
monitoring [1], current measurements [2]–[4], electronic compasses [5]–[7] and proximity detection systems [8]. Fluxgate
sensors are typically used to measure low values of magnetic
field, such as the Earth magnetic field [5] or the magnetic
field induced by low currents. When fluxgate magnetic sensors
are used for current measurements, the electronic interface
circuit plays an important role, since it must guarantee high
linearity, low-power consumption (for portable applications),
reliable results and high magnetic noise rejection. In this
paper we present a complete low-voltage and low-power
CMOS interface circuit for fluxgate sensors. The proposed
circuit allows us to excite sensors with different values of the
excitation coil resistance and to process the sensor signals.
The chip consists of three different modules, namely a timing
block, an excitation block and a read-out chain. The interface
circuit, whose block diagram is shown in Fig. 1, has been
implemented with two different excitation circuits, operating
at 5 V and 3.3 V, respectively, without any high-voltage stage
[9]. The read-out circuit allows us to retrieve the information
on the external magnetic field from the sensing coil signal. In
the interface circuit, we included also a 13 bit ADC [10],
Biasing
+
Timing
Excitation
5V
+
Excitation
3.3 V
Readout Chain
Fig. 1.
+
ADC
System block diagram
to provide the measured magnetic field value as a digital
word. The timing block provides control signals for both
excitation and sensing. In the considered current measurement
application, we have used a fluxgate sensor with an excitation
coil featuring 140 Ω resistance and 4 µH inductance, which
needs to be excited with a 16 mA current signal with odd
symmetry at 100 kHz [11].
II. 3.3-V Excitation Circuit
Fig. 2 shows the excitation circuit operating at 3.3 V. It
consists of an H-Bridge, which exploits an external inductance
to generate a triangular current excitation signal starting from
a square-wave voltage signal. In order to achieve the desired
excitation signal, the value of the external inductance is
380 µH. The external 3.3-V, 400-kHz clock, after frequency
division by 4, drives the H-Bridge with two opposite square
waves at 100 kHz. Fig. 3 shows the excitation signal obtained.
III. 5-V Excitation Circuit
Fig. 4 shows the excitation circuit with 5-V power supply.
It consists of a triangular wave generator, a voltage driven
current generator, a current mirror, and a H-Bridge. The
triangular wave generator provides a 130-mV signal around
2.5 V, obtained by integrating the frequency-divided clock and
level-shifting it around the proper average value. It is possible
to modulate the amplitude of the signal changing the value
Vdd
Vdd
T
S1
Q
M3
M4
M2
M5
C1
Vref2
Clk
R2
CLK
R4
S2
Q
R6
R7
R1
S1
R3
-
Vref
M6
-
+
Vdd
2
S1
Vref2
R5
-
S2
+
Vdd
2
M7
+
+
M1
-
Rs
S1
Rrif
M8
S2
M9
gnd
Vdd/2
Fig. 4.
Ls
5V Excitation Circuit
b0
Vdd
b1
b2
X60
Vdd
CLK
T
Q
Vdd
T
Q
Rs
S1
CLK
DEMUX
S1
Lext
Ls
X1..100
ADC
out 0
SallenKey
Filter
out 12
S2
CLK
Q
Q
Offset Control
S2
Offset Control
Fig. 5.
Offset Control
Block diagram of the read-out chain
gnd
IV. Read-out Chain
Excitation Current [mA]
Fig. 2.
Schematic of the 3.3-V excitation circuit
20
10
0
–10
–20
–30
0.0
30
Time [ µs ]
60
90
Fig. 3. Waveform obtained in simulation at the output of the 3.3-V excitation
circuit
of Vref . The Wilson current mirror amplifies the current by
K = 10, thus leading to a 20-mA peak current signal. The 200kHz H-Bridge driving signals are used to switch alternatively
the direction of the current flowing into the excitation coil of
the sensor. In particular, when signal S 1 is high and signal
S 2 is low, transistors M3 and M7 are switched-on, while
transistors M6 and M5 are switched-off. During this period,
the current flowing through the sensor is KIref . By contrast,
when signal S 1 is low and signal S 2 is high, transistors M6
and M5 are switched-on, and the excitation current flowing
through the sensor is −KIref . As a result, the sensor is excited
with a 40-mA peak-to-peak current, as required. Without the
H-Bridge, this behavior would have been possible only with
a symmetric supply voltage (±5 V).
Fig. 5 shows the block diagram of the read-out chain. In the
presence of an external continuous magnetic field, the even
harmonics of the signal obtained from the sensing coils of
the sensor are not zero. In particular, the spectral component
carrying most of the information is the second harmonic of
the excitation signal. Therefore, the read-out chain performs a
synchronous demodulation of the even harmonics, in order to
extract the value of the external magnetic field component.
A. First Gain Stage
The first gain stage of the read-out chain is an instrumentation amplifier, which amplifies by 35 dB the differential signal
obtained from the sensing coils of the fluxgate sensor. An
instrumentation amplifier is required to eliminate commonmode coupling signals. Fig. 6 shows the schematic of the
instrumentation amplifier. Each operational amplifier used in
this circuit consumes 168 µW, with a supply voltage of 3.3 V,
achieving a bandwidth of 133 MHz with a phase margin of
76◦ .
B. Coherent Orthogonal Demodulator
Since the information on the external magnetic field in
the fluxgate sensor output signal is contained in the even
harmonics of the excitation signal, it is necessary to downconvert this information to the base-band. The demodulation
is performed by multiplying the sensing signal with a 200kHz square-wave. In order to avoid degradation of the downconversion process in the presence of unwanted phase shifts
V1
R1
V2
R2
R2
+
C2
R4
+
R3
+
R3
Vo
R1
Vi
R2
-
+
C1
Vo
-
R4
R4
R3
Fig. 6.
Schematic of the instrumentation amplifier
Fig. 8.
Schematic of the Sallen-Key filter
TABLE I
Programmable-Gain Amplifier Settings
b0
0
0
0
0
1
1
1
b1
0
0
1
1
0
1
1
b2
0
1
0
1
1
0
1
Gain
1
2
5
10
20
50
100
D. Programmable Gain Amplifier
Fig. 7.
Demodulation error
between the sensor signal and the 200-kHz square-wave, we
used a quadrature demodulator. In particular, the sensing signal
is multiplied by a couple of orthogonal square-waves. Ideally,
at the output of the quadrature demodulator, the signal should
be reconstructed using
q
2
2
Vout = Vout,0
(1)
◦ + Vout,90◦ ,
where Vout,0◦ and Vout,90◦ are the output signals of the two
quadrature paths. To simplify the circuit, the reconstruction
has been implemented simply by adding Vout,0◦ and Vout,90◦ ,
according to
Vout = Vout,0◦ + Vout,90◦ .
(2)
The error between the two algorithms depends on the value
of the phase displacement, as shown in Fig. 7. If the phase
displacement is constant, as it is in the fluxgate sensor, this
leads to a gain error, that can be compensated.
The Sallen-Key filter is followed by a final gain stage,
which amplifies the dc signal proportional to the magnetic
field, obtained at the output of the filter, with a programmable
gain. Setting the value of digital signals b0 , b1 and b2 , it is
possible to program the gain of the stage, according to Tab. I.
E. Incremental ADC
The schematic of the 13-bit A/D converter [10] is shown
in Fig. 9. The A/D converter, based on the switched-capacitor
(SC) technique, consists of a fully-differential resettable integrator and a comparator, whose output is connected to a 13-bit
counter. The ADC can be enabled or disabled, according to the
desired power consumption. The allowed input signal range is
variable, with a maximum of about 1 V.
V. Experimental Results
The proposed interface circuit has been integrated in a 0.35µm CMOS process. Fig. 10 shows the micro-photograph of
the chip. Fig. 11 shows the maximum linearity error of the
system (fluxgate sensor and interface circuit) normalized to
C. Sallen-Key Filter
The output of the demodulator contains the useful information at dc, but it also contains several spurs. Therefore, we
introduced a filter for removing any high-frequency unwanted
signal components. The filter implements a second order Butterworth transfer function with a cut-off frequency of 550 Hz
and a Q factor of 0.707, to guarantee a maximally flat response
in the base-band. Fig. 8 shows the schematic of the circuit.
Fig. 9.
Schematic of the 13-bit incremental ADC
25
20
15
Output Voltage [mV]
10
5
0
−5
−10
−15
Fig. 10.
−20
Microphotograph of the interface circuit chip
−25
−50
−37.5
−25
−17.2
−5.5
0 5.5
B [µT]
17.2
25
37.5
50
4
Fig. 12. Transfer characteristic of the system for a full-scale magnetic field
of 100 µT (±50 µT)
3.5
3
Linearity [%]
0.8
2.5
0.6
2
0.4
1.5
e [%]
0.2
1
0
0.5
11
22.2
34.4
50
75
100
B [T]
125
150
175
200
225
−0.2
Fig. 11. Maximum relative linearity error of the system as a function of the
full-scale magnetic field
−0.4
−0.6
0
2
4
6
8
10
12
14
Sample
the full-scale of the applied magnetic field as a function of the
full-scale magnetic field itself. In order to obtain this curve,
we applied the magnetic field with a couple of Helmholtz
coils. The axis of the Helmholtz coils has been oriented
perpendicular to the Earth magnetic field, to avoid undesired
contributions to the applied magnetic field. The maximum
linearity error degrades for large values of the full-scale
magnetic field, because of the saturation in the ferromagnetic
material of the considered fluxgate sensor, as expected. Fig. 12
shows the transfer characteristic of the system for a full-scale
magnetic field of 100 µT (±50 µT). Finally, Fig. 13 shows the
relative linearity error with the same full-scale magnetic field.
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