An X-Band Reconfigurable Bandpass Filter Using Phase Change
RF Switches
Muzhi Wang, Feng Lin, and Mina Rais-Zadeh
Department of Electrical Engineering and Computer Science, University of Michigan, Ann Arbor,
MI, 48109, USA
Abstract — In this paper, we report on a reconfigurable
bandpass filter for X-band applications. The filter is composed
of coupled λ/2 microstrip line resonators. Center frequency
tuning of each resonator is achieved using an RF switch based
on a phase change chalcogenide compound, germanium
telluride, which switches in and out a loading capacitor. A
combination of electric and magnetic coupling between the
resonators realizes a near constant absolute-bandwidth as the
filter is tuned. The center frequency of the filter is switched
between 7.45 and 8.07 GHz, with a 3-dB bandwidth of ~ 500
MHz, insertion loss of less than 3.2 dB, and return loss of
better than 18 dB. The measured third-order intermodulation
intercept point (IIP3) and 1-dB power compression point
(P1dB) are better than 30 dBm at frequency offset of 1 kHz and
25 dBm, respectively. Measured and simulated results are in
good agreement. To our best knowledge, this is the first time
implementation of a tunable filter using germanium telluride
based phase change switches.
Index Terms — Germanium telluride, phase change
materials, MEMS, switches, tunable filters, X band.
(a)
Zea, Zoa
Zeb, Zob, θ0
Port 1
Port 2
PC_switch
PC_switch
Ze1 ,Zo1, θ1
Zstub, θstub
Z2, θ2
I. INTRODUCTION
Compact and low-loss tunable filters are needed for a
number of applications including cognitive radios, modern
transceivers, and anti-jamming communication systems, to
name a few [1]. There have been several designs reported
for implementing tunable or reconfigurable filters. The
frequency tuning for such devices is achieved electronically
by
solid-state
varactors
or
switches,
RF
microelectromechanical systems (MEMS) switches or
capacitors, variable dielectric capacitors, or magnetically
using yttrium-iron-garnet (YIG). Solid state tuning
elements have relatively high losses at high frequencies
(>10 GHz) and linearity issues limiting their application
areas. MEMS electrostatic switches are proven to have high
power handling capability at the cost of large DC voltages
to hold them in the down state. Recently, RF switches using
phase change materials have gained a lot of attention as
they offer fast switching speed (<10 µs), large cut-off
frequency (> 1 THz), high OFF/ON resistance change ratio
(amorphous/crystalline resistivity ratio of > 104), small size
(in few µm2 range), near-zero in-state power consumption,
high power handling capability (P1dB > 25 dBm), and high
linearity (IIP3 > 50 dBm) [2-4]. However, the focus so far
has been on implementing the switching element and only
a handful of reports exist on exploiting phase change
switches in functional RF modules.
C
(b)
Port 1
Port 2
Z2, θ0
Port 1
Yin
Y0
C_stub
C_stub
C_stub
Zea, Zoa
Zeb, Zob, θ0
Z 2, θ 1
Ze1, Zo1, θ1
Z 2, θ 2
Z 2, θ 2
(c)
(d)
Fig. 1. Schematics of (a) the filter structure, (b) filter circuit model,
(c) its coupled-resonator section, and (d) circuit model of the
resonator with external coupling circuit.
In the X band, there are many examples of planar tunable
filters using MEMS switches [5], ferroelectric bariumstrontium-titanate (BST) capacitors [6], or vanadium
dioxide (VO2) switches [7]. As a proof of concept, in this
work, we utilize germanium telluride (GeTe) phase change
switches for the first time to realize a reconfigurable Xband filter.
II. TUNABLE BANDPASS FILTER DESIGN AND FABRICATION
The filter is designed for X-band military communication
applications with uplink and downlink frequency bands of
7.9 – 8.4 GHz and 7.25 – 7.75 GHz, respectively. The target
insertion loss of the filter is less than 4 dB. The filter
consists of two λ/2 microstrip line open-loop resonators
(i.e. no vias) each loaded with a switchable capacitor (opencircuited stubs) at the end. External coupling is achieved by
a coupled feed line (Fig. 1(a)).
passband. Since there are two signal paths: inductive
coupling between the resonator and capacitive source-load
coupling, two transmission zeros are introduced in the S21
response, enhancing the near band rejection of the filter.
A. Filter with Constant Absolute Bandwidth (CABW)
0.055
Yine , o  Y2
Yine  Yino
Yine  Yino
, Y12 
2
2
 , o  jY2 tan  0
Yine
 , o tan  0
Y2  jYine
 , o  Ye1, o1
Yine
jY2 tan  2  jYe1, o1 tan 1
.
(1a)
(1b)
Ye1, o1  Y2 tan 1 tan  2
To achieve a constant bandwidth, the Y-matrix of the circuit
in Fig. 1(c) should satisfy the following resonance and
coupling coefficient conditions across the entire tuning
range [8]:
Im Y11 0   0 ,
where b 
Im Y12 0 
b


g1 g 2
 k12 , (2)
0  Im Y11 0 
, ω0 is the resonant angular
2

center frequency. Using this configuration (Ze1=37Ω,
Zo1=23.3Ω, Z2=33Ω, θ0=55°, θ1=23°, θ2=95° and ∆=6.1% at
8.15 GHz), the required and calculated coupling co-efficient
k12 to achieve a constant bandwidth of 500 MHz at different
center frequencies of coupled resonators is plotted in Fig. 2.
In addition, to maintain constant absolute bandwidth, the
external quality factor Qext of the filter should satisfy (3) [8]
Qext 
b 0  Im Yin 0  g1 g 2
,


Y0 2Y0


Coupling coefficient (k12)
Calculated k12 for CABW of 500 MHz
0.050
0.045
0.040
0.035
7.0
7.2
7.4
7.6
7.8
8.0
Frequency (GHz)
8.2
Fig. 2. Desired and achieved coupling coefficient as a function of
frequency.
B. Phase Change Switch Design and Its Integration into the
Microstrip Filter
The structure of the GeTe phase change switch used in
the tunable filter is shown in Fig. 3 inset. Simulations of the
GeTe switch is performed using Agilent HFSS. The switch
consists of a single GeTe layer connecting two RF terminals
laterally. An embedded heater line runs below the GeTe
layer. The heater is electrically isolated from the RF signal
path by a silicon nitride (Si3N4) layer. The GeTe layer has
a thickness of approximately 150 nm, and the RF electrodes
have a width of 30 μm and a separation of 1.5 μm. The
GeTe layer can be thermally transitioned between
amorphous (OFF) and crystalline (ON) states by applying
proper current pules to the heater. Simulated S-parameters
of the phase change switch are presented in Fig. 3. In
simulations, the conductivity of GeTe at on state is assumed
to be 4.3×104 S/m with OFF/ON resistance ratio of 4.3×103.
These values are extracted from resistivity measurements.
0.0
-10
-0.2
Switch OFF
-0.4
(3)
where Yin is the input admittance of the circuit including the
external coupling element (see Fig. 1(d)). Capacitance C in
Fig. 1(b) represents a source-load coupling between the
feed lines and can be adjusted by changing the gap (g1)
between the feed lines to enhance the rejection close to the
8.4
-20
-30
-0.6
-40
Switch ON
-0.8
0
2
4
6
8
10 12 14
Frequency (GHz)
|S21| (dB)
Y11  jC_ stub 
Desired k12 for CABW of 500 MHz
|S21| (dB)
Fig. 1 (b) shows the equivalent-circuit model of the filter.
To design a two-pole 7.5/8.15 GHz filter with a 3-dB
absolute bandwidth of 500 MHz, the low-pass Butterworth
prototype is first designed with elements gi, i=0, 1, 2, 3.
Given the specifications including the center frequency and
the 3-dB fractional bandwidths (∆), the desired values of
external quality factors (Qext) and coupling coefficients (k12)
of the microstrip resonators are obtained (Equations (2) and
(3)). To calculate the coupling coefficients, the Y-matrix of
the two-port coupled-resonator section in Fig. 1(c) is firstly
derived as
16
18
-50
20
Fig. 3. Simulated S-parameter of the GeTe switch. Inset shows the
cross-section schematic of the switch.
When switches are ON, a lower frequency passband of 7.45
GHz (simulation: 7.5 GHz) is achieved with an insertion
loss of 3.2 dB, 3-dB bandwidth of 482 MHz, and return loss >
30 dB. When the switches are OFF, the filter is switched to
the higher frequency band of 8.07 GHz (simulation: 8.15
GHz) with insertion loss of 2.6 dB, bandwidth of 520 MHz,
and return loss >18 dB (Fig. 6). The Qu of microstrip
resonators with GeTe switches estimated using Eq. (4) [10]
is 59/73 in the ON/OFF states.
n
I.L (dB)  4.343
i 1
gi
.
FBWQui
(4)
0
OFF
ON
-10
|S| (dB)
The filter is built on a high-resistivity (ρ > 20 k.cm)
silicon substrate with εr = 11.7 and h = 0.5 mm. The filter
layout is optimized using the HFSS full-wave
electromagnetic simulation tool. The width of the
microstrip resonators (0.9 mm) is chosen to get a small
resonator impedance of 33 Ω and a high unloaded quality
factor (Qu). To achieve a value close to ideal for k12, the
distance (s) between the two resonators and the length (l2)
of the coupled line are optimized, while the required Qext is
realized by changing the width (w2) of the feed lines and the
gap (g0) between the feed lines and resonators. To facilitate
ground-signal-ground (GSG) probe measurement, two vialess microstrip to coplanar-waveguide (CPW) transitions
[9] are connected at the input and output RF ports. Final
dimensions of the filter are as followings: w0=0.4, w1=0.9,
w2=0.1, w3=0.55, l0=0.8, l1=1.42, l2=1, l3=2.7, l4=0.47,
g0=0.015, g1=0.077, s=0.057 (all in mm). Fig. 4 shows an
optical micrograph of a fabricated filter. The filter area is
10.8×3.9 mm2.
S11
S21
-20
-30
-40
EM simulated
Measured
-50
-60
6.0
6.5
7.0
7.5
8.0
8.5
Frequency (GHz)
9.0
9.5
10.0
Fig. 6. Measured and simulated S-parameters of the filter in both
switch states.
Fig. 4. An image of the fabricated phase change X-band filter.
C. Fabrication Process
Double-side polished high-resistivity silicon is chosen as
the substrate for its high dielectric constant, high thermal
conductivity, low substrate loss, and compatibility with
GeTe switch processes [2]. The process flow is shown in
Fig. 5.
(a) Deposition of
back side gold
layer
(b) Deposition of the
SiO2 passivation
layer
B. Linearity and Power Handling Measurement
The IIP3 is measured at the center frequency of the pass
band. The extracted IIP3 at 1 kHz offset is better than 30
dBm for both states (Fig. 7). Measured results also show
that the tunable filter can handle more than 25 dBm of input
power at both states of the switch before 1-dB compression
occurs (Fig. 8).
40
40
20
0
-20
-40
2f1-f2
-60
(c) Patterning of the
TiN heater layer
Output power (dBm)
Reference
plane
Output power (dBm)
Reference
plane
f1
5
10
0
-20
-40
2f1-f2
-60
-80
0
20
15
20
25
30
Input power (dBm)
35
40
f1
-80
-10 -5
0
5 10 15 20 25
Input power (dBm)
(a)
35
40
Fig. 7. Measured IIP3 at the (a) ON and (b) OFF states of the
switches.
(f) Microstrip line
patterning
Fig. 5. Fabrication process flow of the phase change X-band filter.
0
0
-10
-10
-20
-40
III. RESULTS
A. S-Parameter Response
The DC resistance of GeTe switches at the ON and OFF
states is measured to be ~ 5 Ω and 0.4 MΩ, respectively.
-50
-60
6.0
-20
-5 dBm
5 dBm
15 dBm
20 dBm
25 dBm
-30
6.5
7.0
7.5
8.0
8.5
Frequency (GHz)
-5 dBm
5 dBm
15 dBm
20 dBm
25 dBm
|S21| (dB)
(e) Patterning of
GeTe layer
|S21| (dB)
(d) Patterning of
the Si3N4 isolation
layer
30
(b)
-30
-40
-50
9.0
9.5
10.0
-60
6.0
6.5
7.0
7.5 8.0 8.5
Frequency (GHz)
9.0
9.5
10.0
(a)
(b)
Fig. 8. Measured S21 at varying input power levels when the
switches are (a) ON and (b) OFF.
TABLE I
PERFORMANCE COMPARISON OF THE REPORTED SECOND-ORDER TUNABLE FILTER
Ref.
[5]
[6]
[7]
This
work
Tuning
elements
MEMS
switch
BST
capacitor
VO2
switches
GeTe
switches
BW3dB
(% of fc)
5.8
15
IL
(dB)
2.62.9
2.02.7
Range
(GHz)
Estimated
Qu
Bias
voltage
(V)
Size
(mm2)
Tuning
Speed
Power
Consumption
IIP3
(dBm)
12-15
77-81
60
~5×4
~10 µs
Near 0
N/A
1010.56
< 40
30
3.1×6.9
N/A
N/A
N/A
12-13
5
8.6-9.2
< 20
60
9×7
<10µs
1.8 W (pulse)
N/A
6.4-6.5
2.63.2
7.458.07
59-73
15 to 20
3.9×
10.8
< 6 µs
0.5~1.5W
(pulse)
30
IV. CONCLUSION
A tunable bandpass filter using GeTe switches was
introduced for the first time. Table I compares the
performance specifications of this filter with reported filters
in the X band. As shown, the proposed filter offers
competitive performance. Compared to MEMS tunable
filters, it shows a faster response time and requires smaller
voltage pulses for actuation. Compared to ferroelectric
filters, it shows a small loss for a much higher filter Q and
finally, compared to VO2 type filters, it offers better
linearity and does not consume static power to stay in the
ON or OFF state.
ACKNOWLEDGEMENT
This work is supported by the defense advance research
project agency (DARPA) RF-FPGA program and by the
National Science Foundation (NSF) EARS and CAREER
programs.
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