A Class of Single-Step High-Voltage DC

Highlights of the Chapter 3

1. A buck converter is a voltage step-down circuit. It has a voltage source type of input and a current sink type of load. A buck converter is less efficient in DCM than in CCM operation. In large power applications, a CCM operation is preferred. The buck converter presents an almost self- load regulation when operated in CCM, but there is required a large variation range of the dutycycle for load regulation in DCM operation. The conduction power losses in the power stage increase with the load current, but the losses in the control circuit remain the same even when the converter works with a very light load, affecting the efficiency at the two ends of load range. An input filter is often necessary to cope with the otherwise pulsating input current. In CCM operation, the ripple current in the capacitor is small, being approximately equal with the ripple in the inductor current. To get a small output voltage ripple, a capacitor of small value is needed in the output circuit of a buck converter operated in CCM, but of rather large value if operated in

DCM. Neither the source, nor the gate of the MOSFET in the power stage are referenced to the ground, a driver circuit with additional circuitry being necessary. The buck converter has an inherent capability to cope with output fault conditions. At start-up, it is easy to limit the increase in duty-cycle to avoid any dangerous transient increase in the input current. A voltage-mode control is suitable for buck converters. It is easy to design the controller because the open-loop control-to-output transfer function contains no right-half plane zero. The value of the left-half plane zero compared with the switching frequency value will dictate the actual type of the feedback loop compensators for getting good unity gain bandwidth and large phase margin.

2. The boost converter can be seen as the dual of the buck converter. It is a voltage step-up circuit, exhibiting a current source type of input and a voltage sink type of output. It presents a non-pulsating input current, but a very large ripple current in the capacitor,

Power Electronics and Energy Conversion Systems: Volume 1. Fundamentals and Hard-switching Converters, First Edition.

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© 2013 John Wiley & Sons, Ltd. Published 2013 by John Wiley & Sons, Ltd.

approximately equal to the ripple in the diode current. For all the duration of the inductor charging, the capacitor alone has to supply the load with energy. This is why, a relatively much larger capacitor is needed in a boost converter than in a buck power stage. In

CCM operation, the graphic of the dc voltage gain versus duty-cycle shows that the output voltage decreases to zero after the dutycycle reaches a point where the dc voltage gain is maximum. The capacity of stepping up the input voltage many times by increasing substantially D is further hindered by the fact that the conduction losses in the power stage increase drastically with the duty-cycle.

The maximum duty-cycle at which a boost converter can be designed is limited. When designing a boost converter for DCM operation, the inductor will result in a small value, causing an increase in the maximum inductor current at the end of its charging process, implying more current stress in the transistor and diode, and more ripple current in the capacitor. These elements will have to be overdesigned in comparison with their counterparts intended to CCM operation. This is why, in relatively larger power applications, we prefer to operate a boost converter in CCM. However, some of the switching losses are smaller in DCM, because the switch turns-on with ZCS and the diode turns-off also with ZCS, with no reverse recovery current. The conduction losses in both CCM and DCM operation increase for larger load. The driver circuit of the transistor is simple because the gate is referenced to ground. The control open-loop transfer function presents a right-half plane zero, which gives difficulty in designing the closed loop with good bandwidth and good phase margin. The boost converter has poor ability to react to output fault conditions To avoid hazardous failures, it needs additional circuitry for protection in case of load short-circuit or no-load operation . Protection circuit to limit the input current at start up is also necessary. Current-mode control is more suitable for boost converters , as voltage-mode control would be too slow and will not offer necessary protection in limiting dangerous currents at input or load.


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3. A buck-boost converter supplies energy to a voltage sink type of load from a voltage source through a current source type of element (inductor). Depending on the value of the duty-cycle ,it can step-down or step-up the input voltage. The polarity of the load voltage is opposite to that of the input voltage. If a non-inverting polarity is required, a four-synchronous switch buck-boost converter can be considered. The buck-boost converter is the simplest topology to be used in applications like those powered by batteries where the required load voltage is within the range of the input voltage. The price paid by the multi-functionality of the buck-boost converter is that it takes the shortcomings of both the buck and boost converters : very pulsating input current , like in a buck power stage ,requiring an input filter in many applications. And, like in a boost converter, a very large capacitor ripple current yielding in a capacitor of larger value, and a right-half plane zero, requiring special caution in designing the controllers for assuring good margins, and thus closed-loop stability. Like in a boost converter, the maximum duty-cycle is limited : the output voltage inches to zero for very high values of the duty-cycle due to the resistive parasitic in the power stage , and the efficiency worsens seriously for large values of the duty-cycle. The voltage stresses across the switches are larger for a buck-boost converter than for a boost converter of similar input and output voltages , requiring transistors and diodes with larger voltage rating, therefore the buck-boost converter is more lossy. The efficiency decreases for large load both in CCM and DCM operation. It is lower in DCM operation, because the current stresses in the power stage elements are larger in DCM. However, similar to what happens in a boost converter in DCM, some of the switching losses are smaller in a buck-boost converter operated in DCM, because the switch turns-on with ZCS, and the diode turns-off also with

ZCS, with no reverse recovery current. In DCM operation, the duration of the second switching stage , D

2

T s

is independent of the value of the duty-cycle D, what is characteristic for a buck-boost power stage. Like in a buck converter, neither the source, nor the gate


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of the MOSFET are referenced to the ground. Some applications use a low-side switch for solving this problem, but then the output voltage is referenced to the input voltage, and not to the ground.

4. The Ćuk converter converts the energy from a current source to an output of current sink

characteristic, through a voltage type element ( a capacitor). Similarly with a buck-boost converter, the Ćuk converter is an inverting power stage, providing a load voltage of opposite

polarity to that of the input voltage, and offers both a step-down and step-up of the line voltage,

depending on the value of the duty-cycle. The step-up capability of the converter is limited due

to the parasitic resistances in the power stage, but the dc voltage conversion gain is slightly

larger at the same value of the duty-cycle than that of a buck-boost converter. The main advantage of the Ćuk converter is it non-pulsating input and output currents. But it comes to a cost : four passive elements ,two inductors and two capacitors, are needed in the power stage.

They form two equivalent low-pass filter cells, that in some conditions can be decoupled one from the other. . In order to find the formula of the dc gain, we have to write two equations representing a volt-second balance on each inductor. A current equal to the sum of the input and output currents flows through each switch. Both switches are subjected to a large voltage. The ripple current in the energy transferring capacitor is very large, the current ripple in the output capacitor is very small. The open-loop small-signal transfer functions are of fourth order. The


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parasitic dc resistances of the two inductors assure that the zeros of the open-loop control-to- output transfer function are situated in the left-half plane, causing a minimum-phase dynamic response, like that of a buck converter. The open-loop line-to-output transfer function presents two zeros , a left-half plane one due to the equivalent series resistance of the output capacitor and another one of very high frequency due to parasitic resistances of the transistor and diode , which can also be situated in the left-half plane, depending on the actual values of the parasitic resistances associated with the transistor. The driver circuit of the transistor is simple because the gate is referenced to ground. The converter can operate in CCM, or DCVM ( discontinuous capacitor voltage mode) ,or DICM (discontinuous inductor current mode). In DCVM ,the voltage on the energy-transferring capacitor drops to zero during the transistor conduction, creating a third switching topology in which both switches conduct. In DICM, the sum of the inductor currents drops to zero during the conduction of the diode, resulting in a switching topology in which both the switches are in off-state. Only in a particular case ,DICM operation implies a discontinuity in the conduction of the two inductors. For a certain converter supplying a given load, DICM occurs at relatively low duty-cycle and DCVM for relatively high duty-cycle values.

For a designed converter supplying a variable load, DCVM can appear at high load, when

DICM can appear at light load. A Ćuk converter is purposely designed for a DCVM by choosing


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a sufficient small value of the energy-transferring capacitor. It can be purposely designed for a

DICM by choosing a sufficient small value to an equivalent inductance of the two inductors. The voltage across the energy-transfer capacitor in DCVM is high, yielding a large voltage stress on each switch (double than that in a CCM operation ,and higher than in a DICM) ; the current stress in each switch is low, similar to that in a CCM operation. The voltage stress of each switch is low in DICM, similar to that in a CCM operation, but the switches have to be rated for a higher current. Operation in DCVM is favorable in low-voltage high-current applications, when

operation in DICM is preferred for low current, high voltage applications. A Ćuk converter in discontinuous conduction mode is favorite in power factor correction applications. By using a coupled-inductor in the structure of a Ćuk converter , smaller ripples in the input and/or output currents can be obtained , even up to the point that the ripple in one of them is reduced to zero.

It also exists an isolated version of the Ćuk converter, which allows both dc-dc isolation and multiple-outputs, when keeping the non-pulsating character of both the input and output currents.

The magnetizing current of the isolation transformer takes both positive and negative values,

what is an advantage compared to the forward converter. If the forward converter makes use of two diodes, the Ćuk converter contains only one rectifier diode, but with a larger current rating.

The isolated Ćuk converter is not suitable for high current, low voltage applications because of


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the series capacitor in the secondary side high current path. An integrated magnetic structure of the isolated Ćuk converter can be obtained by wounding the input and output inductors on the transformer core.

5. The SEPIC converter is current-driven , it converts the energy from a current type of source to an output with the characteristic of a voltage sink . Similarly with the buck-boost converter and the Ćuk converter , it offers a capability of both stepping- down and stepping-up the input voltage, However, unlike the former converters, the SEPIC converter does not invert the voltage : the polarity of the load voltage is kept the same as that of the input voltage. This renders the SEPIC converter as the ideal regulator for applications in which the required load voltage is within the range of the input voltage. But this advantage comes with its price : the SEPIC converter has an output part of boost type, i.e. it presents a very pulsating output current, requiring a much larger output capacitor than that needed in a Ćuk converter, and features a non-minimum-phase transient response. Indeed, the small-signal control-to-output transfer function in CCM operation presents one real right-half plane zero like a boost converter, due to the fact that in the on-topology, the inductor L

1

in a charging process from V in

is separated from the load, and two more complex right-half plane zeros , due to the fact that the inductor L

2

,in a charging process from C

1

, is also disconnected from the load in the on- switching topology . Capacitor C

1 in a

SEPIC converter is submitted to the input voltage in both CCM and DCM operation, i.e. at a lower voltage than that to which is submitted C

1 in a Ćuk converter. The SEPIC converter also has two types of discontinuous conduction modes : DCVM and DICM .

The main characteristics of the SEPIC and Ćuk converters are the same in DCM: Table 3.2.


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Table 3.2

Main characteristics of Ćuk and SEPIC converters in DCM

DCVM k

DCVM



2 RC

1 f s

DICM k

DICM



2 L eq

R f s

M

DICM



V out

V in



I in

I out



D k

DICM

M

DCVM



I in

I out



V out

V in

 k

DCVM

1



D

R in , eq



( 1



D )

2

2 C

1 f s usually for high duty-cycle usually at heavy load lower switch current stress larger voltage stress application: low voltage, high current

R in , eq



2 L eq f s

D

2 usually for low duty-cycle usually at light load larger switch current stress lower voltage stress application : high voltage, low current

Both the SEPIC and Ćuk converter in DCM operation are very suitable in power factor regulators ,because of their low content of harmonics in the input current , their input current following naturally the sinusoidal input voltage. Both of them, in DCM operation,


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are also very suitable as regulators after solar cells ( or wind sources of energy), because they easy allow for extracting maximum power from solar panels. By adjusting the duty cycle until the input resistance of the converter becomes equal to the equivalent resistance of the photovoltaic panel , the maximum power transfer is assured.

6. The Zeta converter is also a non-inverting dc-dc converter able of decreasing or increasing the input voltage to the desired regulated output voltage by simply varying the duty-cycle. Unless the SEPIC converter that performs the same function, the Zeta converter features a current-sink output characteristic. So, the Zeta converter is the ideal power supply for applications requiring a constant input current, like battery chargers ,or as LED lamps supply. The non-pulsating output current of the Zeta converter allows for the use of small output capacitors for satisfying the load voltage ripple requirement. The very pulsating input current requires the use of an input capacitor. The switch has neither its gate, nor its source referenced to ground. In practice, either a p-channel type of

MOSFET or a n-channel MOSFET with bootstrapped circuit are used. By replacing the usual Schottky diode with a synchronous switch, we can increase the efficiency and reduce the circuit footprint. The use of a coupled inductor instead of two separate inductors helps in reducing the inductor currents ripple and improving the transient response, by allowing a setting of the crossover frequency in the closed-loop design at a higher value. The small-signal open-loop control-to-output transfer function does not present a right-half zero , what it is different from the case of SEPIC converters. This allows for a simpler compensation circuit in the feedback loop, achieving a wider loop bandwidth, and so a better transient response with a smaller output capacitance. The Zeta converter, similar to the SEPIC converter, will enter DICM at light load, or at small values of the duty-cycle. And, the duration of the second switching stage, D

2

T s

will also be independent of the value of the duty-ratio D .


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The switches voltage and current stresses, and the voltage across the energy-transferring capacitor in the converters Ćuk , SEPIC and

Zeta are given in Table 3.3 :

Table 3.3 Voltage and current stresses in Ćuk , SEPIC and Zeta converters

Ćuk converter

SEPIC converter

Zeta converter

V

V in

V

V

Smax in in

+ V out

+

+

,

V

V

V

Dmax out out

I

I

I

Smax

I

L 1 max

L

L

1

1 max max

, I

+ I

L 2 max

+

+

Dmax

I

I

L

L

2

2 max max

V

V

V

C in

1 out

V in

+ V out

7. The forward converter is a buck converter that has incorporated a high-frequency transformer. It is an isolated converter, featuring dc isolation between the input source and the load. Isolation is required in many industrial applications, for example in off-line utilities, where the converter input is connected to the rectified ac voltage of the power grid .Or ,when several power supplies in a system are tied together , isolation can eliminate ground loops and noise interference. The benefits of isolation come with several penalties in the operation of the forward converter compared to a buck converter. Firstly, the transformer induces more losses, in the leakage inductance and core .Secondly, for resetting the core in each switching cycle, a tertiary winding has to be added to the transformer. It allows the return of the magnetizing energy to the source, avoiding its dissipation as heat. The need of giving time for the purpose of discharging the magnetizing inductance limits heavily the range of the duty-cycle, and thus the line and load regulation capability of the converter. During the discharging of the magnetizing inductance, additional voltage stress is imposed on the switch. There is a


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trade-off between increasing the range of the duty-cycle and increasing the voltage stress on the transistor. At the best, we can reach a compromise of limiting the duty-cycle just under the value 0.5 and having a voltage stress on the transistor of 2 V in

( double than that met in a buck converter). The reset circuit also contains an additional diode which is submitted to a large voltage stress (2 V in

in the case that the number of turns of the tertiary winding equals the number of turns of the primary winding). And we have to trade-off between the transistor voltage stress and its current stress. Even if we optimize the design at the low-end range of the input voltage

( we calculate the maximum duty-cycle and consequently the input-to-output turns ratio at this value), the converter will operate inefficiently at the high-end of the line voltage range. Alternatively, a two-transistor solution can be used, in which case the duty-cycle can be maximum 0.5, but the switches are submitted only to V in

.Would the core not reset, the current through the magnetizing inductance will increase from a cycle to another one, ending in saturating the transformer. For proper operation, the duty-cycle has to be prevented of surpassing its maximum designed limit, even if the need of line or load regulation would require a larger value. This is why we design it for the worst-case operation conditions. Otherwise, the core will not be reset. In order to operate the converter at higher than 0.5 duty-cycle values, without a very large voltage overstress on the switch, dissipative ( RCD ) or non-dissipative active clamping circuits , or non-dissipative resonant circuits can be used. The RCD clamping solution is applicable in low input voltage applications. The resonant clamping circuit can be incorporated either in the primary side or secondary side of the converter. The active clamping solution requires an additional active switch, with its driver and control, but it offers the advantage of almost constant switch voltage stress over the whole range of the line voltage. Thirdly, the ringing due to the leakage inductance adds more voltage stress on the switch. Fourthly , one more diode (without counting the clamping circuit) is required in a forward converter compared


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with the buck converter, reducing further its efficiency. Despite these disadvantages, the forward converter is frequently used in applications requiring isolation in the power range from 50 W to 500 W . Unlike in a buck converter, the transistor in a forward converter can be moved to an advantageous position, like in a boost converter, allowing for a simple driver. The design of forward converter is similar to that of a buck converter .After choosing the transformer for maximum allowable duty-cycle, we calculate its turns ratio, the result is rounded and used in the final calculation of the range of the duty-cycle .Then, most design steps are identical to those for the buck converter for both CCM and DCM operation. As in any converter, the design of the output capacitor has to also take into account the hold-up time requirement for step-load response The incorporation of a transformer in the converter structure allows for realization of multi-outputs. However , the regulation of multiple outputs ( “cross-regulation”) is a complex matter .The recent versions of the forward converter use MOSFET synchronous rectifiers , and operate at high switching frequencies, usually up to

500 KHz , or more if soft-switching is also implemented. The presence of a synchronous rectifier prevents the converter from operating in DCM.

8. The flyback converter is a buck-boost converter that has incorporated a coupled-inductor for getting dc isolation between the line voltage and load. As the role of the coupled-inductor is to store energy in the first switching stage, its core has to be realized with an air gap, to enhance the energy-storing capability. The air gap allows the coupled-inductor to store more energy before the core gets saturated. No core reset mechanism is necessary, as it was the case in forward converters, because the energy stored in the magnetizing inductance is naturally transferred to the load in the second switching stage. The flyback converter is the simplest structure among all the isolated converters , containing the lowest parts count. As no output inductor is necessary, a multiple-output version is realized in a


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very economic way, with only one capacitor and diode in the rectifier circuit of each one of the multiple secondary windings. The transistor is connected with its source terminal to the primary-side ground, so a simple driving circuit can be used. The flyback converter is often used in battery-powered and off-line low power applications, up to 200 W. It is not suitable for a higher power level, as it has a poor efficiency. Like the buck-boost converter, it presents large input and output currents ripple . The output capacitor has to be large to filter the output ripple current. For storing energy, the air gap has to be large. At higher power, the air gap would become very large , i.e. the magnetizing inductance would have to be very small for storing more energy, increasing very much the primary current peak, limiting further the maximum power level of the flyback converter applications. If designed to operate in DCM, the magnetizing inductance will be smaller, and the converter will be able to handle more power. Or, at the same output power, the core will be smaller , as the energy storage is maximized in DCM operation. But the penalty is higher magnetizing current peak, implying also more current stress for the transistor and diode. There is a trade-off in the design of L m

between larger power handling capability and more current stress. The rectifier diode current ripple, and consequently the output capacitor ripple current is larger in DCM operation than in CCM .The output capacitor will be larger in a DCM design. A larger magnetizing current ripple also yields more core loss. In DCM operation, the core loss rather than the core saturation gives the design limits. Like for a boost or buck-boost converter, the closed-loop design is more difficult for a CCM operation ,due to the right-half plane zero in the open-loop small-signal control-to-output voltage transfer function. This zero occurs at a large frequency for a converter designed for DCM operation, affecting much less a stable closed-loop design. In DCM design, the feedback loop can be designed for stability at a wide bandwidth. As in

DCM operation the duty-cycle varies with both the input voltage and load, a wider duty-cycle range is required for a DCM design than


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a CCM design. Typically for a flyback converter, the magnetizing current has only one direction, i.e. the core is energized in a single direction. In DCM , there is the third switching interval when the core is not energized. The dc voltage gain is proportional to the duty-cycle in DCM . At a constant value of the load, the duration of the second switching stage is independent of the duty-cycle, depending only on the value of the reflected-to-secondary magnetizing inductance, load resistance , and switching frequency ( the duty-cycle can change following a disturbance in the line voltage, even if the load remains unchanged). Similar to the buck-boost,

Ćuk and SEPIC converters, when operated in DCM, the flyback converter emulates an input impedance of resistive type , i.e. its input current follows the input voltage naturally, without the need of an additionally control loop. As a result, it behaves like an ideal power factor corrector. Due to its low elements count, dc isolation and multiple output possibility, and despite its pulsating input and output currents, and more core loss, the flyback converter operated in DCM is often used in low-power ac-dc rectification. Mitigating its simplicity, the flyback converter presents an important disadvantage : the non-negligible leakage inductance of its coupled-inductor.

This stray inductance causes voltage spike and ringing with the switch output parasitic capacitance at the turn-off of the transistor. The resulting voltage overstress across the switch requires the use of higher voltage rating transistors, with higher conduction losses.

Different methods, like the dissipative RCD snubber can alleviate the problem, but the additional loss of energy caused by the leakage inductance limits further the use of the flyback converter at higher powers. Use of clamping circuits for recycling the leakage inductance energy allows for the use of the flyback converter up to powers of 500 W level. A flyback converter should always be operated with a load, otherwise, in the second switching stage, the coupled-inductor energy which is transferred to the output may lead to the breakdown of the output capacitor.


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9. For larger power applications, isolated converters, containing transformers for which the entire core B – H loop is used , are preferred. These transformers present a higher power density. The push-pull converter appears in two variants : voltage-driven ( buck characteristic) and current-driven ( boost characteristic). The first one can be seen as a combination of two forward converters operated in anti-phase. Both the transistors are connected with the gate referenced to the ground, allowing for a simple driver circuit.

Never the two transistors are turned on simultaneously in order to avoid a “shoot-through” current spike .They are submitted to a voltage stress equal to two times the input voltage. The core magnetizing current takes both positive and negative values during a switching cycle. Even if the magnetizing current is not reset to zero in each cycle, there is no problem , because its average value is zero over a switching period due to the balanced bidirectional flux . However, in practical transformers, the upper and lower windings are not perfectly similar. Flux imbalances can occur, generating a dc component in the magnetizing current, that can saturate the core.

A current-mode control can deal with this problem. The magnetizing current is small because the transformer is built just for transferring energy, i.e. with a large magnetizing inductance. The push-pull converter operates with four switching topologies per cycle , with half-cycle symmetry , i.e. in the first half-cycle one of the forward circuits acts, and in the second half-cycle, the second forward circuit acts. The duty-cycle is limited to values lower than 0.5. The push-pull converter can operate either in CCM or in DCM , the characteristics of the two operation modes being the same as those discussed for the buck and forward converters. The currentdriven converter has the inductor inserted in series with the input source, yielding in a boost type of voltage gain. The input inductor is charged in the switching stages in which both primary-side switches are turned on. This converter presents a non-pulsating input current. However, the output diodes are submitted to two times the load voltage. For more accuracy in the design phase, the conduction


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losses on the primary-side switches and rectifier diodes can be accounted for when writing the volt-second balance on the converter’s inductor.

10. The half-bridge converter can be seen as composed of two forward converters that operate alternatively, the active switch in each forward component having a different position with reference to the input source. The lower-side transistor is connected with the gate referenced to ground, allowing for the use of a simple driver. However the upper-side transistor is connected with neither the source, nor the gate referenced to the ground, requiring a more complicated driver. Both the primary-side transistors are subjected to a low voltage stress equal to the input voltage, but they have to handle an rms current that is double that to be carried by the active switches in a push-pull converter with the same input current. Thus, the half-bridge converter , with its low count of elements, is often used in off-line applications of relatively high voltage and medium power levels, for which MOSFETs with sufficient voltage and current rating are readily available at low cost. The half-bridge converter presents a good utilization of the core and primary winding of its transformer. In a center-tapped rectifier topology, the center-tapped secondary winding is not well utilized. Even, in the freewheeling stages, when no energy is transferred through the transformer, there is a load current circulation through the secondary winding ,that causes conduction loss. The maximum duty-cycle of each transistor has to be slightly lower than 0.5, to avoid cross-conduction of the primary-side switches ,which would cause a shoot-through the input source. As also it is the case with other converters using synchronous rectifiers, a pre-bias soft-start feature has to be added to the regulator if a pre-bias condition is likely to occur , what is typically if the converter makes part of a system of redundant ( parallel) supply modules. For example, Texas Instruments provides the

PWM controller UCC28250 ,with 1 MHz capable switching frequency, to be used in half-bridge, full-bridge, push-pull or interleaved


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forward converters, one of its features being pre-bias soft start. The presence of the large capacitors C

1

and C

2

in the primary-side circuit prevents the circulation of a dc component of the magnetizing current. This is why, a simple voltage-mode control is not excluded for half-bridge converters. Oscillations between the transformer leakage inductance and rectifier diode parasitic capacitance at the diodes turn-off occur, soft-switching methods can avoid voltage overstresses due to such ringing. Even if we design the output inductor such that to discharge completely to the load in each half-cycle, pointing to a DCM operation, strictly speaking the converter will not enter DCM until the stored energy in the magnetizing inductance was not completely discharged to the load. And if the actual value of the load does not allow for a complete discharge of the magnetizing inductance until the end of each half-cycle, it means that the current through the output inductor does not drop to zero, i.e. the converter will not operate in a true DCM. The current-driven halfbridge converter is obtained as the dual of the buck-type half-bridge converter. It is a boost-derived converter : the primary inductors are charged when both primary-side switches are turned on; the line-to-load energy transfer through the transformer takes place in the switching stages when one of the primary-side transistors is turned off. The leakage inductance energy is naturally transferred to the load. This converter is used in voltage step-up applications with low-input voltage high-input current ,because the voltage stress on the switches is higher than that in the voltage-driven half-bridge power supply, but the rms currents through the primary-side transistors and the current through the primary winding have lower values. The ripple in the input current is low, i.e. this converter absorbs a non-pulsating current from the source. This is why, the current-driven half-bridge converter is a good candidate to be used as an interface between alternative energy cells and their load. .


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11. The full-bridge converter offers the highest power levels among the converters studied up to now. It is used for medium-to-high power levels, from 750 W up to 5 kW, very often in off-line applications. It presents a combination of minimum switch voltage stress

( similar to a half-bridge power supply) and minimum switch current stress ( similar to a push-pull converter). It offers an output voltage double that of a half-bridge converter. However, the full-bridge converter presents the highest count of elements ( switches and their drivers) compared to the half-bridge and push-pull topologies. And its upper transistors are referenced with neither the source, nor the gate to ground. The primary-side switches can be implemented with MOSFETs, or, at large voltage values, with IGBTs or modern thyristors. The two switches pairs of the primary-side circuit are operated with equal conduction periods, such that the converter presents half-cycle symmetry in its operation. As a result, the entire core B – H loop is used, providing a good utilization of the transformer core. For the voltage-driven ( buck-type) full-bridge converter, the maximum duty-cycle of each transistor has to be slightly lower than 0.5, to avoid cross-conduction of the primary-side switches .The rectifier circuit can be realized either in a fullwave center-tapped topology , when the load current in the energy-transfer switching stages flows through only one diode , or in a full-wave diode bridge circuit, when the load current flows through two diodes in series in the energy-transfer switching topologies .

The diodes in the bridge type of rectifier are subjected to half of the voltage stress that the diodes in the former type of rectifier are submitted to. There are almost no conduction losses in the secondary winding of the bridge-type of rectifier during the freewheeling stages. The transformer core and the primary winding are well utilized in both variants of the rectifier circuit. The bridge type of rectifier is used in large voltage applications, when the center-tapped rectifier is preferred in applications with a low output voltage.

The energy stored in the leakage inductance is returned to the supply source at the beginning of the freewheeling stages. Oscillations


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between the leakage inductance and the parasitic capacitances of the rectifier diodes at their turn-off can be avoided with modern softswitching solutions. If the secondary diodes are implemented by synchronous rectifiers, a pre-bias soft-start feature has to be added to the regulator if a pre-bias condition is likely to occur. A current-mode control can prevent imbalances due to the fact that the transistors used in the primary-side switch pairs are never identical. Additionally, the current-mode control provides start-up and over-load protection, by limiting the inrush or the overload currents. In practice, conditions for the existence of a DCM operation rarely occur, as the full-bridge converters are designed for a higher level of power and operated at high switching frequencies. A phase-shift control rather than a duty-cycle control can be used : each one of the primary-side switches conducts for half a cycle. The conduction time of the lower switches is restricted to slightly less than half a cycle, to avoid cross-conduction. Each switch belonging to the right-leg turns on/off with a delay φ with respect to the left-leg switch on its diagonal. The energy is transferred from line to load through the transformer in the switching stages in which the primary-side switches on one of the diagonals conduct, and the switches on the other diagonal are turned off. The freewheeling stages occur when either the upper-side or the lower-side switches of the two legs are turned off. The phase-shift control proves to be the key for developing the modern ZVS full-bridge converters, allowing their use to power levels of 5 kW. The full-bridge converters come in two variants : voltage driven ( buck type) and current-driven ( boost-type). In current-driven full-bridge converters, the input inductor is charged in the switching stages in which all the primary-side transistors are turned on. . The energy is transferred from line to load through the transformer in the switching stages in which the primary-side switches on one of the diagonals conduct, and the switches on the other diagonal are turned off. The presence of the input inductor gives a non-pulsating character to the input current. The diodes in the rectifier circuit are submitted to the large output voltage.


Adrian Ioinovici.


Imbalances in the voltages applied across the primary winding in the two energy-transfer stages due to the fact that the transistors used as primary-side switches are not perfect identical are not dangerous : the dc component of the magnetizing current which is the result of such imbalances is limited by the input inductor.

The characteristics of the buck-type push-pull, half-bridge and full-bridge converters are summarized as in Table 3.4 :

Table 3.4 Characteristics of the buck-type push-pull, half-bridge and full-bridge converters

Push-pull

V

V out in

V

DS max

Number primary switches

N

2 D

N p s

2 V in

2

Halfbridge

Fullbridge

N

D s

V

N in p

N s

2 D

N p

V in

2

4

D max

< 0.5

< 0.5

< 0.5

Switches referenced to ground both

1 (the lower one)

2 (the lower ones) switch stress low high low cr.

The characteristics of the center-tapped rectifier and bridge type rectifier for the full-bridge converter are summarized in Table 3.5

(The voltage stress V

D

on the rectifier diodes in the half-bridge converter is half of the following values ) :


Adrian Ioinovici.


Table 3.5 C haracteristics of the center-tapped and bridge type rectifier of full-bridge converter.

Utilization of secondary winding

No of diodes

(synchronous rectifiers)

Voltage drop on diodes in energy transfer stage

V

D

Center-tapped bad 2 V

F

Load voltage suitability

N s

2

N p

V in low voltage

Bridge type good 4 2 V

F

N

N p s V in high voltage


Adrian Ioinovici.


A Class of Single-Step High-Voltage DC

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