UPTEC F 15013
Examensarbete 30 hp
Maj 2015
Performance evaluation of IQ-modulator ADL5375
at 5.8 GHz and its effect on transmitter
performance in a telecommunications system
Alexander Bergslilja
Abstract
Performance evaluation of IQ-modulator ADL5375 at
5.8 GHz and its effect on transmitter performance in
a telecommunications system
Alexander Bergslilja
Teknisk- naturvetenskaplig fakultet
UTH-enheten
Besöksadress:
Ångströmlaboratoriet
Lägerhyddsvägen 1
Hus 4, Plan 0
Postadress:
Box 536
751 21 Uppsala
Telefon:
018 – 471 30 03
Telefax:
018 – 471 30 00
Hemsida:
http://www.teknat.uu.se/student
Because of the tough competition in
the telecom business there is a
constant push for higher capacity and
data rates and the companies producing
the telecommunications equipment need
more cost effective products to stay
ahead of competitors. It is therefore
interesting to evaluate the
possibilities to use unlicensed
frequency bands at higher frequencies
as a complement to the traditional
lower frequency bands. This study is
focusing on the 5.8 GHz band, which is
mainly used for WLAN applications. A
key component in most transmitter (TX)
designs is is the quadrature
modulator, which upconverts the
information signal to desired carrier
frequency. In this study an attempt to
evaluate the commercially available
quadrature modulator ADL5375 at 5.8
GHz. An AWR Visual System Simulator
(VSS) model based on measurements of
key parameters of ADL5375 is
constructed. An attempt is made to see
whether a TX design can pass the
specifications set by 3rd Generation
Partnership Project (3GPP) for the
Long Term Evolution (LTE) standard. To
test this an LTE signal source was
also constructed. No certain
conclusions can be drawn without
putting the modulator in a complete
(TX) design but the results indicate
that it might be possible to use it in
a (TX) design for the 5.8 GHz band.
Handledare: David Scafe
Ämnesgranskare: Uwe Zimmermann
Examinator: Tomas Nyberg
ISSN: 1401-5757, UPTEC F15 013
Populärvetenskaplig sammanfattning
På grund av den hårda konkurrensen på telekom-marknaden finns det
ett hårt tryck för att öka capaciteten (antal användare telekomnäten kan
hantera) och datahastigheten i dagens nät. Detta innebär att man behöver
mer bandbredd. De frekvensband som idag är licensierade för telekommunikation börjar bli överpopulerade, vilket gör det svårt att höja prestandan
på näten. Detta gör att man i branschen tittar på olika sätt att utöka
frekvensspektrumen som används för 3G och LTE. Ett sätt att utöka är
att använda det olicensierade spektrumet på 5.8 GHz och uppåt. Men om
frekvensen i systemen ökas så sjunker prestandan. Detta ställer högre krav
på designen av dessa system och det blir svårare att klara av de designspecifikationer som utfärdas av standardiseringsorganisationer som t.ex. 3GPP.
Den här rapporten siktar på att testa en delkomponent i en sändarkrets,
IQ-modulatorn. I all radiokommunikation så konverteras den analoga signalen som innehåller informationen man vill överföra upp till en bärvågsfrekvens (fc ). Den uppgiften utförs av IQ-modulatorn. IQ-modulatorn
består internt av någon form av icke-linjära elektriska komponenter, som
dioder eller transistorer. De icke-linjära egenskaperna används för att få
den önskade uppkonverteringen till bärvågsfrekvensen. För att se om en
sändare kan byggas med kommersiellt tillgängliga komponenter så testades
i denna studie IQ-modulatorn ADL5375 från Analog Devices. Den är gjord
för att användas inom frekvensspannet 400 MHz – 6 GHz.
För att undersöka hur ADL5375 presterar på 5.8 GHz och kunna utvärdera
huruvida den är lämplig i en sändadesign behöver den testas med bredbandiga signaler av typen som används när exempelvis LTE-standarden
används. För att göra detta möjligt utan att designa en hel sändare skapades istället en simuleringsmodell av ADL5375 i simuleringsprogrammet
AWR – Visual System Simulator. Simuleringsmodellen baserades på mätningar av vissa nyckelparametrar som är typiska för modulatorer.
Specifikationerna som sätts upp av 3GPP för basstationssändare i LTEsystem utgår från en komplett sändardesign. För att då kunna utvärdera
huruvida ADL5375 klarar av kraven som ställs behöver den sättas i en komplett sändardesign. På så sätt kan dess effekt på sändaren utvärderas. För
att möjliggöra bredbandig signalgenerering så skapades utöver en modell
av ADL5375 även en LTE-signalkälla.
Resultaten av projektet är en simuleringsmodell av ADL5375. Utöver
det skapades även en LTE-signalkälla att använda för kompletta sändardesignsimuleringar. Resultaten av projektet ger inga entydiga svar på huruvida ADL5375 lämpar sig för 5.8GHz-bandet, men under vissa antaganden
kan det vara möjligt.
För att kunna dra några vidare slutsatser behöver en mer komplett
simuleringsmodell utvecklas som tar hänsyn till alla steg i en sändardesign.
i
Foreword
I would like to thank my supervisor, David Scafe, for his support and his neverfading smile. A big thank you goes to the Ericsson TX design department
and especially Wojciech Mudyna for giving me the opportunity to work with
this project. An extra thank you goes to Erik Vedin, Theodor Berg, Jimmy
Andersson and Mathias Augustsson for their patience and willingness to answer
all my questions.
I would also like to thank Andreas Tenggren on Keysight for the oppurtunity to
borrow measurement equipment.
Finally I would like to thank my wife Eva for her endless support and for always
picking me up when I’m down.
ii
Contents
Foreword . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1 Introduction
ii
1
1.1
Background . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1
1.2
Purpose . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
2
2 Theory
3
2.1
Modelling Transmit Signals . . . . . . . . . . . . . . . . . . . . .
3
2.2
General Transmitter Design . . . . . . . . . . . . . . . . . . . . .
4
2.3
Upconversion . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
5
2.3.1
Mixer . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
5
2.3.2
Upconverting Using a Mixer . . . . . . . . . . . . . . . . .
6
2.4
Quadrature Modulator . . . . . . . . . . . . . . . . . . . . . . . .
7
2.5
Modulator Characteristic parameters . . . . . . . . . . . . . . . .
8
2.5.1
LO Leakage . . . . . . . . . . . . . . . . . . . . . . . . . .
8
2.5.2
Sideband Suppression . . . . . . . . . . . . . . . . . . . .
9
2.5.3
Conversion Gain . . . . . . . . . . . . . . . . . . . . . . .
9
2.5.4
1 dB Compression Point . . . . . . . . . . . . . . . . . . .
9
2.5.5
Second and Third Order Intercept Points . . . . . . . . .
9
3 Hardware
3.1
12
Evaluation Board . . . . . . . . . . . . . . . . . . . . . . . . . . .
4 Simulation
4.1
4.2
12
13
AWR’s Visual System Simulator . . . . . . . . . . . . . . . . . .
13
4.1.1
Simulation of ADL5375 in VSS . . . . . . . . . . . . . . .
13
4.1.2
Wideband Signal Source . . . . . . . . . . . . . . . . . . .
15
Simulate Wideband Behaviour . . . . . . . . . . . . . . . . . . .
16
5 Measurements
18
iii
5.1
Necessary Measurements . . . . . . . . . . . . . . . . . . . . . . .
18
5.2
Measurement equipment . . . . . . . . . . . . . . . . . . . . . . .
19
5.2.1
DAC as stimulus . . . . . . . . . . . . . . . . . . . . . . .
19
5.2.2
81160A and N5242A PNA-X . . . . . . . . . . . . . . . .
23
5.2.3
Calibration . . . . . . . . . . . . . . . . . . . . . . . . . .
25
6 Results and Conclusions
6.1
27
Measurement Results . . . . . . . . . . . . . . . . . . . . . . . . .
27
6.1.1
CW Tone Measurements. . . . . . . . . . . . . . . . . . .
27
6.1.2
IM Measurements . . . . . . . . . . . . . . . . . . . . . .
29
6.1.3
Simulation Model . . . . . . . . . . . . . . . . . . . . . . .
30
6.1.4
ACLR Measurements . . . . . . . . . . . . . . . . . . . .
31
6.2
Discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
34
6.3
Summary of Results . . . . . . . . . . . . . . . . . . . . . . . . .
36
6.4
Future Work . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
36
A Appendix
38
A.1 Plots . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
iv
38
Glossary
IP2 Second Order Intercept Point. 12, 13, 25, 26, 31, 36
IP3 Third Order Intercept Point. 11–13, 25, 26, 31, 36
3GPP 3rd Generation Partnership Project. 2, 17, 36
ACLR Adjacent Channel Rejection Ratio. 17, 18, 32–34, 36
BB baseband. 5, 6, 8, 19, 20, 22–25, 28, 31, 32, 35, 37
BPF bandpass filter. 5
BS Base Station. 18
CE complex envelope. 4, 14
CW continuous wave. 19, 22, 32
DAC Digital-to-Analog Converter. 5, 6, 20, 21, 37
DDS Direct Digital Synthesis. 8
DPD Digital Pre-distortion. 36
E-UTRA Evolved Universal Terrestrial Radio Access. 17
EVM Error Vector Magnitude. 18
GPIB General Purpose Interface Bus. 23, 25
IF intermediate frequency. 5, 6, 8, 17, 32
IM intermodulation. 12, 26
LO local oscillator. 6–9, 13–16, 19, 24, 26, 27, 35, 36
LPF lowpass filter. 5
LSB lower sideband. 6, 19, 25
LTE Long Term Evolution. 2, 3, 17, 18, 32, 34, 36
NA network analyser. 20, 24
P1dB 1 dB Compression Point. 10, 13, 19, 20, 31
PA power amplifier. 5, 36, 37
PCB Printed Circuit Board. 3, 22
RF radio frequency. 2, 5, 6, 8–10, 13–15, 17, 20, 26
v
SBS sideband suppression. 9, 28, 29, 31, 32, 38
SSB Single Sideband. 8
TX transmitter. 2, 3, 5, 6, 17, 18, 20, 36, 37
UMTS Universal Mobile Telecommunications Systems. 2
USB upper sideband. 6, 19, 25
VI Virtual Instrument. 23, 24
VSA vector signal analyser. 20
VSS Visual System Simulator. 14–17, 19, 20, 37
WCDMA Wideband Code Division Multiple Access. 3, 17
vi
1
1
1.1
INTRODUCTION
Introduction
Background
In the telecom business there is tough competition. The business constantly
pushes for higher capacity and data rates and the companies producing the
telecommunications equipment need more cost effective products to stay ahead
of competitors. Both Ericsson (Ericsson, 2014) and Qualcomm (QUALCOMM,
2013) project a substantial increase in data traffic over the coming years. At the
same time, the commercial spectrum used for Universal Mobile Telecommunications Systems (UMTS) and Long Term Evolution (LTE) ranging between 700
MHz to 2.6 GHz is getting overcrowded as can be seen in Figure 1.
Figure 1: Spectrum allocation between different network operators in Sweden (PTS,
2014).
Interest to use unlicensed spectrum with carrier aggregation as a supplement
to increase capacity and data rates can be seen by Qualcomm and T-Mobile
among others (Tammy Parker, 2013; Monica Alleven, 2014). 3rd Generation
Partnership Project (3GPP) has started work to make specifications on how
unlicensed spectrum can be used in LTE. Dino Flore (2015) states that focus
will be put on implementing LTE in the 5 GHz band. Specifically, the frequency
range above 5.8 GHz is of interest due to the fact that this is in many regions
(the US and China) unlicensed spectrum.
Working with higher frequencies puts higher demand on the hardware in transceiver
circuitry. With higher frequencies one must overcome performance degradation
such as reduced range and linearity. It also makes the radio frequency (RF)
design more complex due to shorter wavelengths and transmission line effects.
To cope with these issues one needs to carefully evaluate the performance degradation of all components in the communications system chain. One of these
components is the mixer or modulator. Most telecommunications systems use
complex modulation schemes and must therefore use a quadrature modulator.
Therefore it is of interest for Ericsson’s transmitter (TX) design department
to test the modulator stage for transmission at higher frequencies. To make a
design commercially usable it needs not only to perform well but also be cost effective. This project will investigate the performance of a commercially available
IQ-modulator, ADL5375 from Analog Devices (Analog Devices Inc., 2014). To
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1.2
Purpose
1
INTRODUCTION
use said modulator at the 5.8 GHz band would be to push it to its upper limit.
At the same time it is interesting to see if it is possible to make a TX design with
ADL5375 to examine the possibility to keep it inside performance requirements
that a given communications protocol require. This would require testing of
the components in a TX design and fine-tuning link budget parameters. Many
aspects of this process are very time consuming, such as going from schematic to
a physical Printed Circuit Board (PCB) design, redesign for different scenarios,
generating different types of signals etc. An alternative approach that is less
time consuming is to make a sufficiently adequate simulation model describing
the ADL5375 and insert it into a customizable TX chain in a simulation software. Analog Devices provide documentation for ADL5375 of key performance
parameters up to 5.8 GHz. This documentation and complementary reference
measurements gives a good base for constructing a simulation model which in
turn can be inserted into a TX chain. This can be used to evaluate performance
issues on a system level. It also makes it easy to trim different system parameters
and apply scenarios with different broadband signals, such as LTE or Wideband
Code Division Multiple Access (WCDMA).
1.2
Purpose
The purpose of this project is to :
ˆ make detailed measurements of performance parameters on ADL5375,
ˆ design a simulation model for ADL5375,
ˆ use the simulation model to evaluate performance of wideband communications system on the 5.8 GHz band and
ˆ give suggestions for link budget optimizations for high frequency applications with ADL5375.
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2
2
THEORY
Theory
In this section all relevant theory for the project will be presented. Starting in
section 2.1 with how signals with complex modulation is modelled and analysed,
moving on to describing general transmitter design in section 2.2 and covering
the process of upconversion with mixers and modulators in section 2.3.
2.1
Modelling Transmit Signals
As stated by Goldsmith (2005) all physical signals that are transmitted are real
signals. This is because all modulators and mixers are built using oscillators
and circuitry that generate real sinusoidal signals. When modelling wireless
channels mathematically using a complex frequency response, it is done purely
for analytical simplicity. The wireless channel is only introducing a phase lag
and amplitude change at each frequency, so that the received signal is also a
real signal. Modulated and demodulated signals are often represented as the
real part of a complex signal. The purpose of this is to make the analysis of
signals in channels with complex properties possible. The transmitted signal is
modelled as a bandpass signal being modulated by a carrier frequency fc
n
o
s(t) = < u(t)ej2πfc t
= vI (t)cos(2πfc t) − vQ (t)sin(2πfc t)
(2.1.1)
where
u(t) = vI (t) + jvQ (t)
(2.1.2)
is a complex baseband signal, also called the signals complex envelope (CE).The
in-phase component is
< {s(t)} = vI (t)cos(2πfc t) − vQ (t)sin(2πfc t)
(2.1.3)
and the quadrature component is
= {s(t)} = vI (t)sin(2πfc t) + vQ (t)cos(2πfc t).
(2.1.4)
Equation 2.1.1 is a standard representation for bandpass signals such as the one
seen in Figure 2.
Figure 2: Bandpass signal S(f ).
This means that any high frequency bandpass signal s(t) has an equivalent lowpass signal u(t). As a consequence it is easier to work with bandpass signals
because it is possible to work with the equivalent lowpass signal instead. This
greatly simplifies the process of applying signal processing algorithms because
of the much decreased needed sampling rates and thus decreased data rates
(Proakis and Salehi, 2008).
The process of going from passband to baseband as presented in equation 2.1.1
is further described in Proakis and Salehi (2008). It can be implemented by a
system called a modulator and will be thoroughly explained in section 2.4.
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2.2
2.2
General Transmitter Design
2
THEORY
General Transmitter Design
The TX part of a radio communications system is a device that is fed with a
signal containing information which it modulates onto a relatively high carrier
frequency and then amplified to be broadcast from an antenna. In modern
systems, the data is created digitally and then converted to an analog signal
called a baseband (BB) signal. BB signals are low frequency signals centred
around 0 Hz with bandwidths depending on the type of communications system
(Leon W. Couch, 2013). As stated above, no communication is conducted at
BB frequencies. To avoid interfering with other communication, the signal is
upconverted to a carrier frequency, fc . The upconversion is made by a frequency
converting device such as a mixer or a modulator (described in depth in section
2.3). Before the signal is broadcast by an antenna it is generally amplified to
a power level that is required by the communication protocol that the TX is
designed for.
(a)
(b)
Figure 3: (a) Schematic representation of a a complex intermediate frequency (IF)
TX design. After the Digital-to-Analog Converter (DAC) there is a bandpass filter (BPF). Before the final power amplifier (PA) stage there is another BPF. (b)
Schematic representation of a zero IF design. After the DAC stage there is a lowpass
filter (LPF) stage. After upconversion to the RF there is a BPF and a PA.
Two common TX designs is depicted in Figure 3. These two designs are the ones
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2.3
Upconversion
2
THEORY
that are intended to be simulated in this project. In most of today’s systems, the
data is produced digitally in the form of quadrature BB signals and converted in
a DAC before it can be upconverted to a carrier frequency. There are different
types of TX designs, which employ different techniques to get from digital BB
signals to a transmitted signal on a carrier frequency. In the design in Figure
3(a) the BB signal is upconverted to the carrier frequency in two steps. First it
is upconverted digitally to an IF. After that it is converted to an analog signal
by the DAC after which it is upconverted to an RF. In Figure 3(b) the BB
signal is converted by the DAC and then directly upconverted to an RF. The
difference between the two designs is that in the Zero IF design you can use a
lower sampling speed, but you will get the local oscillator (LO) leakage in the
middle of your signals bandwidth. If using the Complex IF design you will need
a higher sampling frequency but will separate the LO from your signal which
can be beneficial (Razavi, 2011).
2.3
Upconversion
In section 2.3.1 the process of upconverting a signal to a carrier frequency by
using a mixer will be presented. The mixers role in frequency conversion when
inside of a modulator will be explained in section 2.4.
2.3.1
Mixer
As stated by Pozar (2012) a mixer is a three port device that uses a non-linear
device for either up or down conversion of a signals frequency. Figure 4 shows a
functional diagram of the upconverting process.
Figure 4: Upconversion with a mixer
If the two input signals are
v1 (t) = sinω1 t and v2 (t) = sinω2 t
(2.3.1)
and the mixer is an ideal, lossless mixer, the output will be
1
y(t) = sin(ω1 t)sin(ω2 t) = [sin(ω1 − ω2 )t + sin(ω1 + ω2 )t].
2
(2.3.2)
These two frequencies, ω1 − ω2 and ω1 + ω2 , are called upper sideband (USB)
and lower sideband (LSB) (Pozar, 2012). One of these sidebands are used for
communications and the other is filtered out.
In practice, a mixer uses the non-linear properties of either a diode or a transistor
to produce the desired mixing products (Pozar, 2012). Therefore the mixer
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2.3.2 Upconverting Using a Mixer
2
THEORY
output can be modelled using the junction diode equation
I(V ) = Is (eqV /nkT − 1)
(2.3.3)
where q is the electron charge, k is the Boltzmann’s constant, T is temperature,
n is the ideality factor and Is is the saturation current. These parameters are
dependant on the type of diode that is considered.
The output of a physical mixer deviates from that of the ideal mixer due to the
non-linearities that are introduced in 2.3.3. If the input voltage is expressed as
V = V0 + v
(2.3.4)
where Vo is a DC bias and v is a small AC signal voltage. According to Pozar
(2012) 2.3.3 can be expanded in a Taylor series about V0 as
1 2 d2 I dI + v
+ ....
(2.3.5)
I(V ) = I0 + v
dV V0 2 dV 2 V0
This derives to
v2 0
(2.3.6)
G + ...
2 d
where Gd is the dynamic conductance of the diode. This is called the Small
signal approximation of the diode and is useful when analysing the diode mixer.
I(V ) = I0 + vGd +
2.3.2
Upconverting Using a Mixer
Given the voltage signals
vIF = VIF cos ωIF t
(2.3.7)
vLO = VLO cos ωLO t
(2.3.8)
and
as the inputs of a diode mixer and using the small signal approximation in
equation 2.3.6, the total mixer current is
i(t) = I0 + Gd (vLO + vIF ) +
G0d
(vLO + vIF )2 + ....
2
(2.3.9)
The DC term I0 is easily blocked and the second term is a replication of the input
signals where the vIF term can easily be filtered out. vLO is an in-band signal
and is difficult to filter out. It is one of the contributions to the modulator’s
LO leakage and will be further explained in section 2.5. The third term can be
rewritten using trigonometrical identities to
G0d
(VIF cos ωIF t + VLO cos ωLO t)2
2
G0 2
2
= d [VIF
(1 + cos 2ωIF t) + 2VLO VIF cosωLO cosωIF + VLO
(1 + cos2ωLO t)
4
1
= [2VIF VLO cos(ωLO − ωIF )t + 2VIF VLO cos(ωLO + ωIF )t]
4
2
2
+ VLO
(1 + cos 2ωLO ) + VIF
(1 + cos 2ωIF )].
(2.3.10)
i(t)quadratic =
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2.4
Quadrature Modulator
2
THEORY
This yields several signal components that are unwanted. The DC terms are
again blocked and the 2ωIF and 2ωLO are filtered out (given that they are
sufficiently far away in frequency to ωLO ). This leaves the last two terms,
and
1
vRF1 (t) = VIF VLO cos(ωLO − ωIF )t
2
(2.3.11)
1
vRF2 (t) = VIF VLO cos(ωLO + ωIF )t
2
(2.3.12)
which are consistent with (2.3.2). One of these two frequencies are used as the
carrier frequency, fC . **
2.4
Quadrature Modulator
Not only is the process of modulating the BB signals made in the digital domain
in high-end communications systems, but also the act of upconverting it to an
IF fIF , by using Direct Digital Synthesis (DDS) (Cushing, 2000). The final
upconversion to an RF is done in the analog domain. It is then possible to use
a similar method as when implementing a Single Sideband (SSB) mixer (Pozar,
2012) to suppress one of the sidebands. this enables more spectrum effective
transmissions. By using a quadrature modulator, such as the one schematically
represented in Figure 5 it is possible to obtain similar suppression. In the same
way as when implementing a SSB mixer, both the two IF signals and the LO
signals fed to their respective mixer core need to be 90◦ out of phase to each other
if maximum suppression is to occur. When the signal is digitally upconverted to
an IF and then converted to an analog signal, the two signals are on the form
of 2.1.3 and 2.1.4 and thus 90◦ out of phase. The two signals can be fed into a
quadrature modulator as the one in Figure 5.
Figure 5: Schematic representation of quadrature modulator (Agilent Technologies,
2007).
If similar assumptions are made about the low frequency mixing products as
in section 2.3 it is sufficient to calculate the quadratic term in equation 2.3.9.
When rewritten using the same trigonometric identities for the two inputs gives
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2.5
Modulator Characteristic parameters
2
THEORY
in-phase component
o
i2
G0 h n
iin−phase (t) = d < u(t)ej2πfIF t + vLO (t)
2
G0d =
− 2VLO VQ sin(ωIF − ωLO )t − 2VLO VQ sin(ωIF + ωLO )t
2
+ 2VLO VI cos(ωIF − ωLO )t + 2VLO VI cos(ωIF + ωLO )t
2
cos 2ωLO t
− VQ2 cos 2ωIF t − 2VQ VI sin 2ωIF t + VI2 cos 2ωIF t + VLO
2
2
2
+ VLO + VQ + VI
(2.4.1)
and quadrature component
o
i2
G0 h n
iquadrature (t) = d = u(t)ej2πfIF t + vLO (t)
2
G0d 2VLO VQ sin(ωIF − ωLO )t − 2VLO VQ sin(ωIF + ωLO )t
=
2
− 2VLO VI cos(ωIF − ωLO )t + 2VLO VI cos(ωIF + ωLO )t
2
+ VQ2 cos 2ωIF t + 2VQ VI sin 2ωIF t − VI2 cos 2ωIF t − VLO
cos 2ωLO t
2
+ VLO
+ VQ2 + VI2
(2.4.2)
as quadrature component. If the two are summed together as in Figure 5 the
output of the modulator is
isum (t) = G0d 2VLO VI cos(ωIF + ωLO )t − 2VLO VQ sin(ωIF + ωLO )t
2
+ VLO
+ VQ2 + VI2 .
(2.4.3)
As can be seen, only one of the two sidebands is left in the output signal, ωIF +
ωLO , but all information is preserved.
Described above is the theoretical operation of an ideal quadrature modulator.
There is however a number of circumstances that degrade the performance of
the modulator. To make the performance of the modulator measurable there are
a number of important parameters that characterizes it. These will be presented
in section 2.5.
2.5
Modulator Characteristic parameters
Contrary to the situation in above sections, where a schematic modulator is
considered, there are imperfections internally that create asymmetries and nonidealities. This will affect the output of the modulator (Nash, 2009).
2.5.1
LO Leakage
LO leakage, as described by Razavi (2011), refers to the situation when the LO
signal is somehow leaking to the output of the mixer. It can be caused by the
non-linear properties of the mixer, as seen in equation 2.3.6, where a replication
of the input signal is produced. It can also be caused by device capacitances
between the LO and RF port or by emissions from the substrate to the output
pad. Both of the latter causes what is referred to as self-mixing (Nash, 2009). It
can be caused by the finite isolation either between the LO port and one or both
of the I and Q ports of the modulator or between the LO port and the actual
RF output. It can also occur when a DC offset is present on the input of the
mixer.
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2.5.2 Sideband Suppression
2.5.2
2
THEORY
Sideband Suppression
The sideband suppression (SBS) of a quadrature modulator is defined as
SBS =
PDSB
PSSB
(2.5.1)
which is the ratio between the desired sideband, PDSB , and the suppressed
sideband, PSSB . Consider the event where the gain in the I channel is greater
then that of the Q channel. This could be caused by any number of reasons such
as input mismatch, internal I/Q gain imbalance etc., but the effect would be the
same. It would cause degradation of the sideband suppression of the modulator.
2.5.3
Conversion Gain
The conversion gain specifies the power transfer of the modulator. A modulator
is generally a lossy component, but can have a small gain in certain designs. The
most general definition of gain is
Gconv = Pout /Pin
(2.5.2)
where Pout is the output signal power and Pin is the input signal power. There
is no convention for the definition of Pout when talking about modulators. Most
logical is to define it as the power confined inside the desired sideband of the
output signal, since this is the only part of the output that is interesting. Similarly there is no convention for Pin either. You could either define it as the sum
of the I and Q input powers, or just one of the two channels power.
The definition that is used in this report for all conversion gain calculations is
that Pin is the power of one of the two input channels and Pout is the power of
the desired sideband. It has been established that this is the same definition used
in the datasheet for ADL5375 by reverse engineering the values for conversion
gain and input voltage levels presented in Analog Devices Inc. (2014).
2.5.4
1 dB Compression Point
Many RF circuits have non-linear properties and thus exhibits a compressive
behaviour when they are driven to hard. The power level of the output signal
is compressed which can create unwanted distortion. It is therefore of interest
to quantify how severe the compression of a component is. As described by
Pozar (2012) and Razavi (2011) it is common to present the point at which the
output power of a device (e.g. amplifier or mixer) has compressed to a point 1
dB below the ideal linear behaviour, the 1 dB Compression Point (P1dB). It can
be specified either as the input power (IP1dB) or the output power (OP1dB),
usually the one giving the highest value. It is calculated by extrapolating the
linear response as depicted in Figure 6.
2.5.5
Second and Third Order Intercept Points
As discussed previously, the modulator is a device that takes advantage of nonlinear properties of either a diode or a transistor to produce certain frequency
shifted output signals. It is clear in section 2.3 that there are several unwanted
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2.5.5 Second and Third Order Intercept Points
2
THEORY
signal components at other frequencies than the wanted. To be able to quantize
the effects of said non-linearities the concepts of second and third order intercept points are widely used (Razavi, 2011; Proakis and Salehi, 2008). If using
equations (2.3.6) – (2.3.9) you can measure the intercept points. To do this the
inputs needs to be changed to two signals with equal amplitude according to
v1 (t) = V cos ω1 t
(2.5.3)
v2 (t) = V cos ω2 t
(2.5.4)
and
with frequencies ω1 and ω2 close to each other. Simplifying the non-linear model
in equation 2.3.6 to
i(t) = I0 + α1 v + α2 v 2 + α3 v 3 .
(2.5.5)
If v is substituted to v( t) + v2 (t) we get
i(t) = I0 + α1 (V cos ω1 t + V cos ω2 t) + α2 (V cos ω1 t + V cos ω2 t)2
+ α3 (V cos ω1 t + V cos ω2 t)3 .
(2.5.6)
Discarding everything else but the third order terms we get, after simplification
1
1
3 3
3 3
i(t) = α3 V
cos ω1 t + cos 3ω1 t + α3 V
cos ω2 t + cos 3ω2 t
4
4
4
4
3
3
3 3
+ α3 V
cos ω2 t + cos(2ω1 − ω2 )t + cos(2ω1 + ω2 )t
2
4
4
3
3
3 3
+ α3 V
cos ω1 t + cos(2ω2 − ω1 )t + cos(2ω2 + ω1 )t
(2.5.7)
2
4
4
The interesting terms are 2ω1 − ω2 and 2ω2 − ω1 . If ω1 and ω2 are close to each
other then 2ω1 − ω2 and 2ω2 − ω1 will be in the vicinity of these frequencies
as well. This causes a distortion of the signal of interest in the situation of a
broadband signal that is impossible to filter out without distorting the signal
of interest. It is easily seen, when looking at equations 2.5.6 and 2.5.7, that as
the input voltage V increases the third order terms increase as V 3 . This means
that for small voltages the third order terms will be very small but will increase
rapidly but will increase rapidly as the input voltage increases. In Figure 6 we
can see that at one point the power of the linear terms will intersect the power
of the third order terms. This is called the Third Order Intercept Point (IP3 ).
It is also apparent from Figure 6 that this point is a strictly theoretical point,
because it occurs well above the 1 dB compression point.
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2.5.5 Second and Third Order Intercept Points
2
THEORY
Figure 6: A diagram of the third order intercept point and the 1 dB compression
point in log-log scale (Pozar, 2012).
Looking at the second order terms in equation 2.4.1 it is clear that one can define
a Second Order Intercept Point (IP2 ) in much the same manner. The increase of
amplitude of the second order terms is as the square of the input voltage V . To
calculate these points extrapolation of a fundamental tone and both second and
third order intermodulation tones is needed. The point where the extrapolated
plot of the intermodulation (IM) products intersects the extrapolated plot of the
fundamental tone is the IP2 and IP3 . The process of extrapolating demands
that long measurement series are performed, which can be very time-consuming
or not possible for other measurement technical reasons. It is then possible to
estimate the IIP3 . Starting with the scenario that the input power is at the level
of PIIP3 . If the input power then is lowered to an arbitrary level Pin1 , it will then
have lowered with 10logPIIP3 − 10logPin . On a log-log scale the IM3 product
power will fall with a slope of 3 and the fundamental will fall with a slope of 1.
This means that the difference between the two plots increase with a slope of 2.
This can be concluded by
10logPf − 10logPIM3 = 2(10logPIIP3 − 10logPin ) = ∆P
(2.5.8)
where PIM3 is the power of the strongest IM3 product and Pf is the power of
the fundamental tone. This gives
10logPIIP3 =
∆P
+ 10logPin
2
(2.5.9)
and more generally
∆P
+ 10logPin .
(2.5.10)
x−1
Since there can be dynamic non-linearities, this is only an estimate and not as
accurate as the method of extrapolation, but can be used as an approximation
(Razavi, 2011).
10logPIIPX =
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3
3
3.1
HARDWARE
Hardware
Evaluation Board
The center of this project is the ADL5375-05 (Analog Devices Inc., 2014). It
is a broadband quadrature modulator, designed to work in the frequency range
of 400 MHz to 6 GHz. The I and Q baseband input ports are both differential
and the LO input port is single-ended. The output RF port is single-ended
and matched to 50 Ω. The baseband input needs a 500 mV biasing for optimal
performance. Some of the key parameters of the modulator can be seen in table
1. All tests and measurements were made on the evaluation board (rev. B
Table 1: Performance Parameters for ADL5375@5.8 GHz (Analog Devices Inc.,
2014).
Parameter
Value
Modulator Voltage Gain
Output P1dB
-5.3
Unit
dB
4.9
dBm
Output IP2
39.1
dBm
Output IP3
14.6
dBm
Noise Floor
-153.0
LO Leakage
-19.5
dBm
-3.2
dBc
Sideband Suppression
dBm/Hz
(Analog Devices Inc., 2014)) supplied from the manufacturer of ADL5375. The
board has two possible RF output paths; one path with a RF drive amplifier
(ADL5320 (Analog Devices Inc., 2013)) and another with a direct path from the
modulator RF output. All I/O connections are of the type SMA 3.5 mm.
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4
4
SIMULATION
Simulation
To be able to make a more detailed performance analysis of the modulator and
see how it performs when using a broadband communication standard in a transmitter design it is convenient to develop a simulation model of the modulator.
This gives the possibility to evaluate the board in a complex and more in-depth
manner, without the need of the time consuming process of designing hardware,
verifying the design etc. It can be very useful in the pre-development stage when
a new product for the intended spectrum is to be investigated. In the next sections the simulation software suite used will be presented, along with how the
modulator simulation model was constructed.
4.1
AWR’s Visual System Simulator
AWR’s Visual Systems Simulator is a software suite which is directed towards
system design of todays modern communications systems (National Instruments,
2015a). It is suitable for the purpose of this project, because it comes with a
comprehensive library of RF component models (system blocks) which ranges
from analog devices such as mixers and amplifiers to logical operators and signal
processing blocks. A complete and more detailed descriptionis given in the Visual
System Simulator (VSS) Getting Started Guide (National Instruments, 2014b).
VSS uses the CE seen in equation 2.1.2 as representation of a signal whenever
it is possible. The reason for this is that even though a signal is modulated
around a high carrier frequency, the actual data is represented by the lowpass
CE. This is sufficiently modelled with a sampling frequency orders of magnitude
lower than is needed for proper sampling of the upconverted signal. Therefore
VSS propagates a center frequency through the simulation which the CE is
centred around. This reduces the sampling frequency and thus the simulation
time considerably (National Instruments, 2014b).
4.1.1
Simulation of ADL5375 in VSS
Properties that a simulation model should adequately simulate are:
ˆ Non-linearities
ˆ Conversion gain
ˆ Gain compression
ˆ I/Q imbalance
These properties should preferably be simulated over a broad frequency range.
To model the internal structure of ADL5375 properly the block diagram depicted
in Figure 7(a) was used as a starting point. It consists of drivers for the two
baseband inputs, a quadrature phase splitter for the LO input, two mixer cores
for upconversion and an RF combiner to sum the I and Q component. To
model the mixer cores the system block MIXER B2 was used. It implements
a behavioural model of a non-linear double-balanced mixer with an equivalent
circuit as in Figure 7(a).
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4.1.1 Simulation of ADL5375 in VSS
4
(a)
SIMULATION
(b)
Figure 7: (a) Block diagram of ADL5375 internal design (Analog Devices Inc., 2014).
(b) VSS system block MIXER B2 equivalent diagram (National Instruments, 2015b).
MIXER B2 uses a set of user specified parameters to model a mixer. This makes
it suitable for creating a model based on measurement data. It is also possible to
specify a vector of frequency values to model a wideband frequency dependant
behaviour. The frequency dependency is valid for the baseband frequency range.
Certain key parameters were specifically chosen to characterize the modulator
(table 2), all of which can use the frequency dependency setting, except for the
Noise Figure parameter.
Table 2: Parameters used to model ADL5375 in MIXER B2.
Name
Description
GCONV
Conversion gain of the mixer.
P1DB
The 1 dB compression point of the mixer.
IP3
3rd order intercept point of the mixer.
IP2
2nd order intercept point.
LO2OUT
The LO leakage between the LO input and the RF output.
NF
Noise Figure of the mixer.
FREQS
Vector for frequency dependent settings.
Apart from the parameters in Table 2 it is also possible to model temperature
dependencies, impedance mismatch on input and output, etc. To conserve time
these aspects were chosen not to be implemented in this study. The non-linear
amplifier in Figure 7(b) is similiar to another system block, AMP B2 (National
Instruments, 2015b). Since measurements of the mixer cores is not done individually it makes more sense to implement the non-linearities on the whole modulator by using AMP B2. The LO generation along with the quadrature phase
splitter was realized with the combination of system blocks TONE, SPLITTER
and PHASE in the manner seen in Figure 8.
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4.1.2 Wideband Signal Source
4
SIMULATION
Figure 8: LO signal generation and quadrature phase splitter implementation in
VSS.
TONE generates a sinusoidal tone at a specified frequency. SPLITTER splits
the incoming signal power according to
Vin
Vout = √ .
2
(4.1.1)
After that the PHASE block shifts the phase in one of the LO paths with 90◦ .
Here it is also possible to insert a phase error to model a sideband that is not
perfectly suppressed. The upconverted I and Q signals are then summed together
with an ADD block, which sums together the inputs and outputs the result. The
whole implementation can be seen in Figure 9(a).
(a)
(b)
Figure 9: (a) VSS realization of ADL5375. (b) ADL5375 implemented as a subcircuit element.
For ease of use the modulator model was implemented as a sub-circuit element
as in Figure 9(b) to make the model more compact.
4.1.2
Wideband Signal Source
To make a performance analysis of ADL5375 and put it in a communications
system perspective one needs to feed it with a wideband signal that corresponds
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4.2
Simulate Wideband Behaviour
4
SIMULATION
to a communications standard used in telecommunications system, such as LTE
or WCDMA. Then it is possible to make measurements that is comparable with
the limits that is specified in standard specifications set by 3GPP (European
Telecommunications Standards Institute, 2015). For this to be possible, it is
necessary to be able to feed the modulator model with a wideband LTE signal.
there is a system block in VSS, LTE DL TSIG, which is a LTE downlink signal
source (National Instruments, 2014a). It is constructed to comply with European Telecommunications Standards Institute (2008), and is therefore of interest
to use to create test signals for measurements. To create a test signal that corresponds to the signal components presented in equation 2.1.3 and 2.1.4 in the
digital domain and enable for upconversion to an IF, one needs to perform the
same multiplication with a complex exponential. This is done by first splitting
the complex baseband signal into two real data streams and then producing the
signal components by using ideal mixers to convert the signals to an IF as in
Figure10(a).
(a)
(b)
Figure 10: (a) Creation of quadrature and in-phase component in AWR. (b) Internals of LTE signal generation sub-circuit.
This was then inserted as a sub-circuit block inside the LTE sub-circuit block
(Figure 10(b)). Using this method to create a baseband envelope at an IF
requires that you sample at a higher sampling speed, to make sure that the
Nyquist Criteria is met(Goldsmith, 2005).
4.2
Simulate Wideband Behaviour
To evaluate the impact of using ADL5375 in the 5.8 GHz band it is necessary to
put it in a TX design perspective. When design of a TX is considered you need
to adhere to the limits specified for the region that is intended for the product.
All specification limits in this report will be taken from European Telecommunications Standards Institute (2015), where all minimum RF characteristics and
minimum performance requirements of Evolved Universal Terrestrial Radio Access (E-UTRA) base station in Europe are specified, which is the air interface
for the LTE standard. Some of the parameters for base station transmitters
that is regulated in European Telecommunications Standards Institute (2015)
are presented in Table 3.
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4.2
Simulate Wideband Behaviour
Base Station (BS)
Property
Rated Output Power
Error Vector Magnitude
(EVM) Requirements
4
Config
SIMULATION
Limit
Wide Area BS
< +38 dBm
Medium Range BS
< +24 dBm
Local Area BS
Home BS (One TX Antenna) < +20 dBm
QPSK
17.5 %
16QAM
12.5 %
64QAM
8.0 %
256QAM
3.5 %
Table 3: Some of the requirements for LTE Downlink base station found in European
Telecommunications Standards Institute (2015).
There are several TX specific limits specified by 3GPP. The ones that are the
most interesting when examining the modulator is the Adjacent Channel Rejection Ratio (ACLR) and Rated Output Power. The ACLR is the ratio between
the transmitted power in one channel to the power in an adjacent channel. This
adjacent channel is usually defined as an identical channel with equal bandwidth
as the source channel, but can be specified differently. In European Telecommunications Standards Institute (2015) there are specifications for many different
source and adjacent channel combinations depending on what type of spectrum
that is measured on (unpaired/paired, contiguous/non-contiguous). An example
of this is presented in Figure 11.
Figure 11: Base station ACLR in paired spectrum (European Telecommunications
Standards Institute, 2015).
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5
5
MEASUREMENTS
Measurements
In order to characterize the performance of ADL5375 a number of key parameters
needed to be measured. The parameters measured for characterization were
chosen to conform to the measurement diagrams that are available in Analog
Devices Inc. (2014) to make a comparison possible. In the coming sections the
equipment used for measuring and the measurements that were performed will
be presented.
5.1
Necessary Measurements
The measurements were chosen to comply to the simulation model and to be
comparable to the ADL5375 datasheet (Analog Devices Inc., 2014). Measurements were performed with two types of stimulus; continuous wave (CW) and
two-tone signals. The former type is used when measuring the intermodulation
properties of the modulator, while CW stimulus is used for all other measurements. In Table 4 the parameters that can be extracted when using CW tones
as stimulus are shown.
Swept
Parameter
BB Frequency
BB Power
LO Frequency
LO Power
BB Phase
Measured
Parameter
vs.
vs.
vs.
vs.
vs.
USB Power
LSB Power
LO leakage
LO Gain
USB Power
LSB Power
LO leakage
LO Gain
USB Power
LSB Power
LO leakage
LO Gain
USB Power
LSB Power
LO leakage
LO Gain
USB Power
LSB Power
LO leakage
LO Gain
Performance
Parameter
Sideband Suppression
Conversion Gain
LO leakage
VSS
Parameter
GCONV
LO2OUT
FREQS
Sideband Suppression
Conversion Gain
P1DB(scalar)
LO leakage
Sideband Suppression
Conversion Gain
LO leakage
Sideband Suppression
Conversion Gain
LO leakage
Sideband Suppression
Conversion Gain
LO leakage
Table 4: Acquired VSS specific parameters and general parameters when measuring
using CW as stimulus.
The measurement data gathered when sweeping BB frequency with a CW input
gives frequency dependent values to the parameters GCONV, LO2OUT and
FREQS and can easily be imported into VSS and used in the simulation model.
Sweeping BB input power gives a value of P1dB by using the method presented
in section 2.5.4. This only generates a scalar value for P1dB and is thus only valid
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5.2
Measurement equipment
5
MEASUREMENTS
for a specific BB and LO frequency. To be able to calculate a value that describes
a frequency dependency of P1dB, you would need to sweep BB input power
for every step of BB frequency. This would require either manually changing
frequency or the construction of a script to automate measurements. To conserve
time, this was not done. Instead, a scalar value was used for P1dB.
Table 5 shows parameters acquired when using a two-tone signal as stimulus and
sweeping various parameters are presented. The measurements performed make
it possible to calculate characteristic parameters in accordance with formulas
found in section 2.5.
Table 5: Acquired VSS specific parameters and general parameters when measuring
using a two-tone signal as stimulus.
Swept
Parameter
BB
Frequency
BB Power
LO
Frequency
5.2
Measured
Parameter
LO leakage
Tone 1 Power
vs. Tone 2 Power
IM2 Power
IM3 Power
LO leakage
Tone 1 Power
vs. Tone 2 Power
IM2 Power
IM3 Power
LO leakage
Tone 1 Power
vs. Tone 2 Power
IM2 Power
IM3 Power
Performance
Parameter
VSS
Parameter
2nd Order Intercept Point IP2
3rd Order Intercept Point IP3
2nd Order Intercept Point
3rd Order Intercept Point
2nd Order Intercept Point
3rd Order Intercept Point
Measurement equipment
To be able to perform RF measurements in the frequency domain it is common
to use either a vector signal analyser (VSA) or a network analyser (NA). The RF
output of the ADL5375 is single-ended, which is standard on most instruments
and the modulator has a 50 Ω interface (Analog Devices Inc., 2014). Problems
arose when trying to supply the modulator with a stimulus. At first, no signal
generation equipment was available that had two differential output channels.
Alternative methods for signal generation were therefore examined, which will
be described in the coming section.
5.2.1
DAC as stimulus
Another approach to create stimulus is to use an RF DAC along with a digital
signal generator, as is done in a real TX design. The first DAC that was considered was the DAC3482 from Texas Instruments (Texas Instruments Inc., 2013).
It is a 16-bit dual channel DAC with a high sampling rate, 1.25 GSPS, making
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5.2.1 DAC as stimulus
5
MEASUREMENTS
it suitable to create broadband BB signals up to the Nyquist frequency. This
would enable a theoretical upper frequency limit of 1.25 GSPS/2 = 612.5 GSPS,
which is more than enough bandwidth for the measurements needed. It also
comes with a lot of integrated functions such as on-chip mixing, interpolation
filters from 2x - 16x, Digital I and Q correction etc. It is available in the form
of a development board, DAC3482 EVM REV F, together with signal generator
TSW1400 REV D, seen in Figure 12(a).
(a)
(b)
Figure 12: (a) Ti signal generator TSW1400 REV D and DAC3482 EVM REV F
development board. (b) Necessary biasing network for interface between DAC3482
and ADL5375.
During this study it was soon realized that it would be difficult to bias ADL5375
properly to 500 mV. To achieve this a biasing network is needed, seen in Figure
12(b), to change the common mode on the DAC output. To implement the network with mentioned setup, unwanted intrusions needs to be made on the DAC
development board due to the lack of pads that comply with above schematic.
A more suitable DAC, in the sense of biasing, is AD80255A. It is a 16-bit, dual
channel DAC with a sampling rate of 500 MSPS, which gives a theoretical upper band limit of 250 MHz according to the Nyquist theorem. For creation of
baseband signals, this bandwidth would suffice. although it might be a too small
bandwidth for extraction of bandwidth parameters from ADL5375, such as flatness properties etc. The DAC is available on the development board AD80255A
EBZ REV B. Digital signals are produced with DPG2 (Analog Devices, 2014),
which is a benchtop signal generator specially made to interface with DAC’s from
Analog Devices. The AD80255A is designed to comply with the bias needed on
the inputs of ADL5375. It has a full-scale current of 20 mA which gives a midrange current of 10 mA. Therefore only two 50 Ω resistors to ground is needed
as depicted in Figure 13 to give a bias of 500 mV.
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5.2.1 DAC as stimulus
5
MEASUREMENTS
Figure 13: Bias network needed for the interface between AD80255A and ADL5375
(Analog Devices Inc., 2014).
The development board for AD80255A is equipped with baluns with the purpose
of converting the differential output signals to single-ended signals. This made
it incompatible with ADL5375, which had differential BB inputs. In this study
this was solved by first desoldering the baluns. Then phase-matched semi-rigid
coaxial cables with 3.5 mm SMA connectors were soldered in directly on the
PCB to create a differential interface for ADL5375 (Figure 14(a)). In Figure
14(b) the complete setup with ADG2 as signal generator, AD80225A converting
to analog differential signals that is fed via 3.5 mm SMA connectors to ADL5375
is shown.
(a) Phase matched semi-rigid cables for differential interfacing.
(b) Complete measurement setup.
To create signals with DPG2 there is a GUI called DPGDownloader (Figure 15)
where predefined signals can be used, such as CW, multi-tone, etc. In accordance
with section 5.1, it is sufficient to create CW tones and multi-tone signals. To
automate the frequency incrementation process, or for loading of signal vectors,
there is a possibility to interface via MATLAB or LabVIEW. This was deemed
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5.2.1 DAC as stimulus
5
MEASUREMENTS
too time-consuming for this study so frequency incrementing was performed
manully.
Figure 15: Picture of DPGDownloader GUI.
The measurements were made with the spectrum analyser FSQ8 from Rohde
& Schwarz (Rohde & Schwarz, 2011). It has a frequency range of 20 Hz – 8
GHz, which covers the frequency range of interest. By manually setting markers
to frequency values of interest measurement values were acquired. A script
was made in VEE which wrote the marker values to a text file via the standard
General Purpose Interface Bus (GPIB) interface. The incrementing of BB power
was done manually by changing the gain settings for the I and Q channels in the
Virtual Instrument (VI) seen in Figure 16 that is supplied by Analog Devices.
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5.2.2 81160A and N5242A PNA-X
5
MEASUREMENTS
Figure 16: VI GUI for AD80225A.
The gain of AD80225A can be incremented between 0 – 15 on the I and Q
channel individually. If biasing ADL5375 is done properly this gives a voltage
range of 881.6 mV – 1630 mV, with increments of approximately 50 mV.
5.2.2
81160A and N5242A PNA-X
Another measurement setup was tried that was supplied from Keysight; N5242A
PNA-X NA (Keysight, 2014c) along with 81160A Pulse Function Arbitrary Noise
Generator (Keysight, 2014a). A beta software application for N5242A PNA-X,
called DIFF-IQ, was available. The application is especially developed for measurements on devices with differential inputs and frequency converting properties
(such as modulators e.g.). DIFF-IQ gives the user the possibility to specify which
parameters to sweep, such as BB frequency, BB power, LO frequency, LO power
and I/Q phase. It is also possible to specify which frequencies measurements
are to be performed on. Frequencies can be specified by using variables, thus
enabling measurements on frequencies that has an offset relative to the input
frequency. An example of this is presented in Table 6.
The generation of an LO signal was done using N5242A’s internal signal source,
while BB signals were generated externally with 81160A. The measurement setup
is depicted in Figure 17.
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5.2.2 81160A and N5242A PNA-X
5
MEASUREMENTS
Table 6: Table showing which measurement were made when sweeping BB frequency
and at what frequencies the measurements were made.
Swept
Parameter
BB Frequency
Frequencies
Measured
Parameter
Relative
frequencies
F1 −10Hz − 450MHz(BB)
F2 −5.8GHz(LO)
USB Power
LSB Power
LO leakage
LO Gain
F1 +F2
F1 −F2
F2
F2
Figure 17: Measurement setup. 81160A above and N5242A below, connected together with a LAN cable.
The two devices can be connected together using LAN, USB or GPIB. This
way N5242A can control 81160A and thus automate the whole measurement
process. This is very time saving and also enables a high density of data points.
81160A has a maximum sampling speed of 2 GSPS for arbitrary waveforms and
a specified upper frequency limit of 500 MHz for the built in sine waveform. The
sampling speed is highly superior the sampling speeds of previously presented
setups. The biasing, which was difficult with previous setups, is a changeable
parameter in the interface of 81160A.
To measure the parameters IP2 and IP3 a two-tone signal needs to be generated,
as in Figure 18, to give the necessary outputs. To do this with the 81160A there
is a software suite from Keysight called Agilent BenchLink Waveform Builder.
With the software it is possible to produce different types of waveforms that can
be downloaded to the 81160A.
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5.2.3 Calibration
5
MEASUREMENTS
Figure 18: Two-tone input signal needed to be able to measure the parameters IP2
and IP3 .
However, this procedure does not make it possible to automate the incrementing
process of a two-tone signal. Therefore this needed to be done in another way
during this study. One way of achieving this would be to terminate one of the
inputs and feed only one mixer core, as described in Tektronix (2013). Assuming
symmetry between the mixer cores in the sense of non-linear properties, this
method produces exact measurements of intermodulation products. To set up
this in a way that enables automation of the procedure with the N5242A, the two
sources available on 81160A were set to two different frequencies with a 5 MHz
spacing. An external RF combiner combined the two signals to generate a twotone stimulus signal for one of the input ports on ADL5375. This method gave
bad results due to the lack of high performance combiners and the measurements
were unusable. Therefore, as a last resort, another method was tried. It was seen
that an output spectrum with intermodulation products was produced when the
I input of ADL5375 was fed with one of two tones and the Q input with the
other tone. Using the schematic representation of a modulator from Figure 7(a)
this would theoretically not generate intermodulation products, since none of
the mixer cores are fed with a two-tone signal. IM products were present in the
spectrum which would suggest that there are some sort of non-linear element
after the internal combiner, such as an amplifier circuit.
5.2.3
Calibration
One of the biggest wins using this setup is the possibility to make precise calibrations on the ports of the N5242A. This enables for precise power measurements
on the RF side and exact output power levels on the LO interface. The input calibration is made using Keysight’s E-Cal kit N4691, seen in picture 19(a)
(Keysight, 2014b).
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5.2.3 Calibration
5
(a)
MEASUREMENTS
(b)
Figure 19: (a) E-cal kit N4691, used for calibration of N5242A. (b) External power
meter E4419B used for output power calibration.
It automates the whole process of calibration over the frequency range that
measurements are to be performed in. The power calibration for the output
source is made by connecting an external power meter, in this case the E4419B
(Figure 19(b)), to the N5242A. N5242A then uses E4419B to automatically
calibrate the output power for the whole range of LO frequencies specified in the
measurements.
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6
6
RESULTS AND CONCLUSIONS
Results and Conclusions
6.1
6.1.1
Measurement Results
CW Tone Measurements.
Measurements made at fLO −5.8 GHz will be presented in this section. To evaluate the performance at said frequency the results will be compared to measurements found in the datasheet for ADL5375 that are made when fLO =5.8 GHz.
The measurements in this project were performed under similar circumstances,
seen in Table 7. PBB = 1 dBm is equivalent to VBBp-p = 709 mV if a 50 Ω load is
assumed. In the datasheet Vbias −500mV, fBB −1MHz and VBBp-p −1V. Figure
Table 7: Conditions for measurements performed in this work at fLO =5.8 GHz.
Parameter
Value
VS
TA
PLO
PBB
Vbias
fBB
5V
25◦ C
0 dBm
1 dBm
530 mV
10 MHz
20 shows measurements of PUSB , PLSB , SBS, PLO,leak and GCONV over a BB
frequency span of 10 kHz to 450 MHz where PUSB is the upper sideband power,
PLSB is the lower sideband power, SBS is the sideband suppression, PLO,leak is
the LO power and GCONV is the conversion gain. It shows that the conversion
gain is flat, and varies slightly around a mean value of -6.1 dBm. Comparing
the tabulated values from Table 8, one can see a deviation of only 0.6 dB. The
sideband suppression is ranging from -27.14 dBc to -22.96 dBc with an average
of -25.21 dB. This shows a relatively flat response, with a small decrease over
frequency. A comparison of values for SBS from Table 8 shows a 11.1 dB lower
value in the datasheet compared to measured data. It also show a significantly
higher LO Leakage of -19.5 dBm, which is 12.7 dB higher then measured data
8. PLSB is increasing slightly over frequency from -32.29dBm to -28.17dBm.
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6.1.1 CW Tone Measurements.
6
RESULTS AND CONCLUSIONS
Figure 20: Measurements of PUSB , PLSB , SBS, PLO,leak and GCONV over a frequency
span of 10 KHz to 450 MHz.
Figure 21 shows measurements of PUSB , PLSB , SBS, PLO,leak and GCONV vs.
PBB , ranging from -20 dBm to 13 dBm. PUSB is seen to increase linearly to a
point were it starts to saturate. This is reflected directly on the GCONV , which
is directly proportional to PUSB . PLSB is increasing in a in a similar manner but
with a less steep slope. This can be seen in that the SBS curve is decreasing
slowly.
Figure 21: Measurements of PUSB , PLSB , SBS, PLO,leak and GCONV when incrementing PBB from -20 dBm to 13 dBm.
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6.1.2 IM Measurements
6
RESULTS AND CONCLUSIONS
Extrapolation of PUSB was done to measure OP1dB. OP1dB was found to be at
-1.43 dBm (Figure 22), which lies 11.43 dB under the value that is found in the
datasheet.
Figure 22: PUSB with extrapolated data from linear region vs. PBB .
6.1.2
IM Measurements
Figure 23 shows measurements taken when sweeping a two-tone input, as specified in section 5.2.2. The two tones are swept between 10 and 15 MHz to 450
and 465 MHz respectively. The IM3 product has an average of -43.05 dBm and
is decreasing slightly. The IM2 fluctuates drastically.
Figure 23: Plot of Ptone1 , Ptone2 , PIM2 , PIM3 vs. BB frequency.
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6.1.3 Simulation Model
6
RESULTS AND CONCLUSIONS
In Figure 24 we can see how these results affect the OIP2 and OIP3 , which are
inversely proportional to PIM2 and PIM3 respectively.
Figure 24: Plot of OIP2 and OIP3 vs. BB frequency where OIP2 is the output
second order intercept point and OIP3 is the output third order intercept point.
Table 8 shows that measured data for OIP2 has a value of 22.8 dBm and OIP3
has a value of 6.0 dBm. These values are 16.3 dB and 8.6 dB lower then the
values presented in the datasheet for ADL5375. A comparison of values for SBS
shows a 11.1 dB lower value in the datasheet compared to measured data. It also
show a significantly higher LO Leakage of -19.5 dBm, which is 12.7 dB higher
then measured data.
Table 8: (a) Parameter values from the datasheet of ADL5375 for fLO =5.8 GHz. All
parameter measurements were made under the conditions presented in Table ??. (b)
Parameter values from the measurements performed in this work for fLO =5.8 GHz.
All parameter measurements were made under the conditions presented in Table 7.
Parameter
Value
Conversion Gain
Output P1dB
LO Leakage
Sideband
Suppression
Output IP2
Output IP3
Parameter
-5.3 dB
4.9 dBm
-19.5 dBm
Coversion Gain
Output P1dB
LO Leakage
Sideband
Suppression
Output IP2
Output IP3
-38.2 dBc
39.1 dBm
14.6 dBm
(a)
6.1.3
Value
-5.9 dB
-1.4 dBm
-32.2 dBm
-27.1 dBc
22.8 dBm
6.0 dBm
(b)
Simulation Model
To verify the simulation model, control measurements were made of PUSB , PLSB
and PLO,leak while making a sweep of the BB frequency between 10 kHz and 450
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6.1.4 ACLR Measurements
6
RESULTS AND CONCLUSIONS
MHz with CW tones as input signals. The conditions were chosen according to
Table 9 to match the conditions in Table 7 as close as possible.
Table 9: Parameter conditions for simulation model fBB =5.8 GHz.
Parameter
Value
PLO
PBB
Vbias
fBB
0 dBm
1 dBm
not needed
10 MHz
This was then used to calibrate the model so the output would correspond to
the measurements in Figure 20 as close as possible. In Figure 25 PUSB , PLSB
and PLO,leak and SBS is plotted against fBB , both for the simulated data and
the measured data. PUSB,sim follows the measured data well until fBB ≈272.3
MHz, where it declines to a level approximately 1.7 dB below the measured
data, still following it. PLSB,sim differs from PLSB,sim by approximately 26.69
dB. This is reflected on the SBS, which is constant at 26.7 dB. When comparing
PLSB,sim and PLSB,meas there is a deviation between the results. PLO,leak,sim
follows PLO,leak,meas exactly, except for a few spikes (-33.6@357.8 MHz eg.) where
it deviates.
Figure 25: Measurements of PUSB , PLSB , PLO,leak from simulation and measurements when sweeping BB frequency.
6.1.4
ACLR Measurements
To evaluate how the modulator is performing in a broadband system, simulations
were made with a LTE signal source outputting a signal with 5 MHz bandwidth
at an IF of 20 MHz. The ACLR was then measured with the adjacent channel
at ±5 MHz and ±10 MHz offset. The bandwidth was the same as the reference
signal, 5 MHz.
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6.1.4 ACLR Measurements
6
RESULTS AND CONCLUSIONS
In Table 10 the ACLR for simulations made with measured data and data taken
from the datasheet with fLO =5.8 GHz is presented. It shows that ACLR measurements for all offsets only differs with maximum 1.9 dB.
Table 10: Simulated ACLR measurements based on measured data and the
datasheet of ADL5375. Measurements with ±5 MHz and ±10 MHz offset.
Adjacent channel Offset
-10 MHz
-5 MHz +5 MHz +10 MHz
ACLR based on
measurements
55.9 dB
50.0 dB
51.5 dB
56.0 dB
ACLR based on
datasheet
55.7 dB
50.1 dB
53.4 dB
55.7 dB
The spectrums depicted in Figure 26 are the ones that ACLR measurements were
performed on. The input power levels was backed of until the intermodulation
almost disappeared in the noise floor.
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6.1.4 ACLR Measurements
6
RESULTS AND CONCLUSIONS
(a)
(b)
Figure 26: (a) Spectrum of output from simulation based on measured data. (b)
Spectrum of output from simulation based on datasheet. LTE signal with BW=5
MHz and fIF =20 MHz as input.
The input and output levels for the ACLR measurements can be seen in Table
11.
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6.2
Discussion
6
RESULTS AND CONCLUSIONS
Table 11: Input and output power for sideband that was measured on when performing ACLR measurements.
Input Power
Output Power
Results based on
measurements
-19.2 dBm/channel
-5.8 dBm
Results based on
datasheet
-19.1 dBm/channel
-14.4 dBm
6.2
Discussion
The purpose of the parameter measurements made on ADL5375 was to evaluate
the performance on a LO frequency of 5.8 GHz. The measurements were then to
be used to construct a simulation model based on the data to gain the possibility
to test the performance from a system design perspective.
Measurement results presented in Figure 20 shows that the performance is relatively stable over frequency. The only parameter that shows an unstable behaviour when the frequency varies is the LO leakage. Plots of LO Leakage found
in the datasheet for ADL5375 show no similar behaviour. There are only plots
up to fLO =3.5 GHz as a highest frequency. It is not likely that a fluctuating
behaviour should arise because of the increased frequency. Then again, the plots
in the datasheet only covers up to 100 MHz in BB frequency. Looking at the
first 100 MHz of the collected data in this project, all parameters are extremely
stable, even the LO leakage. Comparing values in Table 8 one can see that the
parameter that differs least is the Conversion Gain. It only differs by 0.3 dB. The
difference in Sideband Suppression is likely explained by the fact that phase and
gain errors have been trimmed down in the datasheet measurements. The LO
Leakage is measured to a lower level than expected in the BB Frequency sweep.
This is likely a faulty result which can be seen if comparing to measurements
when sweeping BB Input Power, LO Frequency and LO Power (Appendix A.1).
The levels for the LO Leakage in these plots when conditions are the same as in
Table 7 are close to -18 dBm respectively. These values being so close together
and so much higher questions the validity of the value for LO leakage presented
in Table 8(b).
The values of Output IP 2 and Output IP 3 are likely not accurate due to the inaccurate measurement method used. The results show a more severe distortion
due to intermodulation in measured values than in datasheet values. That the
values would differ was predicted, but not that the non-linearities would appear
at such high power levels. It would have been more probable that the measurements showed a more linear behaviour. Since the output from the measurement
method used in (Section 5.2.2) only would show effects coming from non-linear
elements after the mixer cores and after summation of the I and Q branches, it
was expected to exhibit lower levels on the intermodulation products. Since this
is not the case it might indicate that a big source of the non-linear properties
of the modulator is situated after the combination of the I and Q branch. Measurements might not be as wrong as predicted. No conclusions can be drawn
without making more measurements and comparing the method with the more
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6.2
Discussion
6
RESULTS AND CONCLUSIONS
conventional method of feeding a two-tone quadrature signal to the inputs when
measuring.
Figure 21 shows a very early saturation compared to results available in the
datasheet. The value in Table 4 is lower than that in Table 8(a). This is however
consistent with the lower values of Output IP 2 and Output IP 3. This again
points to the possible validity of the measurements made when using the method
described in Section 5.2.2. Conclusions can not be drawn without making more
detailed comparisons between the methods. Since there is a possibility that
the non-linear properties are dynamic it would be better to use the method of
extrapolation to calculate IP 2 and IP 3, instead of the approximation presented
in Section 2.5.5.
The interesting aspect of the performance degradation is how much it will affect
the performance of a real TX system and if it can pass the limits set by 3GPP
for LTE as described in section 4.2. The most significant negative effect that a
modulator contributes with to the performance in a system is by adding nonlinearities and out of band distortion. Based solely on parameter values for
sideband suppression, LO leakage, IP2 , IP3 etc. it is difficult to say much about
how well the modulator will perform based solely on that information. Only
in comparison with a different modulator with other specifications or data that
is applicable for a different LO frequency it is possible to evaluate the actual
performance. Still it is not possbile to determine for sure whether or not it
is going to pass the limits set by a specific standard, only speculate. As a
consequence it is more interesting to analyse how the parameter values affect the
ACLR of the modulator and compare this to the LTE specification for ACLR
that can be found in European Telecommunications Standards Institute (2015),
that is seen in Figure 11. The Table shows that the same level, 45 dB, is set for
all adjacent channel center frequencies. Table 10 shows that the ACLR is higher
than 45 dB at all offsets that have been measured on. The ACLR gets better
when increasing the offset to ±10 MHz. The 3GPP specifications are valid for
the output of the antenna port of a base station. So to draw any conclusions
about if it could pass the requirements, one needs to at least take the PA into
account. The PA severely degrades the ACLR. Even though the ACLR lies below
the 3GPP level after the modulator, this is going to degrade when adding a PA.
There is usually some sort of Digital Pre-distortion (DPD) which compensates
for this, but the PA is still going to add non-linear effects.
It is of interest to see how hard the modulator can be driven without showing
to severe non-linear effects. In Figure 26 LTE spectrums are shown when using
measured values and values from the datasheet as parameters in the simulation
model. The input level has been backed off to decrease non-linear effects on
the output to a reasonable level. The PA stage of the TX is going to add a
lot of non-linearities. it is not uncommon to use DPD to compensate for this.
If the modulator is driven too hard and start to saturate the DPD will be less
efficient. Table 11 shows the output powers of the modulator. If these levels
are pre-amplified and then amplified by the PA there might be enough output
power for short range base stations, such as Home Base Stations or Local Area
Base Stations (Table 3). On the other hand the propagation of electromagnetic
Alexander Bergslilja
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6.3
Summary of Results
REFERENCES
waves is poor at such high frequencies as 5.8 GHz. This makes it more likely
that a design using that frequency band is intended for low range applications.
But is difficult to draw any conclusions based on the model. Other aspects that
need to be taken into account are the back-off needed for production variations,
variations because of temperature fluctuations, ageing effects etc.
The possibility to specify frequency dependant parameters in the AWR system
model MIXER B2 can be concluded to be unnecessary when modelling a modulator. Since no drastic behaviour can be seen at under 200 MHz BB signals,
there is no gain in using vector valued parameters. This simplifies the characterization process greatly by reducing the amount of measurements needed to
be performed. If a frequency dependency for some reason is required, it is easily
done by just measuring on a few points. VSS has a built in interpolation function
that interpolates the desired values.
6.3
Summary of Results
ˆ Detailed performance measurements were made of ADL5375.
ˆ A simulation model of ADL5375 was constructed, which takes non-linear
effects into account.
ˆ A wideband signal source was constructed with quadrature outputs that
follows the theory for quadrature signals.
ˆ The results collected after simulation with wideband signals suggest that
it might be possible to use ADL53575 in a 5.8 GHz band design.
6.4
Future Work
To improve the simulation model there are a number of things that can be
done. First of all the model would generate a more adequate response if the
measurements were remade in the correct way, described in section 2.5.5. It
would be beneficiary to implement a DAC in AWR to properly model the signal
generation of the TX. To add a model of a PA would make it possible to more
accurately predict whether the design could pass design specifications.
The only way to be absolutely certain whether the modulator is possible to use
is to build a physical prototype. The model should only be used as a guideline.
References
Agilent Technologies (2007). Agilent - I/Q Modulation Considerations for PSG
Vector Signal Generators.
Analog Devices (2014). DPG2 Wiki. url: http://wiki.analog.com/resources/
eval/dpg/dpg2 (visited on 03/02/2015).
Analog Devices Inc. (2013). ADL5320 Data Sheet.
— (2014). ADL5375 Data Sheet.
Cushing, Rick (2000). Single-Sideband Upconversion of Quadrature DDS Signals
to the 800-to-2500-MHz Band.
Dino Flore (2015). IEEE 802 Interim Session - 3GPP & unlicensed spectrum.
Alexander Bergslilja
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May 13, 2015
REFERENCES
REFERENCES
Ericsson (2014). Ericsson Mobility Report - November 2014.
European Telecommunications Standards Institute (2008). 3GPP TS 36.211 version 8.4.0 Release 8.
— (2015). ETSI TS 136 104 V12.6.0.
Goldsmith, Andrea (2005). Wireless Communications.
Keysight (2014a). 81160A Data Sheet.
— (2014b). Keysight Electronic Calibration - N4691.
— (2014c). N5242A Data Sheet.
Leon W. Couch, II (2013). Digital and Analog Communications System.
Monica Alleven (2014). “Confirmed: T-Mobile to launch unlicensed LTE at 5
GHz, possibly next year”. In: FierceWirelessTech.
Nash, Eamon (2009). AN-1039 - Correcting Imperfections in IQ Modulators to
Improve RF Signal Fidelity.
National Instruments (2014a). LTE Downlink Signal Source. url: https : / /
awrcorp.com/download/faq/english/docs/VSS%5C_System%5C_Blocks/
LTE % 5C _ DL % 5C _ TSIG . htm # lte % 5C _ dl % 5C _ tsig % 5C _ ref1 (visited on
03/05/2015).
— (2014b). Visual System Simulator: Getting Started Guide. url: https://
awrcorp . com / download / faq / english / docs / Getting % 5C _ Started /
Getting%5C_Started.htm (visited on 02/25/2015).
— (2015a). Visual System Simulator. url: http://www.awrcorp.com/products/
visual-system-simulator (visited on 02/23/2015).
— (2015b). Visual System Simulator. url: ttps://awrcorp.com/download/
faq/english/docs/VSS%5C_System%5C_Blocks/MIXER%5C_B2.htm (visited
on 02/26/2015).
Pozar, David M. (2012). Microwave Engineering.
Proakis, John G. and Masoud Salehi (2008). Digital Communications.
PTS (2014). Rapport av uppdrag att samla in statistik om tillgången till mobila
kommunikationsnät.
QUALCOMM (2013). Rising to Meet the 1000x Mobile Data Challenge.
Razavi, Behzad (2011). RF Microelectronics.
Rohde & Schwarz (2011). FSQ8 Data Sheet.
Tammy Parker (2013). “Qualcomm’s unlicensed LTE could crush carrier Wi-Fi’s
momentum”. In: FierceWirelessTech.
Tektronix (2013). App. Note - Baseband Response Characterization of I-Q Modulators.
Texas Instruments Inc. (2013). DAC3482 Data Sheet.
Alexander Bergslilja
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May 13, 2015
A
A
A.1
APPENDIX
Appendix
Plots
Figure 27: PUSB , PLSB , PLO,leak and GCONV vs. fLO .
Figure 28: Plot showing PUSB , PLSB , PLO,leak , SBS vs. PLO,input .
Alexander Bergslilja
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A.1
Plots
A
APPENDIX
Figure 29: Plot of Ptone1 , Ptone2 , PIM2 , PIM3 vs. LO frequency.
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