IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 38, NO. 2, MARCH/APRIL 2002
523
Stationary-Frame Generalized Integrators for Current
Control of Active Power Filters With Zero
Steady-State Error for Current Harmonics
of Concern Under Unbalanced and
Distorted Operating Conditions
Xiaoming Yuan, Senior Member, IEEE, Willi Merk, Herbert Stemmler, and Jost Allmeling
Abstract—The paper proposes the concepts of integrators for
sinusoidal signals. A proportional-integral (PI) current controller
using stationary-frame generalized integrators is applied for current control of active power filters. Zero steady-state error for the
concerned current harmonics is realized, with reduced computation, under unbalanced utility or load conditions. Designing of the
PI constants, digital realization of the generalized integrators, as
well as compensation of the computation delay are studied. Extensive test results from a 10-kW prototype are demonstrated.
Index Terms—Active power filter, current control, generalized
integrator, resonator.
I. INTRODUCTION
R
ESEARCH on current control for power converters has
been one of the most intensive activities recently [1].
When the reference current is a direct signal, as in the dc
motor drive, zero steady-state error can be secured by using
a conventional proportional-integral (PI) controller. When the
reference current is a sinusoidal signal, as in the ac motor
drive, however, straightforward use of the conventional PI
controller would lead to steady-state error due to finite gain at
the operating frequency. A synchronous-frame PI controller
was then proposed which guarantees zero steady-state error in
a balanced system [2], [3]. For an unbalanced system, a second
reference frame rotating in the opposite direction would also be
needed [4], [5] in order that the negative sequence component
is tracked with zero steady-state error.
When the reference current is a nonsinusoidal signal, as in
active power filters, a hysteresis or predictive controller is often
Paper IPCSD01-081, presented at the 2000 Industry Applications Society Annual Meeting, Rome, Italy, October 8–12, and approved for publication in the
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the Industrial Power Converter Committee of the IEEE Industry Applications Society. Manuscript submitted for review April 1, 2000 and released for publication November 13, 2001.
X. Yuan is with General Electric Corporate R&D-Shanghai, 200233
Shanghai, China (e-mail: xiaoming.yuan@geahk.ge.com).
W. Merk is with the Electrical Engineering Department, Burgdorf School of
Engineering, University of Applied Sciences Bern, 3400 Burgdorf, Switzerland
(e-mail: willi.merk@isburg.ch).
H. Stemmler and J. Allmeling are with the Power Electronics and
Electrometrology Laboratory, Swiss Federal Institute of Technology
Zurich, ETH-Zentrum/ETL, CH-8092 Zurich, Switzerland (e-mail:
stemmler@lem.ee.ethz.ch).
Publisher Item Identifier S 0093-9994(02)02666-X.
deemed a viable solution [6]. While the hysteresis controller is
simple and robust, it has major drawbacks in variable switching
rate, current error of twice the hysteresis band, and high-frequency limit-cycle operation [1]. Performance of the predictive
controller, on the other hand, is subject to accuracy of the plant
model as well as accuracy of the reference current prediction
[7], [8].
Recent contributions have been applying the synchronousframe PI controller for current control of active power filters [9],
[10]. The limitation consists of significant computation arising
from the need for multiple reference frames. To deal with unbalanced conditions, the number of reference frames and therefore
the computation must be doubled.
Revisiting the adaptive filter technique [11], which was recently introduced to converter control [12], [13], the core of the
filter is really an indirectly implemented integrator for a single
sinusoidal signal. Direct realization of the integrator allowing
for reduction of computation was already detailed in [14]. The
direct realization was also used in [15] for reference current
prediction in a synchronous frame dead-beat controlled active
power filter.
From the stationary-frame equivalent transfer matrix given in
[2], a synchronous-frame PI controller, without regard to the PI,
can be deemed an indirectly implemented integrator, however,
only for a positive sequence sinusoidal signal. Directly implementing the transfer matrix shall reduce the computation. The
present paper will further prove that, without the cross-coupling
terms [16]–[18], the new transfer matrix will represent an integrator for either balanced or unbalanced sinusoidal signals.
The paper proposes the concept of integrators for sinusoidal
signals. The concepts of ideal integrator for a single sinusoidal
signal and a stationary-frame ideal integrator for positive or negative sequence sinusoidal signals are explored. The concepts
of generalized integrator for a single sinusoidal signal and stationary-frame generalized integrator for balanced or unbalanced
sinusoidal signal are also clarified.
The paper will further report a PI current controller using
the stationary-frame generalized integrators for current control
of active power filters. Designing the PI constants, digital realization of the generalized integrators, as well as compensation of the computation delay will be studied. The instanta-
0096-9994/02$17.00 © 2002 IEEE
524
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 38, NO. 2, MARCH/APRIL 2002
neous-reactive-power (IRP) theory [19] will be used for reference current generation. The problem in IRP related to nonideal
point-of-common-coupling (PCC) voltage will be resolved by
using a sequence filter.
II. STATIONARY-FRAME IDEAL INTEGRATOR AND THE
STATIONARY-FRAME GENERALIZED INTEGRATOR
A. Ideal Integrator for a Single Sinusoidal Signal and the
Stationary-Frame Ideal Integrator
Similar to the direct signal case, for a sinusoidal signal
, the amplitude integration of this signal can be
. Defining further an auxiliary
written as
, the Laplace transforms of the three
signal
signals are
(1)
(2)
(3)
Then an ideal integrator for a single sinusoidal signal can be
configured as shown in Fig. 1(a), where is the resonant frequency of the integrator corresponding to the signal frequency.
,
Notice that, for an input signal with frequency deviation of
the corresponding output is given in Fig. 1(b). As long as
is sufficiently small and
can be approximated by
, the integration function holds. Using the ideal integrator
on the -axis signal and the -axis signal, respectively, a stationary-frame ideal integrator is then built as shown in Fig. 1(c),
which works without regard to the sequence between the -axis
and the -axis signals.
For a system where the -axis signal leads the -axis signal
by 90 degrees, as in a positive sequence system [2], [3], a
positive sequence ideal integrator can be established as shown
in Fig. 2(a). However, for a system where the -axis signal
always lags the -axis signal by 90 degrees, as in a negative
sequence system, the signs for the cross-coupling terms must
be exchanged to build a negative sequence ideal integrator, as
shown in Fig. 2(b). Fig. 2(c) corresponds to positive sequence
signal passing through the negative sequence ideal integrator,
while Fig. 2(d) corresponds to a negative sequence signal
passing through the positive sequence ideal integrator, each
producing negligible output.
B. Generalized Integrator for a Single Sinusoidal Signal and
the Stationary-Frame Generalized Integrator
Consider then the generalized integrator for a single sinusoidal signal [15], [17], [18], as shown in Fig. 4(a). The integrator output contains not only the integration of the input,
but also an additional negligible component. The corresponding
stationary-frame generalized integrator is shown in Fig. 4(b),
Fig. 1. (a) An ideal integrator for a single sinusoidal signal. (b) Sinusoidal
signal with frequency deviation of ! passing through the ideal integrator for
a single sinusoidal. (c) The corresponding stationary-frame ideal integrator.
1
which works without regard to the sequence between the -axis
signal and the -axis signal.
Observing that the stationary-frame generalized integrator in
Fig. 4(b) is the superposition of the positive sequence ideal integrator in Fig. 2(a) and the negative sequence ideal integrator
in Fig. 2(b), the cross-coupling terms in one are canceled by
the corresponding terms in the other. Observe further that the
output of an ideal integrator of a given sequence resulting from
the signal of the opposite sequence is negligible. Then a typical converter current control system using the stationary-frame
generalized integrator [17], as shown in Fig. 5(a), can be decomposed into a positive sequence signal system and a negative
sequence signal system, as shown in Fig. 5(b)/(d) and (c)/(e), reor a
spectively. Similarly, looking from a counter-clockwise
rotating reference frame, the positive sequence
clockwise
system or the negative sequence system is equivalent to a direct signal system as shown in Fig. 5(f) [2], [3] and (g), respecreference frame, the
tively.In a counter-clockwise rotating
equivalents of Fig. 2(a) and (c) can be represented by Fig. 3(a)
and (c), respectively [18]. Meanwhile, in a clockwise rotating
reference frame, the equivalents of Fig. 2(b) and (d) can
be represented by Fig. 3(b) and (d), respectively. Notice that in
and
are the proportional and integral constants,
Fig. 5
respectively. is the converter voltage gain, and and are
load parameters [17].
YUAN et al.: STATIONARY-FRAME GENERALIZED INTEGRATORS FOR CURRENT CONTROL OF ACTIVE POWER FILTERS
525
Fig. 3. (a) Positive sequence ideal integrator equivalent and (c) negative
sequence ideal integrator equivalent in the counter-clockwise (!) rotating
reference frame. (b) Negative sequence ideal integrator equivalent and (d)
positive sequence ideal integrator equivalent in the clockwise ( ! ) rotating
reference frame.
0
Fig. 2. (a) Positive sequence signal passing through a positive sequence ideal
integrator. (b) Negative sequence signal passing through a negative sequence
ideal integrator. (c) Positive sequence signal passing through a negative
sequence ideal integrator. (d) Negative sequence signal passing through a
positive sequence ideal integrator.
III. CURRENT CONTROL OF ACTIVE POWER FILTERS USING
STATIONARY-FRAME GENERALIZED INTEGRATORS
A. PI Current Controller for Active Power Filters Using the
Stationary-Frame Generalized Integrators
In the case of current control for active power filters, the
current error signal is nonsinusoidal which contains multiple
Fig. 4. (a) The generalized integrator for a single sinusoidal signal
[17]. (b) The corresponding stationary-frame generalized integrator. The
stationary-frame generalized integrator works without regard to the sequence
between the -axis signal and the -axis signal.
current harmonics. For each current harmonic of concern, a
corresponding stationary-frame generalized integrator must be
526
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 38, NO. 2, MARCH/APRIL 2002
Fig. 5. (a) Typical converter current control system using the stationary frame generalized integrator. (b) Stationary-frame positive sequence signal system
decomposition with the original input. (c) Stationary-frame negative sequence signal system decomposition with the original input. (d) Stationary-frame positive
sequence signal system decomposition with the positive sequence input. (e) Stationary-frame negative sequence signal system decomposition with the negative
sequence input. (f) Stationary-frame positive sequence signal system equivalent in the counter-clockwise rotating reference frame. (g) Stationary-frame negative
sequence signal system equivalent in the clockwise rotating reference frame.
YUAN et al.: STATIONARY-FRAME GENERALIZED INTEGRATORS FOR CURRENT CONTROL OF ACTIVE POWER FILTERS
527
Fig. 7. (a) Typical shunt active filter system with a ripple filter. (b) Current
control system using the proposed PI current controller.
Fig. 6. PI current controller for active power filters using the stationary-frame
generalized integrators.
installed. When multiple current harmonics are of concern, the
corresponding multiple integrators can be installed as shown in
Fig. 6. Resonant frequencies of the stationary-frame generalized
integrators correspond to the frequencies of the concerned current
harmonics.
B. Current Control System for Active Power Filters Using the
Proposed PI Current Controller
Fig. 7(a) shows a typical shunt active power filter
is connected to the
system. A voltage source inverter
point-of-common-coupling (PCC) through an interfacing
. The PCC is typically loaded with a nonlinear
inductor
.
represents the utility
diode or thyristor rectifier
short-circuit impedance. A ripple filter is installed consisting
. Several other ripple filter options are found
of , , and
in [22].
Fig. 7(b) shows the corresponding current control system
using the proposed PI current controller. The Instantaneous-Reactive-Power (IRP) theory [19] is applied for generation of
the utility current reference. Inverter dc-link voltage control is
realized by regulating the active power component in the utility
current through the IRP. LP is a low-pass filter for measuring
the dc-link voltage and for decoupling the dc-link voltage
control.
Notice that, instead of direct inverter current control that requires load current and inverter current measurements, the proposal uses a direct utility current control. Only the utility current
measurement is needed as a result.
Fig. 8. Sequence filter for extracting the sinusoidal positive sequence voltage
component from the unbalanced or distorted PCC voltage.
C. Sequence Filter for Utility Reference Current Generation
Under Unbalanced and/or Distorted PCC Voltage Conditions
It is known that the IRP is not able to work correctly under
unbalanced or distorted PCC voltage [20], [21]. The solution
proposed is a sequence filter as shown in Fig. 8. The filter is
connected in the control system as shown in Fig. 7(b).
Notice that the shaded part in Fig. 8 is a positive sequence
ideal integrator as previously given in Fig. 2(a). The filter always delivers the sinusoidal positive sequence component from
the PCC voltage. With this component fed to the IRP, the utility
current reference generated by the IRP will contain only a sinuin Fig. 8
soidal positive sequence component. The constant
controls the bandwidth and the response speed of the filter and
will not be detailed.
D. Identifying the Control Loops in the Control System
The current control system given in Fig. 7(b) contains the
following control loops:
528
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 38, NO. 2, MARCH/APRIL 2002
Fig. 10.
Spectrum of the load current (phase B).
3) Slow Utility Current Control Loop (Spontaneous): This
, and an output
loop has an input of the utility voltage
. Different from the first loop, this
of the utility current
loop involves also the second loop. Details of this loop
will be left for a future report.
E. Designing the PI Constants
Considering now the first loop, similar to the conventional direct signal control system, PI constants can be formulated based
on the open-loop transfer function. When the global load model
, this open-loop transfer
in Fig. 7 is simplified to
function can be written as
(4)
Fig. 9. Magnitude and phase characteristics of the open-loop transfer function
of the fast utility current control loop when the 5th, 7th, 11th, and 13th
harmonics are concerned. Proportional constant K = 0:06. Inverter voltage
gain G = 400. Global load model R = 10 , L = 5 mH. Integral constant:
(a) K = 0, (b) K = 1:2, and (c) K = 4:8.
1) Fast Utility Current Control Loop: This loop has an input
, an output of the
of the utility current reference
, with a perturbation from the utility
utility current
. As this is the fastest loop in the system, the
voltage
utility current reference is assumed independent of the
remaining system.
2) Slow DC-Link Voltage Control Loop: This loop has an
, and an
input of the dc-link voltage reference
output of the inverter dc-link voltage. The loop is always
made decoupled from first loop, by setting properly the
cut-off frequencies of the low-pass filters in the dc-link
voltage measurement, as well as well as in the IRP [19].
The magnitude and phase characteristics of the function when
the 5th, 7th, 11th, and 13th harmonics of concern are shown in
Fig. 9(a)–(c).
The following observations are recognized.
1) Comparing to the simple proportional system, as shown
in Fig. 9(a), adding the generalized integrators radically
changes both the magnitude and phase characteristics at
the concerned harmonic frequencies and the vicinities, as
shown in Fig. 9(b). At other frequencies, however, the
characteristics are not changed.
2) Infinite gains as well as phase leads or lags of up to 90
degrees are created at the concerned harmonic frequencies and the vicinities, adding to the characteristics of the
simple proportional system, as shown in Fig. 9(b). When
the net phase delays are less then 180 degrees, the system
is stable at these frequencies and zero steady-state error
is secured for the current harmonics at these frequencies.
3) Size of the integral constant determines the bandwidths
centered at the concerned harmonic frequencies during
which the characteristics are changed, as shown in
Fig. 9(c). For applications potentially with certain fundamental frequency variation, the integral constant may be
oversized accordingly.
4) Size of the proportional constant decides: 1) stability
of the simple proportional system; 2) order of current
harmonics that can be regulated without violating the
stability limits; 3) cross-over frequency and dynamic
response; and 4) reduction of the current harmonics at
other frequencies. Size of the proportional constant must
also observe the PWM constraint in term of the reference
signal spectrum in relation to the switching frequency
[17].
YUAN et al.: STATIONARY-FRAME GENERALIZED INTEGRATORS FOR CURRENT CONTROL OF ACTIVE POWER FILTERS
529
Fig. 11. Experimental waveforms (phase B) of the PCC voltage, load current, inverter current, and utility current as well as the utility current spectrum in the
cases of: (a), (b) no integrators is used; (c), (d) 1st, 5th, and 7th integrators are used; (e), (f) 1st, 5th, 7th, 11th, and 13th integrators are used; and (g), (h) 1st, 5th,
7th, 11th, 13th, 17th, and 19th integrators are used.
F. Digital Implementation of the Generalized Integrators and
Compensation of the Computation Delay
For digital implementation, the following discrete forms of
the relevant algorithms are used (sampling time ) [15]:
(5)
(6)
Various delays in the current control loop (mainly the computation delay) have been found to deeply affect the filtering
performance and system stability [15], [23]. The problem will
be resolved when the reference current can be predicted [15].
Different approaches for prediction are being pursued also in
predictive current control [7], [8].
530
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 38, NO. 2, MARCH/APRIL 2002
Rather than reference current prediction, this paper uses a
compromising scheme predicting only the harmonic components of the concerned frequencies in the modulating signal. As
the output of each generalized integrator is always sinusoidal,
it is thus easy to predict this output by moving the zero of the
transfer function in the -plane with the following formula for
one- or two-step prediction:
-
-
(7)
-
Fig. 12. Experimental waveforms (phase B) of the PCC voltage, load current,
inverter current, and utility current when R = 39 . The sequence filter is not
used.
-
(8)
Usually two-step prediction is needed. For steady state, as
the proportional path does not contribute to the harmonic components of the concerned frequencies in the modulating signal,
phase lags at the concerned frequencies resulting from the computation delay are thus compensated. For the dynamic state,
however, as the error signal contains also the harmonic components of the concerned frequencies, the effectiveness of the compensation will be reduced. Notice that phase lags at other frequencies during either steady state or dynamic state are not compensated. Stability conditions at these other frequencies must be
met by appropriately choosing the proportional constant.
IV. TEST OF THE ACTIVE POWER FILTER USING THE PROPOSED
PI CURRENT CONTROLLER
Fig. 13. Experimental waveforms (phase B) of the PCC voltage, load current,
inverter current, and utility current when the wire of phase A of the load is open.
It is also noticed that an additional controller for the neutral
potential of the NPC inverter used as an active power filter is
also implemented, which will not be detailed.
B. Experimental Results
A. Prototype Description
A 10-kW active power filter prototype was built in the laboratory using a neutral-point-clamped (NPC) inverter with a dc-link
V. A brake-chopper is connected to each
voltage of
of the two dc-link capacitor banks. Each capacitor bank consists of four 680- F/500-V electrolytic capacitors in parallel.
The Siemens 1200 V/50 A IGBTs are used, switching at 6 kHz
driven by the CONCEPT IHD280AN gate drivers. The inverter
mH. Utility inductor
interfacing inductor
mH. A diode rectifier with an inductor
mH in series
at the dcside is used to simulate a
with a resistor
current source type load [24]. No ripple filter is installed at the
moment.
The PI current controller proposed in Fig. 6 is used with the
and
. A conventional PI conPI constants
troller for direct signal with a proportional constant of 4 and an
integral constant of 4 is used for dc voltage control. Before the
dc voltage controller is a first-order low-pass filter with a cut-off
frequency of 125 rad/s. A fifth-order Butterworth low-pass filter
with a cut-off frequency of 125 rad/s is used in the IRP. Constant in the sequence filter is set to 10. The control system is
implemented in a TMS320C40 DSP using a dSPACE real-time
system. The rate of sampling is 167 S.
Fig. 10 shows the spectrum of the load current, and Fig. 11
shows the experimental waveforms of the PCC voltage, load
current, inverter current, and utility current as well as the utility
current spectrum in cases when different current harmonics are
concerned. When only proportion is used, as shown in Fig. 11(a)
and (b), very limited reduction of the load current harmonics is
obtained. On the other hand, when a specific current harmonic
is the focus and the corresponding integrator is installed, this
specific current harmonic will disappear from the utility current
spectrum, as demonstrated in Fig. 11(c)–(h). Effectiveness of
the proposed PI current controller in ensuring zero steady-state
error for the concerned current harmonics is verified. The results also show a very desirable feature in selective harmonic
elimination [25] for reducing the inverter rating while fulfilling
the relevant harmonics standard.
Fig. 12 shows the PCC voltage, load current, inverter current, and utility current when the sequence filter is not used
. In comparison to Fig. 11(g), the utility curand
rent becomes clearly degraded. This result proves the function
of the sequence filter in solving the problems of the IRP operating under distorted PCC voltage.
Fig. 13 shows the experimental waveforms of the PCC
voltage, load current, inverter current, and utility current under
YUAN et al.: STATIONARY-FRAME GENERALIZED INTEGRATORS FOR CURRENT CONTROL OF ACTIVE POWER FILTERS
531
REFERENCES
Fig. 14. Experimental waveforms (phase B) of the PCC voltage, load current,
inverter current, and utility current during: (a) no-load to full-load and (b)
full-load to no-load step change conditions. In either case, the transient lasts
for about 10 mS.
load unbalance conditions. The wire of phase A of the load
is open, representing an extreme unbalance of the load. It is
observed that the utility current is still sinusoidal. The operation
of the stationary-frame generalized integrator under unbalanced
conditions is verified.
Fig. 14 shows the PCC voltage, load current, inverter current,
and utility current under load transient conditions. The transient
in either case lasts for about 10 mS.
V. CONCLUSION
1) The concepts of integrators for sinusoidal signals offer
novel means for designing a converter control system involving nondirect signals.
2) The PI current controller using the stationary-frame generalized integrators ensures zero steady-state error for the
current harmonics of concern with reduced computation.
3) The sequence filter based on the positive sequence ideal
integrator resolves the problems of the IRP under distorted or unbalanced PCC voltage conditions.
4) The PI current controller using the stationary-frame generalized integrators can work under either balanced or unbalanced operation conditions.
ACKNOWLEDGMENT
The authors would like to acknowledge advice from
P. Daehler, G. Linhofer, Dr. P. Steimer of ABB Industrie AG,
M. Rahmani, Dr. D. Westermann of ABB High Voltage Technologies, Ltd., and Dr. T. Aschwanden of BKW, Switzerland.
[1] R. D. Lorenz, T. A. Lipo, and D. W. Novotny, “Motion control with
induction motors,” Proc. IEEE, vol. 82, pp. 1115–1135, Aug. 1994.
[2] C. D. Schauder and R. Caddy, “Current control of voltage-source inverters for fast four-quadrant drive performance,” IEEE Trans. Ind. Applicat., vol. IA-18, pp. 163–171, Mar./Apr. 1982.
[3] T. M. Rowan and R. J. Kerkman, “A new synchronous current regulator
and an analysis of current-regulated PWM inverters,” IEEE Trans. Ind.
Applicat., vol. IA-22, pp. 678–690, Mar./Apr. 1986.
[4] P. Hsu and M. Behnke, “A three phase synchronous frame controller for
unbalanced load,” in Proc. IEEE PESC, 1998, pp. 1369–1374.
[5] C. B. Jacobina, M. B. R. Correa, T. M. Oliverira, A. M. N. Lima, and
E. R. C. da Silva, “Vector modeling and control of unbalanced electrical
systems,” in Proc. IEEE IAS Annu. Meeting, 1999, pp. 1011–1017.
[6] S. Buso, L. Malesani, and P. Mattavelli, “Comparison of current control
techniques for active power filters,” IEEE Trans. Ind. Electron., vol. 45,
1998.
[7] F. Kamran and T. G. Habetler, “An improved deadbeat rectifier regulator
using a neural net predictor,” IEEE Trans. Power Electron., vol. 10, pp.
504–510, July 1995.
[8] D. G. Holmes and D. A. Martin, “Implementation of a direct digital
predictive current controller for single and three phase voltage source
inverters,” in Proc. IEEE IAS Annu. Meeting, 1996, pp. 906–913.
[9] C. D. Schauder and S. A. Moran, “Multiple reference frame controller
for active power filters and power line conditioners,” U.S. Patent
5 309 353, May 1994.
[10] M. Bojyup, P. Karlsson, M. Alakula, and L. Gertmar. A multiple rotating
integrator controller for active filters. presented at Proc. EPE Conf.. [CDROM]CD-Rom
[11] J. R. Glover, Jr., “Adaptive noise canceling applied to sinusoidal interferences,” IEEE Trans. Acoust., Speech, Signal Processing, vol. ASSP-25,
pp. 484–491, Dec. 1977.
[12] V. Blasko, “Adaptive filtering for selective elimination of higher harmonics from line currents of a voltage source converter,” in Proc. IEEE
IAS Annu. Meeting, 1998, pp. 1222–1228.
[13] S. Fukuda and S. Sugawa, “Adaptive signal processing based control
of active power filters,” in Proc. IEEE IAS Annu. Meeting, 1996, pp.
886–890.
[14] M. Padmanabhan, K. Martin, and G. Peceli, Feedback-Based
Orthogonal Digital Filters: Theory, Applications, and Implementation. Norwell, MA: Kluwer, 1996.
[15] O. Simon, H. Spaeth, K. P. Juengst, and P. Komarek, “Experimental
setup of a shunt active filter using a super-conducting magnetic energy
storage device,” in Proc. EPE Conf., 1997, pp. 1. 447–1. 452.
[16] Y. Sato, T. Ishizuka, K. Nezu, and T. Kataoka, “A new control strategy
for voltage-type PWM rectifiers to realize zero steady-state control
error in input current,” IEEE Trans. Ind. Applicat., vol. 34, pp. 480–486,
May/June 1998.
[17] D. N. Zmood and D. G. Holmes, “Stationary frame current regulation of
PWM inverters with zero steady state error,” in Proc. IEEE PESC, 1999,
pp. 1185–1190.
[18] D. N. Zmood, D. G. Holmes, and G. Bode, “Frequency domain analysis
of three phase linear current regulators,” in Proc. IEEE IAS Conf., 1999,
pp. 818–825.
[19] H. Akagi, Y. Kanazawa, and A. Nabae, “Instantaneous reactive power
compensating comprising switching devices without energy storage
components,” IEEE Trans. Ind. Applicat., vol. IA-20, pp. 625–630,
May/June 1984.
[20] S. Bhattacharya, D. M. Divan, and B. Banerjee, “Synchronous frame
harmonic isolator using active series filter,” in Proc. EPE Conf., 1991,
pp. 3.30–3.35.
[21] J. Sebastian, J. W. Dixon, G. Venegas, and L. Moran, “A simple frequency-independent method for calculating the reactive and harmonic
current in a nonlinear load,” IEEE Trans. Ind. Electron., vol. 43, pp.
647–653, Dec. 1996.
[22] S. Bhattacharya, T. M. Frank, D. M. Divan, and B. Banerjee, “Parallel
active filter system implementation and design issues for utility interface
of adjustable speed drive systems,” in Proc. IEEE IAS Annu. Meeting,
1996, pp. 1032–1039.
[23] W. Le Roux and J. D. Van Wyk, “Effectivity reduction of PWM converter
based dynamic filter by signal processing delay,” in Proc. IEEE APEC
Conf., 1999, pp. 1222–1227.
[24] F. Z. Peng, “Application issues of active power filters,” IEEE Ind. Applicat. Mag., pp. 21–30, Sep./Oct. 1998.
[25] J. Svensson and R. Ottersten, “Shunt active filtering of vector-current
controlled VSC at a moderate switching frequency,” in Proc. IEEE IAS
Annu. Meeting, 1998, pp. 1462–1467.
532
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 38, NO. 2, MARCH/APRIL 2002
Xiaoming Yuan (S’97–M’99–SM’01) received the
B.Eng. degree from Shandong University, China,
the M.Eng. degree from Zhejiang University, China,
and the Ph.D. degree from Federal University of
Santa Catarina, Brazil, in 1986, 1993, and 1998
respectively, all in electrical engineering.
He was with Qilu Petrochemical Corporation,
China, from 1986 to 1990, where he was involved
in the commissioning and testing of relaying and automation devices in power systems, adjustable speed
drives, and high-power UPS systems. From 1998 to
2001, he was a Project Engineer at the Swiss Federal Institute of Technology
Zurich, Switzerland, where he worked on flexible-ac-transmission-systems
(FACTS) and power quality. Since February 2001, he has been with GE Corporate R&D and is Manager of the Low Power Electronics Laboratory based
in Shanghai, China. His research interests are power electronics converters,
controls, and the applications.
Dr. Yuan received the first prize paper award from the Industrial Power Converter Committee of the IEEE Industry Applications Society in 1999.
Willi Merk was born near Schaffhausen, Switzerland, in 1945. He completed his studies in electrical
engineering at the University of Applied Sciences in
Winterthur in 1969.
He joined with Brown Boveri & Cie, Baden, a
former Asea Brown Boveri Company. From 1972
until 1981, he was a leading engineer in HVDC valve
design. Since 1982, he has been a Lecturer at the
University of Applied Sciences Berne, Switzerland,
for control systems, digital signal processing, and
industrial electronics.
Herbert Stemmler received the Dipl.-Ing. degree
in automation from the Techniche Hochschule
in Darmsdadt, Germany, in 1961 and the Ph.D.
degree in power electronics from the Technische
Hochschule in Aachen, Germany, in 1971.
He worked with Brown Bovery and ASEA-Brown
Bovery in Baden, Switzerland, from 1961 to 1991,
in the field of power electronics. From 1971 to 1991,
he was head of the department for development, engineering, test, and commissioning of large power electronics systems. In 1987 he was appointed Vice-President of this department. During these years, he worked with converter and inverter locomotives, 50–16 2/3 Hz interties, all kinds of large ac drives, reactive
power compensators, HVDC transmissions, low-power electronics, and standardized and tailor-made electronic control units. Since 1991, he has been Professor of Power Electronics at the Swiss Federal Institute of Technology Zurich,
Switzerland, and Head of the Power Electronics and Electrometrology Laboratory. At the time of this writing, 15 doctoral projects have been completed
and 7 projects ongoing in traction, motor drives, flexible-ac-transmission-systems (FACTS), solar energy systems, uninterruptable power supplies, matrix
converters, and fuel cell vehicles.
Jost Allmeling was born in Hamburg, Germany, in
1972. He received the Dipl.-Ing. degree in electrical
engineering from the University of Technology
(RWTH) Aachen, Germany, in 1996 and the
Ph.D. degree from the Swiss Federal Institute of
Technology (ETH) Zurich, Switzerland, in 2001.
Since 1996, he has been employed as a Research
Assistant at the Power Electronics Laboratory and the
Power Systems Laboratory of ETH Zurich. His main
research interests are active filters for the medium
voltage grid and the simulation of power electronics
systems.
Dr. Allmeling was awarded a second prize in 1992 in the nationwide youth
research competition Jugend-forscht having investigated the security of smartcards.