Inductive-Boost Switched-Capacitor DC/DC
Converter for Maximum Power Point Tracking
Photovoltaic Systems
Jul-Ki Seok
Ali Gandomkar
Senior Member, IEEE
Student Member, IEEE
Power Conversion Lab.
Yeungnam University
Gyeongsan, Korea
doljk@ynu.ac.kr
http://yupcl.yu.ac.kr
Abstract – With the advent of the photovoltaic (PV) energy
source that has a variable and low dc output voltage, a high
step-up dc/dc converter is necessary for grid-connected power
conditioning systems. These power conversion systems are
required to comply with strict technical and regulatory criteria
to ensure a safe, reliable and efficient operation of the overall
system. In this regard, this paper proposes an isolated and
non-isolated high gain switched capacitor dc/dc conversion
system. This converter can offer high efficiency with the
possibility of the adjustable impedance matching for PV
systems. It can also achieve the high voltage gain without
requiring high values of duty cycles. Soft-switching techniques
are adopted to achieve minimal switching losses and maximum
system efficiency. The presented approach can be applied to
switched capacitor converters with any number of modules to
improve the functionality of the closed loop control system.
Furthermore, a capacitor soft charging function has been
taken into account to suppress an inrush current at the start of
the operation, as well as to avoid an extra voltage and current
stress for power electronic components.
Index Terms-- Capacitor soft charging, galvanic isolation, high
voltage gain, inductive-boost switched-capacitor converter, PV
applications, soft switching techniques.
I.
INTRODUCTION
Photovoltaic (PV) power supplied to the utility grid is
becoming increasingly important and prevalent to meet the
world’s power demands. For low voltage generation systems
equipped with PV arrays, a power conditioning system
(PCS) that complies specific design criteria is desired to
convert a variable and low dc voltage to a higher dc voltage
before making it to an ac grid voltage, as shown in Fig. 1 [13]. A matching device is required to assist the system
operation stays at a maximum power point, which varies
with the atmospheric conditions. The impedance matching
PCS design with a voltage-transfer function depending on a
duty cycle is indispensable to achieve the maximum power
transfers from the PV array to the load [4]. For safety
considerations, most PCSs have a galvanic isolation, either in
the dc/dc converter or on the ac output side [5, 6]. An
embedded transformer in a high-frequency dc/dc converter
978-1-4799-5776-7/14/$31.00 ©2014 IEEE
makes the grounding of the PV modules easier, smaller, and
more efficient. The low voltage provided by the PV array is
always associated with the high current in the primary part of
the PCS converter. These high currents lead to high power
losses in the semiconductors and low efficiency. Therefore,
an isolated step-up dc/dc converter with a high gain and
efficiency is needed to provide a stable dc link to the power
inverters.
Switching power converters have the potential to achieve
a desired voltage conversion ratio with high efficiency [7].
Capacitive means and inductive means are two topological
classes of switching converters. Both topologies are
applicable and mandatory for maximum energy extraction in
the renewable energy sources. These topologies form the
backbone and bases of a vast proliferation of dc/dc
converters, all purporting to offer various advantageous
features and attributes. The classical switched-inductor (SL)
boost converter is theoretically qualified to attain an ultimate
voltage gain as the duty ratio approaches unity. Although it
can be thought of as a preferred choice to achieve the high
voltage gain and perform the impedance matching operation,
the maximum gain is still practically constrained by different
circuit imperfections. An extreme duty cycle has a
destructive effect on the dynamic performance and causes
serious reverse recovery problems [5].
Consumption on
Utility grid
Consumption
meter
Photovoltaic
panels
Production
meter
dc/dc/ac
Transformer
converter
Power conditioning system
Fig. 1. Grid-connected photovoltaic power system.
Dc/dc switched capacitor (SC) circuits can provide a high
voltage gain based on the two variable orientations of
5296
switches and capacitors [8-10]. Although the performance of
the SC converters is found to be superior to the conventional
SL converters for medium to high conversion ratios [8], they
still suffer from the high current spikes and conduction loss.
In order to alleviate the above-mentioned limitations, the
resonant SC (RSC) converter was proposed [10]. However,
the voltage transfer function is independent of the duty cycle,
and the dc voltage gain is usually predetermined by the
circuit structure [11]. These constraints are further
aggravated by a limited regulation for changes either in the
load value or line voltage. Therefore, existing RSC
converters may not be an adjustable impedance matching
circuit because of its poor output voltage regulation similar
to that of SC converters [12, 13]. For both introduced SC and
RSC converters, a high inrush current flows through the
circuit to charge the discharged capacitors when the power
supply unit is switched on. In most publications on SC and
RSC converters, it is usually supposed that they operate at
the steady-state condition without any speculation on the
process of the pre-charging capacitors and its impact [14].
The pre-charging methods for the converter capacitors were
presented in [14-16]. These methods need an independent
control of the switches, or require an extra circuit for the precharging process. One possible improvement might be the
inclusion of the magnetic devices in SC converters.
A.
IB-SC CONVERTER CONFIGURATION
General Topology
The general circuit topology for a multiple module-stacks
IB-SC converter and its overall control block diagram are
shown in Fig. 2(a). The input stage comprises of an input
voltage source (Vin), two switches with a time variable turn
ON/OFF period, and a series inductor. A series inductor (Li)
can be used to convert a variable voltage source into a
constant dc voltage source.
Di
Sn S1S1
input
stage
Cr D1
SPB
SPB SPT
S1
Delay
Logic
3
S1
C
Sn
Cr
1
(b) Commercial
IGBT module.
3
Dn
2
1
s
Sn
Ki
Kp
Crn
1
Lr
Do
Co
Vo
RLoad
IGBT
nth
module
stack
3
2
output
stage
Io
Vo*
Vo
IGBT
module
stack
D2
2
_
II.
Li
SPT
Vin
+
This paper presents an inductive-boost switchedcapacitor (IB-SC) dc/dc converter. The proposed topology
takes the chance to gain advantages from RSC and SL
converters. This converter aims to achieve a high voltage
gain and enhancement of output voltage regulation with a
capacitor soft-charging process. General topology, details of
the circuit operation, and control strategy of the three-stage
IB-SC topology are described. A newfound duty ratio control
method for the regulation of the output voltage is introduced
to improve the performance and increase the output voltage
gain. A soft-switching technique with this duty cycle control
strategy is also developed to increase the efficiency. The
drawback of the conventional boost converter can be
overcome by the proposed topology which has a wide turn
OFF period and high voltage step-up ratio. Once the input
inductor is replaced by a 1:1 transformer in the non-isolated
design, an isolated IB-SC converter can be derived without
using any rectifying stage. Both proposed designs integrate
an intrinsic capacitor charging function without either
auxiliary equipment or special control implementation. The
feasibility of the two-stage IB-SC converter is validated on a
laboratory test bench.
Iin
v_sen
1
(c) IGBT module
equivalent circuit.
(a) General circuit topology
with closed loop controller.
Fig. 2. Topology of the proposed IB-SC converter
The input stage is interfaced through silicon-controlled
rectifiers or diodes to the insulated gate bipolar transistor
(IGBT) module stacks. The module stack enables the
converter components modular, which makes it easier to
replace the stack rather than the repair of a full-cabinet
converter.
The proposed converter has a voltage/current transfer
function, relating the input voltage Vin and current Iin to the
output voltage Vo and current Io. Analysis to derive the
voltage transfer function is based on the input inductor
voltage of Li over one switching period Ts under steady-state
duty cycle D, and input/output conditions.
A PI controller is implemented in the IB-SC converter to
provide desired output voltage regulation under input voltage
and load variation conditions. To achieve the maximum duty
cycle and zero-current-switching (ZCS), a delay time is
required in the module switches pulse-width-modulation
(PWM) operation.
Fig. 2(b) and (c) show the layout of a commercial IGBT
module and its equivalent circuit symbol, respectively. The
equivalent circuit symbol represents that the high power
density and conversion ratio can be achieved with a simple
and compact structure (see blue dotted frames in Fig. 2(a)).
B. Principles of Operation
The proposed two-module stack IB-SC converter is
illustrated in Fig. 3(a). Each module (see green dotted frames
in Fig. 2(a)) stack is composed of one pair of IGBT modules,
5297
one capacitor, and single diode in a stacked arrangement.The
individual modules are connected in parallel.
The input inductor (Li) on the input stage side plays the
same role as in the traditional SL converter. The capacitor
voltages of Cr can be charged higher/lower than the input dc
voltage (Vin) by Li. This paper focuses on the boost mode
operation, that is, power flows from Vin (low voltage side) to
the high voltage dc bus (Vo) to inverter(s). Under the boost
mode, each low-voltage switch (SPB and SPT) is controlled as
an active switch, while the diode (Di) of the input stage
conducts as a complementary switch. The second module
stack is connected in parallel with the output stage.
Therefore, Crs are discharged in series, stacking on the
output capacitor Co.
In this paper, subscripts “B” and “T” denote the
corresponding variables to the circuit components at the
bottom and top sides, respectively. To develop the proposed
model, the following assumptions are made:
1) The switching frequency is less than the resonant
frequency to achieve the ZCS;
2) The input voltage (Vin) is an ideal dc voltage source
and the load is modeled by a resistor (RLoad).
Fig. 3 and 4 show the circuit diagram and the key circuit
waveforms, respectively. The steady-state analysis of the IBSC converter is performed as follows:
1) Mode I [t0~t1] [see Fig. 3(b)]
In the beginning of this mode (t=t0), SPB and SPT are
turned ON. Di is revers-biased by Vin, whereas D1, D2, and
D3 are reveres-biased by VCr, 2VCr, and 3VCr, respectively
During this period, the inductor current of Li increases
linearly as
V
i L i ( t ) = in t.
(1)
Li
Three Crs in series with Lr and Co from the former mode
are discharged to the output stage with the half-cycle
resonant shape. Then, the current through Lr is dropped to
zero after the half-resonant period (refer to Fig. 4(g)). At t=t1,
S1T and S2T become OFF under the zero-current condition.
2) Mode II [t1~t2] [see Fig. 3(c)].]
In this mode (t=t1), S1T and S2T are turned OFF whereas
SPB and SPT are still ON (see Fig. 4(a), (b), and (c)). During
this period, the inductor current of Li is still increasing and
Di is still reverse-biased by Vin, while D1, D2, and D3 are
reverse-biased by VCr. The capacitor voltages of Crs are
unchanged and Co supplies the power to the load as shown in
Fig. 4(i), and 4(j).
3) Mode III [t2~t3] [see Fig. 3(d)]
At t=t2, a gate signal is applied to trigger S1B and S2B
(shown in Fig. 4(b)), and the voltage across these switches
drops to zero. However, there is no current flowing through
the switches because D1, D2, and D3 are still reverse-biased
and open circuited, as mentioned in Mode II. During this
period, the inductor current is increased to the peak value
(refer to Fig. 4(d)) and the load voltage is supplied by Co.
Two pairs of switches (S1T, S2T and S1B, S2B) are operated in
a complementary clock signal with Ts 2 .
4) Mode IV [t3~t4] [see Fig. 3(e)]
In this mode, SPB and SPT are turned OFF at t3=DTs (see
Fig. 4(a) and (e)). The inductor current diverts from SPB and
SPT to the IGBT module stack, and forces the diodes to
conduct. S1B and S2B are naturally turned ON and start to
conduct in the zero-voltage condition. A so-called quasi
zero-voltage-switching (ZVS) operation is created and the
switching losses are minimal in this mode (see Fig. 4(b) and
(f)). The hatched area in Fig. 4(f) highlights the nonoverlapping voltage and current waveforms for the efficient
switching operation. Under this situation, the capacitor
voltages of Crs are applied across Li in the opposite polarity.
Accordingly, the inductor current declines in a nonlinear
manner as
1
(2)
i L i ( t ) = − ∫ v C r dt.
Li
Then, (2) can be rewritten as
i Li ( t ) = −
1
VC r ( t − DTs ).
Li
(3)
When the inductor current is dropped to zero at the end
of D1Ts < t 4 , the accumulated energy is completely
transferred to Crs, as shown in Fig. 4(d). At t=t4, the diodes,
S1B, and S2B are turned OFF under the zero-current condition
since an appropriate phase shift is always applied to the
switching mechanism to provide the ZCS.
In this period, Crs are connected in parallel to Li for
Ts 2 and each Cr is charged to the boosted voltage higher
than Vin. The diode currents of D1, D2, and D3 are determined
by the circuit equation of the parallel connection as follow:
1
i Dn ( t ) =
iL
(4)
n +1 i
where n is the number of the IGBT module stack.
5) Mode V [t4~t5] [see Fig. 3(f)]]
In this mode, all the switches and diodes are OFF and the
capacitor voltages of Crs are unchanged. The output
capacitor voltage is discharged to the load.
6) Mode VI [t5~t6] [see Fig. 3(g)]
At t=t5, S1T and S2T are turned ON while other switches
are OFF. It can be seen from Fig. 4(g) that the current
through S1T and S2T is increased by a soft-switching
operation with the half-cycle resonant shape. It should be
noted that Crs and Co are connected in series for Ts 2
resonating with Lr. In this mode, Do is responsible for the
positive resonant current conduction and opposite energy
transmission prevention.
5298
SPT
Vin
Li IL
i
ISPT
Di
Vin
SPT
ISPB
ICr SPB
C r D1
Di
C r D1
ID2
1
3
2
ISPT
D3
Vin
Cr
S2B
1
Co
ILr
Lr
Do
RLoad
D3
2
Cr
Cr
S2B
1
ILr
Lr
Do
Vo
(c) Mode II [t1~t2].
Li IL
i
ISPT
Di
Vin
SPT
SPB
3
ICr
S1T
ID2
SPB
S1B
1
3
S2T
ID3
vin
Li IL
i
SPT
Di
ISPB
C r D1
S1B
S2T
ID3
3
Cr
D3
Cr
1
ILr
Lr
Do
Co
IL r
Lr
Do
RLoad
_
(e) Mode IV [t3~t4].
(f) Mode V [t4~t5].
Fig. 3. Circuit diagram and topological states of two-module stack IB-SC converter.
Vo
(g) Mode VI [t5~t6].
+
Vo
ID3
S2B
+
Vo
+
RLoad
D2
2
Cr
RLoad
ID2
1
3
D3
Co
Cr D1
2
Cr
1
5299
S1T
D2
S2B
ILr
Lr
Do
ICr SPB
ID2
2
Cr
Co
Di
1
3
D3
1
ISPT
2
Cr
S2B
Li IL
i
3
D2
2
(d) Mode III [t2~t3].
ISPB
Cr D1
ILr
Lr
Do
Co
Vo
S1B
S2T
ID3
Vo
2
_
(a) Proposed two-module stack IBSC converter.
D3
ISPB
+
Vo
Cr
1
3
RLoad
iDi
S1T
D2
RLoad
SPT
ICr
ID2
RLoad
(b) Mode I [t0~t1].
ID3
3
+
Cr
S2T
ID3
+
D2
S1B
Co
+
S2T
Co
Cr D1
2
Cr
1
I Lr
Lr
Do
S1T
D2
S2B
3
2
S1B
Cr
1
SPB
ICr
ID2
2
_
S1T
SPB
Di
Cr D1
1
3
D3
S2B
_
ICr
S2T
ID3
2
ISPB
L i IL
i
SPT
ISPB
_
Vin
Vin
2
S1B
1
3
S2T
Li ILi
SPT
Di
3
_
ISPT
S1T
Cr
S1B
SPB
ICr
ID2
D2
2
ISPT
ISPB
3
S1T
Li IL
i
_
ISPT
D1Ts
DTs
mode
IV
Ts
V
~ ~ ~
(a)
(b)
(c)
I II III
VI
SPT&SPB
t
t
t
S1B&S2B
S1T&S2T
IS1B
t
~
(e) ISPB &SPT
~
ILi
(d)
t
ZCS
Turn-on
&S2B
VCE
t
~
(f) &
VS1B &S2B
ZCS
Turn-off
(g) IS1T &S2T
~
t
VCr
(j)
Vo
~
(i)
t4 t5
t6
Fig. 4. Key current and voltage waveforms of the two-module stack IB-SC
converter.
Applying the charge balance principle to Co leads to
−∫
Ts 2
0
Io dt = −∫
Ts
Ts 2
(iL r ( t ) − Io ) dt.
(5)
Equation (5) can be rewritten as
i Lr (t) = i Cr = −πIo sin(ωr t)
(6)
where ωr (ωr=2π/Ts) is resonant frequency ( 1 / C r L r ).
From (3) and (6), the total capacitor current of Cr yields
⎧
0
0 ≤ t ≤ DTs
⎪⎪ 1
i C r (t ) = ⎨
VC r ( t − DTs ) DTs ≤ t ≤ (D + D1)Ts . (7)
⎪ (n + 1)Li
− πIo sin(ωr t )
(D + D1 )Ts ≤ t ≤ Ts
⎩⎪
The capacitor voltage of Cr can be obtained by applying
the volt-sec balance condition as
1
⎛ R D2Ts ⎞ 2
⎟ .
= ⎜ Load
(8)
Vin ⎜⎝ 2(n + 1)L ⎟⎠
Therefore, the output voltage gain can be written as
VC
r
Vo = (n + 1)VCr .
C. Isolated IB-SC Converter
Grid connection is the best way to fully exploit the
renewable energy at distribution level. However, the voltage
generated by the PV generators and fuel cells cannot be
directly connected to the grid because the fuel cell and PV
array operate in the voltage range from 25 to 45 Vdc and the
grid voltage is 230 Vrm. The PCS must provide a galvanic
isolation between the full cell or PV array terminals and the
utility grid in order to guarantee the system safety. As
shown in Fig. 5, an isolated IB-SC converter is derived if Li,
Di, and SPB are replaced by the transformer in the nonisolated IB-SC converter. In this circuit, the magnetizing
inductance (Lm) plays the same role as Li does in the nonisolated circuit. The secondary winding of the transformer is
connected to the module stacks by the reverse polarity to
prevent the magnetic core saturation. This connection also
provides a current direction and voltage polarity for the
proposed circuit to operate at the six steady-state operating
modes, as explained in section II.
D. Feedback Control Mechanism of the System
Fig. 6 shows an overview of the feedback control
mechanism used in this paper. The constructed control signal
is fed into the positive input of a comparator and the negative
input is connected to a triangular wave to provide a variable
PWM for the SBP (and SPT in the non-isolated circuit). On the
other part, the same triangular wave is compared with the
constant value C=0.5 to create two complementary pulses for
two pairs of switches (S1T, S2T and S1B, S2B). An appropriate
phase shift is always applied to these complementary pulses.
This phase shift increases the capability of converter
operation with a high duty cycle, and also distributes the
currents of S1B and S2B between the pulse margins to achieve
ZCS and ZVS. The amount of phase shift is determined by
the controller, which is directly proportional to the duty cycle
variation.
~
t0 t1 t2 t3
circuit parameters and operating conditions. The operation
limit can be extended by increasing the number of module
stack and the duty ratio.
III. SIMULATION AND EXPERIMENTAL RESULTS
A. Performance Evaluation
The two-module stack IB-SC converter performance was
simulated
in
PLECS
Blockset
within
the
MATLAB/Simulink environment. Table I lists the circuit
component values of the proposed converter. The simulated
waveforms of the switching patterns and the
inductor/capacitor currents are depicted in Fig. 7(a), (b), (c),
and (d) from the top to bottom. From the comparison of Fig.
7(b) and 7(d), it can be observed that corresponding
switches (S1B, S2B and S1T, S2T) can be turned ON and OFF
under the zero-current condition.
(9)
From (8) and (9), the voltage gain of the proposed
converter depicts that the output voltage increases as the duty
ratio D increases. This gain is also a nonlinear function of
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TABLE I.
CIRCUIT PARAMETERS AND COMPONENTS VALUE OF
PROPOSED TWO-MODULE STACK IB-SC CONVERTER.
ISP
Vin
SP
Lm
1: 1
Cr D1
ICr
S1T
3
ID2
D2
2
Cr
S1B
S2T
Parameters
Symbol
Value
Unit
Input voltage
Vin
25
V
Input inductor
(magnetizing inductor)
Resonant capacitor
Output capacitor
Resonant inductor
Load
Resonant frequency
Switching frequency
Li
(Lm)
Cr
Co
Lr
RLoad
fr
fs
58
µH
400
200
70
200
2100
2000
µF
µF
µH
Ω
Hz
Hz
ILm
1
3
1
ID3
0
D3
2
1
Cr
S2B
1
Co
SPT& SPB
(a)
(b)
ILr
Lr
Do
S1B& S2B
S1T& S2T
0
(c)
50
ILi [A]
0
RLoad
(d)
+
Vo
_
Fig. 5. Circuit diagram of the proposed isolated two-module stack IB-SC
converter.
Constant
value
C
Delay
logic
Sn
o
Vo
Ki
1
s
ILr [A]
Time [200µs/div]
Fig. 7. Current waveforms of two-module stack IB-SC converter.
Triangular
wave
V*
0
3
(e) 1.5
0
S1
S1
ICr[A]
16
27
SPB
SPT
Vin[V]
(a) 25
23
84.0
Kp
VCr [V]
(b) 83.3
Fig. 6. Derived feedback control mechanism for the IB- SC converter.
Therefore, the switching losses can be minimal in both ON
and OFF instants. Fig. 7(e) shows the current through Lr
flows to the output stage during the half-resonant period. Fig.
8(a), (b),and (c) show the voltage waveforms of the input
voltage, resonant capacitors, and the output from the top to
bottom.
The controlled output voltage is set to 250 V and the low
ripple voltage is observed. It can be seen from the
comparison of Fig. 7 (c) and 8(b) that the capacitor voltages
of Crs increases while the inductor current drops from the
peak to zero. The output voltage increases and capacitor
voltages of the series-connected Crs are discharged as shown
in Fig. 8(b) and (c).
82.6
251
Vo [V]
(c) 250
249
Time [200µs/div]
Fig. 8. Voltage waveforms of two-module stack IB-SC converter.
B. Initial Charging Process
In the presented converter, Crs and Co are charged to the
desired voltage level without the inrush current during the
initial start-up period. Fig. 9 shows the viable topological
states of the initial charging when Crs and Co have no charge.
Then, the converter continues the operation in the similar
way as the steady state operation shown in Fig. 3. The
capacitor charging process is divided into three steps as
followings:
5301
ISPT
SPT
Vin
Li ILi
ISPT
Di
Vin
iSPB
ICr
S1T
SPB
Cr D1
S2T
IDi
ICr
3
S1T
ID2
SPB
S2T
Co
0
VGE of
SPB & SPT (c)
[5 V/div]
0
Cr D1
D2
ID3
D3
Cr
1
ILr
Lr
Do
IS PB& SPT (d)
[15 A/div]
0
IC r (e)
[5 A/div]
0
IL r (f)
[0.6 A/div]
0
1
3
S2B
IL r
Lr
Do
Co
RLoad
Fig. 11. Experimental current waveforms of two-module stack IB-SC.
RLoad
_
_
(a) Mode I, II, and III [t0~t3].
Vo
+
+
Vo
VGE of
S1B & S2B (b)
[5 V/div]
2
Cr
1
ISPB
Cr
D3
S2B
0
ID2
S1B
ID3
2
VGE of
S1T & S2T (a)
[5 V/div]
2
Cr
1
3
Li IL
i
Di
3
D2
2
S1B
SPT
(b) Mode IV [t4~t4].
Fig. 9. Pre-charging states of two-module stack IB-SC converter.
40
ISPB[A]
Fig. 12. Experimental voltage waveforms of two-module stack IB-SC.
(a) 20
0
40
IDi [A]
(b) 20
0
40
VC r[V]
(c) 20
Fig. 13 Experimental waveforms of the Proposed precharging method. (a)
switch currents of SPB and SPT. (b) capacitore voltage of Cr.
0
90
60
(d)
30
0
Vo[V]
0
1
2
3
Time [ms]
Fig. 10. Transient waveforms of the IB-SC converter. (a) Switch currents of
SPB and SPT. (b) Diode current of Di. (c) Capacitor voltage of Cr. (d) Output
voltage.
1) Energy Transfer Period [see Fig. 9(b)]
When SPB and SPT are turned OFF at t=DTS, the energy
transfer period begins without the occurrence of the inrush
current and transient over voltage. The inductor or diode
current, as shown in Fig. 10(b), continues to decrease and the
accumulated energy is transferred to the capacitors, thereby
increasing the capacitor voltages (refer to Figs. 10(c)).
1) Energy Build-up Period [see Fig. 9(a)]
When the power is first turned ON, SPB and SPT are
closed and the energy build-up period starts. Due to the
polarity of Cr, diodes Di, D1, D2, and D3 are turned OFF
(anodes are grounded by SPB). A linearly-increasing current
flows through Li and active switches as in (1) (see Fig.
10(a)). The energy is piled up in Li with the increasing
current.
2) Recess Period
When the diode current is reduced to zero, the
accumulated energies are completely transferred to the
capacitors. Then, the diodes are turned OFF while S1T and
S2T are flipped to transfer the energy to the load in order to
increase the output voltage, as shown in Fig. 10(d). By
repeating the above operational cycle as shown in Fig. 3, the
charging voltage will be gradually increased along with the
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elevation of the capacitor voltages. This can be never
achieved in existing SC converters.
C. Laberatory test results
To evaluate the performance of the proposed converter, a
prototype IB-SC system was built. The selected passive
component values are the same as those of the simulation.
Figs. 11(a), (b), and (c) are captured to show the gateemitter voltages of three pairs of switches (S1T, S2T, S1B, S2B,
and SPB, SPT) respectively. Figs. 11(d), (e), and (f) show the
switch current of SPB and SPT, capacitor currents of Crs, and
inductor current of Lr from the top to bottom. When Crs are
discharged to Co, the currents through S1T and S2T rise and
fall in a sinusoidal manner. As a result, it is worth
mentioning that two pairs switches (S1T, S2T and S1B, S2B) are
turned ON and OFF under the soft switching techniques.
Fig. 12 shows the experimentally measured voltage
waveforms of the capacitors and load at the nominal output
power. Fig. 12(b) illustrates that Vo has the low voltage
ripple, as mentioned in simulation. It can be seen that the
experimental results closely match the analysis and
simulation results of the proposed topology in Section II and
sub-section III-A, respectively.
The pre-charging process is proved in Fig. 13, when the
converter operates at any desired duty cycle and nominal
load. As shown in Figs. 13(a) and (b), the capacitors voltages
Crs increase proportionally and overcurrent/voltages are
rarely found in any component of the converter.
In this test (n=2), the output voltage gain is set to 10
times of Vin. The voltage gain can be increased more by the
selection of the converter component values and a higher
duty cycle as (8) and (9).
IV. CONCLUSIONS
This paper proposes an isolated/non-isolated inductiveboost switched-capacitor converter for PV applications. This
converter can be a preferred choice for the impedance
matching and MPPT operation by closed loop output voltage
regulation. Soft switching techniques are presented by the
proper design of the duty cycle control strategy. A high
voltage gain can be achieved by the wide turn off period of
this converter compared to the conventional SL converter.
Moreover, the high inductor current is divided by the number
of module stacks. This current division alleviates the diode
conduction losses and equivalent series resistance effect of
the components when the converter operates with the high
duty cycle. The inductor size is significantly decreased
compared to the conventional boost converter. This paper
offers a cost-effective capacitor soft charging method to
prevent the inrush current during the start-up period, leading
to enhancement of the transient stability and increase of the
component lifetime. The presented converter offers the
potential to replace the module stacks shortly when the
individual module is failed. Both the theory and simulation
are substantiated by the practical results.
ACKNOWLEDGMENT
This work was supported by the National Research
Foundation of Korea (NRF) grant funded by the Korea
government (MSIP) (2010-0028509).
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REFERENCES
J. P. Lee, B. D. Min, T. J. Kim, D. W. Yoo, and J. Y. Yoo, “A novel
topology for photovoltaic dc/dc full-bridge converter with flat
efficiency under wide PV module voltage and load range,” IEEE
Trans. Power Electron., vol. 55, no.7 pp. 2655-2663, Jul. 2008.
S. Y. Park, C. C. Liang, and J. S. Lai, “A wide-range active and
reactive power flow controller for a solid oxide fuel cell power
conditioning system,” IEEE Trans. Power Electron., vol. 23, no. 6,
pp. 2703-2709, Nov. 2008.
W. Y. Choi, J. S. Yoo, and J. Y. Choi, “High efficiency dc-dc
converter with high step-up gain for low PV voltage sources,” Proc.
of IEEE ECCE Asia Conf. (ICPE & ECCE), 2011, pp. 1161-1163.
K. H.Hussein, I. Muta, T. Hoshino, and M. Osakada, “Maximum
photovoltaic power tracking: an algorithm for rapidly changing
atmospheric conditions," Proc. of IET, vol. 142, no. 1, pp.59,64, Jan
1995.
W. Li, J. Liu, J. Wu, and X. He, “Design and analysis of isolated ZVT
boost converters for high-efficiency and high-step-up applications,”
IEEE Trans. Power Electron., vol. 22, no. 6 pp. 2363-2374, Nov.
2007.
M. Nymand and M. A. E. Andersen, “High-efficiency isolated boost
DC-DC converter for high-power low-voltage fuel-cell applications,”
IEEE Trans. Power Electron., vol. 57, no. 2 pp. 505-514, Feb. 2010.
R. W. Erickson, and D. Maksimovic. “Fundamentals of power
electronics,” Springer, 2001.
M. D. Seeman and S. R. Sanders, “Analysis and optimization of
switched-capacitor DC-DC converters,” IEEE Trans. Power
Electron., vol. 23, pp. 841-851, Jul. 2006.
L. Gu, K. Jin, X. Ruan, M. Xu, and F. C. Lee, “A family of switching
capacitor regulators,” IEEE Trans. Power Electron., vol. 29, no. 2
pp. 740-749, Feb. 2014
A. Parastar, A. Gandomkar, J. MingGuo, and J. K. Seok, “High
power solid-state step-up resonant Marx modulator with continuous
output current for offshore wind energy systems,” Proc. of IEEE
ECCE Conf. , 2013, pp.1709,1716.
P. K. Peter and V. Agarwal, “Analysis and design of a ground
isolated switched capacitor DC-DC converter,” Proc. of IEEE ISIE,
2010.
L. Zhigang, A. Q. Huang, and G. Rong, “High efficiency switched
capacitor buck-boost converter for PV application,” Proc. of IEEE
APEC, 2012, pp. 1951-1958.
K. Kesarwani and J. T. Stauth, “A comparative theoretical analysis of
distributed ladder converters for sub-module PV energy
optimization,” Proc. of IEEE COMPEL, 2012, pp.1,6.
A. M. Y. M. Ghias, J. Pou, V. G. Agelidis, and M. Ciobotaru, “Initial
capacitor charging in grid-connected flying capacitor multilevel
converters,” IEEE Trans. Power Electron., vol. 29, no. 7, pp.
3245,3249, Jul. 2014.
H. Sepahvand, M. Khazraei, K. A. Corzine, and M. Ferdowsi, “Startup procedure and switching loss reduction for a single-phase flying
capacitor active rectifier,” IEEE Trans. Ind. Electron., vol. 60, no. 9,
pp. 3699,3710, Sep. 2013.
Y. Zhang and L. Sun, ‘‘An efficient control strategy for a five-level
invertercomprising flying-capacitor asymmetric H-bridge,’’ IEEE
Trans. Ind. Electron., vol. 58, no. 9, pp. 4000---4009, Sep. 2011.