A ZVT-PWM single stage PFC converter with an active snubber1
Wei Gu, Jaber Abu-Qahouq, Shiguo Luo, Issa Batarseh
Electrical and Computer Engineering Department
University of Central Florida
Email: wg02128@pegasus.cc.ucf.edu
Abstract-A single-stage power factor correction
converter with soft-switching is proposed in this paper.
Besides the ZVT turn-on performance of the main
switch, the converter has ZCS for the auxiliary switch
with an active snubber. High frequency operation of the
proposed converter makes the ac-dc power supply
possible to be minimized in size and weight. A 50-W 500kHz prototype has been built in the laboratory to
experimentally verify the analysis.
II. PROPOSED CONVERTER AND ITS OPERATION
The proposed soft-switching converter with an
active snubber shown in Fig 1 is based on the hard-switching
single-stage PFC converter with two bulk capacitor proposed
first in [3]. Besides achieving zero-voltage transition (ZVT),
the circuit also using a transformer as a snubber to lower
voltage and current stresses of switches.
The main waveforms and the equivalent circuits for
the modes of operation are shown in Fig. 2 and Fig 3.
I. INTRODUCTION
As the use of power supplies continues to increase,
more distorted mains current is drawn from the line,
resulting in lower power factor and high total harmonic
distortion. Power factor correction (PFC) is becoming more
and more common in single-phase off-line switching-mode
power supplies, not only because low power factor limits the
maximum available power drawn from mains, but also
agency regulation requires that the harmonic current of the
line current of mains-connected equipment remains below
certain limits. For years a great deal of effort has been made
to development efficient and cost-effective power factor
correction schemes. As a branch of active PFC techniques,
the single-stage technique receives particular attention
because of its low cost implementation [1] [2].
Moreover, with the residential industry and defense
industry continuously demanding for even higher power
density, switching mode power supply operating at highfrequency is required because at high switching frequency,
the size and weight of circuit components can be remarkably
reduced. But with the increasing of switch frequency, the
switching loss becomes intolerable, resulting in very low
conversion efficiency. Furthermore, the presence of leakage
inductance in the high-frequency transformer and junction
capacitance in the semiconductor devices causes the power
devices to turn-off and turn-on with more energy loss and
noise. Because of this reason, switch frequency of the
traditional SMPS is limited within 100kHz. To boost the
switching frequency, the soft-switching technique [4], [5],
was introduced to alleviate the switching loss.
Li
Di
Dao
S
Lp1
Lap
Ds
Las
Cs
VAC
Cp2
Dp
Lak
Da
Sa
Lp2
Ca
Cp1
Fig 1 Basic circuit
S
Sa
Vcs
I Lak
I Di
ILp1
t0
t1 t 2
t 3t 4
t5
t6
Fig 2 Circuit waveforms
1
This work is supported by NSF (contract number NSF 99 608 03)
Ls
Lk1
Lk2
Do
Co
Ro
Li
Di
Li
Dao
Lap
Ds
S
Las
Dp
Lak
Dao
Cp2
Lp1
Cs
VAC
Di
Do
S
Ls Co
Lk1
Da
Ro
Ca
Cp1
Lk2
Ca
Cp1
Lk2
Di
Dao
Do
Las
Dp
Co
S
Ro
Cp1
Ca
Dp
Da
Lk2
Sa
Ls
Lk1
Lak
Lp2
Da
Las
Cs
VAC
Cp2
Lp1
Lap
Ds
Ls
Lk1
Lak
Li
Cp2
Lp1
Cs
Sa
Do
Co
Lp2
Ca
Cp1
Lk2
(f) Mode 6
(b) Mode 2
Fig 3 Modes of operation
Li
Di
Dao
Lap
Ds
S
Cp2
Lp1
Las
Cs
VAC
Dp
Lak
Cp1 Lk2
Di
S
Mode 2: t1 < t < t2
Dao
Las
Dp
Cs
VAC
Cp2
Lp1
Lap
Ds
Lak
Sa
Ls
Lk1
Da
Lp2
Ca
Cp1
(d) Mode 4
At t = t0 , the auxiliary switch S a is turned on.
conduct when auxiliary switch is on because the energy is
transferred from the primary winding to the secondary
winding. After that, S is ready to be turned on at ZVS.
(c) Mode 3
Li
Ro
C s , C a , and Lrak form a resonant tank as can been in
Fig. 2a. At the end of this mode (t = t0 ), the capacitor
voltage of the main switch S hits zero. Dao starts to
Lp2
Ca
Mode 1: t0 < t < t1
Do
Ls Co
Lk1
Da
Sa
Ro
(e) Mode 5
Lap
VAC
Co
Lp2
Sa
Dao
Ds
Do
Ls
Lk1
Da
Di
S
Dp
Lak
(a) Mode 1
Li
Las
Cs
VAC
Lp2
Sa
Lp1
Lap
Ds
Cp2
Lk2
Do
Co
Ro
After turning S on at t = t1 , the diode Di
conducts and the source voltage is applied to the input choke
inductor L, causing the current through the inductor
increasing linearly. During this period, energy is transferring
from the source to the choke inductor. The period ends when
the leakage inductor current I Lak reaches zero and Dao is
turned off. The choke current continues to increase linearly.
Mode 3: t2 < t < t3
The equivalent circuit for this freewheeling mode is
shown , The auxiliary switch is turn off in this period.
Ro
Mode 4: t3 < t < t4
At t = t3 , S is turned off. The main switch output
Using the component values from table1, the
schematic (Fig 4) produces the result shown in Fig 5, which
verifies the theoretical analysis.
capacitor C s is quickly charged up to 2V p1 by the current
iL1 . Under the constraint of KCL, both the storage
capacitors, C p1 and C p 2 are being charged by current
iL1 + iL 2 during this operation period. With the inductor
current i Li decreasing linearly, magnetic energy stored in
the choke is being converted into electric energy and being
stored into the storage capacitors. Thus the energy loss of the
storage capacitors during Mode 2 is being recovered.
Mode 5: t4 < t < t5
The choke inductor current, I L
continues to
decrease linearly. Owing to the existence of diode Do , the
primaries of the transformer present very high impedance
with the currents through the windings can be negligible.
This period ends when the choke inductor current reaches
zero.
Fig 4 Simulation schematic
Mode 6: t5 < t < t6
This is a free-wheeling stage for regulation
purpose.
III Soft-switching condition
Soft-switching is maintained for wide line and load
range, which is a unique feature for ZVT. In mode 1, the
differential voltage across the main switch is
VCs = V0 cos(
1
t)
Lak C s
V0 is the Vcs at t = 0 . For ZVS, Vcs drops to zero
before the mode 2 begins. So we get
t1 − t0 >
π
Lak C s
2
The right side of the above inequality is a constant, which
means for this converter, soft-switching operation will be
ensured for the whole load and line range as long as we
choose proper values for the time delay between the gate
signals of main and auxiliary switches and the leakage
inductance of the snubber transformer.
IV. COMPUTER SIMULATION
Fig 5 Simulation result
As can be noted, the simulated waveforms are the
same as theoretical waveforms. Also, the ZVS and ZCS are
achieved for the main switch and auxiliary switch,
respectively.
V. EXPERIMENT RESULTS
A 50-W 500-kHz prototype has been built in the
laboratory to experimentally verify the analysis. It agrees
well with the computer simulation. In fig 6, the upper
waveform is the drive signal of the power MOSFET and the
lower waveform is the voltage across the MOSFET. It’s
shown that drive signal starts rising after the voltage across
the MOSFET drops down to zero, which is zero voltage
switching. However, from fig 6, we can see the noise, which
should be reduced by using a transformer with a smaller
parasitic capacitance and inductance.
Fig 6 Experiment result
6 Conclusions
Due to high frequency operation, the proposed
converter makes the ac-dc power supply possible to be
minimized in size and weight. At the same time, the active
single-stage PFC techniques help to reduce the component
count and cost. Therefore, for the low power application,
this proposed converter could be considered as a strong
competitor.
ACKNOWLEDGMENT
The authors of this paper would like to thank Dr. Peter
Kornetzky and Dr. Huai Wei, for providing the advice.
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PWM
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