Discovery ANALYSIS The International Daily journal ISSN 2278 – 5469 EISSN 2278 – 5450 © 2015 Discovery Publication. All Rights Reserved Switching method based approach towards improvement in dynamic range of Instrumentation Amplifier for biomedical applications Publication History Received: 23 August 2015 Accepted: 20 September 2015 Published: 14 October 2015 Page 151 Citation Akshay A Mane, Mahesh N Parihar. Switching method based approach towards improvement in dynamic range of Instrumentation Amplifier for biomedical applications. Discovery, 2015, 43(199), 151-157 Switching method based approach towards improvement in dynamic range of Instrumentation Amplifier for biomedical applications Akshay A. Mane Mahesh N. Parihar Instrumentation Department Ramrao Adik Institute of Technology Navi Mumbai, India [email protected] Instrumentation Department Ramrao Adik Institute of Technology Navi Mumbai, India [email protected] I. INTRODUCTION We know that the passive components like Resistors & Capacitors have its own variable tolerances in their values. In biomedical or cardiac measurements using different sensors, A.C coupling is widely used for blocking the noise . An ‘Instrumentation Amplifier’ provides features as high input impedance, low bias and offset currents, high C.M.R.R, high slew rate, low noise, low power consumption which gives the preference for sensor circuit interface. Consider the inverting amplifier with Rom and its compensating network as shown in fig A. The amplifier has been nulled at room temperature 250C i.e the effects of Vio (input offset voltage) and Iio (input offset current) have been reduced to zero by use of potentiometer Ra arrangement shown in fig A. The values of Vio and Iio drift (change) with temperature. According to the equation 1.1, any change in the Voot (total output offset voltage). Therefore, the total output offset voltage will not be zero at any temperature other than room temperature (250C). Voot (1 R f / R1 )Vio ( R f ) I io (1.1) The average change in total output offset voltage by per unit change in temperature, is given by equation 1.2 Voot / T (1 R f / R1 )(Vio / T ) ( R f )(I io / T ) (1.2) The maximum possible change in total output offset voltage ∆Voot as the error voltage denoted by Ev is given by equation 1.3, EV (1 R f / R1 )(Vio / T )T ( R f )(I io / T )T (1.3) 152 Keywords— common mode rejection ratio; RC coupling; offset voltage; component tolerance; dynamic range. Fig. A Compensating Inverting Amplifier Page Abstract— This paper proposes various switching methods for the Instrumentation Amplifier to be used in biomedical applications to lower the effects of passive component tolerances in its values. In sensor based measurements having offset potentials, it can be of several orders of magnitude, AC coupling consisting of resistor & capacitor has been prominently used which limits the performance of these amplifiers due to early saturation. Offset voltages larger than the amplitudes of the signals of interest, reduces the admissible gain of an amplifier used in the system. To get rid of this, a high-input resistance stage requires ac input coupling at front of the circuit. This can be achieved by series electrolytic capacitors, however in order to provide a bias path to ground, grounded resistors are usually included, which decreases the common mode rejection ratio (CMRR). The schemes introduced here to improve dynamic range, CMRR, minimize the effect of offset voltages, temperature effects etc. These schemes have been described in this paper for which circuits have been designed and implemented on hardware on real time basis. The results obtained are fairly close to the expected results and defends the efficacy of the new designs. The results obtained show that the proposed schemes are able to give CMRR in the range of 50 to 90 dB. The use of simple analog switch has been implemented by a transistor operated as switch by a toggling square wave input of +5V input with variable frequency from 100Hz to 1MHz. In sensor interface biomedical applications like EEG, human body skin impedance measurement, heart throbbing detection, etc the overall performance of the system is limited by the offset and frequency noise of the input amplifiers as it may override the actual biomedical signal level being very small in amplitude. Sensor offset potentials can be several orders of magnitude larger than actual amplitude of the signal which limits the admissible gain of the amplifier to prevent early amplifier saturation . If finite offset potential exists then output may saturate Op-Amp used in the signal conditioning circuits. Hence, amplifier gain is limited to moderate values which results in low common mode rejection ratio-C.M.R.R. Also as the first stage gain is reduced, subsequent stages are required to obtain the required gain, which lead to increase number of R&C components and therefore high power consumption . It should be noted that operational amplifiers (Op-Amp) to be used in such precise applications should have very low offset & hence Op-Amp family like OP07A or OP327 or LM308 etc may be preferred. Consider following RC based A.C coupling used generally in conventional instrumentation amplifier circuit. Fig, 2 Actual circuit arrangement for implementing basic configuration. Efficiency of this coupling network depends ultimately on the component tolerance used in it. However, as component can never be identical to another due to manufacturing factors, various errors, change in ambient temperature, etc. Here both type of the signals i.e. common mode (both i/ps of the amplifier are same) as well as differential mode (both i/p are different either in amplitude or phase or both) are used as activating inputs to these set of circuits to conclude with valid results in the context of bio medical signals & its conditioning. If we use the components with 10% or less tolerances in worst case conditions, we can find that differential output voltage (Vout differential) is not equal to Vi/p differential input voltage (Vab) and hence it needs separate provision to nullify the difference in these voltages. In laboratory, various case studies are implemented on hardware consisting of all the required active & passive components for improving dynamic range of Instrumentation Amplifier to be used in biomedical applications to facilitate high impedance requirement in human engineering/ instrumentation organization of paper. II. CASE STUDY 1 USING SWITCHING METHOD 1 Above circuit can be modified in such a way so that it does not require any grounded resistor, achieving a large C.M.R.R. It has been validated by the experimental reading as mentioned in this paper. This circuit provides R.C ground coupling for differential signals as well as D.C path for amplifier bias current which then drained to ground through common electrode used in the biomedical equipment . As shown in the figure 3 below, let R8 & R9 are the resistance in series with the switches Q1 & Q2 respectively which are normally in closed states of the switches. These analog switches connected in A.C coupling are opened & closed at regular interval at lower frequency range in commensurate with human pulsation frequency maximum upto 2MHz at input of instrumentation amplifier. Case Study1 for Method 1circuit diagram Switches Q1 & Q2 offer infinite resistances when they are opened and hence input sensor biomedical signals of interest can pass through capacitors C1 & C2. At the same time these 153 In this circuit diagram the resistor R10 has major impact on the C.M.R.R of this amplifier and C.M.R.R increases or decreases with that of R10. In order to achieve infinite C.M.R.R, R10 is required to be infinite but it cannot be infinite value practically. This is due to the fact that OPAMP based amplifier always needs a bias path for the current to the ground . Page Fig-1 Basic configuration of RC coupled Instrumentation Amplifier In this advanced circuit, the semiconductor switching devices like diode, transistors, MOSFETS can be used along with R8 & R9 of basic circuit as shown above in fig no 3, in order to offer infinite resistances when they are kept open & zero resistance when they are closed. These semiconductor switches like diode, transistors or MOSFETS have very stiff resistance values in ON & OFF states with smaller values of tolerance in these resistance values as compared to its passive counter parts. It is undesirable phenomenon & has to be prevented in low voltage or biomedical signal as inputs. During the changeover of the switches Q1 and Q2, a spike may be generated in the output which should be eliminated using a Low Pass Filter ( shown in the circuit diagram) with proper pass band gain of 2dB & cut off frequency as determined from that of noise, at the output of amplifier. In this method, Switches Q1 and Q2 may not operate simultaneously, in order to obtain synchronized switching, semiconductor based pulse generating circuit or Pulse Width Modulated circuit or a single resistor for one of the switch may be used. This method also finds that the switch inherent/Miller capacitance becomes pre-dominant at high frequencies & affects the dynamic range of the amplifier. III. CASE STUDY 2 USING SWITCHING METHOD 2 capacitors block D.C voltages or noise associated with small input or sensor signal. Due to this if a common signal is applied, no current flow through the network so that AC voltage between point Va and Vb is zero/minimum or ideally Va and Vb are at same potential. This absence of potential difference due to common mode inputs gives an ideally infinite C.M.R.R. Here it is considered that the total resistance of R10 with infinite resistance of opened switches Q1 or Q2 amounts to infinite resistance as x∞ if x = y + ∞ & y is constant. Component tolerances do not have any effects on output as differential output voltage is equal to differential input voltage. Also there is no discharge path available for the bias current when the switches Q1 and Q2 are opened. Bias current from OP-Amp flow through the capacitor resulting in a D.C offset, thus may saturate the OP-Amp earlier. This can be reduced by using a high value of capacitor which will have large time-constant giving rise to large time for charging or by using OP-Amp with low offset voltage & low bias current or even using of a simple R-C filter at output as shown in the figure 3. After a finite time or frequency, these switches Q1 and Q2 should be closed for proper normal operation. When they are closed, it provides a discharge path for bias current present in OP-AMP. This in turn prevents the development of voltage due to bias current which may saturate Op-AMP early. When Q1 and Q2 are opened, they offer infinite resistance in the circuit. Therefore, input signals of interest can pass through capacitors only and prevent D.C. voltages that are associated with input voltage to reach OP-AMP. Op-Amps with very low bias current having range in nA to pA is mostly preferred. As a result of it, if common mode input voltage is applied, very small current flow through the circuit so that circuit nodes achieve almost same potential giving rise to infinite C.M.R.R. This indicates that output differential voltage (Vout) is equal to input differential voltage (Vab) which indicates that component tolerances do not have any effect on output. Thus in order to improve or enhance the dynamic range or C.M.R.R of instrumentation amplifier, it is practical to block D.C signal or offset associated with sensor or bio potential electrode or A.C sensor signal. Therefore, an A.C. coupling using conventional resistor and capacitor is required. The RC component tolerances put the limitations on the dynamic range of circuit as differential signal and C.M.R.R comes down. Thus above semiconductor based switching methods 1 and 2 makes effective resistance in RC coupling infinite, thus improving dynamic range as well as CMRR of instrumentation amplifier. 154 Fig-4 Actual circuit arrangement for implementing case study 1 When closed, switches Q1 and Q2 offer negligible resistance & provide a discharge path for bias current to ground directly and hence signal is prevented from reaching the Op-Amp. Page Fig-3 Circuit diagram for case study 1 In method 1 above, we have found from experimental results that there exists high voltage drop across the capacitors which is due to flow of bias current of Op-Amp through the capacitors. Also high value of C is not feasible for high frequency applications because of limitation in terms of capacitive impedance. In such cases, the following method is appreciated as shown in figure 5. Capacitors C1 and C2 are connected in series with finite value resistors R8 and R9 respectively acting as the current limiting resistors. Switches Q1 and Q2 are opened and closed at the regular interval/frequency that are used in A.C coupling at the input of instrumentation amplifier. the efficacy of these methods, these switching circuit was realized on hardware consisting of all required passive components in analog electronics laboratory, the transistors BC547 as switch, operational amplifier OP07A having low DC offset voltages. Fig-5 Switching method 2 for instrumentation amplifier Real time hardware implementation of conventional RC coupling networks as well as the proposed method was done in laboratory. It is then found that the behaviour of the circuit was found to be independent on values of R and C, source impedances as well as on tolerance of R and C as shown above (Figure 1, 3 and 5) respectively with conventional RC circuits. It was also observed that conventional instrumentation amplifier works as per design intent & its performance is not affected by additional switching circuit used for DC offset voltage compensation. From the frequency response of this amplifier as shown, it is seen that it has not degraded in terms of low frequency response, differential gain or CMRR as shown in graph A from Table 1. TABLE I. OBSERVED OUTPUT FOR BASIC CONFIGURATION OF INSTRUMENTATION AMPLFIER WITH CONVENTIONAL RC COUPLING AS SHOWN IN FIGURE NO 1 VIN (COMMON MODE INPUT) =10MV. DIFFERENTIAL GAIN, AD OF THE AMPLIFIER CKT = 11 Observed Output Voltages Frequency For the conventional & classical instrumentation amplifier as shown in the figures, the differential o/p voltage is given by Vout (1 2 R f / R g )Vid Differenatil Gain Ad = Voutput (differential) Vinput (differential) Ad Vout / Vid (1 2 R f / R g ) For the figure no 1 instrumentation amplifier using AC coupling, the above equation can be rewritten as Differential Gain, Ad (1 2 R2 / R1 ) (12 1000200) 11 from fig no1 R2 2 K ohms & R1200 Ohms Also Common mode rejection ratio can be determined as CMRR= 20 log (Ad/Acm) where Acm is common mode gain of an instrumentation amplifier as used in the column in observation tables below. V. RESULTS In order to implement the aforesaid methods for finding A= 20log(Acm) C.M.R.R = 20 log (Ad /Acm) in dB 100Hz 0.562mV 0.0562 -25.021 45.833 1KHz 0.562mV 0.0562 -25.021 45.833 2KHz 0.562mV 0.0562 -25.021 45.833 5KHz 0.562mV 0.0562 -25.021 45.833 10KHz 0.562mV 0.0562 -25.021 45.833 100KHz 0.562mV 0.0562 -25.021 45.833 200KHz 0.618mV 0.0618 -24.180 45.008 500KHz 0.337mV 0.0337 -29.445 50.275 700KHz 0.220mV 0.0220 -32.967 53.979 1MHz 0.129mV 0.0129 -37.049 58.616 2 MHz 0.129 mV 0.0129 -37.049 58.616 It should be noted that above results are obtained for a standard differential input of 10mV AC signal with band of frequency from 100Hz to 2 MHz in line with the various biomedical signals. It should be noted that the common mode input voltage applied to this circuit should ideally give rise to zero output ideally or as minimum as possible practically, meaning that the effective common mode gain Acm will be zero or as low as possible respectively. With this understanding, it can be interpreted that from the above table1 that there exist a finite output voltage due to offset & mismatch in Op-amps & its value reduces as the frequency decreases. As input is common mode type, the o/p will be lesser than the input & hence the common mode gain is negative as indicated in the tables. 155 IV. CALCULATIONS Acm = Vout (cm) / Vin Page Fig-6 Actual circuit arrangement for implementing case study 2 Vout (common mode o/p) It also indicates that CMRR is fairly constant over wide frequency range & increases with the increase in frequency of the signal for common mode signals. Graph A showing the variation of CMRR versus frequency of the signal for method involving basic RC coupling. TABLE II. OBSERVED OUTPUT FOR THE CONFIGURATION USING METHOD 1/CASE STUDY 1 OF INSTRUMENTATION AMPLIFIER AS SHOWN IN FIGURE NO 3. VIN (COMMON MODE INPUT) =10MV. DIFFERENTIAL GAIN, AD OF THE AMPLIFIER CKT = 11 Observed Output Voltages Frequency Acm = Vout (cm) / Vin Vout (common mode o/p) A= 20log(Acm) C.M.R.R = 20 log (Ad /Acm) in dB 100Hz 0.26 0.026 -31.70 52.528 1KHz 0.28 0.028 -31.06 51.884 2KHz 0.30 0.030 -30.46 51.285 5KHz 0.28 0.028 -31.06 51.884 10KHz 0.28 0.028 -31.06 51.884 100KHz 0.28 0.028 -31.06 51.884 200KHz 0.28 0.028 -31.06 51.884 500KHz 0.28 0.028 -31.06 51.884 700KHz 0.28 0.028 -31.06 51.884 1MHz 0.28 0.028 -31.06 51.884 Observed Output Voltages Frequency Vout (common mode o/p) Acm = Vout (cm) / Vin Subhead A= 20log(Acm) C.M.R.R = 20 log (Ad /Acm) in dB 100Hz 0.0224mV 0.00224 -21.116 73.795 1KHz 0.0562 mV 0.00562 -44.980 65.836 2KHz 0.0674mV 0.00674 -49.789 64.252 5KHz 0.0674mV 0.00674 -49.789 64.252 10KHz 0.0674mV 0.00674 -49.789 64.252 50KHz 0.0787mV 0.00787 -42.086 62.914 100KHz 0.0787mV 0.00787 -42.086 62.914 200KHz 0.0787mV 0.00787 -42.086 62.914 500KHz 0.0787mV 0.00787 -42.086 62.914 700KHz 0.0787mV 0.00787 -42.086 62.914 1 MHz 0.0787mV 0.00787 -42.086 62.914 2 MHz 0.0787mV 0.00787 -42.086 62.914 Graph C plotted from Table III TABLE IV. OBSERVED OUTPUT FOR THE CONFIGURATION USING METHOD 1 OF INSTRUMENTATION AMPLIFIER AS SHOWN IN FIGURE NO 3. VIN (DIFFERENTIAL MODE INPUT) =10MV COMMON MODE GAIN, AC OF THE AMPLIFIER CKT = 0.028(AVERAGE VALUE TAKEN FROM TABLE NO 2) Observed Output Voltgaes 0.028 -31.06 51.884 A= 20log(Acm) C.M.R.R = 20 log (Ad /Acm) in dB It can be seen from table2 that CMRR is constant over wide dynamic range & is better as compared to previous set of observation for a circuit with conventional AC coupling for the common mode signal. 100Hz 2V 200 46 77 1KHz 2V 200 46 77 2KHz 2V 200 46 77 Graph B plotted from Table No II 5KHz 2V 200 46 77 10KHz 2V 200 46 77 50KHz 2V 200 46 77 100KHz 2V 200 46 77 200KHz 2V 200 46 77 500KHz 2V 200 46 77 700KHz 2V 200 46 77 1 MHz 2V 200 46 77 2 MHz 2V 200 46 77 TABLE III. OBSERVED OUTPUT FOR THE CONFIGURATION USING METHOD 2/ CASE STUDY 2 OF INSTRUMENTATION AMPLIFIER AS SHOWN IN FIGURE NO 5 FOR COMMON MODE SIGNAL. VIN (COMMON MODE INPUT) =10MV. DIFFERENTIAL GAIN, AD OF THE AMPLIFIER CKT = 11 156 0.28 Acm = Vout (cm) / Vin Page 2 MHz Frequency Vout (common mode o/p) Graph D plotted from Table IV having finite tolerances are used for RC coupling which causes a portion of common mode signal appears as differential signal and CMRR falls down. This is highly undesirable in the design of biomedical instrumentation. This paper introduces a new approach of making resistance in R & C coupling infinite, thus improving dynamic range and CMRR for both types of i/p signals. The new scheme involving switching technique has been designed & implemented on hardware on lab scale and later compared with conventional method. It indicates that it enhances these characteristics which are highly desirable in the design of biomedical instruments having low amplitude & frequency input signals. References It can be seen from graph or experimental reading that in case of conventional RC coupling, there is much distortion at lower frequencies since coupling capacitors offer high impedance i.e high attenuation at lower frequencies. This low frequency distortion is totally removed in switching method. Due to temperature variation, the drift in offset voltage places a stringent requirement on designer or equipment design. Practically it is difficult to get such a large CMRR in biomedical applications. In order to improve dynamic range, it is necessary to remove DC signal or offset associated with AC signal from human body or sensors, RC coupling using resistor and capacitor is required. The resistors and capacitors Enrique mario spinelli, Ramon pallàs - areny, Miguel angel mayosky, “Ac-coupled front-end for Biopotential measurements” IEEE transactions on biomedical engineering, vol. 50, no. 3, march 2003.  Oscar Casas, Enrique Mario Spinelli, and Ramon Pallàs-Areny, “Fully Differential AC -Coupling Networks:A Comparative Study” IEEE transactions on instrumentation and measurement, vol. 58, no. 1, January 2009.  Operational Amplifier and Linear Integrated Circuits by Ramakant Gayakwad 4th Edition.  Oscar Casas, Enrique Mario Spinelli, and Ramon Pallas –Arney, “ Fully Differential AC-Coupling Networks: A comparative study”, IEEE Transaction on Instrumentation and Measurement, Vol. 58, No.1, January 2009, pp. 94-98.  Charles Kitchin and Lew Counts, “ A Designer’s Guide to Instrumentation Amplifier”, 3rd ed., Analog Devices, 2006,pp 2-3.  R. Pallàs-Areny and J. G.Webster, “Common mode rejection ratio in differential amplifiers,” IEEE Trans. Instrum. Meas., vol. 40, pp. 669– 676, Aug. 1991.  Fu Qiang, Xiao Yan, Tan Kai, Liu Xiaowei, Shan Qiang “ Analysis and Design of Instrumentation Amplifier Based On Chopper Technology” IEEE Laser Physics and Laser Technologies (RCSLPLT) and 2010 Academic Symposium on Optoelectronics Technology (ASOT), 2010 10th Russian-Chinese Symposium on, 2010, pp.318.  Anton Bakker, Kevin Thiele and Johan H. Huijsing. “A CMOS NestedChopper Instrumentation Amplifier with 100-nV Offset,” IEEE Journal of Solid-State Circuits. VOL. 35, NO. 12, DECEMBER 2000,pp.18771883.  www.analog.com/static/imported-files/design_handbooks/ 5812756674312778737 Complete_In_Amp.pdf  S.Grimnes, “Impedance Measurement of Individual Skin Surface Electrodes”, Medical and Biomedical Engineering and Computing, Nov1983, vol 21, pp 750-755  Enrique Mario Spinelli, Ramon Pallas –Arney and Miguel Angel Mayosky, “AC-Coupled Front-End for Biopotential Measurements”, IEEE Transaction on Biomedical Engineering, Vol. 50, No.3, March 2003, pp. 391–395.  Enrique Mario Spinelli, Nolberto Martínez, Miguel Angel Mayosky, Ramon Pallas –Arney, “A novel Fully Differential Biopotential Amplifier with DC Supression”, IEEE Transaction on Biomedical Engineering, Vol. 51, No.8, August 2004, pp. 1444-1448. 157 VI. CONCLUSION From the observation & result as mentioned above in this paper, it is concluded that common mode noise is cancelled out due to differential configuration leading to enhanced CMRR due to use of active switching transistorized arrangement. Also the characteristics of instrumentation amplifier which are very much vital in biomedical applications like differential gain and CMRR, are related to frequency of signal as well as that of the switching. The tolerance in the values of R and C used in conventional RC coupling circuits affects these characteristics significantly. The switching methods proposed in this paper acts as a technique for recovery of dynamic range of instrumentation amplifier for both common as well as differential signals to be used in biomedical applications.  Page It can be again seen from the above table 4 that CMRR is constant over wide entire dynamic range & is better & enhanced as compared to previous set of observation for a circuit with conventional AC coupling as well as both of the switching method using transistors.