# Full Text - Discovery Publication

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ANALYSIS
The International Daily journal
ISSN 2278 – 5469
EISSN 2278 – 5450
Switching method based approach towards improvement
in dynamic range of Instrumentation Amplifier for
biomedical applications
Publication History
Accepted: 20 September 2015
Published: 14 October 2015
Page
151
Citation
Akshay A Mane, Mahesh N Parihar. Switching method based approach towards improvement in dynamic range of Instrumentation Amplifier
for biomedical applications. Discovery, 2015, 43(199), 151-157
Switching method based approach towards
improvement in dynamic range of Instrumentation
Amplifier for biomedical applications
Akshay A. Mane
Mahesh N. Parihar
Instrumentation Department
Navi Mumbai, India
[email protected]
Instrumentation Department
Navi Mumbai, India
[email protected]
I.
INTRODUCTION
We know that the passive components like Resistors &amp;
Capacitors have its own variable tolerances in their values. In
biomedical or cardiac measurements using different sensors,
A.C coupling is widely used for blocking the noise [1]. An
‘Instrumentation Amplifier’ provides features as high input
impedance, low bias and offset currents, high C.M.R.R, high
slew rate, low noise, low power consumption which gives the
preference for sensor circuit interface.[2]
Consider the inverting amplifier with Rom and its
compensating network as shown in fig A. The amplifier has
been nulled at room temperature 250C i.e the effects of Vio
(input offset voltage) and Iio (input offset current) have been
reduced to zero by use of potentiometer Ra arrangement shown
in fig A. The values of Vio and Iio drift (change) with
temperature. According to the equation 1.1, any change in the
Voot (total output offset voltage). Therefore, the total output
offset voltage will not be zero at any temperature other than
room temperature (250C).[3]
Voot  (1  R f / R1 )Vio  ( R f ) I io
(1.1)
The average change in total output offset voltage by per
unit change in temperature, is given by equation 1.2
Voot / T  (1  R f / R1 )(Vio / T )  ( R f )(I io / T )
(1.2)
The maximum possible change in total output offset
voltage ∆Voot as the error voltage denoted by Ev is given by
equation 1.3,
EV  (1  R f / R1 )(Vio / T )T  ( R f )(I io / T )T
(1.3)
152
Keywords— common mode rejection ratio; RC coupling; offset
voltage; component tolerance; dynamic range.
Fig. A Compensating Inverting Amplifier
Page
Abstract— This paper proposes various switching methods for
the Instrumentation Amplifier to be used in biomedical
applications to lower the effects of passive component tolerances
in its values. In sensor based measurements having offset
potentials, it can be of several orders of magnitude, AC coupling
consisting of resistor &amp; capacitor has been prominently used
which limits the performance of these amplifiers due to early
saturation. Offset voltages larger than the amplitudes of the
signals of interest, reduces the admissible gain of an amplifier
used in the system. To get rid of this, a high-input resistance
stage requires ac input coupling at front of the circuit. This can
be achieved by series electrolytic capacitors, however in order to
provide a bias path to ground, grounded resistors are usually
included, which decreases the common mode rejection ratio
(CMRR). The schemes introduced here to improve dynamic
range, CMRR, minimize the effect of offset voltages, temperature
effects etc. These schemes have been described in this paper for
which circuits have been designed and implemented on hardware
on real time basis. The results obtained are fairly close to the
expected results and defends the efficacy of the new designs. The
results obtained show that the proposed schemes are able to give
CMRR in the range of 50 to 90 dB. The use of simple analog
switch has been implemented by a transistor operated as switch
by a toggling square wave input of +5V input with variable
frequency from 100Hz to 1MHz.
In sensor interface biomedical applications like EEG, human
body skin impedance measurement, heart throbbing detection,
etc the overall performance of the system is limited by the
offset and frequency noise of the input amplifiers as it may
override the actual biomedical signal level being very small in
amplitude[4]. Sensor offset potentials can be several orders of
magnitude larger than actual amplitude of the signal which
limits the admissible gain of the amplifier to prevent early
amplifier saturation [1]. If finite offset potential exists then
output may saturate Op-Amp used in the signal conditioning
circuits. Hence, amplifier gain is limited to moderate values
which results in low common mode rejection ratio-C.M.R.R.
Also as the first stage gain is reduced, subsequent stages are
required to obtain the required gain, which lead to increase
number of R&amp;C components and therefore high power
consumption [4]. It should be noted that operational amplifiers
(Op-Amp) to be used in such precise applications should have
very low offset &amp; hence Op-Amp family like OP07A or
OP327 or LM308 etc may be preferred.
Consider following RC based A.C coupling used generally
in conventional instrumentation amplifier circuit.
Fig, 2 Actual circuit arrangement for implementing basic
configuration.
Efficiency of this coupling network depends ultimately on the
component tolerance used in it. However, as component can
never be identical to another due to manufacturing factors,
various errors, change in ambient temperature, etc. Here both
type of the signals i.e. common mode (both i/ps of the
amplifier are same) as well as differential mode (both i/p are
different either in amplitude or phase or both) are used as
activating inputs to these set of circuits to conclude with valid
results in the context of bio medical signals &amp; its conditioning.
If we use the components with 10% or less tolerances in
worst case conditions, we can find that differential output
voltage (Vout differential) is not equal to Vi/p differential input
voltage (Vab) and hence it needs separate provision to nullify
the difference in these voltages.
In laboratory, various case studies are implemented on
hardware consisting of all the required active &amp; passive
components for improving dynamic range of Instrumentation
Amplifier to be used in biomedical applications to facilitate
high impedance requirement in human engineering/
instrumentation organization of paper.
II. CASE STUDY 1 USING SWITCHING METHOD 1
Above circuit can be modified in such a way so that it does
not require any grounded resistor, achieving a large C.M.R.R.
It has been validated by the experimental reading as
mentioned in this paper. This circuit provides R.C ground
coupling for differential signals as well as D.C path for
amplifier bias current which then drained to ground through
common electrode used in the biomedical equipment [8].
As shown in the figure 3 below, let R8 &amp; R9 are the
resistance in series with the switches Q1 &amp; Q2 respectively
which are normally in closed states of the switches. These
analog switches connected in A.C coupling are opened &amp;
closed at regular interval at lower frequency range in
commensurate with human pulsation frequency maximum upto 2MHz at input of instrumentation amplifier.
Case Study1 for Method 1circuit diagram
Switches Q1 &amp; Q2 offer infinite resistances when they are
opened and hence input sensor biomedical signals of interest
can pass through capacitors C1 &amp; C2. At the same time these
153
In this circuit diagram the resistor R10 has major impact on
the C.M.R.R of this amplifier and C.M.R.R increases or
decreases with that of R10. In order to achieve infinite
C.M.R.R, R10 is required to be infinite but it cannot be infinite
value practically. This is due to the fact that OPAMP based
amplifier always needs a bias path for the current to the
ground [5].
Page
Fig-1 Basic configuration of RC coupled Instrumentation
Amplifier
In this advanced circuit, the semiconductor switching
devices like diode, transistors, MOSFETS can be used along
with R8 &amp; R9 of basic circuit as shown above in fig no 3, in
order to offer infinite resistances when they are kept open &amp;
zero resistance when they are closed. These semiconductor
switches like diode, transistors or MOSFETS have very stiff
resistance values in ON &amp; OFF states with smaller values of
tolerance in these resistance values as compared to its passive
counter parts.
It is undesirable phenomenon &amp; has to be prevented in low
voltage or biomedical signal as inputs.
During the changeover of the switches Q1 and Q2, a spike
may be generated in the output which should be eliminated
using a Low Pass Filter ( shown in the circuit diagram) with
proper pass band gain of 2dB &amp; cut off frequency as
determined from that of noise, at the output of amplifier. In
this method, Switches Q1 and Q2 may not operate
simultaneously, in order to obtain synchronized switching,
semiconductor based pulse generating circuit or Pulse Width
Modulated circuit or a single resistor for one of the switch
may be used. This method also finds that the switch
inherent/Miller capacitance becomes pre-dominant at high
frequencies &amp; affects the dynamic range of the amplifier.
III. CASE STUDY 2 USING SWITCHING METHOD 2
capacitors block D.C voltages or noise associated with small
input or sensor signal. Due to this if a common signal is
applied, no current flow through the network so that AC
voltage between point Va and Vb is zero/minimum or ideally
Va and Vb are at same potential. This absence of potential
difference due to common mode inputs gives an ideally
infinite C.M.R.R. Here it is considered that the total resistance
of R10 with infinite resistance of opened switches Q1 or Q2
amounts to infinite resistance as x∞ if x = y + ∞ &amp; y is
constant.
Component tolerances do not have any effects on output as
differential output voltage is equal to differential input
voltage. Also there is no discharge path available for the bias
current when the switches Q1 and Q2 are opened. Bias current
from OP-Amp flow through the capacitor resulting in a D.C
offset, thus may saturate the OP-Amp earlier. This can be
reduced by using a high value of capacitor which will have
large time-constant giving rise to large time for charging or by
using OP-Amp with low offset voltage &amp; low bias current or
even using of a simple R-C filter at output as shown in the
figure 3. After a finite time or frequency, these switches Q1
and Q2 should be closed for proper normal operation. When
they are closed, it provides a discharge path for bias current
present in OP-AMP. This in turn prevents the development of
voltage due to bias current which may saturate Op-AMP early.
When Q1 and Q2 are opened, they offer infinite resistance
in the circuit. Therefore, input signals of interest can pass
through capacitors only and prevent D.C. voltages that are
associated with input voltage to reach OP-AMP. Op-Amps
with very low bias current having range in nA to pA is mostly
preferred.
As a result of it, if common mode input voltage is applied,
very small current flow through the circuit so that circuit
nodes achieve almost same potential giving rise to infinite
C.M.R.R.
This indicates that output differential voltage (Vout) is
equal to input differential voltage (Vab) which indicates that
component tolerances do not have any effect on output.
Thus in order to improve or enhance the dynamic range or
C.M.R.R of instrumentation amplifier, it is practical to block
D.C signal or offset associated with sensor or bio potential
electrode or A.C sensor signal. Therefore, an A.C. coupling
using conventional resistor and capacitor is required. The RC
component tolerances put the limitations on the dynamic range
of circuit as differential signal and C.M.R.R comes down.
Thus above semiconductor based switching methods 1 and 2
makes effective resistance in RC coupling infinite, thus
improving dynamic range as well as CMRR of
instrumentation amplifier.
154
Fig-4 Actual circuit arrangement for implementing case
study 1
When closed, switches Q1 and Q2 offer negligible
resistance &amp; provide a discharge path for bias current to
ground directly and hence signal is prevented from reaching
the Op-Amp.
Page
Fig-3 Circuit diagram for case study 1
In method 1 above, we have found from experimental
results that there exists high voltage drop across the capacitors
which is due to flow of bias current of Op-Amp through the
capacitors. Also high value of C is not feasible for high
frequency applications because of limitation in terms of
capacitive impedance. In such cases, the following method is
appreciated as shown in figure 5. Capacitors C1 and C2 are
connected in series with finite value resistors R8 and R9
respectively acting as the current limiting resistors. Switches
Q1 and Q2 are opened and closed at the regular
interval/frequency that are used in A.C coupling at the input of
instrumentation amplifier.
the efficacy of these methods, these switching circuit was
realized on hardware consisting of all required passive
components in analog electronics laboratory, the transistors
BC547 as switch, operational amplifier OP07A having low
DC offset voltages.
Fig-5 Switching method 2 for instrumentation amplifier
Real time hardware implementation of conventional RC
coupling networks as well as the proposed method was done
in laboratory. It is then found that the behaviour of the circuit
was found to be independent on values of R and C, source
impedances as well as on tolerance of R and C as shown
above (Figure 1, 3 and 5) respectively with conventional RC
circuits. It was also observed that conventional
performance is not affected by additional switching circuit
used for DC offset voltage compensation. From the frequency
response of this amplifier as shown, it is seen that it has not
degraded in terms of low frequency response, differential gain
or CMRR as shown in graph A from Table 1.
TABLE I.
OBSERVED OUTPUT FOR BASIC CONFIGURATION OF
INSTRUMENTATION AMPLFIER WITH CONVENTIONAL RC COUPLING AS
SHOWN IN FIGURE NO 1
VIN (COMMON MODE INPUT) =10MV.
DIFFERENTIAL GAIN, AD OF THE AMPLIFIER CKT = 11
Observed Output Voltages
Frequency
For the conventional &amp; classical instrumentation amplifier
as shown in the figures, the differential o/p voltage is given by
Vout  (1  2 R f / R g )Vid
Differenatil Gain Ad = Voutput (differential)  Vinput
(differential)
Ad  Vout / Vid  (1  2 R f / R g )
For the figure no 1 instrumentation amplifier using AC
coupling, the above equation can be rewritten as
Differential Gain,
Ad  (1  2 R2 / R1 )
 (12 1000200)  11 from fig no1
R2  2 K ohms &amp; R1200 Ohms
Also Common mode rejection ratio can be determined as
CMRR= 20 log (Ad/Acm) where Acm is common mode gain
of an instrumentation amplifier as used in the column in
observation tables below.
V. RESULTS
In order to implement the aforesaid methods for finding
A=
20log(Acm)
C.M.R.R = 20
dB
100Hz
0.562mV
0.0562
-25.021
45.833
1KHz
0.562mV
0.0562
-25.021
45.833
2KHz
0.562mV
0.0562
-25.021
45.833
5KHz
0.562mV
0.0562
-25.021
45.833
10KHz
0.562mV
0.0562
-25.021
45.833
100KHz
0.562mV
0.0562
-25.021
45.833
200KHz
0.618mV
0.0618
-24.180
45.008
500KHz
0.337mV
0.0337
-29.445
50.275
700KHz
0.220mV
0.0220
-32.967
53.979
1MHz
0.129mV
0.0129
-37.049
58.616
2 MHz
0.129 mV
0.0129
-37.049
58.616
It should be noted that above results are obtained for a
standard differential input of 10mV AC signal with band of
frequency from 100Hz to 2 MHz in line with the various
biomedical signals.
It should be noted that the common mode input voltage
applied to this circuit should ideally give rise to zero output
ideally or as minimum as possible practically, meaning that
the effective common mode gain Acm will be zero or as low as
possible respectively. With this understanding, it can be
interpreted that from the above table1 that there exist a finite
output voltage due to offset &amp; mismatch in Op-amps &amp; its
value reduces as the frequency decreases. As input is common
mode type, the o/p will be lesser than the input &amp; hence the
common mode gain is negative as indicated in the tables.
155
IV. CALCULATIONS
Acm =
Vout
(cm) /
Vin
Page
Fig-6 Actual circuit arrangement for implementing case
study 2
Vout
(common
mode o/p)
It also indicates that CMRR is fairly constant over wide
frequency range &amp; increases with the increase in frequency of
the signal for common mode signals.
Graph A showing the variation of CMRR versus frequency of the signal for
method involving basic RC coupling.
TABLE II.
OBSERVED OUTPUT FOR THE
CONFIGURATION USING METHOD 1/CASE STUDY 1 OF INSTRUMENTATION
AMPLIFIER AS SHOWN IN FIGURE NO 3.
VIN (COMMON MODE INPUT)
=10MV. DIFFERENTIAL GAIN, AD OF THE AMPLIFIER CKT = 11
Observed Output Voltages
Frequency
Acm =
Vout
(cm) /
Vin
Vout
(common
mode o/p)
A=
20log(Acm)
C.M.R.R = 20
in dB
100Hz
0.26
0.026
-31.70
52.528
1KHz
0.28
0.028
-31.06
51.884
2KHz
0.30
0.030
-30.46
51.285
5KHz
0.28
0.028
-31.06
51.884
10KHz
0.28
0.028
-31.06
51.884
100KHz
0.28
0.028
-31.06
51.884
200KHz
0.28
0.028
-31.06
51.884
500KHz
0.28
0.028
-31.06
51.884
700KHz
0.28
0.028
-31.06
51.884
1MHz
0.28
0.028
-31.06
51.884
Observed Output Voltages
Frequency
Vout
(common
mode o/p)
Acm =
Vout (cm) /
Vin
A=
20log(Acm)
C.M.R.R =
/Acm) in dB
100Hz
0.0224mV
0.00224
-21.116
73.795
1KHz
0.0562 mV
0.00562
-44.980
65.836
2KHz
0.0674mV
0.00674
-49.789
64.252
5KHz
0.0674mV
0.00674
-49.789
64.252
10KHz
0.0674mV
0.00674
-49.789
64.252
50KHz
0.0787mV
0.00787
-42.086
62.914
100KHz
0.0787mV
0.00787
-42.086
62.914
200KHz
0.0787mV
0.00787
-42.086
62.914
500KHz
0.0787mV
0.00787
-42.086
62.914
700KHz
0.0787mV
0.00787
-42.086
62.914
1 MHz
0.0787mV
0.00787
-42.086
62.914
2 MHz
0.0787mV
0.00787
-42.086
62.914
Graph C plotted from Table III
TABLE IV.
OBSERVED OUTPUT FOR THE
CONFIGURATION USING METHOD 1 OF INSTRUMENTATION AMPLIFIER AS
SHOWN IN FIGURE NO 3.
VIN (DIFFERENTIAL MODE INPUT) =10MV
COMMON MODE GAIN, AC OF THE AMPLIFIER CKT = 0.028(AVERAGE VALUE
TAKEN FROM TABLE NO 2)
Observed Output Voltgaes
0.028
-31.06
51.884
A=
20log(Acm)
C.M.R.R = 20 log
It can be seen from table2 that CMRR is constant over wide
dynamic range &amp; is better as compared to previous set of
observation for a circuit with conventional AC coupling for
the common mode signal.
100Hz
2V
200
46
77
1KHz
2V
200
46
77
2KHz
2V
200
46
77
Graph B plotted from Table No II
5KHz
2V
200
46
77
10KHz
2V
200
46
77
50KHz
2V
200
46
77
100KHz
2V
200
46
77
200KHz
2V
200
46
77
500KHz
2V
200
46
77
700KHz
2V
200
46
77
1 MHz
2V
200
46
77
2 MHz
2V
200
46
77
TABLE III.
OBSERVED OUTPUT FOR THE CONFIGURATION USING
METHOD 2/ CASE STUDY 2 OF INSTRUMENTATION AMPLIFIER AS SHOWN IN
FIGURE NO 5 FOR COMMON MODE SIGNAL.
VIN (COMMON MODE INPUT)
=10MV. DIFFERENTIAL GAIN, AD OF THE AMPLIFIER CKT = 11
156
0.28
Acm =
Vout
(cm) /
Vin
Page
2 MHz
Frequency
Vout
(common
mode o/p)
Graph D plotted from Table IV
having finite tolerances are used for RC coupling which
causes a portion of common mode signal appears as
differential signal and CMRR falls down. This is highly
undesirable in the design of biomedical instrumentation. This
paper introduces a new approach of making resistance in R &amp;
C coupling infinite, thus improving dynamic range and CMRR
for both types of i/p signals. The new scheme involving
switching technique has been designed &amp; implemented on
hardware on lab scale and later compared with conventional
method. It indicates that it enhances these characteristics
which are highly desirable in the design of biomedical
instruments having low amplitude &amp; frequency input signals.
References
It can be seen from graph or experimental reading that in
case of conventional RC coupling, there is much distortion at
lower frequencies since coupling capacitors offer high
impedance i.e high attenuation at lower frequencies. This low
frequency distortion is totally removed in switching method.
Due to temperature variation, the drift in offset voltage
places a stringent requirement on designer or equipment
design. Practically it is difficult to get such a large CMRR in
biomedical applications. In order to improve dynamic range, it
is necessary to remove DC signal or offset associated with AC
signal from human body or sensors, RC coupling using
resistor and capacitor is required. The resistors and capacitors
Enrique mario spinelli, Ramon pall&agrave;s - areny, Miguel angel mayosky,
“Ac-coupled front-end for Biopotential measurements” IEEE
transactions on biomedical engineering, vol. 50, no. 3, march 2003.
[2] Oscar Casas, Enrique Mario Spinelli, and Ramon Pall&agrave;s-Areny, “Fully
Differential AC -Coupling Networks:A Comparative Study” IEEE
transactions on instrumentation and measurement, vol. 58, no. 1, January
2009.
[3] Operational Amplifier and Linear Integrated Circuits by Ramakant
[4] Oscar Casas, Enrique Mario Spinelli, and Ramon Pallas –Arney, “ Fully
Differential AC-Coupling Networks: A comparative study”, IEEE
Transaction on Instrumentation and Measurement, Vol. 58, No.1,
January 2009, pp. 94-98.
[5] Charles Kitchin and Lew Counts, “ A Designer’s Guide to
Instrumentation Amplifier”, 3rd ed., Analog Devices, 2006,pp 2-3.
[6] R. Pall&agrave;s-Areny and J. G.Webster, “Common mode rejection ratio in
differential amplifiers,” IEEE Trans. Instrum. Meas., vol. 40, pp. 669–
676, Aug. 1991.
[7] Fu Qiang, Xiao Yan, Tan Kai, Liu Xiaowei, Shan Qiang “ Analysis and
Design of Instrumentation Amplifier Based On Chopper Technology”
IEEE Laser Physics and Laser Technologies (RCSLPLT) and 2010
Academic Symposium on Optoelectronics Technology (ASOT), 2010
10th Russian-Chinese Symposium on, 2010, pp.318.
[8] Anton Bakker, Kevin Thiele and Johan H. Huijsing. “A CMOS NestedChopper Instrumentation Amplifier with 100-nV Offset,” IEEE Journal
of Solid-State Circuits. VOL. 35, NO. 12, DECEMBER 2000,pp.18771883. [6] www.analog.com/static/imported-files/design_handbooks/
5812756674312778737 Complete_In_Amp.pdf
[9] S.Grimnes, “Impedance Measurement of Individual Skin Surface
Electrodes”, Medical and Biomedical Engineering and Computing,
Nov1983, vol 21, pp 750-755
[10] Enrique Mario Spinelli, Ramon Pallas –Arney and Miguel Angel
Mayosky, “AC-Coupled Front-End for Biopotential Measurements”,
IEEE Transaction on Biomedical Engineering, Vol. 50, No.3, March
2003, pp. 391–395.
[11] Enrique Mario Spinelli, Nolberto Mart&iacute;nez, Miguel Angel Mayosky,
Ramon Pallas –Arney, “A novel Fully Differential Biopotential
Amplifier with DC Supression”, IEEE Transaction on Biomedical
Engineering, Vol. 51, No.8, August 2004, pp. 1444-1448.
157
VI. CONCLUSION
From the observation &amp; result as mentioned above in this
paper, it is concluded that common mode noise is cancelled
out due to differential configuration leading to enhanced
CMRR due to use of active switching transistorized
arrangement. Also the characteristics of instrumentation
amplifier which are very much vital in biomedical applications
like differential gain and CMRR, are related to frequency of
signal as well as that of the switching. The tolerance in the
values of R and C used in conventional RC coupling circuits
affects these characteristics significantly. The switching
methods proposed in this paper acts as a technique for
recovery of dynamic range of instrumentation amplifier for
both common as well as differential signals to be used in
biomedical applications.
[1]
Page
It can be again seen from the above table 4 that CMRR is
constant over wide entire dynamic range &amp; is better &amp;
enhanced as compared to previous set of observation for a
circuit with conventional AC coupling as well as both of the
switching method using transistors.
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