Safe-commutation principle for direct single-phase
AC-AC converters for use in audio power
amplification
Petar Ljušev, Michael A.E. Andersen
Ørsted • DTU, Automation
Technical University of Denmark
Kgs. Lyngby DK-2800, Denmark
Email: pl@oersted.dtu.dk
Abstract— This paper presents an alternative safe commutation
principle for a single phase bidirectional bridge, for use in the new
generation of direct single-stage AC-AC audio power amplifiers.
As compared with the bridge commutation with load current or
source voltage sensing, in this approach it is not required to do
any measurements, thus making it more reliable. Initial testing
made on the prototype prove the feasibility of the approach.
I. I NTRODUCTION
As the performance of modern power electronics components increases in terms of both switching speed and power
levels, new application fields emerge which have been previously ruled solely by the linear electronics. One of the most
exciting areas with a rapid and significant improvement in
the last few years is certainly the audio power amplification.
Switching-mode Class D audio power amplifiers are becoming
increasingly popular, as more and more commercial products
with appealing efficiency, audio quality, weight and dimensions are emerging on the market. However, its development
is far from being over and there are several issues that still
need to be resolved to gain even larger acceptance.
The extraordinary performance of the linear audio amplifiers
in classes A and B, their improved hybrid AB, as well as
complex B2 and G, expressed as very low total harmonic
distortion + noise (THD+N) levels and supplemented with
simple control techniques, made them the only audiophile
choice for a very long time. However, they are characterized
with poor efficiency, huge power losses, voluminous heat
sinks and therefore high price. The switching approach (ex.
Class D) in audio technology and acoustics is rather new,
since its inevitable nonlinearity, introduction of time delays
(rise, fall and dead times), distortion, EMC problems, need
for output filter as well as limited switching frequency made
it less preferable choice. However, introduction of advanced
control techniques [1] to compensate for the nonlinearities in
the modulation and subsequent amplification phase has even
led to several commercially available products. These products
feature extraordinary high efficiency, i.e. low dissipation at
full power and at idle, leading to smaller heat sinks, less
weight, better integration and thus smaller packages. Due to
this, power supply nominal rating and weight can be reduced,
while in the same time their THD+N is becoming comparable
with some of the high-end linear audio power amplifiers.
In the switching audio amplifiers (Class D) power MOSFETs are used to connect the power supply DC voltage with
a certain polarity across the combination of a filter and a
loudspeaker, as decided by the control algorithm. Therefore,
much of the performance of these switching audio amplifiers depends upon the quality and stiffness of the DC bus
voltage, thus often necessitating the usage of bulky linear
stabilized DC power supplies or switch mode power supplies
(SMPSs). Power supply rejection ratio (PSRR) can be greatly
improved by introduction of different linear and nonlinear
control methods, but this results in substantial increase in
control complexity.
SIngle Conversion stage AMplifier (SICAM) is the next important evolutionary step in designing switching-mode audio
power amplifiers. It is intended for the next generation of light,
highly efficient and cheap audio appliances with satisfactory
audio performance. Its unique structure and control principles
should provide energy efficiency beyond the levels of the
”classical” Class D audio amplifiers, by closely interconnecting the power supply stage and the subsequent audio power
amplification stage (multiplexing the functions of the active
switches has been already seen in many single stage power
factor correction topologies). This dedicated integration should
eventually result in simplification of the power supply, for
example transferring the demand for galvanic isolation to the
audio power amplifier itself. Lowering the demands against
the power supply will inevitably impose some trade-offs in the
level of complexity of the control and modulation methods in
the audio amplification stage. Some performance deterioration
is expected too. Notwithstanding, new topologies and control
methods will introduce lower component count and higher
energy efficiency, since most of the power losses are generated
in the power supply as a result of the dissipation during the
numerous internal electrical energy conversions.
II. SICAM STRUCTURE AND POSSIBLE TOPOLOGIES
SICAM presents, as its name explains, a direct conversion
topology for the AC mains waveform into an AC waveform of
arbitrary shape and frequency, representing some audio signal.
The structure of the SICAM provides galvanic isolation, except
for those application where safety precautions are not needed
due to the complete isolated inclosure of the device. Being a
home appliance with low power consumption, the SICAM is
intended for single phase usage and this is by far the most
stringent limitation we have faced.
In the three-phase case, the topology which is capable
of a single stage power conversion is known as a matrix
converter. Matrix converter is capable of direct transformation of sinusoidal voltage level and frequency without any
energy storage components, but some limitations apply [2].
In fact, maximum output voltage level is dependent on the
input voltage amplitude and use of some reactive elements is
compulsory, in order to obtain input current and output voltage
filtering. The obvious advantages of the matrix converter
are: sinusoidal input and output waveforms, less higher order
harmonics and no subharmonics, bidirectional flow of energy,
minimal size reactive components (for filtering purposes) and
adjustable power factor. This approach was therefore considered for building a non-isolated SICAM. Due to availability of
three phase voltages, three phase matrix converter can provide
continuous power flow from the grid to the load and this can be
controlled very effectively, so that no reactive storage elements
are needed. It is obvious that in the single phase case, the
power transfer can be achieved only at instantaneous input
voltages different than zero [3], making the output voltage
phase locked to the input AC mains voltage. High dynamics
of the audio signal can not tolerate any interruptions in the
power flow and this makes the single-phase matrix converter
useless for the intended application.
The above discussion clearly states that although SICAM
tends to be a single conversion stage topology, it must be
constructed in a way that incorporates some kind of reactive
energy storage elements, i.e. it necessitates a simple power
supply. On the other hand, the voltage output and the structure
of the power supply are not defined and limited, as long as the
subsequent audio amplification stage is successfully integrated
with it and performs satisfactorily, as shown in the principal
block scheme in Fig. 1.
The following SICAM topologies have been identified as
interesting to pursue:
• AC-AC - the power supply creates HF AC voltage in
conjunction with a cycloconverter SICAM or creates
another displaced (quadrature) phase voltage for use in
a two-phase matrix converter;
• DC-AC - simple rectifier and storage capacitor are used
in the power supply providing uninterrupted power flow
to the inverter SICAM, with or without a transformer as
a galvanic barrier; and
• DC-DC - several simple DC-DC converters are connected
differentially across the load to create the desired AC
voltage.
While AC-AC SICAMs look expensive and difficult to
construct at the moment from the power supply perspective
and their performance is questionable, various kinds of isolated
Pulse Width Modulated (PWM) and Pulse Density Modulated
(PDM) DC-AC SICAMs are already investigated, designed
and tested. Although the group of DC-DC SICAMs sounds
extraordinary, the importance of that approach is in its modularity, i.e. DC-DC converters are reconsidered as elementary
building blocks which can be cascaded and anti-paralleled to
yield structures of increased complexity and functionality.
III. PWM/PDM DC-AC SICAM S
The principle block diagram of the DC-AC SICAM is
shown in Fig. 2. Simple power supply is used to rectify
the AC mains voltage, followed by an inverter as an input
stage, an isolation barrier in a form of a high frequency (HF)
transformer and a bidirectional bridge as an output stage.
Depending on the output power level of the audio amplifier
and the desired complexity of the HF transformer, both input
and output sections can be designed as either a push-pull, halfbridge or a full bridge. It is clear from the block diagram that
the purpose of the input stage is to create HF AC voltage
using the rectified mains voltage and transfer the energy
over the isolation barrier, thus keeping the dimensions of the
transformer small. In the output stage, bidirectional bridge
is performing simultaneously rectification and inversion of
the secondary voltage according to the audio signal to be
amplified.
Fig. 2.
Fig. 1.
Block diagram of an isolated DC-AC SICAM
Principal block scheme of the SICAM
Fig. 1 shows how complex and diverse the SICAM project
can be, when one looks at the possible topologies which can
come into play. To make the analysis easier, classification of
the SICAMs was made according to the type of the input
voltage to the power amplification stage provided by the
simple power supply and the type of the output voltage.
A. Modulation of transformer voltages
Looking from a control perspective of the input stage, the
transformer voltages can be made either dependent or totally
independent from the audio signal. DC to low-frequency
inventer in [4] and used later in [5] presents a SICAM topology which uses PWM audio modulated transformer primary
voltages, but the large low frequency (LF) voltage content
which can stretch down to 20 Hz is reduced by toggling
the polarity of each second pulse. This reduces the flux
swing in the transformer core, thus effectively decreasing the
transformer size. In the subsequent output stage the polarity
of the toggled voltage pulses is reversed back to the correct
one, as to resemble the audio signal. The approach is very
interesting and not too complex to implement, though it has
some shortcomings. Apart from the difficulty of providing a
suitable commutation technique for the bidirectional bridge in
the output stage, having the transformer voltages dependent
on the audio signal will make very hard to put any auxiliary
windings on the same transformer core and provide continuous
power flow to the control and driving circuitry. This will
eventually increase the price and reduce the efficiency of the
design, as another converter is to be implemented to derive
the necessary auxiliary voltages.
Approaches where the input stage does not modulate transformer voltages and the whole audio modulation task is
performed solely by the output stage are preferred, since the
transformer flux levels become more predictable and conservative (bulky) design of the transformer can be successfully
avoided. This will make possible not only to put several
auxiliary windings on the main transformer core for powering
up the control and driving circuitry, but will also solve the
problem of transferring the audio reference over the barrier to
the primary high-voltage side and will eventually reduce the
complexity of the design.
B. Commutation of the bidirectional bridge
The output section consists of bidirectional i.e. four quadrant switches (4QSWs) in different configurations. Bidirectional switches can block voltages of either polarity and
can conduct currents of either polarity. Although this sounds
perfect, it is really challenging to perform commutation of
the latter [6]. The goal of the commutation is to displace the
load current from one set of 4QSWs to another set of 4QSWs
without any interruption which results in dangerously high
overvoltages across the inductive part of the load, and in the
same time not to cause any short-circuit on the input voltagesource side.
The bidirectional bridge in the output stage introduced in
the previous section can be also referred as a single-phase to
single-phase matrix converter. This makes the matrix converter
commutation techniques applicable to the aforementioned audio output stage. However, some limitations when moving
from the largely exploited three-phase case [6], [7], [8], [9] to
the single-phase case are observed.
The simplest strategies for commutation of the switches,
which consist of providing a dead-time between the switches
or allowing an overlap, result in disrupted load current or
short circuited source voltage correspondingly, so they are both
theoretically and practically unusable.
The basic commutation strategies are therefore:
• current controlled - the commutation strategy is strongly
relying on accurate determination of the load current
direction, so that a continuous current path is provided
without allowing a short circuit of the source; and
• voltage controlled - the commutation strategy is based on
the input voltage polarity measurement, so that the right
switches are chosen which not result in any violation of
the electrical laws.
Both presented strategies have some practical pitfalls. Current controlled commutation strategies are very popular in the
motor drives community, since the load current is a measured
quantity in order to accomplish field oriented control. However, large current measurement and accurate current zerocrossing detection in a same current sensor are two opposing
goals, since for current measurements (for ex. in motor drives)
a wide range of load currents should be accommodated, while
for zero current detection very low noise and high accuracy
environment with very small currents should be provided.
Therefore, poor results are reported with current controlled
commutation only because of the low sensitivity, high noise
and offset levels of the present state-of-the-art current sensors.
An obvious advantage of this approach is that load current
can change its direction even when the commutation process
has started. In that case, the other current direction will be
usually prohibited, so the load current will settle at zero until
the commutation process has ended.
Voltage controlled strategies are facing the same problems
of inadequate measuring sensors like the current controlled
strategies, despite of the additional volume and price burden
imposed by their installation. However, they have an another
disadvantage - input voltage reversal during a commutation
is not allowed since this can result in having wrong switches
turned on with possible disastrous results.
Combination of both techniques can also be used to alleviate
commutation during uncertain conditions. In these approaches
the alternative strategy is used whenever the first strategy
enters into the uncertainty region where the voltage/current
sign can not be accurately determined. Unfortunately, in the
case of both voltage and current sign being uncertain, further
switching can be catastrophic. To avoid this some retarding
or prohibition techniques are executed where the switching
is avoided as long as the voltage and/or current signs are
uncertain, but the quality of the output voltage and input
current are therefore compromised.
There are some three-phase techniques which can be used
to overcome the pitfalls of the aforementioned strategies, like
replacement [8] and prevention [9]. These consist of a smart
commutation algorithm which changes the order of switching
between the phases where the input line voltage sign is not
certain by putting the third phase for safe commutation in
between. Basically this results in shifting from one uncertain
commutation to two certain, but more lossy commutations.
Since in the case of the single-phase bidirectional bridge for
audio output stage there is no other phase to safely commute
to, another approach must be found.
The only advantage of the bidirectional bridge in a singlephase SICAM over the three-phase matrix converter, is that
the input voltage to the former can be shaped by choosing an
appropriate switching pattern of the input stage. This has been
utilized in [10], where intentional short intervals with zero
input voltage are inserted to allow for the safe commutation
of the bidirectional bridge through overlapping of the 4QSW
conduction times without short-circuit condition. Although
relatively simple to implement, the latter approach requires
a full bridge on the primary side to allow for the 3-level
modulation, i.e. for the switching intervals with zero voltage
at the transformer secondary side, which is usually not price
competitive for the lower power SICAMs.
IV. S AFE COMMUTATION OF THE OUTPUT STAGE
THROUGH MASTER / SLAVE OPERATION
Before presenting the proposed method for performing safe
commutation of the output stage, a short review of the bidirectional switches is needed. 4QSW is constructed by anti-parallel
connection of two voltage 2QSWs, or by anti-series connection
of two current 2QSWs. In the cases where the 4QSW consists
of two anti-parallel voltage 2QSWs, the current paths for
different current directions are clearly separated, and each of
the current directions can be separately controlled. This is not
so obvious for the 4QSW consisting of two current 2QSW and
this important feature can emerge only after combining the two
2QSW together. Therefore, an anti-parallel connection of two
voltage 2QSWs was used as a 4QSW, as shown in Fig. 3.
Fig. 3.
MOSFET bidirectional bridge as audio output stage
The main idea of the proposed approach is to implement
modified voltage controlled commutation of the output stage.
Since the input voltage to the bidirectional bridge is created
by switching the primary side input stage, this process is
completely controlled and the polarity of the input voltage
is known at almost any time. The only problem with voltage
controlled commutation which can appear, as stated before,
is if the output stage receives a command from the audio
modulator to switch the voltage across the load, and in the
same time the input stage is performing a transition. Than,
the sign of the transformer secondary side voltage is uncertain
and wrong commutation sequence can result in catastrophic
malfunction of the converter. The proposed solution consists
of avoiding any simultaneous transition of both stages, so that
a Master/Slave operation of the input/output stage is implemented. The control block diagram is shown in Fig. 4 and
the premises for the operation of this compound input/output
stage are like follows:
1) Each of the stages can be either a master or a slave,
2) Both stages can not be masters in the same moment,
3) It is possible for both stages to reside in a slave mode
at the same time,
4) Transition of one stage from slave to master is done only
if the other stage is in slave mode,
5) Transition back from master to slave of one stage does
not depend on the internal events in the other stage, but
it allows the latter to transfer from slave to master, if it
was waiting for a permission.
Fig. 4.
Master/slave control of isolated PDM/PWM SICAM
So the proposed SICAM can be imagined like having a
single logic ”master/slave” line, which can have two possible
states. If it is implemented with negative logic, than logic zero
(”0”) means that one of the stages is in master mode and
logic one (”1”) means that both stages are slaves. Each of
the stages becomes a master, if premise 4 is satisfied and a
command is issued to make a transition: for the input stage it
means that the voltage across the transformer primary is to be
reversed and for the output stage it means that a commutation
of the bidirectional bridge is to be undertaken in order to
accommodate the audio reference. Whenever a transition is
to be made and the master/slave line is idling at ”1”, the
corresponding stage is pulling the master/slave line down to
”0” occupying it for the time of its transition. Master/slave
line will be released as soon as the transition has ended,
but according to premise 5 this is done without any regards
to the other slave stage. In this way, the ”master” can take
all the time needed to finish the transition safely. This is,
however, done on expense of increased distortion, sacrificing
some performance. Due to the premise 1, both stages can
not be in master mode in the same time, so transitions in
both stages can not occur simultaneously, thus effectively
avoiding any possible dangerous commutation. The rest of
the commutation proceeds according to the voltage-controlled
commutation sequence described in Tab. I and depends solely
upon the input voltage vin polarity and the beginning state.
V. E XPERIMENTAL RESULTS
A prototype implementing master/slave safe commutation
technique was built and tested for performance. Both input
and output stages were implemented as full-bridges, although
TABLE I
VOLTAGE CONTROLLED COMMUTATION SEQUENCE OF THE
BIDIRECTIONAL BRIDGE IN FIGURE 3
beginning state
1,6 & 3,8
2,5 & 4,7
vin
>0
<0
>0
<0
on %
4,7
2,5
3,8
1,6
off &
1,6
3,8
2,5
4,7
on %
2,5
4,7
1,6
3,8
off &
3,8
1,6
4,7
2,5
final state
2,5 & 4,7
1,6 & 3,8
the maximum output power was limited at 100 W. Switching
frequency of the input stage was set to fs1 = 100 kHz
and of the output stage fs2 = 200 kHz. The proposed safe
commutation algorithm was implemented in a bunch of digital
logic TTL gates and the necessary delays were implemented
as RCD elements. This increased the design dimensions and
weight, but the control algorithm is going to be transferred to
a PLD/FPGA chip soon, which should bring the design within
the expected volume. The photo of the prototype is given in
Fig. 5.
Fig. 5. Photo of the prototype master/slave-controlled isolated PWM DC-AC
SICAM
Input stage M/S line driver base voltage, output stage M/S
line driver base voltage, M/S line voltage and input stage
voltage polarity are shown in figure 6. The high levels of
the M/S line driver base voltages correspond to the instants
when the input stage or the output stage are performing the
switching, so they are pulling down the M/S line voltage to
prevent switching of the other stage. It is interesting to notice
that, whenever the audio signal reference stays the same while
the input stage is performing transition, the latter is followed
by an immediate transition of the output stage, just to conform
with the audio signal reference. This can be seen from the base
voltages of the M/S line drivers belonging to the input and the
output stage, which occupy the M/S line immediately one after
another.
The delayed driving signals for the MOSFETs: 1&6, 2&5,
3&8 and 4&7 are given in figure 7. This transition occurs
due to change of the output voltage of the input stage from
positive to negative (upper rail negative), according to Tab. I
Fig. 6. 1) Input stage M/S line driver base voltage, 2) output stage M/S line
driver base voltage, 3) M/S line voltage and 4) input stage voltage polarity
(T1 /T4 driving signal) (all probes 10x)
and starting from 1,6&3,8 turned on.
Fig. 7. 1) MOSFETs 1&6 driving signal, 2) MOSFETs 2&5 driving signal,
3) MOSFETs 3&8 driving signal and 4) MOSFETs 4&7 driving signal for
negative rail voltage (all probes 10x)
Figure 8 is showing the load (loudspeaker) voltage, the
voltage applied across the output filter and a reference signal,
with a frequency of f = 10 kHz and in open-loop control.
Due to the immense noise created by the switching of both the
input and the output stage, the measurement of the reference
signal is corrupted. The output voltage is following the signal
reference, although the distortion is high.
The load voltage, the reference signal and the Fast Fourier
Transform (FFT) of the load voltage, for output powers Pout =
1 W and 10 W with closed-loop control are given in Fig 9 and
10. It is obvious from the diagrams that the switching action
of the prototype results in large quantity of noise injected
in the closed loop, which disrupts the proper operation of
the SICAM. Noise is observed especially at the switching
frequency of the output stage fs2 = 200 kHz, but also at the
switching frequency of the input stage fs1 = 100 kHz. PSRR
Fig. 8.
1) Load voltage, 2) output stage voltage at output filter and
3) reference signal (probes 1 and 2 - 50x, 3 - 10x)
Fig. 10. Pout =10 W: 1) Load voltage, 2) reference signal and M1) FFT
(probe 1 - 50x, probe 2 - 10x)
is also compromised due to the low overall gain of the closed
loop and non-regulated power supply, and this is observed as
a harmonic with a high amplitude at a frequency of 50 Hz.
almost certain that during each switching transition of the
bidirectional bridge there will be a path for the load current
and no overvoltages will be created due to its interruption.
ACKNOWLEDGMENT
The SICAM project is funded under the grant of the Danish
Energy Authority EFP no. 1273/02-0001, in cooperation with
Bang & Olufsen ICEpower a/s in Kgs. Lyngby, Denmark.
R EFERENCES
Fig. 9. Pout =1 W: 1) Load voltage, 2) reference signal and M1) FFT (probe
1 - 50x, probe 2 - 10x)
Although the closed-loop performance of the DC-AC PWM
SICAM amplifier with master/slave operation is moderate,
the measurements have shown that the proposed commutation
principle is capable of performing a safe commutation of the
output stage each time a transition is needed. The work on the
prototype to improve the control and the audio performance
continues.
VI. C ONCLUSION
The paper presented the idea of master/slave operation of
the input and output stage of a SICAM audio power amplifier
for obtaining safe commutation of the bidirectional bridge.
Through analysis and prototyping, the approach proved to
be viable and with a more careful converter/control design
and commutation delay selection can operate safely in a large
range of input voltages and output powers. Therefore it is
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