energies
Article
An Integrated Multifunctional Bidirectional
AC/DC and DC/DC Converter for Electric
Vehicles Applications
Liwen Pan * and Chengning Zhang
National Engineering Laboratory for Electric Vehicles, Beijing Institute of Technology, Beijing 100081, China;
mrzhchn@bit.edu.cn
* Correspondence: panliwen2002@163.com; Tel.: +86-10-6891-2947
Academic Editor: King Jet Tseng
Received: 1 March 2016; Accepted: 22 June 2016; Published: 28 June 2016
Abstract: This paper presents an on-board vehicular battery charger that integrates bidirectional
AC/DC converter and DC/DC converter to achieve high power density for application in electric
vehicles (EVs). The integrated charger is able to transfer electrical energy between the battery pack
and the electric traction system and to function as an AC/DC battery charger. The integrated charger
topology is presented and the design of passive components is discussed. The control schemes are
developed for motor drive system and battery-charging system with a power pulsation reduction
circuit. Simulation results in MATLAB/Simulink and experiments on a 30-kW motor drive and
3.3-kW AC/DC charging prototype validate the performance of the proposed technology. In addition,
power losses, efficiency comparison and thermal stress for the integrated charger are illustrated.
The results of the analyses show the validity of the advanced integrated charger for electric vehicles.
Keywords: electric vehicle; integrated bidirectional charger; AC/DC converter; DC/DC converter
1. Introduction
As the demand for renewable energy increases, alternative and sustainable transportation
methods become more attractive in comparison to conventional transportation. Electric vehicles
(EVs) are efficient solution which draws interest of government, manufacturers and researchers [1–4].
However, the lack of a convenient charging options prevents widespread use of EVs. In order to
overcome this issue and make EVs more competitive, an efficient and reliable, high power density, low
cost, small size and lightweight on-board charger for EVs becomes a valuable topic [5].
EV charging power levels defined by society of automotive engineers (SAE) are summarized and
classified in Figure 1: Level 1 charging is slow charging; Level 2 charging is assumed primary charging
level; and Level 3 is intended for off-board charging infrastructure providing quick charging. In terms
of application, an off-board charger can be designed for high charging rates and is less constrained
by size and weight. On-board battery chargers can reduce user range anxiety by allowing for battery
packs to be charged from any available power outlet [6]. However, on-board charging increases system
volume and weight. A compact, high power density charger is needed to maintain vehicle operation
efficiency. Galvanic isolation from the utility grid is a favorable option in the charger circuits because
of safety reasons, but isolated on-board chargers are usually avoided due to their cost impact on the
system. With a dedicated socket-outlet, earth current monitoring is an optional function as mentioned
in the standard [7]. The electrical continuity of the earth conductor should be permanently monitored
by the charger, and in the case of loss of electrical continuity of the earth conductor, the charger shall
be switched off. In addition, if the traction battery is bonded to the vehicle chassis, the charging
system shall provide a galvanic isolation between the mains and the battery. For the chargers, they
Energies 2016, 9, 493; doi:10.3390/en9070493
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Energies 2016, 9, 493
Energies 2016, 9, 493
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battery.
For the
chargers,
theycharging
are classified
into conductive
charging
inductive
charging
are
classified
into
conductive
and inductive
charging
[8]. Theand
inductive
charging
can[8].
be
The inductivewireless
chargingcharging,
can be implemented
wireless
charging, which has lower efficiency.
implemented
which has lower
efficiency.
Level 1
On-Board Charging
Wheel
Wheel
Level 2
On-Board Charging
Regenerative Power Flow
Braking
(Bidirectional)
Grid
Battery Charger
AC
DC
V2G
Level 3
Off-Boarding Charging
Traction Drive
Battery Pack
DC
Cdc
Power Flow
(Bidirectional)
Motor Drive
AC
Charger
DC
DC
Electric
Motor
Electronic Loads
(Light, Heater, Aux,
etc)
Wheel
Wheel
Figure
1. 1.
Electric
vehicles
by society
societyofofautomotive
automotiveengineers
engineers
(SAE).
Figure
Electric
vehicles(EVs)
(EVs)battery
batterychargers
chargers classification
classification by
(SAE).
..
.. .
.. .
..
An on-board battery charger is an inevitable part of an EV powertrain [9]. From the perspective
Energies
9, 493 battery charger is an inevitable part of an EV powertrain [9]. From the perspective
3 of 23
An2016,
on-board
of power electronics, the on-board charger for an EV consists of an AC/DC power converter and
of power electronics, the on-board charger for an EV consists of an AC/DC power converter and some
some
filter
components.
An
efficient solution
to make
a high power
density
system is
integrating
the
based
on the
operating
conditions.
The
proposed
structure
contains
one electric
motor,
one
input
filter
components.
An efficient
solution
to make
a high
power density
system
is integrating
the
charger
charger
unit
with
motor
drive
converter.
In
the
case
of
using
an
integrated
charger,
as
illustrated
in
inductor,
one electromagnetic
one DC-link
capacitor,
six power
switching
unit
with motor
drive converter. interference
In the case of(EMI)
using filter,
an integrated
charger,
as illustrated
in Figure
2, it is
Figure
2,
it
is
viable
to
share
the
same
power
switching
devices
and
passive
components
because
devices,
six diodes
and power
six set relays,
which
are used
to switch
operating modes.
viable
to share
the same
switching
devices
and passive
components
because motor drive mode
motorThe
drive
mode
and
battery-charging
mode
do
not
happen
at
the
same in
time
[10]. This
configuration
batteries canmode
be charged
the external
supply.
charging
mode,
the motor
and battery-charging
do notby
happen
at the power
same time
[10].When
This configuration
can effectively
can
effectively
reduce
the
size
and
cost
of
the
overall
system.
Meanwhile,
with
the
DC/DC
needs
to
be
disengaged
for
safety
purposes.
Therefore,
relay
J1
can
disconnect
the
motor
and
reduce the size and cost of the overall system. Meanwhile, with the DC/DC buck/boost charging
buck/boost
charging
capability,
it
enables
operating
with
a
wide
charging
range.
In
addition,
converter it
physically
from the battery-grid
side, ensuring
is not powered
in battery-charging
capability,
enables operating
with a wide charging
range.the
In EV
addition,
essential capabilities
such as
essential
capabilities
such as buck
for has
the five
battery
voltage
in motor
drive
mode
andmode,
boost AC/DC
for the
mode.
Proposed
integrated
charger
basic
modes,
namely,
motor
drive
buck for the battery voltage in motor drive mode and boost for the battery voltage in regenerative
battery
voltage
in
regenerative
braking
mode
can
be
achieved
as
well
[11].
Consequently,
the
battery-charging
regenerative
braking
mode, DC/DC
battery-charging
mode and
V2G mode.
braking
mode can mode,
be achieved
as well [11].
Consequently,
the conventional
configuration
consists
of at
conventional
configuration
of at least
two 1.
individual converters for two independent
The energy
flow
in each
modeconsists
are
in Table
least
two individual
converters
forsummarized
two independent
operation modes.
operation modes.
In order to obtain a simple and low cost design,
transformer-less
topology
Bidirectional
AC/DC and DC/DC
Converteris used in the
proposed integrated charger [12]. Compared with traditional design, which has a transformer at the
AC grid side, the transformer-less topology will improve the efficiency, reduce the total weight and
S1
D1 S3
D5
D3 S5
Vmotor
allow for less complicatedJ1control. J6
In this paper, an integrated bidirectional AC/DC and DC/DC
La
converter is proposed to implementL the integration of EV with a household electric system. The
2
Lb
system configuration and five operational
modes
principles
are illustrated
andCdc
theVbatt
pulsating power
C
B
A
L3
Lc
L
s
reduction circuit is presented. The control schemes are described and compared for the
J4
battery-charging mode in order to achieve a better performance. J3
This paper is organized as follows:S2
SectionD22 S4
introduces
the proposed
D4 S6
D6 Cs topology and explains the
modes of operation in details. The utilized control schemes are illustrated in Section 3. The
simulation and experimental results are presented in Section 4. In addition, power loss, system
Ripple energy
J2
efficiency and thermal stress calculation results
are presented
in this
section. Finally, Section 5
storage
circuit
J5
L1
summarizes the study based on the achieved results.
.
.
.
.
... .
.. . ..
.
EMI Filter
2. Proposed Integrated Charger Topology and Operation Modes
Vgrid
Conventional electric drive system contains a DC/AC converter while the battery-charging
Figure
Proposed
integrated
bidirectional
AC/DCand
and
DC/DC converter.
2.2.Proposed
integrated
AC/DC
DC/DC
system includesFigure
an AC/DC
converter.
Thebidirectional
proposed converter
enhances
them into an integrated
multifunctional bidirectional AC/DC and DC/DC converter, which can also charge the battery from
Table 1. Operation modes of the proposed converter.
DC power sources such as micro-grid. Proposed integrated charger topology is shown in Figure 2
Operation Mode
Energy Flow
Mode
Propulsion
AC/DC Battery-Charging
Vbatt to Vmotor
Vgrid to Vbatt
Buck
Boost
Energies 2016, 9, 493
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In order to obtain a simple and low cost design, transformer-less topology is used in the proposed
integrated charger [12]. Compared with traditional design, which has a transformer at the AC
grid side, the transformer-less topology will improve the efficiency, reduce the total weight and
allow for less complicated control. In this paper, an integrated bidirectional AC/DC and DC/DC
converter is proposed to implement the integration of EV with a household electric system. The system
configuration and five operational modes principles are illustrated and the pulsating power reduction
circuit is presented. The control schemes are described and compared for the battery-charging mode in
order to achieve a better performance.
This paper is organized as follows: Section 2 introduces the proposed topology and explains the
modes of operation in details. The utilized control schemes are illustrated in Section 3. The simulation
and experimental results are presented in Section 4. In addition, power loss, system efficiency and
thermal stress calculation results are presented in this section. Finally, Section 5 summarizes the study
based on the achieved results.
2. Proposed Integrated Charger Topology and Operation Modes
Conventional electric drive system contains a DC/AC converter while the battery-charging
system includes an AC/DC converter. The proposed converter enhances them into an integrated
multifunctional bidirectional AC/DC and DC/DC converter, which can also charge the battery from
DC power sources such as micro-grid. Proposed integrated charger topology is shown in Figure 2
based on the operating conditions. The proposed structure contains one electric motor, one input
inductor, one electromagnetic interference (EMI) filter, one DC-link capacitor, six power switching
devices, six diodes and six set relays, which are used to switch operating modes.
The batteries can be charged by the external power supply. When in charging mode, the motor
needs to be disengaged for safety purposes. Therefore, relay J1 can disconnect the motor and converter
physically from the battery-grid side, ensuring the EV is not powered in battery-charging mode.
Proposed integrated charger has five basic modes, namely, motor drive mode, AC/DC battery-charging
mode, regenerative braking mode, DC/DC battery-charging mode and V2G mode. The energy flow in
each mode are summarized in Table 1.
Table 1. Operation modes of the proposed converter.
Operation Mode
Energy Flow
Mode
Propulsion
AC/DC Battery-Charging
DC/DC Battery-Charging
V2G
Regenerative Braking
V batt to V motor
V grid to V batt
V dc to V batt
V batt to V grid
V motor to V batt
Buck
Boost
Buck/Boost
Buck
Boost
2.1. Motor Drive Mode
The motor drive mode is buck operation from the battery to the traction motor. In this mode,
V batt and V motor sequentially become input and output voltage, respectively. Phase A, B and C
use a three-phase inverter as buck circuit. Power flow from the battery to the traction motor
can be transferred by buck chopper. A 30 kW surface-mounted permanent magnet synchronous
motor is used for motor drive operation. The rated rotor speed and rated torque are 3000 rpm and
100 Nm, respectively.
2.2. AC/DC Battery-Charging Mode
The battery pack can be charged by the grid power supply. In boost operation for AC/DC
battery-charging mode, unipolar modulation is selected to control the single-phase converter, as shown
in Figure 3.
Energies 2016, 9, 493
Energies 2016, 9, 493
. .
. .
S1
L1
uac
.
D1 S3
S1
D3
L1
C
A
.
. .
. .
Udc
uac
.
D 1 S3
D 2 S4
(a)
S1
L1
uac
.
(b)
D3
S1
L1
.
. .
Udc
D 2 S4
S2
C
A
D 2 S4
(c)
S1
uac
.
L1
uac
.
C
A
. .
(d)
D3
S1
.
. .
B
S2
D2 S4
(e)
D2 S4
S2
L1
C
A
D4
D3
.
. .
Udc
B
D4
D1 S 3
Udc
D4
D1 S3
B
S2
.
. .
. .
B
D4
D1 S3
D3
C
A
B
S2
4 of 23
4 of 23
Udc
uac
.
D4
. .
D1 S3
D3
C
A
.
. .
Udc
B
S2
D2 S4
D4
(f)
Figure
Unipolarmodulation
modulationprocess:
process:
Capacitor
charge
battery,
grid
and capacitor
charge
Figure 3.
3. Unipolar
(a)(a)
Capacitor
charge
battery,
grid and
capacitor
charge inductor;
inductor;
(b)
Capacitor
charge
battery,
grid
charge
inductor;
(c)
Grid
and
inductor
charge
capacitor
(b) Capacitor charge battery, grid charge inductor; (c) Grid and inductor charge capacitor and battery;
and
battery; (d)
Capacitor
charge
battery and
Capacitor
charge
(f) Grid
(d) Capacitor
charge
battery
and inductor;
(e) inductor;
Capacitor(e)
charge
battery;
and battery;
(f) Grid and
and inductor
and
inductor
charge
capacitor,
inductor
charge battery.
charge
capacitor,
inductor
charge
battery.
The input inductance is calculated based on the current ripple on the inductor. The ripple
The input inductance is calculated based on the current ripple on the inductor. The ripple current
current can be estimated based on the worst case ripple current. The relationship of current ripple,
can be estimated based on the worst case ripple current. The relationship of current ripple, input
input inductance and DC bus voltage can be expressed in Equation (1):
inductance and DC bus voltage can be expressed in Equation (1):
𝑈 sin ω𝑡
𝑈 sin ω𝑡
´ −
¯)
𝑈𝑑𝑐 ×
× (1
𝑈𝑑𝑐
𝑈𝑑𝑐
Usinωt
(1)
𝐼𝑝𝑒𝑎𝑘 = Udc ˆ Udc ˆ 1 ´ Usinωt
Udc
2
×
𝐿
×
𝑓
𝑠
I peak “
(1)
2 ˆ L ˆ fs
2
2
2
2
4
𝑈 𝐼
𝜔 𝐿𝐼
√b
+
(2)
42 I 2 ω24L2 I 4
U
𝐶=
4 ×`
4× ω
2
×
𝑈
∆𝑈
𝑑𝑐
𝑑𝑐
C“
(2)
2 ˆ Udc ˆ ∆Udc ˆ ω
where, Udc is the DC bus voltage, Usin(ωt) is the instantaneous value of gird side voltage and fs is the
where, U is the DC bus voltage, Usin(ωt) is the instantaneous value of gird side voltage and f s is the
switchingdcfrequency.
switching frequency.
In AC/DC battery-charging mode, the integrated converter operates as a single-phase
In AC/DC battery-charging mode, the integrated converter operates as a single-phase pulse-width
pulse-width modulation (PWM) rectifier. H-bridge is dominant topology for Level 2 application low
modulation (PWM) rectifier. H-bridge is dominant topology for Level 2 application low voltage and
voltage and current requirement. Besides the charging capability, high performance, including low
current requirement. Besides the charging capability, high performance, including low output current
output current harmonic injection to the grid, is essential [13,14]. The energy passes the input
harmonic injection to the grid, is essential [13,14]. The energy passes the input inductor and then
inductor and then the H-bridge finally charges the DC capacitor. The input inductor is designed
the H-bridge finally charges the DC capacitor. The input inductor is designed based on the current
based on the current ripple on the inductor. Around 10% of the peak output current is used to
ripple on the inductor. Around 10% of the peak output current is used to determine the inductance.
determine the inductance. The correlation between the DC bus voltage, input inductor and current
The correlation between the DC bus voltage, input inductor and current ripple is described in Figure 4a.
ripple is described in Figure 4a.
Energies 2016, 9, 493
Energies 2016, 9, 493
5 of 23
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(a)
(b)
Figure 4.
4. (a)
(a) Relationship
Relationship of
of input
input inductor,
inductor, current
current ripple
ripple and
and DC
DC bus
bus voltage;
voltage; and
and (b)
(b) Relationship
Relationship
Figure
among
angle
of
the
supply
voltage,
supply
current
and
supply
frequency.
among angle of the supply voltage, supply current and supply frequency.
Considering the DC bus capacitor, the fairly large bus capacitor needed to smooth the DC
Considering the DC bus capacitor, the fairly large bus capacitor needed to smooth the DC voltage
voltage can be expressed by Equation (2). A single-phase PWM converter has a simple structure,
can be expressed by Equation (2). A single-phase PWM converter has a simple structure, higher
higher efficiency, fewer components and lower cost in comparison to a three-phase converter. The
efficiency, fewer components and lower cost in comparison to a three-phase converter. The only
only drawback is the ripple power pulsating with twice the grid frequency [15]. At steady state, the
drawback is the ripple power pulsating with twice the grid frequency [15]. At steady state, the current
current and voltage of capacitor absorbing ripple energy can be expressed as Equations (3) and (4):
and voltage of capacitor absorbing ripple energy can be expressed as Equations (3) and (4):
(3)
𝐼𝐶 = 𝐼̅𝐶 × cos(2ω𝑜 𝑡 + 𝜑 + 𝛾)
IC “ IC ˆ cos p2ωo t ` ϕ ` γq
(3)
−1
̅
̅
̅̅̅̅
𝑉𝐶 = 𝐼𝐶 × sin(2𝜔𝑜 𝑡 + 𝜑 + 𝛾) × (2ω𝑜 𝐿) + 𝐼𝐶 × sin(2𝜔𝑜 𝑡 + 𝜑 + γ) × (´
)¯ + 𝑉𝑑𝑐
2ω
(4)
´𝑜1𝐿 ` V
VC “ IC ˆ sin p2ωo t ` ϕ ` γq ˆ p2ωo Lq ` IC ˆ sin p2ωo t ` ϕ ` γq ˆ 2ω
dc
oL
(4)
= X × sin(2ω𝑜 𝑡 + 𝜑 + γ) + ̅̅̅̅
𝑉𝑑𝑐
“ X ˆ sin p2ωo t ` ϕ ` γq ` Vdc
where, 𝜑 is the phase angle, X is the impendence, VDC is the DC bus average voltage and L is
where,
ϕ is the phase angle, X is the impendence, V DC is the DC bus average voltage and L is
input inductance.
inputThe
inductance.
integrated converter should meet all the demands from IEEE-1547, SAE-J2894 and
The
integrated
converter
meet
the demands
from IEEE-1547,
and
IEC-1000-3-2 standards
[16,17]. should
However,
the all
second-order
harmonic
current andSAE-J2894
corresponding
IEC-1000-3-2
standards
[16,17].
However,
the
second-order
harmonic
current
and
corresponding
ripple voltage will produce harmonic pollution in grid. Based on the above analysis, DC ripple
ripple
will circuit
produce
pollution
in grid.
Based
onrectifier
the above
ripple current
currentvoltage
reduction
onharmonic
single-phase
voltage
source
PWM
willanalysis,
play an DC
important
role to
reduction
circuit
on
single-phase
voltage
source
PWM
rectifier
will
play
an
important
role
to
solve
solve harmonic between converter and the grid. In addition, battery is expected to meet criteria
in
harmonic
between
converter
and
the
grid.
In
addition,
battery
is
expected
to
meet
criteria
in
terms
terms of long cycle life while maintaining most of its capacity at end of life. The 2ω0 pulsating power
of
long
cyclebattery
life while
maintaining
of its capacity
at end of life.
The 2ω
power may
0 pulsating
may
cause
heat
generationmost
imbalance.
The relationship
among
power
factor angles,
2ω0
cause
generation
imbalance.
The in
relationship
factor
2ω0 ripple
ripple battery
energy heat
and supply
frequency
is shown
Figure 4b. among
Several power
methods
haveangles,
been explored
to
energy
andthis
supply
frequency is
shown
in Figure
Several
methodsapplication.
have been explored
to deal with
deal with
second-order
ripple
power
from4b.the
single-phase
These methods
are
this
second-order
ripple power
the single-phase
application.
are summarized
summarized
into inductive
and from
capacitive
energy storage
shown in These
Figuremethods
5.
into inductive and capacitive energy storage shown in Figure 5.
For both single-phase
reduction
W-phaseportion and the ripple
U-phase V-phaserectifier
W-phase a 20 kHz triangular
U-phase
V-phase circuit,
modulation signal
and
are used for the PWM
S1 is generated.
S3
S5The power devises on U-phase
S1
S3 V-phase
S5
operation, where V U denotes U-phase control signal, V V represents V-phase control signal, V tri denotes
L
L
triangular modulation
signal, iU is U-phase current and V
UV represents the effective value of input
voltage. Figure 6 shows the typical waveforms
PWM rectifier. Figure
7 shows
the gate
Cdc Vbatt
Cdc Vbatt of proposed
Ls
Vac
Vac
signal of V U , V V and V tri during
L one switching cycle. The sum of U-phase duty cycle and V-phase
duty cycle is always equal to 1.
Cs
The method using capacitor as ripple energy storage component is illustrated and calculated
S2
S4
S6
in [18]. The inductor
to be
S2 is found
S4
S6 not as good as a capacitor in terms of energy density for integrated
charger application. Therefore, the capacitive ripple energy storage method is selected in this paper.
Constant current-constant
voltage (CC-CV) charging method(b)is selected in this paper.
(a)
Battery-charging typically follows a two-phase approach. In the first phase, constant current is
Figure 5. Inductive and capacitive energy storage classification: (a) Inductive energy storage;
used to charge the battery. When the battery reaches a specified set-point voltage, the charging circuit
.
. .
.
.
.
.
.
.
.
.
.
.
. .
.
.
and (b) Capacitive energy storage.
.
IEC-1000-3-2 standards [16,17]. However, the second-order harmonic current and corresponding
ripple voltage will produce harmonic pollution in grid. Based on the above analysis, DC ripple
current reduction circuit on single-phase voltage source PWM rectifier will play an important role to
solve harmonic between converter and the grid. In addition, battery is expected to meet criteria in
terms of long cycle life while maintaining most of its capacity at end of life. The 2ω0 pulsating power
Energies 2016, 9, 493
6 of 23
may cause battery heat generation imbalance. The relationship among power factor angles, 2ω0
ripple energy and supply frequency is shown in Figure 4b. Several methods have been explored to
deal
withonly
thisenough
second-order
ripple
power
theatsingle-phase
application.
These
methods are
provide
current to
maintain
thefrom
voltage
set-point voltage.
The CC-CV
battery-charging
summarized
intoisinductive
capacitive
energy storage shown in Figure 5.
method profile
shown in and
Figure
8.
.
U-phase V-phase
S1
L
Vac 9, 493
Energies 2016,
S3
. .
W-phase
S5
S1
.
Cdc Vbatt
.
S3
Vac
.
S5
.
.
.
L
L
.
U-phase V-phase W-phase
.
.
Ls
Cdc Vbatt
6 of 23
For both single-phase rectifier portion and the ripple reduction circuit, aCs 20 kHz triangular
modulation signal is generated. The power devises on U-phase
and
V-phase
are used for the PWM
S2
S4
S6
S2
S4
S6
operation, where VU denotes U-phase control signal, VV represents V-phase control signal, Vtri
denotes triangular modulation signal, iU is U-phase current and VUV represents the effective value of
(a)
(b)
input voltage. Figure 6 shows the typical waveforms of proposed PWM rectifier. Figure 7 shows the
Inductive
and
capacitive
energy
storage
classification:
Inductive
energy
storage;
gateFigure
signal
of
U, VV and
Vtricapacitive
during
one
switching
cycle.
The sum(a)
of(a)
U-phase
duty
cycle
and and
V-phase
Figure 5.
5. V
Inductive
and
energy
storage
classification:
Inductive
energy
storage;
(b)
Capacitive
energy
storage.
dutyand
cycle
is
always
equal
to
1.
(b) Capacitive energy storage.
.
. .
.
Du, Dv
Du ,Vu Vtri
.
Dv ,Vv
1.0
0.5
t
0
VUV: fundamental component
0
t
iU
0
t
iW: AC inductor method
0
t
iRECT
idc
0
0
π/4 3π/4 π 5π/4 7π/4 2π
iripple
Figure
Figure 6.
6. Waveforms
Waveforms of
of the
the pulse-width
pulse-width modulation
modulation (PWM)
(PWM) rectifier.
rectifier.
1
Vtri
VU
Vv
Gate Signal for S1
Gate Signal for S2
Du
.
0
0
π/4 3π/4 π 5π/4 7π/4 2π
iripple
Energies 2016, 9, 493
Figure 6. Waveforms of the pulse-width modulation (PWM) rectifier.
7 of 23
1
Vtri
VU
Energies 2016, 9, 493
7 of 23
Vv
The method using capacitor as ripple energy storage component is illustrated and calculated
in [18]. The inductor is found
to be
not as
Gate
Signal
forgood
S1 as a capacitor in terms of energy density for integrated
Dustorage method is selected in this paper.
charger application. Therefore, the capacitive ripple energy
Constant current-constant
voltage
charging method is selected in this paper.
Gate Signal
for (CC-CV)
S2
Battery-charging typically follows a two-phase approach.Dv
In the first phase, constant current is used
to charge the battery. When
the battery
reaches
specified
Switching
sequence
1a a 1b
1c set-point
1b 1a voltage, the charging circuit
provide only enough current to maintain the voltage at set-point voltage. The CC-CV
Figure
7. Gate
Gate
signal 8.
of PWM rectifier.
battery-charging method profile is
shown
in Figure
signal
U
U(t)
I
I(t)
0
Constant current
Constant voltage
t
Figure
Figure 8.
8. Constant
Constant current-constant
current-constant voltage (CC-CV) charging mode.
2.3.
2.3. Regenerative
Regenerative Braking
Braking Mode
Mode
The
limitationofofdriving
driving
range
is the
key restriction
for development
EVs. Regenerative
The limitation
range
is the
key restriction
for development
of EVs.of
Regenerative
braking
braking
is an approach
effective to
approach
to driving
extend the
driving
range
of electric
vehicles. braking
Regenerative
is an effective
extend the
range
of electric
vehicles.
Regenerative
is the
braking
is
the
process
where
some
of
the
kinetic
energy
stored
in
the
vehicle
transfer
into
electric
process where some of the kinetic energy stored in the vehicle transfer into electric energy in the
energy
the battery
during deceleration.
Generally,
the back electromotive
(EMF)
not high
battery in
during
deceleration.
Generally, the
back electromotive
force (EMF)force
is not
high is
enough
to
enough
to
directly
charge
the
battery,
and
a
bidirectional
converter
is
necessary
to
realize
the
energy
directly charge the battery, and a bidirectional converter is necessary to realize the energy transferring
transferring
the the
vehicle
and Regenerative
the battery. braking
Regenerative
braking
operation
is achieved
between the between
vehicle and
battery.
operation
is achieved
through
boost
through
boost
circuit.
Throughout
this
mode,
the
braking
energy
can
charge
the
battery
because
of
circuit. Throughout this mode, the braking energy can charge the battery because of insulated
gate
insulated
gate bipolar
(IGBT)
or metal oxide
semiconductor
field effect
transistor
bipolar translator
(IGBT) translator
or metal oxide
semiconductor
field effect
transistor (MOSFET)
bidirectional
(MOSFET)
bidirectional
conduction
character.
The
boost
operating
mode
during
regenerative
conduction character. The boost operating mode during regenerative braking is similar to the drive
braking
is similar
to the
drive motor
butthe
in reverse
When(SOC)
the battery
state of charge
(SOC)
motor but
in reverse
direction.
When
battery direction.
state of charge
is considered,
it is needed
is
considered,
it
is
needed
to
protect
the
battery
from
overcharging,
which
affects
the
battery
life.
To
to protect the battery from overcharging, which affects the battery life. To protect the battery from
protect
the
battery
from
damage
caused
by
large
charging
currents
during
regenerative
braking,
the
damage caused by large charging currents during regenerative braking, the charging current should be
charging
should be
limited in
circumstances
of braking
under
maximal
feedback
power
and
limited incurrent
circumstances
of braking
under
maximal feedback
power
and
maximal
feedback
efficiency.
maximal
feedback
efficiency.
Meanwhile,
recovering
as
much
braking
energy
as
possible
in
order
to
Meanwhile, recovering as much braking energy as possible in order to improve the energy utilization
improve
the
energy
utilization
efficiency
is
required.
In
order
to
explain
regenerative
braking
efficiency is required. In order to explain regenerative braking operation, the operation of PM motor in
operation,
the operation
PM motor inplane
the four
the torque-speed
the four quadrants
of theof
torque-speed
andquadrants
operation of
modes
are shown inplane
Figureand
9. operation
modes
are
shown
in
Figure
9.
When PM motor is operated in the first and third quadrant, the supplied voltage is greater than
When
PMwhich
motorare
is operated
in the firstand
andreverse
third quadrant,
supplied
voltage but
is greater
than
the back
EMF,
forward motoring
motoring the
modes,
respectively,
the current
the
back flow
EMF,differs.
whichWhen
are forward
motoring
reverse
modes,
respectively,
the
direction
the motor
operatesand
in the
secondmotoring
and fourth
quadrant,
the valuebut
of back
current
direction
flow
differs.
When
the
motor
operates
in
the
second
and
fourth
quadrant,
the
value
EMF generated by the motor should be greater than the supplied voltages, which are the forward
of
back EMF
generated
by the
motorofshould
be greater
than the
voltages,
which as
arewell.
the
braking
and reverse
braking
modes
operation,
respectively,
thesupplied
current flow
is reversed
forward braking and reverse braking modes of operation, respectively, the current flow is reversed
as well. The PM motor is initially made to rotate in clockwise direction under propulsion mode,
but when the speed reversal command is obtained, the control goes into the clockwise
regeneration mode.
Energies 2016, 9, 493
8 of 23
The PM motor is initially made to rotate in clockwise direction under propulsion mode, but when the
Energiesreversal
2016, 9, 493
speed
command is obtained, the control goes into the clockwise regeneration mode. 8 of 23
Speed (n)
Negative torquepositive speed
Forward-braking
QuadrantⅡ
QuadrantⅠ
Regenerative braking
Positive torquepositive speed
Forward-accelerating
Motor driving
Torque (M)
Motor driving
Negative torquenegative speed
Reverse-accelerating
Regenerative braking
Quadrant Ⅳ
Quadrant Ⅲ
Positive torqueNegative speed
Reverse-braking
(a)
Speed (n)
Forward
Braking
I
E
I
V
V
E
E>V
Forward
motoring
V>E
Ⅱ
Ⅰ
Ⅲ
Ⅳ
Torque (M)
Reverse
Motoring
I
V
E
Reverse
Braking
I
V
E
|V|>|E|
|E|>|V|
(b)
Figure 9.
9. Operation
in the
four quadrants
quadrants of
of the
the torque-speed
torque-speed plane:
plane: (a)
(a) Four
Four quadrants
quadrants
Figure
Operation of
of PM
PM motor
motor in
the four
of
operation;
and
(b)
Operation
modes.
of operation; and (b) Operation modes.
2.4. DC/DC Buck/Boost Battery-Charging Mode
2.4. DC/DC Buck/Boost Battery-Charging Mode
A three-phase DC/DC converter is formed by the power devices and passive components. By
A three-phase DC/DC converter is formed by the power devices and passive components.
controlling S1–S6, either a buck DC/DC converter or a boost DC/DC converter can be configured to
By controlling S1–S6, either a buck DC/DC converter or a boost DC/DC converter can be configured
meet the different voltage levels of the power source and batteries requirement. The battery pack can
to meet the different voltage levels of the power source and batteries requirement. The battery pack
be charged by high-voltage DC power from DC micro-grid. A DC/DC buck operation is required
can be charged by high-voltage DC power from DC micro-grid. A DC/DC buck operation is required
between the battery and the DC power source.
between the battery and the DC power source.
The topology has a buck (step-down) characteristic, the DC source voltage should be higher
The topology has a buck (step-down) characteristic, the DC source voltage should be higher than
than the Vbatt. There are two basic working states, as illustrated in Figure 10. S5 keeps conductive
the V batt . There are two basic working states, as illustrated in Figure 10. S5 keeps conductive both in
both in working stage 1 and working stage 2. In working stage 1, only S4 conducts. The inductors in
working stage 1 and working stage 2. In working stage 1, only S4 conducts. The inductors in A-phase
A-phase and C-phase can be used in series as a part of DC/DC converter. The DC source discharges
and C-phase can be used in series as a part of DC/DC converter. The DC source discharges the energy
the energy to the batteries. In working stage 2, S4 stops conducting, while D2 continues conducting.
to the batteries. In working stage 2, S4 stops conducting, while D2 continues conducting. The inductors
The inductors in A-phase and C-phase discharge the energy to the batteries.
in A-phase and C-phase discharge the energy to the batteries.
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S1
S1
Vmotor
Vmotor
S3
S3
..A
A
..
.. ..
...
S2
S2
J2
J2
L1
L1
...BB
.
...
S4
S4
.. J5 ..
. J5 .
S1
S1
S5
S5
Cdc Vbatt
Cdc Vbatt
... .
...
.
S6
S6
C
Ls
C
L
J3 s
J3
Cs
Cs
Vmotor
Vmotor
.. ..
.. .
S5
S5
S3
S3
..A
A
..
S2 D2
S2 D2
J2
J2
L1
L1
EMI Filter
EMI Filter
...BB
.
S4
S4
...
.. J5 ..
. J5 .
Cdc Vbatt
C
Cdc Vbatt
Ls
C
L
J3 s
J3
Cs
Cs
... .
...
.
S6
S6
EMI Filter
EMI Filter
Vgrid
Vgrid
Vgrid
Vgrid
(a)
(a)
(b)
(b)
Figure 10.
10. DC/DC
DC/DC buck
Figure
buckconverter
convertercircuit
circuit operating
operating phases:
phases: (a)
(a)S4
S4 state-on;
state-on; and
and (b)
(b) D2
D2 state-on.
state-on.
Figure 10. DC/DC buck converter circuit operating phases: (a) S4 state-on; and (b) D2 state-on.
When the DC source voltage is lower than the battery voltage, a buck circuit is necessary
the
source
voltage
is lower
thanthan
the battery
voltage,
a buck circuit
iscircuit
necessary
between
When
theDC
DC
source
voltage
lower
battery
voltage,
buck
is maintains
necessary
between
external
power
source
and is
battery
pack, asthe
illustrated
in Figurea11.
Switching
S4
external
power
source
and
battery
pack,
as
illustrated
in
Figure
11.
Switching
S4
maintains
turn on,
between
external
and battery
pack,stage
as illustrated
Figuredischarges
11. Switching
S4 maintains
turn on, and
S2 is power
kept atsource
open-state
in working
1. The DCinsource
the energy
to the
and S2
keptS2
atisopen-state
in working
stage
1. The
DC 1.
source
discharges
the energy to
the
inductor
L .
turn
on,isand
kept atstage
open-state
working
The
DC
source
discharges
energy
to the
inductor
L1. In
working
2, S2 isin
turned
off, stage
and D1
is
turned
on. The
inductor the
L1 discharges
the1
In working
2, S2 isstage
turned
off,
and
D1 isoff,
turned
on.
The
inductor
L1 discharges
the
energy to the
the
inductor
L
1.stage
In
working
2,
S2
is
turned
and
D1
is
turned
on.
The
inductor
L
1 discharges
energy to the battery pack.
battery to
pack.
energy
the battery pack.
S1 D1
S1 D1
Vmotor
Vmotor
.. ..
.. .
S3
S3
..A
A
..
S2
S2
J2
J2
L1
L1
...BB
.
...
S4
S4
.. J5 ..
. J5 .
EMI Filter
EMI Filter
Vgrid
Vgrid
(a)
(a)
S5
S5
S1 D1
S1 D1
....CC . LL
..J3J3.
Cdc Vbatt
Cdc Vbatt
s
s
S6
S6
Cs
Cs
Vmotor
Vmotor
.. ..
.. .
..A
A
..
S2
S2
S3
S3
...BB
.
S4
S4
J2
J2
L1
L1
... .. J5 ...
. J5
S5
S5
Cdc Vbatt
Cdc Vbatt
C
Ls
C
L
J3 s
J3
Cs
Cs
... .
...
.
S6
S6
EMI Filter
EMI Filter
Vgrid
Vgrid
(b)
(b)
Figure 11. DC/DC boost converter circuit operating phases: (a) S2 state-on; and (b) D1 state-on.
Figure 11.
11. DC/DC
Figure
DC/DCboost
boostconverter
convertercircuit
circuitoperating
operatingphases:
phases: (a)
(a)S2
S2state-on;
state-on;and
and (b)
(b) D1
D1 state-on.
state-on.
2.5. Vehicle-to-Grid (V2G) Mode
2.5. Vehicle-to-Grid (V2G) Mode
2.5. Vehicle-to-Grid (V2G) Mode
Bidirectional converter not only charges EVs’ battery pack but also operates EV in V2G mode.
Bidirectional
converter
not only
charges
EVs’
battery the
pack
but also
operates
EV in V2G
The integrated
converter
operates
as an
inverter
to control
battery
feeding
real power
to themode.
grid.
Bidirectional
converter
not only
charges
EVs’
battery pack
but also
operates
EV in V2G
mode.
The
integrated
converter
operates
as
an
inverter
to
control
the
battery
feeding
real
power
to
the
grid.
V2G
is a way ofconverter
connecting
the EVs
powertogrid
properly
and utilizing
power
to meet
The integrated
operates
asto
anthe
inverter
control
the battery
feeding battery
real power
to the
grid.
V2G
is arequirement.
way of connecting
the EVs to
power grid
properly
and utilizing
battery
power
meet
the
It can improve
thethe
efficiency
of power
utilization
by filling
the valley
of to
electric
V2Ggrid
is a way of connecting
the EVs to
the
power grid
properly
and utilizing
battery
power
to
meet
the
grid requirement.
It can
improve
the efficiency
of power
by filling
valley
of electric
power
at night
[19].
In addition,
V2G function
can utilization
supply power
to thethe
grid,
it imposes
to
the griddemand
requirement.
It can
improve
the efficiency
of power
utilization
by filling
the
valley
of electric
power
demand
at night
[19]. In addition,
V2G function can supply power to the grid, it imposes to
follow
standard
1547-2003
power IEEE
demand
at night
[19]. In [20].
addition, V2G function can supply power to the grid, it imposes to
follow
IEEEon
standard
1547-2003
[20]. analysis, the integrated charger with AC/DC battery-charging
Based
the vector
relationship
follow
IEEE standard
1547-2003
[20].
Based
on
the
vector
relationship
analysis, the
integrated
charger with
AC/DC
battery-charging
modeBased
and V2G
mode
is relationship
able to be operated
four-quadrants,
shown
in Figure
12. In V2G
on the
vector
analysis,on
theallintegrated
chargeras
with
AC/DC
battery-charging
mode
and
V2G
mode
is
able
to
be
operated
on
all
four-quadrants,
as
shown
in
Figure
12. In V2G
mode,
particularly,
the
converter
provides
power back
to thein
grid.
mode and
V2G mode
is bidirectional
able to be operated
on all
four-quadrants,
as shown
Figure 12. In V2G mode,
mode, particularly, the bidirectional converter provides power back to the grid.
particularly, the bidirectional converter provides power back to the grid.
Energies 2016, 9, 493
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9, 493
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10 of 23
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23
10
Positive reactive
reactive power
power
Positive
Q
Q
G2V mode
mode
G2V
Inductive
Inductive
Battery charge
charge
Battery
Negative cos
cosψ
ψ
Negative
QuadrantⅡ
QuadrantⅡ
QuadrantⅠ
QuadrantⅠ
SS
Q
Q
ψ
ψ
Negative effective
effective power
power
Negative
G2V mode
mode
G2V
Capacitive
Capacitive
Battery charge
charge
Battery
Negative
cosψ
ψ
Negative cos
Quadrant Ⅲ
Ⅲ
Quadrant
V2G mode
mode
V2G
Inductive
Inductive
Battery discharge
discharge
Battery
Positive cos
cosψ
ψ
Positive
P
P
P
P
Positive effective
effective power
power
Positive
V2G mode
mode
V2G
Capacitive
Capacitive
Battery discharge
discharge
Battery
Positive
cosψ
ψ
Positive cos
Quadrant Ⅳ
Ⅳ
Quadrant
Negative reactive
reactive power
power
Negative
Figure 12.
12. Power
Power control
control in
in all
all four-quadrants.
four-quadrants.
Figure
3. Control
Control
Scheme
3.
ControlScheme
Scheme
Drive Control
Control Strategy
Strategy
3.1. Drive
The control
controlstrategy
strategyforfor
for
motor
drive
is shown
shown
in Figure
Figure
13;
the conventional
conventional
proportional
The
motor
drive
is shown
in Figure
13; the13;
conventional
proportional
integral
control
strategy
motor
drive
is
in
the
proportional
integral
(PI)
controllers
including
DC-link
voltage
out
control
loop
and
ac
side
current
inner
control
(PI)
controllers
including
DC-link DC-link
voltage out
control
andloop
ac side
control
are
integral
(PI) controllers
including
voltage
outloop
control
andcurrent
ac sideinner
current
innerloop
control
loop are
arein
applied
in the
the three-phase
three-phase
inverter circuit.
circuit.
applied
the three-phase
inverter circuit.
loop
applied
in
inverter
dc
iidc
Three-phase
Three-phase
Inverter
iiin Inverter
vvaa
in
vvbb
Vbatt
batt
V
Cdc
dc
C
Vdc**
V
dc
iiaa
vvcc
iibb
iicc
Vdc
dc
V
PI
PI
PM
PM
Motor
Motor
eeaa
eebb
eecc
SVPWM
SVPWM
vvαα
vvββ
αβ
αβ
dq
dq
*
iiqq*
PI
PI
V
q_decompling
Vq_decompling
-ωLi
+e
d
q
V
-ωLid+eq Vd_decompling
d_decompling
PI
PI
ωLiqq+e
+edd
ωLi
iiqq
iidd
dq
dq
abc
abc
*
iidd*
Figure 13.
13. Control
Control
strategy for
for three-phase
three-phase inverter.
inverter.
Control strategy
Figure
3.2. AC
AC Power
Power Charging
Charging Control
Control Strategy
Strategy
3.2.
Line-frequency fluctuations
fluctuations
appear
in
both active
active power
power and
and reactive
reactive power
power when
when the
the DC
DC
Line-frequency
fluctuations appear
appear in
in both
Line-frequency
components in
in the
the AC
AC voltage
voltage and
and AC
AC current
current are
are considered.
considered. The
The line-frequency
line-frequency active
active power
power
components
components
AC
current
are
considered.
fluctuations will
will cause
cause voltage
voltage ripple
ripple in
in the
the DC-link,
DC-link, which
which generate
generate aa second-order
second-order harmonic
harmonic in
in the
the
fluctuations
AC
current.
The
impact
of
DC
components
on
battery-charging
system
is
illustrated
in
Figure
14.
AC current. The
The impact
impact of
ofDC
DCcomponents
componentson
onbattery-charging
battery-chargingsystem
systemisisillustrated
illustratedininFigure
Figure14.14.
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DC component
DC
component
in AC
current
in AC current
Line-frequency
Line-frequency
fluctuation in
fluctuation
AC activein
AC
active
power
power
DC component
DC
component
in AC
voltage
in AC voltage
2nd order
2nd orderin
Harmonic
Harmonic
in
AC current
AC current
Line-frequency
Line-frequency
fluctuation in
fluctuation
DC currentin
DC current
Line-frequency
Line-frequency
fluctuation in
fluctuation
DC powerin
DC power
AC side
AC side
Line-frequency
Line-frequency
fluctuation in
fluctuation
DC voltagein
DC voltage
DC side
DC side
Figure 14.
14. Influences
of DC
DC component
component on
on battery
battery charging
charging system.
Figure
Influences of
of
Figure
14. Influences
DC component
on battery
charging system.
system.
The proposed control scheme based on AC/DC battery-charging mode is shown in Figure 2.
The proposed control scheme based on AC/DC
AC/DC battery-charging
battery-chargingmode
modeisis shown
shown in
in Figure
Figure 2.
2.
The A-phase and B-phase are connected to the AC grid. The C-phase accomplishes the pulsation
The A-phase
A-phaseand
andB-phase
B-phase
are
connected
to
the
AC
grid.
The
C-phase
accomplishes
the
pulsation
are connected to the AC grid. The C-phase accomplishes the pulsation power
power compensation by calculating the 2ω0 pulsation power between the DC-link and the ripple energy
power
compensation
by calculating
the0 2ω
0 pulsation
power
between
DC-linkand
andthe
theripple
ripple energy
compensation
by calculating
the 2ω
pulsation
power
between
thethe
DC-link
storage capacitor Cs. The control block diagram of AC battery-charging mode is given in Figure 15.
storage
block
diagram
of of
ACAC
battery-charging
mode
is given
in Figure
15. 15.
storage capacitor
capacitorCCs.sThe
. Thecontrol
control
block
diagram
battery-charging
mode
is given
in Figure
Vs
Vs
PLL
PLL
L1
L1
ωot
ωot
S1
S1
Vs
Vs
..
S3
S3
D1
D1
..A
D3
D3
S5
S5
..
D2
D2
S4
S4
D5
D5
C
..C
B
B
A
S2
S2
Pulse power
Pulse power
reduction
circuit
reduction circuit
D4
D4
S6
S6
D6
D6
Cdc Vbatt
Cdc Vbatt
Ls
Ls
Cs
Cs
..
Gate drive
Gate
S3 S4drive
S3 S4
S1 S2
S1 S2
not
not
PWM single phase rectifier control
PWM single phase rectifier control
sin
ωot
sin
ωot
idc
idc
not
not
not
not
Vdc
Vdc
-1
-1
Vs
Vs
is
is
S5 S6
S5 S6
Kp4
Kp4
Kp1
Kp1
ics
ics
is*
is*
Decoupling
Decoupling
ics*
ics*
Kp3
Kp3
2ωot
2ωot
PI
PI
Vdc*
Vdc*
Pulse power reduction circuit control
Pulse power reduction circuit control
idc
idc
Kp2
Kp2
Vdc
Vdc
LPF
LPF
sin
sin
Vdc
Vdc
Figure 15. Control block diagram of integrated converter.
Figure
Figure 15.
15. Control
Control block
block diagram
diagram of
of integrated
integrated converter.
converter.
The control block diagram of the single-phase rectifier is shown on the left side of Figure 15. The
The control block diagram of the single-phase rectifier is shown on the left side of Figure 15. The
control is
block
of the
the DC-link
single-phase
rectifier
is shown
on thefactor
left side
of Figure
15.
PWMThe
controller
useddiagram
to control
voltage
through
unity power
operation
at the
PWM controller is used to control the DC-link voltage through unity power factor operation at the
The
PWM
controller
is
used
to
control
the
DC-link
voltage
through
unity
power
factor
operation
AC grid side. A proportional-integral (PI) closed-loop control regulates the DC-link voltage Vdc at
AC grid side. A proportional-integral (PI) closed-loop control regulates the DC-link voltage Vdc at
at
AC grid
side.
proportional-integral
(PI)
closed-loop
control regulates
the
voltage
thethe
required
level.
AA
feedforward
command of
load
power consumption
is added
onDC-link
the closed-loop
the required level. A feedforward command of load power consumption is added on the closed-loop
V
required
level.toAimprove
feedforward
of loadThe
power
consumption
on the
control
output
in order
the Vcommand
dc performance.
phase
informationisisadded
obtained
by
dc at the
control output in order to improve the Vdc performance. The phase information is obtained by
phase-locked-loop (PLL). The compensation current produced by auxiliary circuit is sensed and
phase-locked-loop (PLL). The compensation current produced by auxiliary circuit is sensed and
filtered by a low-pass filter (LPF) to attenuate the switching noise. The generating current command
filtered by a low-pass filter (LPF) to attenuate the switching noise. The generating current command
Energies 2016, 9, 493
12 of 23
12 of 23
output in order to improve the V dc performance. The phase information
is
obtained by phase-locked-loop (PLL). The compensation current produced by auxiliary circuit is
amplitude
then multiplied
by a filter
normalized
wave, which
is acquired
from
to the AC
grid
sensed andisfiltered
by a low-pass
(LPF) tosin
attenuate
the switching
noise.
ThePLL
generating
current
voltage.
The
final
current
command
i
s* ensures the unity power factor operation of the single-phase
command amplitude is then multiplied by a normalized sin wave, which is acquired from PLL to
PWM
The C-phase
designed
for circulating
thethe
reactive
powerfactor
by producing
the ACrectifier.
grid voltage.
The finaliscurrent
command
is * ensures
unity power
operation aofDC
the
voltage
and
a
2ω
0 sinusoidal voltage. The resulting 2ω0 current component and the DC voltage can
single-phase PWM rectifier. The C-phase is designed for circulating the reactive power by producing a
compensate
the 2ω
0 pulsating power. The error of the output current is amplified by a proportional
DC voltage and
a 2ω
0 sinusoidal voltage. The resulting 2ω0 current component and the DC voltage
gain
Kp
1 and the line voltage is added to produce the final voltage reference for the PWM. The
can compensate the 2ω0 pulsating power. The error of the output current is amplified by a proportional
control
is shown on the right side of the Figure 15. The inductor in auxiliary circuit is required
gain Kpblock
1 and the line voltage is added to produce the final voltage reference for the PWM. The control
to
operate
in continuous
conduction
(CCM).
Ininductor
order to in
operate
in CCM
it is essential
to
block
is shown
on the right
side of themode
Figure
15. The
auxiliary
circuitmode,
is required
to operate
determine
the
minimum
inductance
value
ensures
CCM
operation
[21].
The
compensate
current
in continuous conduction mode (CCM). In order to operate in CCM mode, it is essential to determine
produced
by pulse
powervalue
reduction
cs can be constructed by creating a sine wave of
the minimum
inductance
ensurescircuit
CCM ioperation
[21]. The compensate current produced by
frequency
2ω0reduction
with C-phase
shift
andbe
theconstructed
amplitude by
ics. creating
Simple proportional
Kp4 amplifies
the
pulse power
circuit
ics can
a sine wave gain
of frequency
2ω0 with
current
error
and
then
bias
voltage
V
batt is added to produce the final voltage reference of the
C-phase shift and the amplitude ics . Simple proportional gain Kp4 amplifies the current error and then
C-phase
PWM
bias voltage
V waveform.
is added to produce the final voltage reference of the C-phase PWM waveform.
Energies
2016, 9, control
493
closed-loop
batt
4.
4. Simulation
SimulationResults
Results
Space
vector pulse
pulsewidth
widthmodulation
modulation
(SVPWM)
is acquired
to conduct
the permanent
Space vector
(SVPWM)
is acquired
to conduct
on theon
permanent
magnet
magnet
synchronous
motorcontrol.
drive control.
The drive
motormode
drive performance
mode performance
is shown
in Figure
synchronous
motor drive
The motor
is shown
in Figure
16. 16.
400
300
Line Voltage (V)
200
100
0
-100
-200
-300
-400
0
0.01
0.02
0.03
0.04
0.05
Time (s)
0.06
0.07
0.08
0.09
0.06
0.07
0.08
0.09
0.1
(a)
50
40
Inverter Current (A)
30
20
10
0
-10
-20
-30
-40
-50
0
0.01
0.02
0.03
0.04
0.05
Time (s)
0.1
(b)
Figure
16. Simulation
Simulationresult:
result:(a)(a)
Three-phase
inverter
output
line voltage;
(b) Three-phase
Figure 16.
Three-phase
inverter
output
line voltage;
and (b)and
Three-phase
inverter
inverter
output current.
output current.
To evaluate the performance of the proposed integrated charger, AC battery-charging mode
simulation has been performed. The input inductance is selected as 3 mH. To leave some margin, the
root-mean-square (RMS) ripple current is 2.65 A, which is 10% of the peak output current (23.57 A).
In order to show the advantage of the proposed auxiliary circuit method, simulation with ripple
compensation circuit is carried out. The simulation results are shown in Figure 17.
Energies 2016, 9, 493
13 of 23
To evaluate the performance of the proposed integrated charger, AC battery-charging mode
simulation has been performed. The input inductance is selected as 3 mH. To leave some margin, the
root-mean-square (RMS) ripple current is 2.65 A, which is 10% of the peak output current (23.57 A).
In order to show the advantage of the proposed auxiliary circuit method, simulation with ripple
compensation
circuit is carried out. The simulation results are shown in Figure 17.
Energies
2016, 9, 493
13 of 23
(a)
(b)
(c)
(d)
(e)
(f)
(g)
Figure
17. Simulation
Simulationresult:
result:(a)
(a)AC
AC
input
voltage
current;
(b) input
AC input
current
track (c)
error;
Figure 17.
input
voltage
andand
current;
(b) AC
current
track error;
The
(c)
The
system
total
harmonic
distortion
(THD);
(d)
Root
locus;
(e)
System
Bode
plot;
(f)
Maximum
system total harmonic distortion (THD); (d) Root locus; (e) System Bode plot; (f) Maximum AC input
AC
input
current
ripple;
and (g)output
DC-link
output voltage.
current
ripple;
and
(g) DC-link
voltage.
Simulations are carried out for AC battery-charging mode of the proposed integrated charger
Simulations are carried out for AC battery-charging mode of the proposed integrated charger
with the parameters shown in Table 2. As mentioned in the previous section, the auxiliary capacitor
with the parameters shown in Table 2. As mentioned in the previous section, the auxiliary capacitor
Cs is selected as 298 µ F. The auxiliary inductor Ls is selected as 50 µ H. The DC bus capacitor is
Cs is selected as 298 µF. The auxiliary inductor Ls is selected as 50 µH. The DC bus capacitor is
selected
as 50 µ F. By using MATLAB/Simulink as a simulation tool, the ripple energy from the ac
selected as 50 µF. By using MATLAB/Simulink as a simulation tool, the ripple energy from the ac
side of the single-phase rectifier is verified to store in the auxiliary capacitor Cs effectively. Using the
side of the single-phase rectifier is verified to store in the auxiliary capacitor Cs effectively. Using the
conventional method, a 4.665 mF capacitance is selected to meet the DC bus voltage
ripple within 2%
conventional method, a 4.665 mF capacitance is selected to meet the DC bus voltage ripple within 2%
limit requirement. Figure 17a,b shows the AC input voltage and current, respectively. It should be
limit requirement. Figure 17a,b shows the AC input voltage and current, respectively. It should be
noted the input current can track the reference value accurately. The total harmonic distortion is
noted the input current can track the reference value accurately. The total harmonic distortion is 1.92%,
1.92%, as shown in Figure 17c. The root locus of rectifier is given in Figure 17d. The root locus
as shown in Figure 17c. The root locus of rectifier is given in Figure 17d. The root locus indicates
indicates the stability of system. The rectifier Bode plot is shown in Figure 17e. The large crossover
the stability of system. The rectifier Bode plot is shown in Figure 17e. The large crossover frequency
frequency indicates the high frequency harmonic can be filtered at switching frequency. The maximum
indicates the high frequency harmonic can be filtered at switching frequency. The maximum AC input
AC input current ripple is given in Figure 17f. It shows the maximum ripple is in accordance with
current ripple is given in Figure 17f. It shows the maximum ripple is in accordance with unipolar
unipolar modulation theoretical value. The DC-link output voltage is showed in Figure 17g.
modulation theoretical value. The DC-link output voltage is showed in Figure 17g.
Energies 2016, 9, 493
14 of 23
Table 2. Simulation Parameters.
Energies 2016, 9, 493
Symbol
Vs
V batt
L1
fs
Cs
Ls
Cdc
Table 2. Simulation
Parameters.
Parameter
Symbol
Vs
Vbatt
L1
fs
Cs
Ls
Cdc
14 of 23
Value
Input Voltage
Parameter
Value 230 V/50 Hz
Battery
Voltage
Input
Voltage
230 V/50 Hz 320 V
Input Inductance
3 mH
Battery
Voltage
320 V
Switching
Frequency
20
kHz
Input Inductance
3 mH
Auxiliary
Capacitance
298
µF
Switching Frequency
20 kHz
Auxiliary
Inductance
50
µH
Auxiliary Capacitance
298 µ F
DC bus Inductance
Capacitance
50 µF
Auxiliary
50 µ H
DC bus Capacitance
50 µ F
The simulation waveforms of AC battery-charging mode with active ripple energy storage circuit
The simulation
of AC
battery-charging
with activewith
ripple
energy
storage
are presented
in Figurewaveforms
18. Charging
power
is selected tomode
be compatible
SAE
standard
J1772
circuit
are
presented
in
Figure
18.
Charging
power
is
selected
to
be
compatible
with
SAE
standard
level 2 charging rate [22]. The method based on one digital signal control leads the auxiliary inductor
J1772
2 charging
rate [22].
TheIn
method
based
one digital
signaland
control
leads the auxiliary
work
in level
continuous
current
mode.
addition,
it on
provides
a simple
comparatively
low cost
inductor work in continuous current mode. In addition, it provides a simple and comparatively low
solution to implement integrated charger. The battery voltage waveform, auxiliary capacitor current,
cost solution to implement integrated charger. The battery voltage waveform, auxiliary capacitor
and grid side current are grid side voltage are presented in Figure 18.
current, and grid side current are grid side voltage are presented in Figure 18.
When the third leg is engaged, the 2ω0 pulsation power is absorbed by the auxiliary circuit.
When the third leg is engaged, the 2ω0 pulsation power is absorbed by the auxiliary circuit. The
The ripple energy is stored in the auxiliary capacitor. The auxiliary inductor works as a ripple energy
ripple energy is stored in the auxiliary capacitor. The auxiliary inductor works as a ripple energy
transfer
component.
scheme successfully
successfullyreduces
reducesthe
thelow
lowfrequency
frequency
ripple
transfer
component.The
The proposed
proposed control
control scheme
ripple
energy
back
to
the
DC
bus.
The
compensate
circuit
draws
the
low
frequency
current,
the
current
energy back to the DC bus. The compensate circuit draws the low frequency current, the current
flowing
into
the
component.Therefore,
Therefore,
a small
bus
flowing
into
themain
maindcdccapacitor
capacitorcontains
contains high
high frequency
frequency component.
a small
DCDC
bus
capacitor
is
still
needed
to
maintain
a
smooth
DC
bus
voltage.
Figure
18b
shows
the
auxiliary
capacitor
capacitor is still needed to maintain a smooth DC bus voltage. Figure 18b shows the auxiliary
current.
Significant
can0 ripple
be observed.
In Figure 18c,d,
the 18c,d,
AC side
current
0 ripple 2ω
capacitor
current. 2ω
Significant
can be observed.
In Figure
thevoltage
AC sideand
voltage
andare
illustrated.
And
the AC side
Is contains
distortion
due to thedue
input
inductor.
current are
illustrated.
And current
the AC side
current some
Is contains
some distortion
to the
input inductor.
(a)
(b)
(c)
Figure 18. Cont.
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15 of 23
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15 of 23
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15 of 23
(d)
Figure
Simulationresult:
result:(a)
(a)DC-link
DC-link voltage;
voltage;(d)
(b)
current;
(c)(c)
ACAC
Figure
18.18.
Simulation
(b) The
The power
powerdecoupling
decouplingcapacitor
capacitor
current;
grid current; and (d) AC grid voltage.
gridFigure
current;
(d) AC result:
grid voltage.
18.and
Simulation
(a) DC-link voltage; (b) The power decoupling capacitor current; (c) AC
grid current; and (d) AC grid voltage.
High
switching frequency can improve the active ripple energy storage circuit performance
High
switching
frequency
can improve
the simulations
active ripple
storage
circuit
effectively. The results
of the integrated
converter
at energy
20 kHz are
shown
in the performance
upper side
High switching frequency can improve the active ripple energy storage circuit performance
effectively.
The
results
of
the
integrated
converter
simulations
at
20
kHz
are
shown
in the
upper
side
of Figure 19, while the results of the integrated charger simulations at 73 kHz shown
in the
lower
effectively. The results of the integrated converter simulations at 20 kHz are shown in the upper side
of half
Figure
of the integrated
simulations
at 73 kHz
shown inthe
theauxiliary
lower half
of 19,
the while
Figurethe
19.results
As observed
in Figure charger
19, by increasing
switching
frequency,
of Figure 19, while the results of the integrated charger simulations at 73 kHz shown in the lower
of the
Figure 19.
As observed
in
Figure
19,
by increasing
switching
frequency,
the auxiliary
inductance
inductance
is
selected
as
30
µ
H
in
the
system
with
the
73
kHz
switching
frequency.
However,
this
half of the Figure 19. As observed in Figure 19, by increasing switching frequency, the auxiliary
is selected
as
30
µH
in
the
system
with
the
73
kHz
switching
frequency.
However,
this
will
lead
will
lead
to
a
high
current
harmonic
in
auxiliary
capacitor.
inductance is selected as 30 µ H in the system with the 73 kHz switching frequency. However, this to a
highwill
current
harmonic
in auxiliary
capacitor.
lead to
a high current
harmonic
in auxiliary capacitor.
220
210
220
200
200
180
190
170
180
160
170
150
160
140
0.045
150
Uc (V)
Uc (V)
210
190
0.05
0.055
Time (s)
0.06
0.065
220
140
0.045
210
220
0.05
Discharging
200
210
Discharging
Charging
0.055
Time (s)
0.06
Discharging
Charging
Discharging
Charging
0.065
Charging
180
190
Uc (V)
Uc (V)
190
200
170
180
160
170
150
160
140
150
0.045
140
0.045
0.05
0.055
Time (s)
0.05
0.055
Time (s)
(a)
(a)
20
0.06
Charging
Charging
1520
1015
0.065
0.06
0.065
Charging
Charging
Ic (A)
Ic (A)
510
05
-5
0
-5
-10
-10
-15
Discharging
Discharging
-15
-20
0.045
-20
200.045
0.05
0.05
0.055
Time (s)
0.055
Time (s)
Discharging
Discharging 0.06
0.065
0.06
0.065
0.06
0.06
0.065
0.065
20
15
Ic (A)
Ic (A)
15
10
10
5
5
0
0
-5
-5
-10
-10
-15
-15
-20
0.045
-20
0.045
0.05
0.05
0.055
0.055
Time
(s)
Time (s)
(b)
(b)
Figure
voltage; and
and(b)
(b)auxiliary
auxiliarycapacitor
capacitorcurrent.
current.
Figure19.
19.Simulation
Simulationresult:
result:(a)
(a) auxiliary
auxiliary capacitor
capacitor voltage;
Figure 19. Simulation result: (a) auxiliary capacitor voltage; and (b) auxiliary capacitor current.
Energies 2016, 9, 493
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350
350
300
300
250
250
200
200
150
150
100
100
50
50
0
0
-50
0.03
-50
0.03
15.5
15.5
Output
Current
Output
Current
(A)(A)
Inductor
Voltage
Inductor
Voltage
(V)(V)
According
According to
to the
the comparison,
comparison, there
there is
is aa trade-off
trade-off between
between one
one PWM
PWM signal
signal control
control strategy
strategy and
and
According to the comparison, there is a trade-off between one PWM signal control strategy and
two
two PWM
PWM signals
signals control
control strategy
strategy for
for auxiliary
auxiliary circuit.
circuit. For
Forthe
theS5
S5and
and S6
S6 control
control separately,
separately, itit can
can
two PWM signals control strategy for auxiliary circuit. For the S5 and S6 control separately, it can
achieve
keeping the
the value
valueof
ofripple
rippleenergy
energytransfer
transferinductance
inductanceatat
minimum.
addition,
achieve DCM, keeping
thethe
minimum.
In In
addition,
its
achieve DCM, keeping the value of ripple energy transfer inductance at the minimum. In addition,
its
control
scheme
is more
complex
than
PWM
signal
control.
control
scheme
is more
complex
than
oneone
PWM
signal
control.
its control scheme is more complex than one PWM signal control.
The
The steady
steady states
states of DC/DC
DC/DCbuck
buckoperation
operationhave
havealso
also been
been simulated.
simulated. The
Theresults
results for
for the
the DC
DC
The steady states of DC/DC buck operation have also been simulated. The results for the DC
external
external power
power source
source voltage
voltage of
of 400
400 V
V and
and the
the integrated
integrated charger
charger is
is considered
considered able
able to
to operate
operate as
as aa
external power source voltage of 400 V and the integrated charger is considered able to operate as a
DC/DC
in Figure
Figure20.
20.The
Theinductor
inductorvoltage
voltageand
andoutput
outputcurrent
current
are
shown
DC/DCbuck
buckconverter,
converter, are shown in
are
shown
in
DC/DC buck converter, are shown in Figure 20. The inductor voltage and output current are shown
in
Figure
20,
respectively.
It
is
seen
that
the
DC/DC
buck
operation
can
work
properly.
Figure 20, respectively. It is seen that the DC/DC buck operation can work properly.
in Figure 20, respectively. It is seen that the DC/DC buck operation can work properly.
0.0305
0.0305
0.031
Time
(s)
0.031
Time (s)
(a)
(a)
0.0315
0.0315
0.032
0.032
15
15
14.5
14.5
14
14
13.5
0.03
13.5
0.03
0.0305
0.0305
0.031
Time
(s)
0.031
Time (s)
0.0315
0.0315
0.032
0.032
(b)
(b)
Figure
Figure 20.
20. DC/DC
DC/DCbuck
buckoperation:
operation:(a)
(a)Inductor
Inductorvoltage;
voltage;and
and(b)
(b)Output
Outputcurrent.
current.
Figure 20. DC/DC buck operation: (a) Inductor voltage; and (b) Output current.
In this paper, two types of power devices are compared, i.e., silicon (Si) IGBT module and
In this
this paper,
paper, two
twotypes
typesofofpower
powerdevices
devices
compared,
silicon
module
areare
compared,
i.e.,i.e.,
silicon
(Si) (Si)
IGBTIGBT
module
and
silicon carbide (SiC) MOSFET module. The state-of-the-art power devices are investigated for the
and
silicon
carbide
MOSFET
module.
The state-of-the-art
are investigated
silicon
carbide
(SiC) (SiC)
MOSFET
module.
The state-of-the-art
power power
devicesdevices
are investigated
for the
integrated charger application. Wide-bandgap SiC material meet the demand for novel power
for
the integrated
application.
Wide-bandgap
SiC material
the demand
for power
novel
integrated
charger charger
application.
Wide-bandgap
SiC material
meet themeet
demand
for novel
semiconductor devices capable of operating at higher voltage, higher frequency and higher
power
semiconductor
capable
of operating
at higher
voltage,
higherfrequency
frequencyand
and higher
semiconductor
devicesdevices
capable
of operating
at higher
voltage,
higher
temperature. The comparison between Si and SiC material properties is shown in Figure 21. This
temperature. The
Thecomparison
comparison
between
Si and
material
properties
is shown
in Figure
21.
between
Si and
SiCSiC
material
properties
is shown
in Figure
21. This
section will develop the analytical averaged loss model for the power devices shown in Figure 2. The
This
section
will
develop
the
analytical
averaged
loss
model
for
the
power
devices
shown
in
Figure
2.
section will develop the analytical averaged loss model for the power devices shown in Figure 2. The
power loss and the efficiency variation with switching frequency of Si IGBT module and SiC MOSTFE
The
power
lossthe
andefficiency
the efficiency
variation
with switching
frequency
of module
Si IGBT and
module
and SiC
power
loss and
variation
with switching
frequency
of Si IGBT
SiC MOSTFE
module will also be investigated. Device characteristics are extracted from module datasheet.
MOSTFE
module
will
also be investigated.
Device characteristics
arefrom
extracted
from
module datasheet.
module will
also be
investigated.
Device characteristics
are extracted
module
datasheet.
Figure 21. Comparison between Si and SiC material properties.
Figure
Figure 21.
21. Comparison
Comparison between
between Si
Si and
and SiC
SiC material
material properties.
properties.
The switching device power loss can be characterized by the forward voltage drop during
The switching device power loss can be characterized by the forward voltage drop during
conduction
and the turn-on
turn-off
energy
during switching.
For forward
simplification,
thedrop
conduction
The switching
device and
power
loss can
be characterized
by the
voltage
during
conduction and the turn-on and turn-off energy during switching. For simplification, the conduction
characteristic
of the
considered
to be linear,
which
meansFor
on-state
resistancethe
is assumed
as
conduction and
the devices
turn-on is
and
turn-off energy
during
switching.
simplification,
conduction
characteristic of the devices is considered to be linear, which means on-state resistance is assumed as
acharacteristic
fixed value. of
Due
to
the
bidirectional
current
conductivity
of
the
MOSFET
channel,
an
assumption
the devices is considered to be linear, which means on-state resistance is assumed as
a fixed value. Due to the bidirectional current conductivity of the MOSFET channel, an assumption
is made to simplify the calculation that MOSFET conduct all the load current throughout the period.
is made to simplify the calculation that MOSFET conduct all the load current throughout the period.
In fact, the Schottky diode and the body diode share a part of the load current as well when the load
In fact, the Schottky diode and the body diode share a part of the load current as well when the load
Energies 2016, 9, 493
17 of 23
a fixed value. Due to the bidirectional current conductivity of the MOSFET channel, an assumption
isEnergies
made2016,
to simplify
the calculation that MOSFET conduct all the load current throughout the period.
9, 493
17 of 23
In fact, the Schottky diode and the body diode share a part of the load current as well when the
load
current
is negative.
on above
assumption,
the conduction
lossoccur
only occur
in MOSFET.
current
is negative.
BasedBased
on above
assumption,
the conduction
loss only
in MOSFET.
For Si
For
Si
IGBT
and
SiC
MOSFET,
the
switching
loss
includes
turn-on
and
turn-off
losses.
For
diode,
IGBT and SiC MOSFET, the switching loss includes turn-on and turn-off losses. For diode, the
the
switching
only
includes
turn-offloss
lossdissipated
dissipatedininthe
the reverse
reverse recovery process.
switching
lossloss
only
includes
turn-off
process. MOSFET
MOSFET isis
assumed
load
current
during
thethe
negative
current
interval,
but the
loss still
assumedtotoconduct
conductallallthe
the
load
current
during
negative
current
interval,
butswitching
the switching
loss
occur
in theinSchottky
diode.diode.
Due toDue
the to
existence
of dead-time,
load current
flow toflow
the Schottky
diode,
still occur
the Schottky
the existence
of dead-time,
load current
to the Schottky
and
then
flow
to the
MOSFET.
For SiC MOSFET
manufacturers,
it is claimed
reverse
recovery
diode,
and
then
flow
to the MOSFET.
For SiC MOSFET
manufacturers,
it iszero
claimed
zero
reverse
current,
which
means
no reverse
recovery
loss
for SiC loss
module.
Themodule.
total loss
is the
recovery
current,
which
means no
reverse
recovery
for SiC
The
totalsum
lossofisconduction
the sum of
loss
and switching
loss
in all power
The devices
comparison
arecomparison
Si IGBT module
conduction
loss and
switching
loss devices.
in all power
devices.used
The for
devices
used for
are Si
from
(1200 V, 300 A, Semikron,
Nuremberg,
Germany)
and SiCGermany)
MOSFET module
IGBTSKM300GA12V
module from SKM300GA12V
(1200 V, 300
A, Semikron,
Nuremberg,
and SiC
from
CAS300M12BM2
V, 300 A, Cree,
Durham,
NC,
USA).
Figure NC,
22 shows
loss 22
distribution
MOSFET
module from(1200
CAS300M12BM2
(1200
V, 300 A,
Cree,
Durham,
USA).the
Figure
shows the
among
the devicesamong
for rectifier
and inverter
operation
a modulation
0.9 and unityindex
power
loss distribution
the devices
for rectifier
andwith
inverter
operationindex
with of
a modulation
of
factor.
The
fullpower
list of Si-based
0.9 and
unity
factor. IGBT losses and SiC-based MOSFET losses are given in the appendix.
Figure
Figure 22a,b
22a,billustrates
illustrates the
thepower
powerloss
lossdistribution
distributionin
inintegrated
integratedconverter
converterapplying
applyinginverter
inverter
operation
operationand
andrectifier
rectifieroperation,
operation,respectively.
respectively.As
Asshown
shownin
inFigure
Figure22a,
22a,the
theswitching
switchingloss
lossdominate
dominate
60%
60%total
totalpower
powerloss
lossoccur
occurin
inSi
SiIGBT.
IGBT.As
Asshown
shownin
inFigure
Figure22c,d,
22c,d,ititisisassumed
assumedthat
thatall
allpower
powerloss
loss
occur
MOSFET.
It illustrates
that SiC
have a good
in reducing
occuronly
onlyin in
MOSFET.
It illustrates
thatMOSFET
SiC MOSFET
have performance
a good performance
inswitching
reducing
loss.
The efficiency
Si-based of
module
drop
fast compared
SiC-based
with
the switching
switching
loss. Theofefficiency
Si-based
module
drop fast to
compared
tomodule
SiC-based
module
with the
frequency
Figure
22e,f shows
the22e,f
SiC MOSFET
excellent
performance
and attractive
potential
switchingvariation.
frequency
variation.
Figure
shows the
SiC MOSFET
excellent
performance
and
attractive
potential of
in the
of high switching
frequency.
At frequency
above
20 kHz,loss
the
in
the applications
highapplications
switching frequency.
At frequency
above
20 kHz, the
switching
switching to
loss
contributeamount
to a significant
of power
dissipation
[23].
The
results
both Si
contribute
a significant
of poweramount
dissipation
[23]. The
results for
both
Si and
SiCfor
converter
and SiCbreakdown
converter are
volume
breakdown
are shown
in Figure
22g,h.
should
be noted
that the
volume
shown
in Figure 22g,h.
It should
be noted
thatItthe
switching
frequency
is
switching
frequency
is 73
kHz
for SiC
converter
and
kHz
for Sithe
converter.
can be
the
73
kHz for SiC
converter
and
20 kHz
for Si
converter.
As20
can
be seen,
inductorAs
volume
is seen,
reduced
inductor
volume31%.
is reduced
by approximately
As expected,
the with
volume
of thefrequency
converter
by
approximately
As expected,
the volume of 31%.
the converter
decreases
switching
decreasesdue
with
frequency variation
variation
to switching
the good performance
of SiC. due to the good performance of SiC.
(a)
(b)
(c)
(d)
Figure 22. Cont.
Energies 2016, 9, 493
Energies 2016, 9, 493
18 of 23
18 of 23
(e)
(f)
(g)
(h)
Figure 22.
22. Calculated
Calculatedresults:
results:(a)
(a)SiSiIGBT
IGBTand
anddiode
diodeloss
loss
inverter;
Si IGBT
diode
in
Figure
inin
thethe
inverter;
(b)(b)
Si IGBT
andand
diode
lossloss
in the
the
rectifier;
(c)
SiC
MOSFET
loss
in
the
inverter;
(d)
SiC
MOSFET
loss
in
the
rectifier;
(e)
Inverter
rectifier; (c) SiC MOSFET loss in the inverter; (d) SiC MOSFET loss in the rectifier; (e) Inverter efficiency
efficiency with
variation
with frequency;
switching frequency;
Rectifierwith
variation
withfrequency;
switching (g)
frequency;
variation
switching
(f) Rectifier(f)
variation
switching
Volume
(g)
Volume
breakdown
of
Si
IGBT
converter;
and
(h)
Volume
breakdown
of
SiC
MOSFET
breakdown of Si IGBT converter; and (h) Volume breakdown of SiC MOSFET converter. converter.
The junction temperature describes the temperature inside the chip, which is relevant for the
The junction temperature describes the temperature inside the chip, which is relevant for the loss
loss and safe operating area. The power pulsation may cause extra thermal stress to the single-phase
and safe operating area. The power pulsation may cause extra thermal stress to the single-phase PWM
PWM power device and auxiliary circuit power device. In order to evaluate the thermal
power device and auxiliary circuit power device. In order to evaluate the thermal performance of
performance of the integrated converter, the power device junction temperature variation are
the integrated converter, the power device junction temperature variation are simulated. Figure 23a
simulated. Figure 23a shows H-bridge rectifier device power loss at 73 kHz, which indicates at
shows H-bridge rectifier device power loss at 73 kHz, which indicates at frequency above 20 kHz,
frequency above 20 kHz, the switching loss contributes to a significant amount of power dissipation.
the switching loss contributes to a significant amount of power dissipation. Figure 23b–d shows the
Figure 23b–d shows the H-bridge and ripple energy reduction circuit device junction temperature
H-bridge and ripple energy reduction circuit device junction temperature variation. The Foster thermal
variation. The Foster thermal network is used to calculate the junction temperature variation. The
network is used to calculate the junction temperature variation. The heatsink temperature is assumed
heatsink temperature is assumed to be fixed at 80 °C due to its large thermal time constant [24]. For
to be fixed at 80 ˝ C due to its large thermal time constant [24]. For the H-bridge, the IGBT temperature
the H-bridge, the IGBT temperature varies between 97 and 100 °C. When SiC MOSFET is used in
varies between 97 and 100 ˝ C. When SiC MOSFET is used in single-phase, the MOSFET temperature
single-phase, the MOSFET temperature various between 91.3 and 93.3 °C. When the auxiliary
various between 91.3 and 93.3 ˝ C. When the auxiliary compensation circuit is applied, the MOSFET
compensation circuit is applied, the MOSFET temperature varies between 88.8 and 89.7 °C. The larger
temperature varies between 88.8 and 89.7 ˝ C. The larger junction temperature variation may reduce
junction temperature variation may reduce the lifetime of the power device. The thermal design
the lifetime of the power device. The thermal design should ensure the peak temperature is lower than
should ensure the peak temperature is lower than the maximum allowance junction temperature.
the maximum allowance junction temperature.
High power density is one of the key motivations for the integration of power electronics
High power density is one of the key motivations for the integration of power electronics
components. Integrated converters are designed not only to meet the requirement of input and
components. Integrated converters are designed not only to meet the requirement of input and
output, but also to achieve a light weight and a small size. The auxiliary ripple energy reduction
output, but also to achieve a light weight and a small size. The auxiliary ripple energy reduction circuit
circuit can also achieve a high power density, as discussed above.
can also achieve a high power density, as discussed above.
In order to verify the effectiveness of the proposed system, an integrated multifunctional
In order to verify the effectiveness of the proposed system, an integrated multifunctional
bidirectional AC/DC and DC/DC converter for electric vehicles has been implemented. The
bidirectional AC/DC and DC/DC converter for electric vehicles has been implemented.
propulsion and AC/DC charging performances of the proposed topology and control schemes are
The propulsion and AC/DC charging performances of the proposed topology and control schemes are
verified in this paper.
verified in this paper.
Energies 2016, 9, 493
19 of 23
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19 of 23
Energies 2016, 9, 493
19 of 23
(a)
(b)
(a)
(b)
(c)
(d)
Figure 23. Simulation results: (a) SiC MOSFET loss in the rectifier (73 kHz); (b) Rectifier device
(Si IGBT) junction(c)temperature variation; (c) Rectifier device (SiC MOSFET)
(d) junction temperature
variation; and (d) Auxiliary circuit device (SiC MOSFET) junction temperature variation.
Figure 23. Simulation results: (a) SiC MOSFET loss in the rectifier (73 kHz); (b) Rectifier device
Figure 23. Simulation results: (a) SiC MOSFET loss in the rectifier (73 kHz); (b) Rectifier device (Si IGBT)
(Si IGBT) junction temperature variation; (c) Rectifier device (SiC MOSFET) junction temperature
Figure
24a illustrates
the(c)
system
layout
design.
As mentioned
earlier,
power module,
heatsink,
junction
temperature
variation;
Rectifier
device
(SiC MOSFET)
junction
temperature
variation;
and
variation; and (d) Auxiliary circuit device (SiC MOSFET) junction temperature variation.
inputMOSFET)
inductor,junction
DC capacitor
and variation.
sensors are selected in the integrated
(d)controller,
Auxiliary drive
circuitboard,
device (SiC
temperature
converter. The controller is shown in Figure 24b. The photo of the experimental hardware prototype
Figure 24a illustrates the system layout design. As mentioned earlier, power module, heatsink,
platform is shown in Figure 24c. The power module in the main power circuit is SEMIKRON
controller,
board,the
input
inductor,
DCdesign.
capacitor
sensors earlier,
are selected
in module,
the integrated
Figure 24adrive
illustrates
system
layout
As and
mentioned
power
heatsink,
SKM300GA12V with the maximum junction temperature 175 °C.
converter.
The
controller
shown inDC
Figure
24b. The
of the
hardware
prototype
controller,
drive
board,
inputisinductor,
capacitor
andphoto
sensors
areexperimental
selected in the
integrated
converter.
The motor driving operation photo of experimental prototype platform is shown in Figure 25a.
power
in the main power
circuit
is SEMIKRON
The platform
controllerisisshown
shownininFigure
Figure24c.
24b.The
The
photomodule
of the experimental
hardware
prototype
platform
The drive control method is implemented, as shown in Figure 25b. Figure 25b shows the rated speed
SKM300GA12V
with
the
maximum
junction
temperature
175
°C.
is shown
in Figure
24c.rated
Thetorque
powerismodule
thethe
main
power
is SEMIKRON
is 3000
rpm, the
100 Nmin
and
rated
powercircuit
is controlled
as 30 kW,SKM300GA12V
which are the
The motor driving
operation
photo175
of experimental
prototype platform is shown in Figure 25a.
˝ C.
with the
maximum
junction
temperature
same as the theoretical values.
The drive control method is implemented, as shown in Figure 25b. Figure 25b shows the rated speed
is 3000 rpm, the rated torque is 100 Nm and the rated power is controlled as 30 kW, which are the
Heatsink
Driver board
Inductor
same as the theoretical values.
Heatsink
Driver board
Inductor
(a)
(a)
Sensor
(b)
DC capacitor
(c)
Sensor
DC capacitor
Figure 24. Experiment setup built: (a) System layout; (b) Controller board; and (c) System hardware
(b)
(c)
design structure.
The motor driving operation photo of experimental prototype platform is shown in Figure 25a.
The drive control method is implemented, as shown in Figure 25b. Figure 25b shows the rated speed is
Energies 2016, 9, 493
20 of 23
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20 of 23
Energies 2016, 9, 493
20 of 23
Figure
24. torque
Experiment
setup
built:
System
(b)is
Controller
board;
andkW,
(c) System
3000 rpm, the
rated
is 100
Nm
and(a)the
ratedlayout;
power
controlled
as 30
whichhardware
are the same
design
structure.
as theEnergies
theoretical
values.
2016,24.
9, 493
20 of 23
Figure
Experiment setup built: (a) System layout; (b) Controller board; and (c) System hardware
designPMSM
structure.
Integrated charger
Figure 24. Experiment setup built: (a) System layout; (b) Controller board; and (c) System hardware
Integrated charger
PMSM structure.
design
PMSM
Integrated charger
LabView operation interface
(a)
(b)
LabView operation interface
(a)Figure 25. Motor driving operation experimental setup(b)
and result: (a) Setup; (b) Results.
LabView operation
interface
Figure
25. Motor
driving operation experimental setup and result: (a) Setup; (b) Results.
Figure
result: (a) Setup; (b) Results.
(a) 25. Motor driving operation experimental setup and(b)
Figure 26a,b shows AC/DC battery-charging operation experimental setup and results of the
Figure
25.and
Motor
drivingbattery-charging
operation
experimental
andexperimental
result:
(a) is
Setup;
(b)and
Results.
Figure
26a,b
shows
AC/DC
operation
setup
andload
results
of the
DC-link
voltage
26c,d
showsoperation
thesetup
DC-link
voltage
330
V
current
Figure 26a,b
showscurrent.
AC/DC Figure
battery-charging
experimental
setup
andthe
results of
the is
DC-link
and
current.
26c,d
thecontrol
DC-link
voltage
330
V
load
current
10voltage
A,voltage
respectively.
Thus,Figure
the
proposed
charging
strategy
can
retain
3.3and
kWthe
charging
power.
DC-link
and
current.
Figure
26c,d shows
shows
the
DC-link
voltage
is is
330
V and
the
load
current
is is
Figure 26a,b
shows
AC/DC
battery-charging
operation
experimental
setup
andcharging
results ofpower.
the
10 A,
respectively.
Thus,
the
proposed
charging
control
strategy
can
retain
3.3
kW
10 A, respectively. Thus, the proposed charging control strategy can retain 3.3 kW charging power.
DC-link voltage and current. Figure 26c,d shows the DC-link voltage is 330 V and the load current is
10 A, respectively. Thus, the proposed charging control strategy can retain 3.3 kW charging power.
(a)
(b)
(c)
(d)
(a) 26. AC/DC battery charging(b)
(d)Setup;
Figure
operation experimental setup(c)
and results: (a) Setup; (b)
(c)
Current;
and
(b)
Voltage.
(a)
(b)
(c)
(d)
Figure 26. AC/DC battery charging operation experimental setup and results: (a) Setup; (b) Setup;
Figure 26. AC/DC battery charging operation experimental setup and results: (a) Setup; (b) Setup;
(c)
Current;
and (b) Voltage.
Figure
26.
battery charging operation experimental setup and results: (a) Setup; (b) Setup;
(c) Current;
andAC/DC
Voltage.
Figure
27(b)shows
the rectifier device junction temperature variation. With AC/DC battery
(c) Current; and (b) Voltage.
charging
operation,
the Si
IGBT maximum
junction
temperaturevariation.
varies between
98 and 100
°C. The
Figure 27
shows the
rectifier
device junction
temperature
With AC/DC
battery
larger
junction
temperature
variation
may
reduce
the
lifetime
of
the
power
device.
The
thermal
Figure
27
shows
the
device
junction
temperature
variation.
WithWith
AC/DC
charging
operation,
therectifier
Si
maximum
junction
temperature
varies
between
98 AC/DC
andbattery
100 battery
°C.charging
The
Figure
27 shows
theIGBT
rectifier
device
junction
temperature
variation.
˝
design
should
also
make
sure
themay
peak
temperature
does of
not
allowable
operation,
theoperation,
Si IGBT
maximum
junction
temperature
varies
between
98the
and
100The
The
larger
junction
temperature
variation
reduce
the
lifetime
theexceed
power
device.
thermal
charging
the
Si IGBT
maximum
junction
temperature
varies
between
98maximum
and
100C.°C.
Thelarger
junction
temperature,
°C.
design
should
also
make175
sure
the
peak
temperature
does
notofexceed
the
maximum
allowable
junction
temperature
variation
may
reduce
the
lifetime
oflifetime
the power
device.
The
thermal
should
larger
junction
temperature
variation
may
reduce
the
the
power
device.
Thedesign
thermal
temperature,
175
°C.sure the does
designsure
should
peak not
temperature
does
not exceed
the maximum
alsojunction
make
the also
peakmake
temperature
exceed the
maximum
allowable
junctionallowable
temperature,
˝
temperature, 175 °C.
175 junction
C.
Figure 27. Single-phase PWM rectifier (IGBT/diode) junction temperature variation.
Figure 27. Single-phase PWM rectifier (IGBT/diode) junction temperature variation.
Figure 27. Single-phase PWM rectifier (IGBT/diode) junction temperature variation.
Figure 27. Single-phase PWM rectifier (IGBT/diode) junction temperature variation.
Energies 2016, 9, 493
21 of 23
5. Conclusions
A SiC-based high power density integrated multifunctional bidirectional AC/DC and DC/DC
converter for future application in electric vehicles has been proposed in this paper. The integrated
converter infrastructure and operational principles have been described in detail. Additional details on
converter passive components have been discussed. An improved control scheme has been developed
and utilized to achieve better performance compared to DCM control in AC/DC battery-charging
mode. Through the simulations, the functionalities for the three operating modes, i.e., motor
drive mode AC/DC battery-charging mode and DC/DC battery-charging mode, have been verified.
The simulation results show the feasibility of the design. The auxiliary circuit working as a parallel
ripple current filter method achieves good performance. A simulation and 3.3 kW experimental test
have been conducted for verification purpose. Meanwhile, in order to verify the proposed converter
feasibility, the converter power loss, efficiency variation with switching frequency and thermal stress
have been addressed. These results demonstrate the potential improvements in system performance.
Acknowledgments: This work was supported by the International Cooperation Research Program of China
(No. 2015DFE72810), the Higher School Discipline Innovation Intelligence Plan (No. B12022) and the
Beijing Municipal Science and Technology Commission of China (No. Z121100006612006).
Author Contributions: Liwen Pan conceived the topology and performed the simulations; Chengning Zhang
conducted the experiment and analyzed the data; Liwen Pan wrote the paper.
Conflicts of Interest: The authors declare no conflit of interests.
Appendix A Appendix
Si-based IGBT loss calculation:
´
Pcon.loss “
Diode conduction loss
Pcon.loss “
IGBT (diode) switching loss
Pswitch.loss “
¯
´
¯
mcosp ϕq
mcosp ϕq
1
2
I V
3π ¯ ICM R ` ´ 2π `
8
¯ CM CE0
mcosp ϕq
mcosp ϕq
1
2
´ 3π
ICM R ` 2π ´
ICM VCE0
8
f s Vswitch
2
p 2π Vtest q ˆ p πC
2 ICM ` 2BICM ` πAq
1
´8
1
8
IGBT conduction loss
`
where V CE0 denotes the device initial voltage drop, R represents the equivalent resistance to represent
the linear region of device voltage drop, the modulation index is expressed as m, f s is the switching
frequency, ICM denotes the load peak current and ϕ is the power factor angle.
SiC-based MOSFET loss calculation:
SiC MOSFET conduction loss
Diode conduction loss
SiC MOSFET switching loss
Diode switching loss
I2 R
Pcon.loss “ CM
4
0
f
2
Pswitch.loss “ p 2πs VVswitch
q ˆ p πC
2 ICM ` 2B¨ ICM ` πAq
test
0
where, A, B, and C describe the relationship between the switching energy and current. The parameters
above can be identified from the device characteristics given in the datasheet or through
experimental tests.
References
1.
2.
Zhou, X.; Wang, G.; Lukic, S.; Bhattacharya, S.; Huang, A. Multi-function bi-directional battery charger for
plug-in hybrid electric vehicle application. In Proceedings of the 2009 IEEE Energy Conversion Congress
and Exposition, San Jose, CA, USA, 20–24 September 2009; pp. 3930–3936.
Dusmez, S.; Khaligh, A. Compact and integrated multifunctional power electronic interface for PEVs.
IEEE Trans. Power Electron. 2013, 28, 5690–5701. [CrossRef]
Energies 2016, 9, 493
3.
4.
5.
6.
7.
8.
9.
10.
11.
12.
13.
14.
15.
16.
17.
18.
19.
20.
21.
22.
22 of 23
Dusmez, S.; Khaligh, A. A novel low cost integrated on-board charger topology for electric vehicles and
plug-in hybrid electric vehicles. In Proceedings of the 2012 Twenty-Seventh Annual IEEE Applied Power
Electronics Conference and Exposition (APEC), Orlando, FL, USA, 5–9 February 2012; pp. 2611–2616.
Dusmez, S.; Cook, A.; Khaligh, A. Comprehensive analysis of high quality power converter for level 3
off-board chargers. In Proceedings of the 2011 IEEE Vehicle Power and Propulsion Conference, Chicago, IL,
USA, 6–9 September 2011; pp. 1–10.
Zhou, X.; Lukic, S.; Kuang, A. A comparison of converter topologies for vehicle-to-grid applications:
Three-leg converter versus H-bridge converter. In Proceedings of the Industrial Electronics Annual
Conference of IEEE, Porto, Portugal, 3–5 November 2009; pp. 3711–3716.
Trovao, J.P.; Pereirinha, P.G.; Trovao, L.; Jorge, H.M. Electric vehicles chargers characterization: Load demand
and harmonic distortion. In Proceedings of the Electrical Power Quality and Utilisation (EPQU), Lisbon,
Portugal, 17–19 October 2011.
Zhou, X.; Lukic, S.; Bhattacharya, S.; Kuang, A. Design and control of grid-connected converter in
bi-directional battery charger for plug-in hybrid electric vehicle application. In Proceedings of the 2009 IEEE
Vehicle Power and Propulsion Conference, Dearborn, MI, USA, 7–10 September 2009; pp. 1716–1721.
Dusmez, S.; Chen, C.; Khaligh, A. A reduced-part single stage direct AC/DC on-board charger for automotive
applications. In Proceedings of the 2013 Twenty-Eighth Annual IEEE Applied Power Electronics Conference
and Exposition (APEC), Long Beach, CA, USA, 17–21 March 2013; pp. 1791–1797.
Dusmez, S.; Hasanzadeh, A.; Khaligh, A. Comparative analysis of bidirectional three-level DC-DC converter
for automotive applications. IEEE Trans. Ind. Electron. 2015, 62, 3305–3315. [CrossRef]
Vittorias, I.; Metzger, M.; Kunz, D.; Gerlich, M.; Bachmaier, G. A bidirectional battery charger for electric
vehicles with V2G and V2H capability and active and reactive power control. In Proceedings of the
Transportation Electrication Conference and Expo (ITEC), Dearborn, MI, USA, 15–18 June 2014.
Joice, C.S.; Paranjothi, S.R.; Kumar, V.J.S. Digital control strategy for four quadrant operation of three phase
BLDC motor with load variations. IEEE Trans. Inform. 2013, 9, 974–982. [CrossRef]
Bobba, P.B.; Rajagopal, K.R. Modeling and analysis of hybrid energy storage systems used in electric vehicles.
In Proceedings of the 2012 IEEE International Conference on Power Electronics, Drives and Energy Systems
(PEDES), Bengaluru, India, 16–19 December 2012.
Zhang, X.; Wang, Y.; Liu, G.; Yuan, X. Robust regenerative charging control based on T-S fuzzy sliding-mode
approach for advanced electric vehicle. IEEE Transp. Electr. 2016, 2, 52–65. [CrossRef]
Musavi, F.; Eberle, W.; Dunford, W.G. A high-performance single-phase AC-DC power factor corrected boost
converter for plug in hybrid electric vehicle battery chargers. In Proceedings of the IEEE Energy Conversion
Congress and Exposition, Atlanta, GA, USA, 12–16 September 2010.
Cousland, A.W.; Ciaravolo, R.J.; Blieden, G.; Hosseinzadeh, N. Design of a battery charger and charging
management system for an electric vehicle. In Proceedings of the Universities Power Engineering Conference
(AUPEC), Christchurch, New Zealand, 5–8 December 2010.
Bai, H.; Zhang, Y.; Semanson, C.; Luo, C.; Mi, C.C. Modelling, design and optimization of a battery charger
for plug-in hybrid electric vehicles. IET Electr. Syst Transp. 2011, 1, 3–10. [CrossRef]
Musavi, F.; Edington, M.; Eberle, W.; Dunford, W.G. Energy efficiency in plug-in hybrid electric vehicle
chargers: Evaluation and comparison of front end AC-DC topologies. In Proceedings of the IEEE Energy
Conversion Congress and Exposition, Phoenix, AZ, USA, 17–22 September 2011.
Pan, L.; Zhang, C. A high power density integrated charger for electric vehicles with active ripple
compensation. Math. Probl. Eng. 2015. [CrossRef]
Xue, L.; Diaz, D.; Shen, Z.; Luo, F.; Mattavelli, P.; Boroyevich, D. Dual active bridge based battery charger
for plug-in hybrid electric vehicle with charging current constaining low frequency ripple. IEEE Trans.
Power Electron. 2015, 30, 7299–7307. [CrossRef]
Xue, L.; Mattavelli, P.; Boroyevich, D.; Shen, Z.; Burgos, R. Closed-loop control on DC link voltage ripple
of plug-in hybrid electric vehicle charger with sinusoidal charging. In Proceedings of the IEEE Energy
Conversion Congress and Exposition, Denver, CO, USA, 15–19 September 2013.
Kim, S.; Kang, F.S. Multifunctional onboard battery charger for plug-in electric vehicles. IEEE Trans.
Ind. Electron. 2015, 62, 3460–3472.
Awami, A.T.A.; Sortomme, E.; Akhtar, G.M.A.; Faddel, S. A voltage-based controller for an electric vehicle
charger. IEEE Trans. Veh. Technol. 2015, 6, 4185–4196. [CrossRef]
Energies 2016, 9, 493
23.
24.
23 of 23
Yuan, X.; Lovett, A. DC-link capacitance reduction in a high power medium voltage modular wind power
converter. In Proceedings of the 2013 15th European Conference on Power Electronics and Applications
(EPE), Lille, France, 2–6 September 2013; pp. 1–10.
Boroyevich, D.; Chen, A.; Luo, F.; Ngo, K.; Ning, P.; Wang, R.; Zhang, D. High-density system integration for
medium power application. In Proceedings of the 2010 6th International Conference on Integrated Power
Electronics Systems (CIPS), Nuremberg, Germany, 16–18 March 2010.
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