High Efficiency Multilevel
Flying-Capacitor DC/DC Converter
for Distributed Generation Applications
MingGuo Jin
Amir Parastar
Jul-Ki Seok
Student Member, IEEE
Student Member, IEEE
Senior Member, IEEE
Power Conversion Lab.
Yeungnam University
Gyeongsan, Korea
doljk@ynu.ac.kr http://yupcl.yu.ac.kr
Abstract – This paper presents a high efficiency multilevel
flying-capacitor (FC) DC/DC power conversion with the
reduced component rating and device count for distributed
power generation applications. A resonant technique is adopted
for the proposed FC DC/DC converter to achieve the zerocurrent-switching for all the switches. Theoretical analysis is
carried out for the quintuple FC configuration. The proposed
converter is evaluated to the conventional FC converter in terms
of the component rating and count to highlight its advantages
with respect to the power density, power loss, and cost. The
experimental results of a 1 kW prototype FC converter are
presented to validate the theoretical analysis and principles as
well as attest the feasibility of the proposed topologies.
Index Terms—Distributed power generation, flying-capacitor
DC/DC power conversion, high efficiency, power density, zerocurrent-switching.
I.
INTRODUCTION
Distributed power generation is widely promoted around
the world due to the reduction in greenhouse gas emission
and the fast depletion of natural resources. The distributed
power generation can provide a high-quality and low-cost
electric power when it is fully implemented. The distributed
generation is a concept that covers wide spectra of schemes
used for local electric power generation from renewable and
non-renewable energy sources in an environmentally
responsible way [1]. Basic schemes are mainly based on solar
energy, wind energy, fuel cells, and micro-turbine systems.
A fuel cell is potentially the most efficient modern
approach to distributed power generation because it provides
a continuous power in all seasons. The conversion efficiency,
i.e., the ratio of the electrical output to the heat content of the
fuel, could be as high as 65-70% [2]. The generated DC
voltage is usually relatively low (< 50 V for a 5–10 kW fuel
cell stack) [3-4]. To interconnect a low-dc-voltage-producing
fuel cell to the sufficiently high voltage level of AC
residential loads (typically 230-Vac single-phase or threephase 400-Vac), a step-up interface DC/DC converter is
978-1-4799-5776-7/14/$31.00 ©2014 IEEE
required to comply with the imposed standards and
requirements [5].
The design of the step-up DC/DC power conversion is one
of most important concerns for distributed power generation
systems because this stage is the main contributor of
interface-converter efficiency, weight, and overall
dimensions [1]. The relatively low voltage provided by the
fuel cell is associated with large current flows in the primary
side of the DC/DC converter. These large currents lead to
high conduction and switching losses in the switching devices,
which make system inefficient. Furthermore, the large
voltage gain requirement presents a unique challenge to the
DC/DC converter design. Conventional step-up DC/DC
converters with a high voltage gain result in the large input
current ripple and high device voltage rating [3]. Multilevel
converters have solved the problem of the high voltage on
conventional converters using the lower-voltage-rated
devices operating at lower blocking voltage levels [6-8].
Within the multilevel topologies, there are some
configurations which include an array of power
semiconductors and capacitive storage elements, such as
flying capacitors. These DC/DC converters, called multilevel
switched-capacitor (SC) or flying-capacitor (FC) DC/DC
converters, are considered as an attractive solution for
meeting the requirements, such as high-power density and
control simplicity [9-12]. In this regard, a multilevel SC
DC/DC converter based on voltage multiplier concepts was
reported to reach the high voltage gain in [8]. Nonetheless,
the introduced converter requires a large number of
capacitors and high diode conduction losses. A 55 kW 3×
(the output voltage is three times the input voltage) FC
DC/DC converter was introduced with a lower device voltage
rating and fast dynamic response for the hybrid electric
vehicles [12]. Unfortunately, the presented SC and FC
converters suffer from high current spikes, high switching
loss, and severe electromagnetic interference (EMI) noise. In
4269
[13-15], the resonant SC (RSC) and resonant FC (RFC)
converters were investigated, where the inductors were added
to form a sinusoidal shape with the capacitors to perform a
soft switching for all the switches. Furthermore, the resonant
topologies can significantly minimize the switching losses
and reduce EMI noise.
This paper presents a high efficiency multilevel FC DC/DC
converter with the reduced component rating and device
count. A resonant technique is adopted for the proposed FC
DC/DC converter to achieve the zero-current-switching
(ZCS) for all the switches. Therefore, the switching
frequency of power devices can be raised to realize a compact
and efficient RFC converter. The DC voltage source is
connected to the switch rail so as to make two modular top
and bottom cells. Consequently, the lower degree of the total
capacitor voltage rating and control complexity can be
achieved. The proposed converters can be used as the frontend converters to balance the multilevel DC-link voltage. The
proposed converter is evaluated to the conventional FC
converter in terms of the power density, power loss, and cost.
The developed topology has been implemented on a
prototype converter to verify its feasibility.
II.
analysis and results are based on the proposed quintuple RFC
converter with three active switches.
Fig. 1 Power device rails connected to the source and load.
PROPOSED FC CONFIGURATION
A. General FC Topology
Fig. 1 shows three power device rails where the DC
voltage source Vs is connected to the active switch rail. The
output capacitors, Cto and Cbo, are linked to the top and
bottom diode rails. The flying capacitors can be
symmetrically separated in two top and bottom sides
connected to the power devices. Fig. 2 shows two different
quintuple FC DC/DC converters with three and four active
switches, respectively. The proposed conversion system can
work as a step-up DC/DC converter with a voltage gain of
m+n+1, where m and n are the number of capacitors in the
top and bottom sides, respectively. In a symmetric operation
(m=n), the converter provides an odd voltage gain of 2n+1. In
this paper, the subscripts “t” and “b” represent the
corresponding variables to the circuit components at the top
and bottom sides, respectively.
B. Proposed RFC DC/DC Converters
The proposed quintuple RFC converters are shown in Fig.
3. In the top and bottom cells, each resonant capacitor is in
series with one small resonant inductor. Due to the small
value of the resonant inductances, the air-core inductor or
parasitic inductor existing in the circuit can be used to
achieve a higher power density [16]. In this paper, the
Fig. 2 Proposed quintuple DC/DC FC converters: (a) With three active
switches; and (b) With four active switches;
C. Operation Principle of Quintuple RFC Converter with
Three Active Switches
Fig. 4 shows the equivalent circuits of the proposed
quintuple RFC converter with three active switches. It is
assumed that all the switches, diodes, capacitors, and
inductors are ideal, Vs is an ideal DC voltage source, and the
load is modeled by a pure resistor (Rload). The operation of the
circuit can be described in six modes when each switch is
turned ON. Fig. 5 shows the key current waveforms of the
proposed quintuple RFC converter under the steady-state
condition.
4270
1) Mode I [t0, t1] [Fig. 4(a)]
In the beginning of Mode I, the switch S1 is OFF, whereas
S2 and S3 are ON. The charging currents flow through Dt1,
Db3, S2, and S3 as shown in Fig. 4(a). Therefore, the capacitor
Ct1 is charged by Vs and Cb2 is discharged to Cbo (Cb2 was
charged to two times the input voltage in Mode III) in the top
Dt3
Dt3
Cto
Dt2
Dt1
S1
Vs
Db1
Dt2
Dt1
Lt1 Lt2
Ct1 Ct2
S1
Load
S2
S3
Cbo
Cto
Vs
Lt1
Ct1
Lt2
Ct2
S2
Db2
Lb2
Cb2
S4
Db1
Load
Cbo
Db3
Lb1
Cb1
2) Mode II [t1, t2] [Fig. 4(b)]
In this mode, all the switches and diodes are turned OFF.
Therefore, the resonances stop at the resonant loops as shown
in Fig. 4(b). The resonant capacitor voltages of Ct1, Ct2, Cb1,
and Cb2 are unchanged. The output capacitor voltages of Cto
and Cbo are discharged to the load (see the blue dashed line in
Fig. 4(b)).
3) Mode III [t2, t3] [Fig. 4(c)]
In the Mode III, S1 and S3 are turned ON whereas S2 is
OFF. It can be seen from Fig. 5(e) that the currents through
S1 is increased by the ZCS operation with the half-cycle
resonant shape.
S3
Lb1
Cb1
Fig. 5(g). At t=t1, the switch S2 can be turned OFF under the
zero-current condition.
Lb2
Cb2
Db2
Db3
(a )
(b)
Fig. 3 Proposed quintuple DC/DC RFC converters: (a) With three active
switches; and (b) With four active switches;
t
t
t
t
t
t
t
Fig. 5. Key current waveforms of the quintuple RFC converter at the
steady-state. (a), (b), and (c) Switching patterns. (d) Diode currents in the top
cell. (e) and (f) Switch currents S1 and S2. (g) Capacitor currents Ct1 and Ct2.
Fig. 4. Equivalent circuits of the quintuple RFC converter. (a) Mode I [t0,
t1]. (b) Mode II, Mode IV, and Mode VI [t1, t2]. (c) Mode III [t2, t3]. (d) Mode
V [t4, t5].
and bottom cells, respectively. The resonant capacitor current
Ct1 rises and then falls in a sinusoidal manner, as shown in
The resonant capacitors Ct2 and Cb2 are charged to two
times the input voltage level by Vs, Ct1, and Cb1, as shown in
Figs. 4(c) and 5(g). It can be observed from Fig. 5(g) that the
capacitors Ct1 and Ct2 are charged and discharged by a 120º
phase shift with respect to each other. At the time of t3, the
switch S3 become OFF under the zero-current condition and
there are no reverse recovery losses for Dt2 and Db2.
4271
4) Mode IV [t3, t4] [Fig. 4(b)]
This mode is similar to that of Mode II. Therefore, all the
switches and diodes are turned OFF, and the load is supplied
by the output capacitors Cto and Cbo (see Fig. 4(b)).
5) Mode V [t4, t5] [Fig. 4(d)]
At the instant t=t4, S1 and S2 are turned on while S3 is OFF.
The current through S1 and S2 are increased from zero and
then S1 and S2 are turned ON with the zero-current condition.
In this mode, the capacitor Cb1 is charged by Vs whereas Ct2
is discharged to the Cto from two resonant loops (see the red
and green dashed lines in Fig. 4 (d)). Therefore, the capacitor
currents Cb1 and Cb2 are charged and discharged by a 120º
phase shift with respect to each other. Then, S1 and S2 can be
turned OFF with the zero-current condition, and there are no
reverse recovery losses for Dt3 and Db1.
6) Mode VI [t5, t6] [Fig. 4(b)]
This mode is similar to that of Mode II and Mode IV.
Therefore, all the switches and diodes are turned OFF. The
load is supplied by the output capacitors Cto and Cbo (see Fig.
4(b)).
Here, it is assumed that the resonant inductors have the
same values and all the capacitances are equal except for the
output capacitors that are large enough to provide the low
voltage ripple for the load. The state equations of the
associated equivalent circuit to Mode I (Fig. 4(a)) are
di C t1
Vs = L t1
+ v C t1 ,
(1)
dt
dvC t1
i C t1 = C1
,
(2)
dt
di L b 2
(3)
− vC b2 − Lb2
+ v C bo = 0 .
dt
From Fig. 5, by ignoring the impact of the short times
( t1 ≤ t ≤ t 2 , t 3 ≤ t ≤ t 4 , and t5 ≤ t ≤ t 6 ), it can be assumed
iC
b2
(t) = −
3πPo
3
sin( ωs t )
2Vo
2
where ωs is the switching frequency equal to
(7)
2
3 L t1C t1
, Vo
and Po are the output voltage and output power, respectively.
The voltage ripple of Ct1 is
πPo
.
(8)
ΔVC =
t1
Vo C t1ωs
The capacitance value should be chosen according to the
voltage ripple. From the analysis, it can be noticed that the
voltage and current stresses of all the switches are identical.
D. Component Stress Analysis and Comparisons
The proposed quintuple FC converter in Fig. 2(a) is
evaluated against the conventional FC converter to highlight
its advantages for the high voltage gain applications. Table I
lists the voltage and current ratings of the components for
both converters in a voltage gain of 5. It should be noted that
all the voltage and current ratings are average values. From
the Table I, it can be seen that the number of switches is
smaller than that of the conventional FC topology. Smaller
switch count means lower cost and fewer gate drivers.
that t 0 = 0 , t1 = t 2 ≈ Ts / 3 , t 3 = t 4 ≈ 2Ts / 3 , and t 5 = t 6 ≈ Ts .
The initial conditions of (1) and (2) are
πPo
. The solutions of
i L (0) = 0 and VCt1 (0) = Vs −
t1
Vo C rt1ω r
Mode I can be obtained as
πPo
3
v C ( t ) = Vs −
cos( ωs t )
t1
Vo C t1ωs
2
iC (t ) =
t1
vC
b2
3
3πPo
sin( ωs t )
2
2Vo
( t ) = 2Vs +
πPo
3
cos( ωs t )
2
Vo C t1ωs
(4)
(5)
(6)
Fig. 6. Conventional FC converter with the voltage gain of 5.
Furthermore, the switching scheme of the conventional FC
converter is more complicated than that of the proposed FC
converter due to the connection of the capacitors to the diodes
and switches. It can be observed that the total voltage rating
of capacitors for the proposed FC converter can be reduced to
10Vs whereas the conventional FC converter has a total
voltage rating of 15Vs. The total power rating obtained of
active switches is 12VsIo for the proposed FC converter
whereas the conventional FC converter achieves a total power
rating of 20VsIo.
4272
Table I
Voltage and current ratings of the components for the
proposed FC and conventional FC converters
Topology
Proposed FC
converter
Conventional
FC converter
No. of flying capacitors
No. of output capacitors
4
2
Voltage rating of capacitors
Vs, 2Vs, 2Vs
4
1
Vs, 2Vs, 3Vs,
4Vs, 5Vs
Io
Io
3
Vs
4Io
6
Vs
5
Vs
4Io
5
Vs
Current rating of diodes
Io
Io
Total capacitor voltage rating/Vs
Charging/discharging current
rating of capacitors
No. of switches
Voltage rating of switches
Current rating of switches
No. of diodes
Voltage rating of diodes
components. It can be also observed that the proposed FC
converter achieves about 30% reductions in the total losses
and cost, compared to the conventional FC converter. It
should be noted that switching losses are minimal in the
power loss comparisons shown in Fig. 8 owing to the soft
switching technique for both FC topologies.
III. SIMULATION RESULTS
The quintuple RFC converter was simulated by using
MATLAB/Simulink with the PLECS Blockset. The circuit’s
component values are listed in Table II. The proposed FC
converter by itself has a poor regulation property. It only
amplifies its input voltage and achieves a high efficiency with
a fixed duty cycle. As a result, it is mandatory to add an
additional stage that will provide a regulated DC voltage for
the proposed RFC converter and an electrical isolation for
safety reasons. An isolated hard-switched DC/DC converter
with a high frequency 1:1 transformer can fulfill
aforementioned requirements.
200
Proposed FC converter
Conventional FC converter
160
120
+
80
−
40
0
3
5
7
9
11
13
Voltage gain
15
17
Fig. 9. Proposed RFC configuration connected to the isolated DC/DC
converter and renewable energy source.
19
70
Fig. 7. Normalized total capacitor voltage rating versus voltage gain.
(a )
50
70
(b)
VCt1 [V]
60
60
VCb1 [V]
50
120.5
( c ) 120
119.5
120.5
( d ) 120
VCto [V]
VCbo [V]
119.5
301
( e ) 300
Vo [V]
299
Fig. 8. Power density, total power loss, and cost comparisons.
Fig. 7 shows the normalized total voltage rating of
capacitors versus the voltage gain. The total capacitor voltage
rating can be considerably reduced, compared to the
conventional FC converter, particularly for the higher voltage
gains. The power density of the proposed FC converter is 2.5
times higher than that of the conventional FC topology, as
shown in Fig. 8 because of the lower voltage stress of passive
Time[100 μs / div]
Fig. 10. Simulation voltage waveforms of the quintuple RFC converter.
The high frequency 1:1 transformer reduces the voltage
stress of main switches and power dissipation from the
leakage inductance on the secondary side [17]. The isolated
DC/DC converter supplies a regulated DC voltage for the
proposed quintuple RFC converter as shown in Fig. 9.
4273
Fig. 10 shows the simulation voltage waveforms of the
resonant capacitors Ct1 and Cb1, output capacitors Cto and Cbo,
and output from the top to bottom. The controlled output
voltage is set to 300 V. It can be seen that Ct1 and Cb1 are
charged by a 240° phase shift respect to each other. Fig. 11
shows the simulation waveforms for the switching patterns,
switch currents, and diode currents. It can be seen from Fig.
11(d) and (e) that the switches S1 and S2 can be turned ON
and OFF under the zero-current condition. There are no
reverse recovery losses for Dt1 and Dt2, as depicted in Fig.
11(f).
Table II
Simulation parameters
Parameter
Value
Input voltage
60V
Switching frequency
10 kHz
Resonant capacitor
20µF
Resonant inductor
5.6µH
Output capacitor
100µF
Output power
1kW
The experimental results consist of the relevant voltage and
current waveforms at the nominal load.
Fig. 12 presents the experimental waveforms of the
capacitors voltages and output voltages. It can be seen that
the top and bottom capacitors, Cb2 and Ct2, are charged and
discharged with a 120° phase shift respect to each other. The
empirical voltages observed in Fig. 12 agree well with the
simulation results of Fig. 10.
Fig. 12. Experimental results of operating voltages. (a) and (b) resonant
capacitor voltages in the top cell. (c) and (d) resonant capacitor voltages in
the bottom cell. (e) and (f) Output capacitor voltages. (g) Output voltage.
Time [50 μs / div ]
Fig. 11. Simulation current waveforms of the quintuple RFC converter. (a)
Switching patterns. (b) and (c) Switch currents S1 and S2. (d) Diode currents
Dt1 and Dt2.
IV. EXPERIMENTAL RESULTS
A 1 kW laboratory prototype converter was implemented
to confirm the theoretical developments of the proposed
quintuple FC converter. The selected passive component
values are the same as those of the simulation in Section III.
Fig. 13 depicts the experimental waveforms of diode
currents. It is not convenient to measure the current of S1, S2,
and S3 directly due to its package and circuit layout.
Therefore, the diode currents of Dt1, Dt2, Db2, and Db3 are
captured to show the soft-switching technique for all the
switches. Two key observations can be found in Fig. 13:
First, all the diodes are operated under the zero reverse
recovery losses. Second, the switch current S3 is turned OFF
under the zero-current condition because the turned-OFF
instants of Dt2 and Db2 are in phase with that of S3 as
explained in Mode III (see Fig. 13(c) and (e)). As a result, it
is worth mentioning that all the switches in the proposed
converter are turned ON and OFF under the zero-current
condition.
The efficiency of the proposed RFC converter for a
different load was measured and its result is shown in Fig. 14.
The efficiency reaches more than 93%, which is reasonably
high considering that the converter is a prototype. The test
4274
results clearly show that the proposed circuit can enhance the
power density and efficiency owing to the minimum
switching losses.
power device count to highlight its advantages with respect to
the power density, power loss, and cost. The developed
topology has been implemented on a 1-kW prototype
converter to test its feasibility.
ACKNOWLEDGMENT
This work was supported by the National Research
Foundation of Korea (NRF) grant funded by the Korea
government (MSIP) (2010-0028509).
REFERENCES
[1]
[2]
[3]
[4]
[5]
[6]
[7]
Fig. 13. Experimental results of the current waveforms. (a) Gate-emitter
voltages of S1, S2, and S3. (b) and (c) Diode currents in the top cell. (d) and
(e) Diode currents in the bottom cell.
[8]
[9]
[10]
[11]
[12]
[13]
Fig. 14. Efficiency curve with different output power.
[14]
V.
CONCLUSIONS
In this paper, a new multilevel flying-capacitor (FC)
DC/DC converter is proposed for distributed power
generation applications. A resonant technique is adopted for
the proposed FC DC/DC converter to achieve a softswitching scheme for all the switches. The soft-switching
scheme significantly reduces switching losses and minimizes
the overall system volume. The proposed FC converter is
verified by a simulation and evaluated to the conventional FC
DC/DC converter in terms of the component rating and
[15]
[16]
[17]
4275
D. Vinnikov and I. Roasto, “Quasi-Z-source-based isolated DC/DC
converters for distributed power generation,” IEEE Trans. Ind.
Electron., vol. 58, no. 4, pp. 1239–1250, Jan. 2011.
A. F. Zobaa and C. Cecati, ‘‘A comprehensive review on distributed
power generation,’’ in Proc. SPEEDAM, 2006, pp. 514---518.
W. H. Li and X. N. He, “Review of nonisolated high-step-up DC/DC
converters in photovoltaic grid-connected applications,” IEEE Trans.
Ind. Electron., vol. 58, no. 4, pp. 1239–1250, Apr. 2011.
F. Gao, B. Blunier, A. Miraoui, and A. E. Moudni, “A multiphysic
dynamic 1-D model of a proton-exchange-membrane fuel-cell stack for
real-time simulation,” IEEE Trans. Ind. Electron., vol. 57, no. 6, pp.
1853–1864, Jun. 2010.
J. Y. Lee, Y. S. Jeong, and B. M. Han, “An isolated DC/DC converter
using high-frequency unregulated LLC resonant converter for fuel cell
applications,” IEEE Trans. Ind. Electron., vol. 58, no. 7, pp. 2926–
2934, Jul. 2011.
T. A. Meynard and H. Foch, “Multilevel conversion: high voltage
choppers and voltage-source inverters,” in Proc. IEEE PESC, 1992, pp.
397–403.
J. C. Rosas-Caro, J. M. Ramirez, , F. Z. Peng, and A. Valderrabano, “A
DC-DC multilevel boost converter,” IET Power Electron., vol. 3, no. 1,
pp. 129-137. 2010.
S. B. Monge, S. Alepuz, and J. Bordonau, ‘‘A Bidirectional multilevel
boost---buck DC---DC converter’’ IEEE Trans. Power Electron., vol. 26,
no. 8, pp. 2172–2183, Aug. 2011.
Y. Tang, T. Wang, and Y. He, “A switched-capacitor based activenetwork converter with high voltage gain,” IEEE Trans. Power
Electron., vol. 29, no. 6, pp. 2959-2968, Jun. 2013.
F. H. Khan, L. M. Tolbert, and W. E. Webb, “Hybrid electric vehicle
power management solutions based on isolated and nonisolated
configurations of multilevel modular capacitor-clamped converter,”
IEEE Trans. Ind. Electron., vol. 56, no. 8, pp. 3079–3095, Aug. 2009.
J. C. R. Caro, J. M. Ramirez, and P. M. G. Vite, “Novel DC-DC
multilevel boost converter,” in Proc. IEEE PESC, 2008, pp. 2146–2151.
W. Qian, F. Z. Peng, M. Shen, and L. M. Tolbert, “3X DC-DC
multiplier/divider for HEV systems,” in Proc. IEEE APEC, 2009, pp.
1109–1114.
A. Parastar and J. K. Seok, “High-gain resonant switched-capacitor
cell-based DC/DC converter for offshore wind energy systems,” IEEE
Trans. Power Electron., “in press”.
A. Parastar, A. Gandomkar, M. G. Jin and J. K. Seok, “High power
solid-state step-up resonant Marx modulator with continuous output
Current for Offshore Wind Energy Systems”, in proc. IEEE ECCE,
2013, pp. 1709-1716.
K. Sano and H. Fujita, “Performance of a high-efficiency switchedcapacitor-based resonant converter with phase-shift control,” IEEE
Trans. Power Electron., vol. 26, no. 2, pp. 344–354, Feb. 2011.
K. Zou, M. J. Scott, and J. Wang, “A switched-capacitor voltage tripler
with automatic interleaving capability,” IEEE Trans. Power Electron.,
vol. 27, no. 6, pp. 2857–2868, Jun. 2012.
N. P. Papanikolau and E. M. Tatakis, “Active voltage clamp in flyback
converters operating in CCM mode under wide load variation,” IEEE
Trans. Ind. Electron., vol. 51, no. 3, pp. 632–640, Jun. 2004.