A CLOCK INDEPENDENT UWB TRANSMITTER AND FRONT-END
RECEIVER FOR MEDICAL IMAGING
A thesis submitted to
the Department of Electrical and Electronic Engineering
of
Bangladesh University of Engineering and Technology
in partial fulfillment of the requirements
for the degree of
MASTER OF SCIENCE IN ELECTRICAL AND ELECTRONIC ENGINEERING
By,
Mohammad Nahidul Karim
Student ID: 0412062251
BANGLADESH UNIVERSITY OF ENGINEERING AND TECHNOLOGY
March 2014
i
DECLARATION
It is hereby declared that this thesis or any part of it has not been submitted elsewhere for the
award of any degree or diploma.
--------------------------------------(Mohammad Nahidul Karim)
Signature of the Supervisor,
-----------------------------------------(Prof. Pran Kanai Saha)
ii
APPROVAL CERTIFICATE
The thesis titled “A CLOCK INDEPENDENT UWB TRANSMITTER AND FRONT-END
RECEIVER FOR MEDICAL IMAGING” submitted by Mohammad Nahidul Karim, Student ID:
0412062251, Session: April 2012 has been accepted as satisfactory in partial fulfillment of the
requirement for the degree of MASTER OF SCIENCE IN ELECTRICAL AND ELECTRONIC
ENGINEERING on March, 2014.
BOARD OF EXAMINERS
1.
Dr. Pran Kanai Saha
Professor,
Department of EEE, Bangladesh University of Engineering
and Technology, Dhaka-1000
2.
Dr. Pran Kanai Saha
Professor and Head,
Department of EEE, Bangladesh University of Engineering
and Technology, Dhaka-1000
3.
Dr. A.B.M. Harun-ur Rashid
Professor,
Department of EEE, Bangladesh University of Engineering
and Technology, Dhaka-1000
4.
Dr. Md. Anwarul Abedin ,
Professor and Head, Department of EEE,
Dhaka University of Engineering & Technology,
Gazipur, Bangladesh
5.
Dr. Md. Forkan Uddin
Assistant Professor,
Department of EEE, Bangladesh University of Engineering
and Technology, Dhaka-1000
Chairman
(Supervisor)
Member
(Ex-Officio)
Member
Member
(External)
Member
iii
CONTENTS
TITLE
Page
no:
Declaration
Approval Certificate
List of Figures
List of Tables
Acknowledgement
Abstract
Chapter 01
1.1
1.2
1.3
1.3
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Introduction
Motivation
Objective of this thesis work
Literature Review
Thesis Outline
Chapter 02
Medical Imaging and Sensing using UWB
Technology
2.1
Microwave Imaging for Medical Application
2.1.1
Early Stage Cancer Cell Detection
2.1.2
Capsule Endoscopy
2.1.3
Blood Pressure Measurement
2.1.4
Wireless Body Area Network
2.2
Architecture of UWB Transceiver for Medical Imaging
2.2.1
On/In-Body Transmitter
2.2.2
On-Body Receiver
Chapter 03
3.1
Characterization of Impulse Radio Based UWB
Transmitter
UWB Pulses
3.1.1
Gaussian pulse
3.1.2
Gaussian monocycle pulse
3.1.3
Scholtz monocycle
3.1.4
Rectangle monocycle
3.1.5
Sinusoidal monocycle
3.1.6
Pulse train
3.2
UWB Modulation Techniques
3.2.1
Pulse Position Modulation (PPM)
3.2.2
Binary Phase Shift Keying (BPSK)
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3.2.3
Other Modulation Techniques
3.2.3.1
Pulse Amplitude Modulation (PAM)
3.2.3.2
On Off Keying(OOK)
3.2.3.3
Orthogonal Pulse Modulation(OPM)
3.2.3.4
Proposed Modulation Technique
3.3
Characteristics Parameters for UWB Pulses
3.3.1
3.3.2
3.3.3
Pulse Width
Power Spectral Density
Equivalent Isotropically Radiated Power (ERIP)
Chapter 04
Characterization and Design Parameters of UWB
Front End Receiver
4.1
Scattering Parameters(S-Parameters)
4.2
Input Impedance Matching
4.3
Voltage Standing Wave Ratio (VSWR)
4.4
Voltage Gain
4.5
Power Gain
4.6
Available Power Gain
4.7
Transducer Power Gain
4.8
Available Power Gain
4.9
Noises
4.10
Linearity
4.10.1
Gain Compression
4.10.2
2nd and 3rd order Intercept Point (IP2 and IP3)
4.11
Stability
Chapter 05
5.1
5.2
5.2.1
5.2.2
5.3
5.4
Basic theory
Design Methodology of UWB Transmitter Circuit
UWB Transmitter Circuit for Medical Imaging
UWB Transmitter Circuit for Data Communication
Power Dissipation
Pulse width changing by width of Active Devices
Chapter 06
6.1
6.2
6.2.1
6.2.2
6.2.3
6.2.4
Design of a CMOS Impulse Radio UWB Transmitter
Design of UWB front End Receiver
LNA Topology and Design Challenges
Circuit Analysis
Input Matching
Noise Analysis
Linearization
Output Matching
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6.3
6.3.1
6.3.2
6.3.3
6.3.4
6.3.5
LNA Topologies
Classical Noise Matching Technique (CNM)
Balanced Amplifier
Distributed Amplifier
Negative Feedback Wideband LNA
Simultaneous Noise and Input Matching (SNIM) Technique
Chapter 07
7.1
7.1.1
7.1.2
7.1.3
7.2
7.2.1
7.2.2
7.2.3
7.2.4
7.3
7.3.1
7.3.2
7.3.3
7.3.4
7.3.5
7.3.6
7.4
Proposed UWB transmitter
Design Concerns for the Delay Block
Design Concerns for the Charge Pump Stage
Design Concerns for the Data Communication
Simulated Results of the Proposed Transmitter
Pulse Width and Pulse Amplitude
Power Spectral Density
Power Consumption
Pulse Rate
Proposed UWB Front End Receiver
Design Concerns for Filter Design and Cascade
Design concerns against Noise
Design concerns for Cascode structure
Design Concerns for Linearity Improvement
Design Concern for Output Matching
Design Concerns for Overall Power Consumption
Simulated Results for The Proposed LNA Structure
Chapter 08
8.1
Simulated Results and Performance Analysis
Conclusion
Future Work
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References
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Appendix A:
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Appendix B:
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Appendix C:
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List of Figures
Figure no
Figure 1.1
Figure 2.1
Figure 2.2
Figure 2.3
Figure 2.4
Figure 2.5
Figure 2.6
Figure 3.1
Figure 3.2
Figure 3.3
Figure 3.4
Figure 3.5
Figure 3.6
Figure 3.7
Figure 3.8
Figure 3.9
Figure 3.10
Figure 3.11
Figure 3.12
Figure 3.13
Figure 3.14
Figure 3.15
Figure 3.16
Figure 4.1
Title of the Figure
Page
Summary of the measured dielectric constant (a) and conductivity (b)
of normal muscle tissue, malignant and fat at radio and microwave
frequencies
A comprehensive breast model is shown in (a), whereas (b)
demonstrates the position of UWB transceivers for microwave imaging
WBAN topology for medical sensing
UWB Transmitter Block Diagram for microwave imaging
PSD of the Transmitted Gaussian Pulse (5th Derivative & 2nd
Derivative)
UWB Receiver Block Diagram for (a) imaging and (b) for data
communication
Averaged BER Performance Using Different Templates
(a)Gaussian Pulse (b)PSD of Gaussian Pulse
(a) Gaussian Monocycle (b)PSD of Gaussian Monocycle
(a) Scholtz‟s Monocycle (b) PSD of Scholtz‟s Monocycle
(a) Rectangular Monocycle Pulse (b)PSD of a Rectangular Monocycle.
As these pulses have a dc value, the PSD of these pulses are spread
near 0Hz
Sinusoidal Monocycle (b)PSD of Sinusoidal Monocycle
(a)Square Pulse Train (b)Gaussian Pulse (c)First derivative Pulse
(d)Second derivative Pulse
(a) Time domain raised and pseudo raised cosine pulses and (b) their
power spectrum
Basic Communication system
Comparison of pulse position modulation and bi-phase modulation
methods for UWB communication.(a)Unmodulated Pulses (b)PPM
Modulated (c)Bi-phase Modulated signal
Comparison of other modulating techniques (a)Unmodulated Signal
(b)Pulse
Amplitude
Modulation(PAM)
(c)On-Off
Keying
(d)Orthogonal Pulse Modulation(OPM)
The figure at left illustrates the OOK for IR based UWB
communication. To facilitate BPSK modulation with OOK, the figure
on right depicts the proposed modulation technique
An ultra short UWB pulse and its spectrum.
Comparison Between Low And High PSD
UWB spectrum as described by FCC(For GPRs, Wall Imaging, &
Medical Imaging Systems)
UWB spectrum as described by FCC for Indoor Systems
UWB spectrum as described by FCC for hand held devices
A two port network in terms of S-Parameters
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Figure 4.2
Figure 4.3
Figure 4.4
Figure 5.1
Figure 5.2
Figure 5.3
Figure 5.4
Figure 5.5
Figure 5.6
Figure 5.7
Figure 5.8
Figure 6.1
Figure 6.2
Figure 6.3
Figure 6.4
Figure 6.5
Figure 6.6
Figure 6.7
Figure 6.8.1
Figure 6.8.2
Figure 7.1
Figure 7.2
Figure 7.3
Figure 7.4
Figure 7.5
Figure 7.6
Figure 7.7.1
Figure 7.7.2
Figure 7.8
Figure 7.9
Figure 7.10
Input impedance matching of antenna to an amplifier
Variation of output power due to variation in input power
Illustration of inter-modulation behavior
Block Diagram for BPSK modulated IR-UWB Communication
Source follower with RLC network
Data Pulse, delayed inverted data pulse and ultra short pulse
Input triangular pulses feed to the PMOS(red) and NMOS(blue) of the
charge pump stage and the output across load resistance(green).
The proposed IR based UWB transmitter for medical imaging
The wave shapes at different stages of the proposed circuit
The proposed IR based UWB transmitter for data communication
The wave shapes at different stages of the proposed circuit for data
communication
3rd Order T-section Chebyshev Filter
Proposed simplified circuit for wideband LNA and small signal model
for M1 transistor
Noise model for MOS transistor, M1
Cascode amplifier with PMOS IIP3 booster
Schematic of an LNA topology in cascode structure(a) and simplified
small signal model (b) of the LNA for calculating input referred noise
Block diagram of balanced amplifier
Basic Distributed Amplifier
Shunt-series amplifier
Schematic for CS-CG cascode LNA topology in SNIM technique (a)
and small signal model of the structure (b)
Delay Block used in transmitter design
Data bits and delayed data bits(a), triangular and delayed triangular
pulse (b) by delay blocks used in the proposed circuit
The input data bit, delayed data bits to NOR gate and output of the
NOR gate. The output pulse is generated during transition of input data
bit from „1‟ to „0‟
Charge pump stage of the proposed circuit
The effect of output capacitor on the output pulse settlement after
switching „on‟ the circuit. Larger the value of the capacitor result in
larger the pulse amplitude with cost of larger settling time
Shape of the output pulse at proper scale for the proposed circuit of
medial microwave imaging
Shape of the output pulse at proper scale for the proposed circuit of
medial microwave imaging
PSD of the UWB pulse for microwave imaging
PSD of the UWB pulse for high speed data communication and
microwave imaging
Pulse generation @5Gpulse per second. The data type at the input of
this circuit is NRZ data
Pulse generation @100Mpulse per second. The data type at the input of
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Figure 7.11
Figure 7.12
Figure 7.12.1
Figure 7.13
Figure 7.14
Figure 7.15
Figure 7.16
Figure 7.17
Figure 7.18
Figure 7.18.1
Figure 7.18.2
Figure 7.19
Figure 7.20
Figure 7.21
Figure 7.22
Figure 7.23
Figure 7.24
Figure 7.25
Figure 7.26
Figure 7.27
Figure 7.28
Figure 7.29
Figure 7.29.1
Figure 7.30
Figure 7.31
Figure 7.32
Figure 7.33
Figure 7.34
Figure 7.35
Figure: 7.36
Figure 7.37
Figure 7.38
this circuit is NRZ data
Pulse generation @4Gpulse per second. The data type at the input of
this circuit is NRZ data.
Pulse generation @200Mpulse per second. The data type at the input of
this circuit is NRZ data
Monte Carlo Simulation results for proposed circuits
Comparison between the response of a chebyshev and butterworth
filter
Input matching network
Dependency of NF on Re{Z opt} and Re{Zin} derived from analytical
expressions
Demonstrating (a) CS-CS cascode (b) CS-CG cascode and (c) CGCS cascode structures
(a) Cascode amplifier with the folded cascode PMOS IIP3 booster.
(b) Third-order power series coefficients of drain current I sA at DC
Proposed LNA design without buffer stage
(a) The load impedance of the core Amplifier (b) Approximation of (a)
Dependency of noise on width and bias current of the amplifier
Proposed front end LNA for UWB application
Input and output reflection parameters
Input impedance and output impedance of the proposed LNA
Noise Figure of the proposed LNA
Equivalent Noise Resistance
Forward gain and reverse isolation of the proposed LNA
Transducer gain, GT, Available gain, GA and Power gain, GP of the
proposed LNA
The group delay of the proposed circuit
Voltage Standing Wave Ratio at input and output (matched for 50Ω)
Linearity performance of the proposed LNA
Variation in linearity of the Proposed LNA
Linearity performance of the Proposed LNA @ 1.5V supply and
variation of output with respect to different supply voltage
Stability of the proposed LNA
Variation of minimum noise figure and forward gain of the circuit due
to change in supply voltage
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Forward gain of LNA at different process corner
Input matching parameter, S11 at different process corner
Output matching parameter at different process corners
Minimum noise figure for different process corner
Time domain input and output of the front-end receiver
Variation in S21 and minimum noise figure in dB due to random
process variations
S-parameter of UWB microstrip antenna with in-band notches
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Figure B.1
Figure C.1
Figure C.2
Figure C.3
Figure C.4
Figure C.5
Figure C.6
Figure C.7
Figure C.8
Figure C.9
Figure C.10
Figure C.11
Figure C.12
BER for different SNR and line of sights
(a)Illustrates the variation in drain current and transconductance with
respect to gate-source voltage under different body-source voltage of a
typical NFET, (b) shows the DC characteristics of NFET
(a)Illustrates the variation in drain current and transconductance with
respect to gate-source voltage under different body-source voltage of a
typical PFET, (b) shows the DC characteristics of PFET
Input referred noise voltages of NFET and PFET. PFET shows lower
contribution to noise at higher frequencies
(a)Illustrates the variation in drain current and transconductance with
respect to gate-source voltage under different body-source voltage of a
typical LVT_NFET, (b) shows the DC characteristics of LVT_NFET
Variation in quality factor, Q with respect to frequency of a single layer
inductor
Variation in inductance and (b) quality factor, Q with respect to
frequency of a single layer inductor
Variation in capacitance under different voltage and temperature
Variation in capacitance and quality factor with respect to width of the
capacitor models
Variation in capacitance of ncap models under different voltage
Varactor model @ Vg=0V, Vsd=0V
Variation in resistance of a silres model resistance under different
frequency
Variation in resistance of a P+ Polysilicon Resistance model resistance
under different frequency
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List of Tables
Table no.
Table 7.1
Table 7.2
Table 7.3
Table 7.4
Table 7.5
Table 7.6
Table 7.7
Table 7.8
Title of the Table
Power Consumption at different Process Corner
Inductors used in the proposed LNA
List of device sizes used in the design
LNA‟s performance at different Process Corner
Tabulated Transmitter Response
Tabulated LNA Response
Summary of the proposed transmitter and other published work
Summary of the proposed LNA and other published work
Page no.
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ACKNOWLEDGEMENT
First of all thanks to Almighty God, without his favor, nothing is possible, nothing can be
achieved.
I would start with conveying my profound respect to my supervisor Dr. Pran Kanai Saha for
providing me the opportunity to choose and work on this topic and for his guidance throughout
the thesis work. It was his idea on which I was able to work further and prepare this thesis
dissertation. His inspiration and academic guidance has motivated me to work on development of
circuits and systems for medical application.
I would like to show my gratitude to Intel Bangladesh Association for providing me the license
to use IBM 130nm CMOS RF technology and industry standard Cadence Spectre Simulation
Tools for during this thesis work.
I am also grateful to Dr. A.B.M. Harur-ur-Rashid for his relentless help with available software
and valuable time. With his unparallel knowledge in the field of VLSI circuit designing, he has
helped me several times in the lab to perform and interpret toughest simulation results.
Then, I would like to express my gratitude to my classmates and VLSI lab assistant who helped
me a lot along the way last one year.
Finally, I have tried my best to represent this thesis dissertation as appropriate as possible. I am
feeling myself fortunate to finish the thesis on such an interesting topic is really a great
experience.
xii
ABSTRACT
Due to inherent noise like characteristics, Ultra wideband (UWB) radio signals are highly
suitable for less invasive medical applications. So, apart from the short distance high-speed
communication, nowadays Ultra wideband systems are considered a viable alternative for
microwave imaging because of its penetration capability and high precision imaging. In this
thesis work, an IR-based UWB transmitter and front end receiver are proposed for medical
imaging and short distance data communication. The transmitter generates Gaussian monocycle
pulse with only data as its input, thus relaxing the complexity of any synchronization. A new
modulation technique is incorporate to facilitate this transmitter for both imaging and data
communication within FCC unlicensed band (3.1-10.6GHz with EIRP <-41.3dBm/MHz) with a
flexibility of varying data speed from few MHz to 4GHz with low power operation. Simulation
has shown the proposed design consumes 0.68pJ/bit irrespective of data speed. The proposed
source degenerated CS-CG cascode low noise amplifier cascaded with Chebyshev filter shows
better noise performance, forward gain and input matching and lower power consumption. A
buffer stage is employed to increase the forward gain of the circuit. The -3dB bandwidth of the
LNA is 6.2GHz from 4.7-10.9GHz with flat response of 12.1±.3dB from 5.2-9.8GHz. The input
and output reflection co-efficient are below -10dB within the band of interest. The simulated
average noise figure is below 3.2dB. The core amplifier consumes 5.22mW power and all the
transistors are biased within the circuit. The proposed circuits are designed using IBM 130nm
CMOS RF technology and simulated in Cadence Spectre and Hspice RF.
xiii
P age |1
CHAPTER 01
INTRODUCTION
1.1 MOTIVATION
In recent times, Ultra wideband (UWB) imaging technique has been proven attractive for the
early breast cancer detection, cardiology, obstetrics, breath pathways and arteries noninvasively [1][2][3]. By utilizing low power ultra short pulses, UWB technique can be used for
localization of malignant tissue/cells at centimeter level accuracy with high resolution in
medical imaging. Figure 1.1 illustrates the contrast of dielectric constant and conductivity of
human muscle and fat tissues at different frequencies. Owing to this obvious contrast in the
electrical properties of different tissues, microwave imaging via UWB technique has shown
much more potential than other traditional imaging methods like X-ray or MRI[4].
Figure 1.1: Summary of the measured dielectric constant (a) and conductivity (b) of normal
muscle tissue, malignant and fat at radio and microwave frequencies.
The utilization of wireless technology utilizing UWB in traditional medical services provides
patients with enhanced mobility. This has a positive effect in the recovery speed of a person
after major surgical procedures or prolonged illness. Due to inherent noise-like behaviour
(ERIP -41.3dBm/MHz), UWB signals are difficult-to-detect and robust against jamming,
potentially rescinding the need for complex encryption algorithms in tiny transceivers. This
low radiation is safe for human body and could influence the environment around a little,
which are suitable for hospital room[5]. Because of these characteristics, developing low
power UWB transceiver system capable of transmitting impulse and receiving reflected and
scattered signals within transmitted frequency range for achieving good spatial resolution
under FCC regulation is important to apply UWB biomedical imaging methods [6][7].
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Recently developed UWB systems are entirely focused on high speed communication. To
convert these systems for medical imaging, size and power consumption are two major
concerns. Several communication techniques, such as, multi-band orthogonal frequency
division multiplexing (MB-OFDM), direct-sequence spread spectrum (DSSS) and Impulse
Radio for UWB communication. The MB-OFDM and DS-UWB transceivers are complex to
design, power hungry and occupy more die area in the chip. UWB radar system has been tested
for remote monitoring and detection for patient monitoring, but not for imaging. Several
published work on breast cancer detection([1]-[3]) had shown cancer cell detection algorithm
based on mono-cycle ultra-short UWB pulses having broad spectral region. These pulses can
be generated by adopting Impulse Radio-based UWB techniques[3]. IR based UWB
transceiver system does not require any mixing operation at the receiver which relaxes further
complexity for signal processing. These types of transceiver systems are not power hungry,
easier to design and smaller in size. IR-based transceiver system is also suitable for high speed
communication. This can facilitate in designing a wireless body area network in order to
develop a fully non-invasive medical monitoring system.
1.2 OBJECTIVE OF THIS THESIS WORK
The goal of this work is to develop an on-chip clock independent Ultra Wideband transmitter
and front-end in circuit level using IBM 130nm CMOS RF technology for medical microwave
imaging and explore its capability to high speed short distance communication.
The possible outcome of the thesis will be a novel UWB transceiver system in which the
transmitter can be used for both data communication and imaging as well as the front end
receiver satisfies the gain-noise-linearity requirements of a microwave receiver and can be
fabricated using a state of the art industrial RF CMOS process design flow.
1.3 LITERATURE REVIEW
UWB imaging systems can allow a physician to monitor internal organ movements without
invasive surgical procedures. Unlike traditional ultrasound systems, which require direct skin
contact, UWB sensors and imaging systems can operate at a standoff distance. Since UWB
signals are non-ionizing, they do not cause the adverse effects associated with X-ray systems
such as CT scanners [3][4]. UWB imaging may enable physicians to make a preliminary
diagnosis without subjecting the patient to risk or discomfort. In addition, these sensors are
well suited for continuous patient monitoring to identify baseline change.
In order to develop such UWB transceiver system for medical imaging, several issues must be
taken in consideration, such as ERIP emission level defined by the FCC regulation, total power
consumption of the circuit, size of the circuit and complexity of the design.
Several designs for UWB transmitter has been proposed for UWB communication system [8][15]. All these designs are suitable to be used for communication purposes, so constrains on
power consumption and size of the device are relaxed. To implement a wireless body area
network via UWB technique, sensors must consume less than 100µW thus putting restrictions
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on power consumption of the transceiver systems. If these designs are to be used for medical
imaging, it must be ensured that the duty cycle of the UWB pulses are lowered reducing
baseline power consumption. That is why Gaussian mono-cycle based UWB transmitters are
most suitable for this purpose. In [11], an Impulse Radio(IR) based UWB radar has been
demonstrated for wireless body area networks. IR-UWB, directly modulates the data bits with
ultra short-duration pulses which has very strong temporal and space resolving, which is
suitable for the localization and detection in the medical applications [3][16][17].
IR based UWB transceiver systems are purely logical combinational circuits designed with
basic logic gates. There are several topologies for IR based transmitters such as Short
Rectangular Pulse generation technique, utilization of choke inductor, triangular pulse
generation technique and delay based designs [11]-[14][18]. All these designs are suitable for
UWB communication and medical imaging, but the design may include some additional
circuitry like delay locked loop and phase locked loop for synchronization of data bits with
clocks. On the other hand, some of these circuits include inductor and large capacitors in the
circuit design which indicates larger die area for the circuit. For imaging, Bit Error Rate (BER)
is not the major concern to be looked for, but for communication between sensors, BER is an
important parameter to be analyzed. In several literatures antipodal, bi-phase GMP based UWB
communication has shown lower BER for high speed communication [3].
For medical imaging, an UWB pulse is transmitted from outside the body and the scattered
signals are collected from different points and the received signals are analyzed based on the
depth and electrical properties of human tissues. Hence the transmitter for this purpose must be
capable of transmitting UWB pulses at any instances. This requirement relaxes the use of any
clock synchronizing module which is used for data communication system and encourages
designing a clock independent UWB transmitter design.
There are several reported medical implants/sensors that remain inside the body and
communicate with a chip that is outside the body [3][10]. For this type of one way
communication, the transmitter to be designed for the implants or sensors must consume lower
energy as much as possible. That is why size and the energy per pulse are important concerns.
Rather than proposing a new circuit topology, a modulation technique which only carries the
information of rising or falling edge of a data stream can extremely reduce the power
consumption. In this way, the same type of transmitter can be used for both medical imaging
and data communication. But, BER and other performance parameter must be analyzed for this
type of communication. High speed communication between UWB sensors can be motivated
for gathering and exchanging a large quantity of sensory data. Such UWB devices can also be
used for inter chip high and low speed short range wireless communication.
The front end of UWB receiver, Low Noise Amplifier (LNA) must have higher bandwidth,
better input output matching, higher linearity and lower noise figure to collect the reflected
signals for better imaging. Classic LNA design techniques like shunt feedback amplifiers,
balanced amplifiers and other topologies are unable to fulfill all the requirements such as lower
noise figure (NF), lower power consumption and linearity at the same time [19][20].
Multistage LNAs which are designed incorporating transmission line based matching
techniques, inherently consume more space. Moreover, noise and power consumption increase
P age |4
with each additional stage [7][19]. Basic common gate topology and balanced amplifiers
provide better input matching at the cost of degradation of overall noise performance [21].
One of the most useful techniques for designing an LNA is to cascade a filter with an
inductively degenerated common source (CS) amplifier to achieve a pass band filtering, to
ensure wide-band input match and low noise figure. LC ladder filter [22] and multi-section
reactive network can be designed to resonate over a band at high frequency to attenuate outband blockers relaxing the power consumption and linearity constraints of the LNA. The
cascode structures such as CS-CG[21], CS-CS[23] and CG-CS[25] configurations are applied
for the improvement of input–output reverse isolation, stability and frequency response of the
LNA.
However, several other important parameters such as linearity and 1-dB compression point
have not been treated significantly by the evolution of design topology. This has motivated the
researchers to incorporate several linearization techniques with the traditional device
topologies for improving linearity. Until now, the most efficient linearization method for a
CMOS LNA has been the derivative superposition (DS) technique [26] which uses a MOS
transistor to reduce the 3rd harmonic current. In [27], an improved linearization technique is
proposed using an auxiliary current path for 3rd harmonic component of current. In [28], a gmboosted current reuse LNA design is demonstrated for narrow band but the noise figure of this
circuit is strongly dependent on operating frequency.
As noise and power consumption of an LNA are inversely related, there is a tradeoff between
these two parameters. Body biasing techniques [29] are relatively new to address this problem,
but this requires an additional biasing. Each CMOS technology has a specific supply voltage,
but circuits can be designed for lower supply voltage to attain the required transconductance by
varying width in order to improve noise figure and lower power consumption.
In recent times, because of high mobility, most of the high frequency LNAs are designed in
compound materials and expensive processes like SiGe (silicon gremanium)[30], GaAs
(Gallium Arsenide) or MMIC (Monolithic Microwave Integrated Circuit)[31] but silicon
technology such as CMOS is more matured and available for designing microwave low noise
amplifiers used in wireless communication systems. This has motivated IC designers to
develop UWB on-chip transceiver system to be used as medical sensors for a long time in
CMOS technology.
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1.4 THESIS OUTLINE:
The thesis dissertation has a total of ten chapters covering various points of view for
high frequency communication.
CHAPTER ONE provides an overview of the literature related to the field of ultra
wideband microwave imaging is provided to clarify the motivation behind this line of research.
CHAPTER TWO presents the theoretical aspects of microwave imaging based on
published works which are relevant to setting the design parameters and methodology for the
proposed design
CHAPTER THREE addresses the characteristics of an UWB pulse and high frequency
design parameters for an UWB transmitter that need to be considered by a designer before
entering the microwave UWB domain.
CHAPTER FOUR discusses the characterization and design parameters of an UWB front
end receiver.
CHAPTER FIVE represents the topology of the proposed UWB transmitter for imaging
and data communication.
CHAPTER SIX illustrates the design considerations for a Low Noise Amplifier design.
CHAPTER SEVEN presents the simulated performance and summary of the proposed
UWB transmitter and front end receiver.
CHAPTER EIGHT provides the design insight for considering the fabrication of the
proposed design in IBM 130nm CMOS RF technology.
CHAPTER NINE by presenting a summary of the overall research and providing a few
suggestions for future work in this particular field
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CHAPTER 02
MEDICAL IMAGING AND SENSING USING UWB TECHNOLOGY
It is known that the UWB pulse is generated in a very short time period (sub-nano second). So
it has spectrum below the allowed noise level. This feature makes it possible to get Gbps speed
by using 10GHZ spectrum. So UWB is suitable to be used for high-speed over short distances.
Such ―
noise-like‖ feature relies on ultra-short waveforms and does not require IF processing
because they can be operated at baseband. This UWB feature has long been appreciated as key
advantages for medical engineering.
This chapter presents a summary of current researches on medical imaging and sensing, both
invasive and noninvasive. The temporarily-invasive imaging technique for early stage breast
cancer detection, noninvasive BP measurements using UWB radar and capsule endoscopy
using UWB communication link are described here and the architecture for medical imaging
and sensing are discussed.
2.1 MICROWAVE IMAGING FOR MEDICAL APPLICATION
2.1.1 EARLY STAGE CANCER CELL DETECTION
Breast cancer continues being one of the main causes of women death; therefore, early
detection of cancerous tumours increases the possibility of successful treatment and survival.
Although a large number of detection methods are available, X-Ray mammography is currently
the most widely used. Nevertheless, despite its ability to provide high resolution images, this
method suffers from high false-alarm rate and the incapability to distinguish between
malignant and benignant tumours [32]. UWB radar medical imaging involves transmitting an
extremely short pulse through the breast tissues and then records the backscattered signal from
different locations. The basis for detecting and locating a cancerous tumour is the different
dielectric properties of healthy and malignant breast tissue. Healthy tissue is largely transparent
to microwaves, whereas tumours, which contain more water and blood, scatter them back to
the probing antenna array [1].
(a)
(b)
Figure 2.1: A comprehensive breast model is shown in (a), whereas (b) demonstrates the
position of UWB transceivers for microwave imaging [3].
P age |7
Figure 2.1 illustrates the microwave imaging perspectives for breast cancer detection.
Preliminary results have been reported using near-field tomographic image reconstruction
(TIR), confocal microwave imaging, space-time beamforming, generalized likelihood ratio test
based detection, and time-of-arrival (TOA) data fusion method [3]. Until recently, the tumour
detection capabilities of various imaging techniques are evaluated through finite-difference
time-domain (FDTD) methods. Different phantom models for EM simulations have been
proposed in the literature.
UWB positioning technology demands extremely high speed signal processing with significant
power efficiency. Furthermore, medical WBANs require miniature devices at low power
consumption. Finding the best compromise between size, power, and localization accuracy is a
major challenge. The so-called continuous-time binary value (CTBV) signal processing
paradigm has demonstrated high speed sampling (>30 GHz) [3].
2.1.2 CAPSULE ENDOSCOPY
Traditional techniques like the insertion of flexible tubes containing cameras to examine parts
of the digestive tract can examine the upper portion of the digestive tract only, while
colonoscopes help to visualize the lower part (colon). There is a large portion (approx. 6
meters) of the small intestine that cannot be inspected with these traditional techniques.
Capsule endoscopes help to fill this gap with less discomfort for the patient.
State-of-the-art capsule endoscopes are swallowed with water, after which the patient put a
recorder belt on his/her waist. Some hours later (typically eight), medical staff look for
abnormalities by reviewing a video created from the still images transmitted wirelessly from
the capsule endoscope to the recorder belt. Adding the capability to transmit and analyze video
in real time can provide further flexibility and advantages to the current technology. This
additional capability, however, might increase the complexity of the circuitry and hence the
power consumption of the capsule endoscope. The power consumption of a capsule endoscope
must be as low as possible, on the order of 1 mW, with a mandatory small physical size on the
order of 300 cubic millimetres [3]. Transmitting real-time video requires a high data rate
communication link, on the order of 73.8 Mbps for uncompressed VGA data. All these
requirements are hard to achieve using narrowband (NB) systems that operate in the medical
implant communication systems (MICS) frequency band of 402–405MHz. In contrast, UWB
technology has the potential to fulfil them all [3].
2.1.3 BLOOD PRESSURE MEASUREMENT
The use of radar techniques to measure BP may draw upon ideas from apexcardiography,
blood pressure pulse transit time (peripheral locations), as well as from ground penetrating
radar (GPR), yet is sufficiently different to merit a specific approach; in particular, the
complexity of geometry and stronger attenuation compared with early detection of breast
cancer and as for heart and respiration rate measurements, which are essentially based on
shallow reflections.
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Estimating BP using radar techniques is necessarily indirect; pressure only affects propagation
through the geometry unless it affects material EM properties, which is not expected.
2.1.4 WIRELESS BODY AREA NETWORK
The integration of on-body medical sensors, in-body sensors, and UWB radars into a single
network requires a carefully planned architecture in order to guarantee proper operation,
without mutual interference between the different devices. The Federal Communication
Commission (FCC) has allocated the 3.1–10.6 GHz frequency band for UWB communications
in the United States [7]. It is important to notice, however, that a large part of this spectrum is
strictly regulated in Europe and might not be available for new UWB radio systems. According
to European regulations [6], only the 6–8.5 GHz part of the spectrum is readily available for
UWB transmissions with a maximum EIRP spectral density of –41.3 dBm/MHz. In
Bangladesh, 3.1-10.6 GHz bandwidth is still unused, but a band from 2.6-4.4GHz is still in
consideration to be used for telecommunication. This has facilitate a viable communication
configuration of a wireless body area network (WBAN) using UWB radio interfaces in the 4.510.6GHz spectrum in Bangladesh.
Figure 2.2: WBAN topology for medical sensing
2.2 ARCHITECTURE OF UWB TRANSCEIVER FOR MEDICAL IMAGING
As discussed before, the UWB transceiver system must work within a specific band not to
overlap with other communication link. IR based UWB systems are easy to design and power
efficient. That is why, this type of transceiver are more favourable for medical imaging and
P age |9
sensing when it comes power constrain and device size. This section demonstrates a typical
architecture of the transceivers used for UWB microwave imaging and sensing.
2.2.1 ON/IN-BODY TRANSMITTER
Due to limitations at the in/on-body transmitter including power consumption, size, system
cost and complexity, its communication architecture must be as simple as possible (Figure 2.3).
Antenna
Pulse
Modulator
Data
Mapping
Tx Data
Figure 2.3: UWB Transmitter Block Diagram for microwave imaging.
A pulse generator provides the UWB pulse that is subsequently modulated and transmitted.
The shape of the transmitted pulse determines the signal bandwidth. Typical second and fifth
derivative of Gaussian pulses are designed to cover a bandwidth of approximately 1–5 GHz.
The power spectral density (PSD) of the transmitted pulse is shown in Figure 2.4.
-10
Average PSD
5th derivative
2nd derivative
-15
-20
Magnitude (dB)
-25
-30
-35
-40
-45
-50
-55
-60
1
1.5
2
2.5
3
3.5
4
Frequency (GHz)
4.5
5
5.5
6
Figure 2.4: PSD of the Transmitted Gaussian Pulse (5th Derivative & 2nd Derivative)[3].
The imaging is done by sending an UWB pulse to the portion of the body where the imaging is
done and the reflected pulse is received by on-body receiver for further processing. Based on
delay and tissue characteristics, the received pulses are processed for microwave imaging.
The generated data from the medical sensors can be processed for data compression and coding
or can be directly modulated without further processing thus simplifying the transmitter
architecture. The modulation is performed by changing the characteristics of the generated
pulse. The bi-phase pulse modulation scheme, in which the data bits are expressed by the
polarity of the transmitted pulses can be used for wireless communication. The transmitter
P a g e | 10
antenna must cover the entire frequency range with little pulse distortion. But, the design of a
compact UWB antenna for the on/in-body transmitter is a challenging task that has opened a
new field of research activities.
2.2.2 ON-BODY RECEIVER
The block diagram of the receiver is depicted in Figure 2.5.
Figure 2.5: UWB Receiver Block Diagram for (a) imaging and (b) for data communication
The UWB antenna at the receiver can be placed on the skin or at some distance away. The
practical implementation of the receiver antenna requires a special antenna structure since it
must cover a relatively wide body area for imaging. A less invasive receiver can be attained by
placing the antenna at some distance away the skin; in such case, extra free-space path loss is
added and the near-field coupling gain is lost. An alternative approach is to use multiple
receivers positioned at different strategic points for receiving scattered signals. However, a
high-gain antenna can be used to compensate for a portion of these losses.
The low noise amplifier (LNA) increases the power of the received pulses to a suitable level
for signal processing and to overcome noise in subsequent electronic stages.
Based on modes of activity, the receiver can have two variations as shown in fig 2.5. Figure
2.5(a) illustrates the simplest receiver architecture for microwave imaging where only received
signals are analyzed without any additional circuitry like demodulator or correlator. For data
communication, the common practice is to multiply the received signal with a predefined
template and integrate the signal for signal processing. The template generation interval is
dependent on data rate.
The design of the pre-defined template depends on the propagation channel characteristics. The
second derivative of a Gaussian pulse can approximate fairly well the radio channel PSD and
P a g e | 11
therefore can chosen as the pre-defined template. It is important to mention, however, that this
template pulse choice must include the effects of antenna on received signal [3].
The average bit-error-rate (BER) performance for different templates in an additive white
Gaussian noise environment is compared in Figure 2.6. The worst performance is observed
using the fifth derivative of the Gaussian pulse as template. The reduced BER performance
reveals significant distortion of the transmitted pulse while propagating through the body
tissues. The best BER performance among monocycle GMPs is obtained for the second
derivative, which collects more signal energy from the distorted pulses. Using the first and the
third derivatives provide almost similar BER performance. But bi-phase modulation with GMP
yield best performance depicted in the following figure. Only around 5dB SNR is needed for
BER <10-3
0
10
-1
10
Average BER
-2
10
-3
10
Gaussian
1st derivative
2nd derivative
3rd derivative
4th derivative
5th derivative
BPAM
-4
10
-5
10
0
2
4
6
8
10
SNR (dB)
12
14
16
18
Figure 2.6: Averaged BER performance using different templates
P a g e | 12
CHAPTER 03
CHARACTERIZATION OF IMPULSE RADIO BASED UWB
TRANSMITTER
3.1 UWB PULSES
The choice of pulse is a major issue of UWB based technique which has effect on the
performance of the device. Different types of UWB pulses are:
(i)Gaussian pulse
(ii)Gaussian mono-cycle pulse
(iii)Scholtz‘s Monocycle
(iv)Sine Monocycle
(v)Rectangle Monocycle
3.1.1 GAUSSIAN PULSE
A Gaussian pulse is shaped as a Gaussian function and is produced by a Gaussian filter. It
has the properties of maximum steepness of transition with no overshoot and minimum group
delay.
Figure 3.1: (a) Gaussian Pulse (b)PSD of Gaussian Pulse
G ( x)  
1
2 2
exp(
x2
2 2
)
where  = Standard Deviation
(3.1)
P a g e | 13
3.1.2 GAUSSIAN M ONOCYCLE PULSE
There are different types of Gaussian mono-cycle pulses. These pulses are derived from the
derivative of pulses. The figure below illustrates the 1 st derivative of Gaussian pulse or
negative Gaussian mono-cycle pulse. The equation describing a GMP is given by,
1
x2
Gm ( x)   2
x exp( 
)
2
 
2
(3.2)
Figure 3.2: (a) Gaussian Monocycle (b)PSD of Gaussian Monocycle
3.1.3 Scholtz Monocycle
This is the 2nd derivative of Gaussian Pulse. Its equation is given by,
G s ( x)  (
x2 2
2
 1) *
1
2 
exp( 
x2
2 2
)
Figure 3.3: (a) Scholtz‘s Monocycle (b) PSD of Scholtz‘s Monocycle
(3.3)
P a g e | 14
3.1.4 RECTANGLE M ONOCYCLE
Rectangular monocycle pulse has very low pulse width. Its equation is given by,
Gr  1, t1  t  t 2
(3.4)
= 0; Otherwise
(a)
(b)
Figure 3.4 (a) Rectangular Monocycle Pulse (b)PSD of a Rectangular Monocycle. As these
pulses have a dc value, the PSD of these pulses are spread near 0Hz
3.1.5 SINUSOIDAL M ONOCYCLE
by,
It is nothing but a sinusoidal Pulse with a very low time period. Its equation is given
GSin ( x)  A sin(2ft ); t1  t  t 2
= 0; Otherwise
Figure 3.5 (a) Sinusoidal Monocycle (b)PSD of Sinusoidal Monocycle
(3.5)
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Figure 3.6 (a)Square Pulse Train (b)Gaussian Pulse (c)First derivative Pulse (d)Second
derivative Pulse
3.1.6 PULSE TRAIN
One pulse by itself does not communicate a lot of information. Information or data needs to be
modulated onto a sequence of pulses called a pulse train. When pulses are sent at regular
intervals, which are sometimes called the pulse repetition rate or the duty cycle, the resulting
spectrum will contain peaks of power at certain frequencies. These frequencies are the inverse
of the pulse repetition rate.
(b)
(a)
Figure 3.7 (a) Time domain raised and pseudo raised cosine pulses and (b) their power
spectrum.
P a g e | 16
There are few modified sinusoidal pulses like time domain raised cosine and time domain
pseudo raised cosine pulses can be used for short distance UWB communication to ensure
effective use of the communication band. Figure 3.7(a) shows the raised cosine pulses and
figure 3.7(b) shows their PSD demonstrating spectral efficiency.
3.2 UWB M ODULATION TECHNIQUES
Communication can generally be defined as the transmission of information from a source to a
recipient. A basic communication system can be defined as,
Figure 3.8: Basic Communication system[1]
The three basic elements are as follows:
• The transmitter, whose primary task is to group the digital data stream into symbols, to map
these symbols onto an analog waveform, and then to transmit them to the air through an
antenna.
• The channel, which represents the effect of traveling through space, including reflections and
distortions as the electromagnetic pulses impinge on other objects.
• The receiver, which collects the electromagnetic energy from the antenna, takes the extremely
weak signal, reconstructs the pulse shape, and maps it to the appropriate symbols and then to
the binary bit stream.
By far the most common method of modulation in the literature is pulse position modulation
(PPM) where each pulse is delayed or sent in advance of a regular time scale. Thus, a binary
communication system can be established with a forward or backward shift in time. By
specifying specific time delays for each pulse, an M-array system can be created.
Another common method of modulation is to invert the pulse: that is, to create a pulse with
opposite phase. This is known as bi-phase modulation (BPM). In this thesis work, a type of
modulation technique was proposed to reduce modulation complexity. This modulation
technique is the combination of on-off keying and bi-phase modulation.
P a g e | 17
Figure 3.9: Comparison of pulse position modulation and bi-phase modulation methods for
UWB communication.(a)Unmodulated Pulses (b)PPM Modulated (c)Bi-phase Modulated signal
3.2.1 PULSE POSITION MODULATION (PPM)
The important parameter in pulse position modulation is the delay of the pulse. That is, by
defining a basis pulse with arbitrary shape p(t), we can modulate the data by the delay
parameter τi to create pulses si.
si  p(t   i )
(3.6)
Mathematically the data w(t) modulated by PPM can be expressed as,
Y (t )  
j 
j  
w(t  jT  xd j )
Where, w(t) is the pulse waveform to be modulated
(3.7)
T is the bit time,  is a fixed delay and d j is the binary data
Figure 3.9(b) illustrates PPM corresponding data bits. For ‗1‘ the GMP has been shifted to
right and for ‗0‘ shifted to right from their original position demonstrated in figure 3.8(a).
3.2.2 BINARY PHASE SHIFT KEYING (BPSK)
Bi-phase modulation can be defined as a kind of shape modulation. Since phase in a sinusoidal
communication system is associated with the delay of a sine wave.Bi-phase modulation is
easily understood as the inversion of a particular pulse shape; therefore, we take the following
equation
P a g e | 18
si   i p(t ),
 i  1, 1
(3.8)
To create a binary system based on inversion of the basis pulse p(t). The parameter σ is often
known as the pulse weight, but here we will refer to it as the shape parameter. For a binary
system the two resultant pulse shapes s1, s2 are defined simply as s1 = p(t) and s2 = −p(t).
Mathematically BPSK modulated wave can be expressed as,
Y (t ) 
j 
 w(t  jT )  (2d
j 
j
 1)
(3.10)
Where, w(t) is the pulse waveform to be modulated
T is the bit time,  is a fixed delay and d j is the binary data
Figure 3.9(c) illustrates PPM corresponding data bits. For ‗1‘ the GMP remains unchanged but,
for ‗0‘, GMP gets shifted by 1800 from their original position demonstrated in figure 3.9(a).
One of the reasons for the use of bi-phase modulation, especially in comparison with pulse
position modulation (which is a mono-phase technique) is the 3-dB gain in power efficiency.
This is simply a function of the type of modulation method. That is, bi-phase modulation is an
antipodal modulation method, whereas pulse position modulation, when separated by one pulse
width delay for each pulse position, is an orthogonal modulation method.
Another benefit of using BPM is that the mean of σ is zero. This has the important benefit of
removing the comb lines or spectral peaks. This, of course assumes that transmitted bits are
equally likely; however, this is a common and reasonable assumption in most digital
communication systems.
3.2.3 OTHER MODULATION TECHNIQUES
Although the previously discussed PPM and BPM constitute the major approaches to
modulation in UWB communication systems, other approaches have been proposed.
The Unconventional modulation techniques are illustrated below:
P a g e | 19
Figure 3.10: Comparison of other modulating techniques (a)Unmodulated Signal (b)Pulse
Amplitude Modulation(PAM) (c)On-Off Keying (d)Orthogonal Pulse Modulation(OPM)
3.2.3.1 PULSE AMPLITUDE M ODULATION (PAM)
Pulse amplitude modulation (PAM) for UWB can be represented as
si   i p(t ),
i  0
(3.11)
where the pulse shape parameter σ takes on positive values greater than zero.
As an example we can set σi = 1, 2 and obtain the binary pulse set s1 = p(t), s2 = 2p(t).
In general, amplitude modulation is not the preferred way for most shortrange communication.
The major reasons include the fact that, in general, an amplitude-modulated signal which has
smaller amplitude is more susceptible to noise interference than its larger amplitude
counterpart. Furthermore, more power is required to transmit the higher amplitude pulse.
In sinusoidal systems, amplitude-modulated systems are usually characterized by a relatively
low bandwidth requirement and power inefficiency in comparison with angle modulation
schemes. Thus, the major advantage (low bandwidth) can be seen to be anti-ethical to UWB,
and in most UWB applications power efficiency is of high importance.
P a g e | 20
3.2.3.2 ON OFF K EYING (OOK)
On-off keying (OOK) for UWB can be characterized as a type of pulse shape modulation
where the shape parameter s is either 0 or 1,can be represented as,
si   i p(t ),
 i  0,1
(3.12)
The ―
on‖ pulse is created when σi = 1 and the ―
off‖ pulse when σi = 0; thus, s1 = p(t) and s2 = 0.
The major difficulty of OOK is the presence of multipath, in which echoes of the original or
other pulses make it difficult to determine the absence of a pulse.On-off keying is also a binary
modulation method, similar to BPM, but it cannot be extended to an M-ary modulation
method.
3.2.3.3 ORTHOGONAL PULSE M ODULATION (OPM)
Of the three unconventional modulation techniques, orthogonal pulse modulation (OPM) is
simply a subset of general pulse shape modulation with the property that the pulse shapes are
orthogonal to each other. The advantage of using orthogonal pulses is not strictly related to the
modulation, but rather to the multiple access method.
In narrowband sinusoidal communication, orthogonal sine and cosine functions form the basis
for communication. In UWB we can design different pulse shapes that have the property of
being orthogonal to each other.
Unfortunately, a simple pulse shape parameter σ is inadequate to describe the set of pulses
which we may encounter, and here we simply label each pulse as a general p1, p2 . . . pi and
assume that pulses are designed so as to be orthogonal. It can be represented as,
si  { p1 , p2 ........ pi }
(3.13)
3.2.3.4 PROPOSED M ODULATION TECHNIQUE
The proposed modulation technique can be expressed by the equation:
Sj=[pj(t)-pj(t-dj)], pj=sin(2πt/T)*rect(t/T)
x(t )   jj 
 w(t  jd j ).S j
(3.14)
(3.15)
Where pj(t) is the Gaussian mono-cycle pulse, T is the time period for the GMP, w(t) is the
information bit and dj is the bit duration. This type of modulation is a combination of OOK
because it generates pulse on during pulse transition representing presence of an information
bit and BPSK because of pulse polarity. Figure 3.11 illustrates the proposed modulation
technique for UWB IR based communication.
P a g e | 21
Figure 3.11: The figure at left illustrates the OOK for IR based UWB communication. To
facilitate BPSK modulation with OOK, the figure on right depicts the proposed modulation
technique.
This type of modulation technique is helpful for data communication because it bears the
information of transition of pulses with two GMPs with different polarity. This could yield in
lower BER than OOK. This modulation can also reduce design complexity of the transmitter.
The bit error rate calculation for AWGN channel with one line of sight is demonstrated in
Appendix B.
3.3 CHARACTERISTICS PARAMETERS FOR UWB PULSES
According to FCC regulation, Ultra-Wideband (UWB) is a technology for transmitting
information spread over a large bandwidth (>500 MHz). That is why the pulses transmitted in
this system must have a very short duration. The UWB pulses should have an inherent noise
like behavior implying very low ERIP. For long distance communication (1m-100m), the
pulses must have higher peak-peak amplitude. For biomedical imaging, it is important that the
pulses must occupy a large bandwidth for better resolution
3.3.1 PULSE WIDTH
Figure 3.12: An ultra short UWB pulse.
In order to occupy a large bandwidth, the transmitted UWB pulses must have an ultra short
pulse width because of inverse relationship of pulse width to bandwidth. The following figure
P a g e | 22
shows the pulse width of a typical UWB pulse and its spectrum. The pulse width should be
around 200-300ps to occupy a large bandwidth.
3.3.2 POWER SPECTRAL DENSITY
The power spectral density of UWB systems is generally considered to be extremely low,
especially for communication applications. The power spectral density (PSD) is defined as:
PSD 
P
B
(3.16)
Where P is the power transmitted in watts (W),
B is the bandwidth of the signal in hertz (Hz),
and the unit of PSD is watts/hertz (W/Hz).
Since, we know that frequency and time are inversely proportional, sinusoidal systems have
narrow B and long time duration t. For a UWB system the pulses have a short t and very wide
bandwidth C.
Figure 3.13: Comparison Between Low And High PSD[1]
One of the benefits of low-power spectral density is a low probability of detection. This is a
concern for wireless consumer applications, where the security of data for corporations and
individuals using current wireless systems is considered to be insufficient.
Table 3.1 : PSD For Some wireless broadcast and telecommunication system[7]
System
Transmission
Bandwidth[Hz]
Power Spectral
Classification
Power(W)
density[W/MHz]
Radio
50kW
75KHz
666,600
Narrowband
2G Cellular
10mW
8.33kHz
1.2
Narrowband
802.11a
1W
20MHz
0.05
Wideband
UWB
1mW
7.5GHz
0.043
Ultra wideband
P a g e | 23
3.3.3 EQUIVALENT ISOTROPICALLY RADIATED POWER (EIRP)
EIRP the product of the power supplied to the antenna and the antenna gain in a given direction
relative to an isotropic antenna rules, EIRP refers to the highest signal strength measured in
any direction and at any frequency from the UWB device.
The Basic FCC Regulations which are proposed at the first set is given bellow:
• Emissions from supporting digital circuitry is considered separately from the UWB portion,
and is subject to existing regulations, not new UWB rules.
• The frequency of the highest radiated emission occurs, fM, must be within the UWB
bandwidth.
• Other emissions standards apply as cross-referenced in the UWB rules, such as conducted
emissions into AC power lines.
• Emissions below 960 MHz are limited to the levels required for unintentional radiators.
• Within a 50 MHz bandwidth centered on fM, peak emissions are limited to 0 dBm EIRP.
• UWB radar, imaging and medical system operation must be coordinated.
Dates and areas of operation must be reported, except in the case of emergency. These systems
also must have a manual switch (local or remote) to turn the equipment off within 10 seconds
of actuation.
The rules regarding measurement methods, such as determining the frequency range,
measurement bandwidth, type of detector, etc., are not included here. Discussion continues on
UWB measurement methodology and these first rules are likely to change.
P a g e | 24
Figure 3.14: UWB Spectrum as described by FCC(For GPRs, Wall Imaging, & Medical
Imaging Systems)[7]
Figure 3.15: UWB Spectrum as described by FCC (For Indoor Systems)[7]
P a g e | 25
Figure 3.16: UWB spectrum as described by FCC (For Hand held outdoor Systems)[7]
Figure 3.14 demonstrates the FCC regulation for UWC. From 3.1-10.6GHz bandwidth, any
carrier less UWB system must not have emission level >-41.3dBm. Human cells are sensitive
to higher radiation power. That is why for medical imaging, the emission level must be kept
low. If UWB system uses any carrier, it must be within the unlicensed band. Figure 3.15 and
3.16 illustrates the emission level for indoor and outdoor application of UWB systems.
P a g e | 26
CHAPTER 04
CHARACTERIZATION AND DESIGN PARAMETERS OF UWB FRONT
END RECEIVER
Due to design or model parasitic, designing high frequency VLSI circuit is quite challenging,
especially for analog circuits. That is why there are few microwave design parameters that
must be considered for evaluating a circuit‘s performance. These parameters are given below.
4.1 S CATTERING PARAMETERS(S-PARAMETERS)
Motivation for using S-Parameter is to resolve the difficulties of measuring lumped
parameters at high frequencies. CMOS transistors are prone to oscillate when they are opened
or shorted to measure lumped parameters. Moreover, lumped parameter representation may not
be suitable in some cases, for example in the case of transmission line (lumped ‗T‘ or ‗π‘
model are not suitable in microwave frequencies). For these reasons S-parameter
representation is adopted at high frequencies, whose variable is a travelling wave with
associated power with it. Scatter Parameters, also called S-parameters, belong to the group to
two port parameters used in the two port theory. Like the Y or Z parameters, they describe the
performance of the two ports. The difference of S parameters from Y or Z parameters is that
they relate to the traveling waves that are scattered or reflected when a network is inserted into
a transmission line of certain characteristic impedance , Zo.
S-parameters are important in microwave design because they are easier to measure at higher
frequencies than other types of two port parameters. They are conceptually simple, analytically
convenient and capable of providing detailed insight into measurement or modeling problems.
However, it must be kept in mind that like all other two port parameters S-parameters are linear
by default, that is to say they assume linear behavior from the two ports.
Two port S-Parameters has been evaluated below:
Figure 4.1: A two port network in terms of S-Parameters[4]
The following S-Parameters are characterized by the following equations:
Input matching parameter, S11=
𝑏1
𝑎1
; when 𝑎2 =0
P a g e | 27
Output matching parameter, S22=
Forward gain parameter, S21=
𝑏2
𝑎1
𝑏2
𝑎2
; when 𝑎1 =0
; when 𝑎2 =0
Reverse insertion parameter, S12=
𝑏1
𝑎2
; when 𝑎1 =0
In this case the relationship between the reflected, incident power waves and the S-parameter matrix is
given by:
b1   S 11 S 12  a1 
 
 
S
b2   21 S 22  a 2 
(4.1)
Where, a = Incident power on two port network and
b = Reflected power from the two port network
Scattering parameters or S-parameters are used in electrical engineering, electronics
engineering, and communication systems engineering, describing the electrical behavior of
linear electrical networks when undergoing various steady state stimuli by small signals. They
are members of a family of similar parameters used in electronics engineering, other examples
being: Y-parameters, Z-parameters, H-parameters, T-parameters or ABCD-parameters. They
differ from these, in the sense that S-parameters do not use open or short circuit conditions to
characterize a linear electrical network; instead matched loads are used. These terminations are
much easier to use at high signal frequencies than open-circuit and short-circuit terminations.
Moreover, the quantities are measured in terms of power. Many electrical properties of
networks or components may be expressed using S-parameters, such as gain, return loss,
voltage standing wave ratio (VSWR), reflection coefficient and amplifier stability. The term
'scattering' is more common to optical engineering than RF engineering, referring to the effect
observed when a plane electromagnetic wave is incident on an obstruction or passes across
dissimilar dielectric media. In the context of S-parameters, scattering refers to the way in
which the traveling currents and voltages in a transmission line are affected when they meet a
discontinuity caused by the insertion of a network into the transmission line. This is equivalent
to the wave meeting impedance which differs from the line's characteristic impedance.
Although applicable at any frequency, S-parameters are mostly used for networks operating at
radio frequency (RF) and microwave frequencies where signal power and energy
considerations are more easily quantified than currents and voltages. S-parameters change with
the measurement frequency so this must be included for any S-parameter measurements stated,
in addition to the characteristic impedance or system impedance. S-parameters are readily
represented in matrix form and obey the rules of matrix algebra.
P a g e | 28
4.2 INPUT IMPEDANCE MATCHING
The concept for impedance matching was developed from electrical power transfer theorem.
The maximum power transfer theorem yields that if the source impedance matches the load
impedance, maximum power will be transferred to the load. The antenna collects any
electromagnetic signal; maximum power will be transferred to the input of a receiver if the
source impedance of the antenna is equal to the input impedance of the receiver within the
operational frequency band. In most of the cases, for wireless communication, the input
impedance of any RF amplifier must be equal to 50Ω. Operating within a predefined band, any
impedance matching technique must also ensure minimum reactance at input of the receiver.
Figure 4.2 illustrates the input matching of an amplifier where Zs must be equal to Zin for
maximum power transfer.
Figure 4.2: Input impedance matching of antenna to an amplifier
Complex conjugate matching is used in cases in which the source and load are reactive. This
form of impedance matching can only maximize the power transfer between a reactive source
and a reactive load at a single frequency. In this case, Zload= Zsource* where, * indicates the
complex conjugate.
If the signals are kept within the narrow frequency range for which the matching network was
designed, reflections (in this narrow frequency band only) are also minimized. For the case of
purely resistive source and load impedances, all reactance terms are zero and the formula
above reduces to Zlaod= Zsource as would be expected.
4.3 VOLTAGE STANDING WAVE RATIO (VSWR)
The voltage standing wave ratio (VSWR) at a port is a measure of port matching to
return loss and is a scalar linear quantity quantified by the ratio of the standing wave maximum
voltage to the standing wave minimum voltage. It relates to the magnitude of the voltage
reflection coefficient (which is the magnitude of either S 11 for the input port or S22 for the
output port).
Input Port VSWR, Sin =
1+|𝑆11 |
1−|𝑆11 |
Output Port VSWR, Sout =
1+|𝑆22 |
1−|𝑆22 |
(4.2)
(4.3)
P a g e | 29
Measuring SWR is important for installing and tuning transmitting antennas. When a
transmitter is integrated with an antenna by a feed line, the impedance of the antenna and feed
line must match exactly for maximum energy transfer from the feed line to the antenna to be
possible. The antenna's natural resonance at the frequency being transmitted, the antenna's
height above the ground, and the size of the conductors used to construct the antenna vary the
impedance of an antenna. When an antenna and feed line do not have matching impedances,
some of the electrical energy cannot be transferred from the feed line to the antenna. Energy
not transferred to the antenna is reflected back towards the transmitter. It is the interaction of
these reflected waves with forward waves which causes standing wave patterns.
Reflected power has two main implications in radio transmitters: Radio Frequency (RF) energy
losses increase and damage to the transmitter can occur. Matching the impedance of the
antenna to the impedance of the feed line is typically done using an antenna tuner. The tuner
can be installed between the transmitter and the feed line, or between the feed line and the
antenna. Both installation methods will allow the transmitter to operate at a low SWR, however
if the tuner is installed at the transmitter, the feed line between the tuner and the antenna will
still operate with a high SWR, causing additional RF energy to be lost through the feed line.
4.4 VOLTAGE GAIN
The output voltage gain has to be maximized for proper operation. For this reason
forward gain (S21) has to be optimized as the gain is dependent on the forward gain.
4.5 POWER GAIN
The power gain of an electrical network is the ratio of the output power to the input
power. Three important parameters are defined in relation to the power gain of a circuit,
namely average power gain, transducer power gain and available power gain.
4.6 AVAILABLE POWER GAIN
The average power gain of a two-port network, GP, is defined as,
Gp 
Pload
Pin
(4.4)
Where Pload = the average power delivered to the load
Pin
= the average power entering the network
4.7 TRANSDUCER POWER GAIN
The transducer power gain of a two-port network, GT, is defined as:
GT 
Pload
Psource.max
(4.5)
P a g e | 30
Where Pload = the average power delivered to the load
Psource,max = the maximum average power available at the source
4.8 AVAILABLE POWER GAIN
The available power gain of a two-port network, GA, is defined as:
GA 
Pload,max
Psource.max
Where
(4.6)
Pload,max = the maximum average power available at the load
Psource,max = the maximum average power available at the source
4.9 NOISES
In physics and analog electronics, noise is a mostly unwanted random addition to a signal. The
noise level in a signal is characterized by SNR (signal to noise ratio) for analog standards,
which is defined as the ratio of the signal power to the noise power. Highest possible SNR is
always desirable for analog communication.[4]
There are several types of noises:
(i)Thermal Noise
(ii)Shot Noise
(iii)Flicker

THERMAL NOISE
Thermal noise is produced due to the randomness of electron motion and increases with rising
temperature. Thermal noise is also influenced by increasing resistance and the mean square
value of the noise voltage is given by:
V 2  4kTRf
(4.8)
Where, k = Boltzman Constant
T = Absolute Temperature
R = Resistance
f = Bandwidth (Hz)

SHOT NOISE
Shot noise is present in diodes and transistors due to the presence of junctions in these devices.
Shot noise is generated as a result of randomness of potential energy of the charge carriers
which cross the junction.
P a g e | 31
i 2  2qI DC B
(4.9)
where q, IDC and B are charge of electrons, DC current and the bandwidth in Hz respectively.

FLICKER NOISE
Flicker Noise is known as 1/f noise is inversely proportional to the frequency. Can be
expressed as,
i 2  KI DC

B
f
(4.10)
Where, K and α are constants which depends on the nature of the device
f and B are frequency and bandwidth in Hz respectively
Flicker noise is usually insignificant in higher frequencies.
Noise Figure (NF) : Noise figure can be defined as ratio between input SNR and
Output SNR,
NF 
SNRin
SNRout
(4.11)
Where, SNRin = Signal to Noise ratio at input
SNRout = Signal to Noise ratio at input
4.10 LINEARITY
RF devices are non-linear in operation. When a receiver receives a signal of wideband, the
amplifier will generate undesired inter-modulated spurious signal. Linearity of the amplifier
will decide the dynamic range of desired and spurious signal. If an amplifier is driven hard
enough the output power will begin to roll off resulting in a drop of gain known as gain
compression. The measurement of gain compression is given by the 1dB gain compression
point. The linearity of LNA is defined in terms of 2 parameters(a) Gain Compression
(b) 2nd and 3rd order Intercept Point (IP2 and IP3)
4.10.1 GAIN COMPRESSION
The output power of an amplifier typically exhibits a linear correspondence to the input
power as it changes the gain, i.e. the output power/input power quotient remains constant. If
the power of the input signal is raised, starting at a certain point the output power no longer
P a g e | 32
corresponds exactly to the input power. There is an increasing deviation, resulting an saturation
of output power. This is commonly known as gain compression.
The most common measurement of amplifier compression is the 1-dB compression point. This
is defined as the input power (or sometimes the output power) which results in a 1-dB decrease
in amplifier gain (relative to the amplifier's small-signal gain).The 1-dB compression point
parameter in also another measure of the linearity of a device. The figure 4.3 demonstrates 1dB compression point with respect to input power.
Figure 4.3: Variation of output power due to variation in input power.
4.10.2 2ND AND 3RD ORDER INTERCEPT POINT (IP2 AND IP3)
In RF communications, second and third-order intercept point (IP2 and IP3) are measure for
nonlinear systems and devices, for example receivers, linear amplifiers and mixers. It is based
on the idea that the device nonlinearity can be modeled using a low order polynomial, derived
by means of Taylor series expansion. The third-order intercept point relates nonlinear products
caused by the 3rd order term in the nonlinearity to the linearly amplified signal. The intercept
point is a purely mathematical concept, and does not correspond to a practically occurring
physical power level.
Figure 4.4 illustrates the behavior of the various inter-modulation products with input
amplitude. With the input and output amplitudes plotted on a log scale, the inter-modulation
product amplitudes follow straight line trajectories with slopes given by the order of the
products. By extrapolating, intercept points can be found that serve as figures of merit for the
linearity of the amplifier. These points can be referred to the input or output of the amplifier, as
desired. In a differential implementation, the second-order distortion is cancelled. Thus, in
practice, second-order intercept points are typically much higher than third-order intercept
points
P a g e | 33
Figure 4.4: Illustration of inter-modulation behavior[20]. Both curves are extended with
straight lines of slope 1 and n (3 for a 3rd order intercept point). The point where the curves
intersect is the intercept point. It can be read off from the input or output power axis, leading to
input or output intercept point, respectively
4.11 STABILITY
Stability, in referring to amplifiers, refers to an amplifier's immunity to causing spurious
oscillations. The oscillations can be full power, large-signal problems, or more subtle spectral
problems.
Unconditional stability refers to a network that can find any possible impedance on the Smith
chart from the center to the perimeter (up to Γ=1.0) at any phase angle. Γ< 1 means that the
real part of the impedance is positive. Note that any network can oscillate if it sees real
impedance that is negative, so if any system goes outside the normal Smith chart all bets on
stability are off.
If K-factor is greater than 1, it yields that the amplifier is unconditionally stable. Below is the
equation for K-factor (in two parts):
2
2
1  S11  S 22  
K
2 S11S 22
  S11S22  S12S22
2
(4.12)
(4.13)
Equation 4.12 and 4.13 indicates that the stability of a circuit will improve if the circuit is
properly matched to its input and it has better isolation between input and output.
Conditional stability refers to a network that is stable when its input and output find the
intended characteristic impedance Z0 (usually 50 ohms, sometimes 75 ohms), but if the
P a g e | 34
application presents a mismatch, there is a region of either source or load impedances that will
definitely cause it to oscillate.
P a g e | 35
CHAPTER 05
DESIGN OF A CMOS IMPULSE RADIO UWB TRANSMITTER
Transmission of pulses of ultra-short duration with a large fractional bandwidth is the typical
characteristic of Impulse Radio(IR) based Ultra wideband(UWB) systems. The challenge in the
circuit implementation are to achieve large bandwidth, low power consumption, smaller size
and less circuit complexity which are also important requirements for IR-UWB wireless
interconnects and battery powered sensor networks [19]. In this section, an IR-UWB
transmitter for microwave medical imaging and UWB communication between battery
powered sensor networks are proposed. The simulated results are also shown here to
demonstrate proposed circuits‘ functionality.
5.1 B ASIC THEORY
UWB signals can support high data rates due to the large bandwidth available and low power
due to the use of narrow pulses in time. There are presently two main competing technical
approaches to the development of UWB systems:


Multi-band (MB) OFDM UWB, and
Impulse radio (IR) UWC.
The MB-OFDM approach has been primarily used for applications such as streaming video
and wireless USB with data rates of 480Mb/s. Because of the high-performance electronics
required to operate a MB-OFDM UWB radio, these systems generally are not amenable to
energy-constrained applications. IR-UWB radios, however, can be designed with relatively
low-complexity and low power consumption. They have therefore found a niche in energyconstrained, short-range wireless applications including personal-area-networks, low-power
sensor networks, and wireless body-area-networks. Because of the bandwidths that can be
achieved with IR-UWB radios, they are also used in precise location systems and for dedicated
high-data-rate communication links [20]. So, we will focus on Impulse Radio-UWB (IR-UWB)
systems because of their suitability for many different applications, including sensor networks,
ad-hoc networks, cognitive radio, home networking [29]. Steps of designing IR-UWB systems
will also be discussed.
The basic block diagram for IR-UWB wireless communication with BPSK modulation is
shown in Figure 5.1:
Figure 5.1: Block Diagram for BPSK modulated IR-UWB Communication.
P a g e | 36
The main problem in this technique is that it requires an external clock, so it involves
additional complicacy of data and clock synchronization.
To design an UWB pulse generator, two most common schemes are:

Monocycle Pulse Generation with RLC network for oscillation for PPM and On-off
keying modulation.

Generation of Triangular Pulses and overlapping these pulses in different time to
produce BPSK, PPM and On-off Keying modulated
The first scheme uses a RLC network which oscillates when a Short Rectangular Pulse (SRP)
is feed at the gate of a source follower. Figure 5.2 shows the basic configuration of that circuit:
Figure 5.2: Source follower with RLC network
The main limitation for this type of circuit is that the pulse width is controlled by the time of
oscillation determined by the RLC network which does not generate pulses of sub-nano
second. Besides, the pulse amplitude is low for this kind of technique. But the main advantage
of this scheme is the implementation of this circuit is very easy. Generating triangular pulse
and overlapping them timely to produce a Gaussian mono-pulse is another technique to
produce UWB pulses of sub-nano second with higher peak-to-peak amplitude.
Generation of triangular pulses is complicated to implement in CMOS circuits. Due to nonlinear effects, a very complex topology is needed to be followed.
An alternative option for the fore mentioned schemes is to produce a pulse by comparing a
data bit with its delayed data bit to produce an ultra short pulse. This can be done by
implementing simple logic operations. The shape of the pulse can be controlled by controlling
rise and fall time of that pulse. Rise and fall time can be tuned by changing MOS transistor
width. This technique will not only avoid complexity in triangular pulse generation, but also
reduce the number of MOS transistors in the circuit. Figure 5.3 shows the shape of a triangular
pulse generated from rectangular pulse.
P a g e | 37
Figure 5.3: Data pulse(red), delayed inverted data pulse(blue) and ultra short pulse(green).
Figure 5.4: Input triangular pulses feed to the PMOS(red) and NMOS(blue) of the charge pump
stage and the output across load resistance(green).
These short pulses are delayed around ~100ps and feed to a charge pump stage consecutively
to get a Gaussian mono-pulse at output. This is similar to the distributed wave generation
technique [22]. A charge pump stage is composed of one NMOS and PMOS transistor and a
series network of capacitor and resistor as load. Two triangular pulses with opposite polarity
delayed by sub-nano second are feed to NMOS and PMOS and output is seen across a load
resistance. To filter out any DC component capacitor is introduced. The sequence by which
pulses are feed to the input of a charge pump and the derived output, a Gaussian mono-pulse
are shown in Figure 5.4. The operation and circuit design of a charge pump are explained later
in this chapter.
P a g e | 38
To adopt BPSK modulation, the sequence at which the pulses are feed is altered to get a
reverse Gaussian mono-pulse. Thus a BPSK modulated UWB pulse generator can be
implemented. This output can be integrated to a monopole antenna for wireless transmission
followed by wideband impedance matching. To convert the output voltage across the load to a
differential output, a single input differential output amplifier can be implemented. Differential
input is essential for dipole antenna.
5.2 DESIGN M ETHODOLOGY OF UWB TRANSMITTER CIRCUIT
As IR based UWB transceiver system is a carrier less transmission system, the proposed
technique greatly reduces the system complexity and the overall power consumption and
device size [33]. A combination of logic gates can be used to design IR based UWB transmitter
capable of generating ultra-short Gaussian Mono-cycle Pulses(GMP). Modulation can be
performed via simple logic operations. The proposed transmitter circuit can generate GMP
during the transition of data pulses from either ‗0‘ to ‗1‘ or ‗1‘ to ‗0‘. In this scheme, the pulse
generation will be dependent only on the transition of input data pulses, thus the pulse rate can
be varied from several Mbps to 8Gbps within FCC specified unlicensed band (3.1-10.6GHz).
For microwave imaging, best bit error rate (BER) performance was seen for 1 st and 2nd
derivative of GMP which yields collection of more signal power from distorted signals [3]. For
short range data communication, the efficient pulse modulation technique is needed to be
applied for low BER.
In this section, the architecture of the proposed IR based UWB pulse transmitter will be
discussed. The circuit is implemented and simulated in CADENCE and HSPICE.
5.2.1 UWB TRANSMITTER CIRCUIT FOR M EDICAL IMAGING
Figure 5.6 demonstrates a simplest circuit design for implementing IR based UWB transmitter
for microwave imaging. The proposed circuit can generate a GMP during the transition of data
pulse from high to low. Thus a pulse can be generated and transmitted within any pre-defined
time window for imaging. A data pulse is delayed and fed to a 2-input NOR gate with the data
pulse to generate an ultra short pulse (~100ps) during the transition from ‗1‘ to ‗0‘ of the data
pulse.
Delay blocks are used for delaying the pulses up to hundred of picoseconds. The rising time
and width of the pulse is dependent on the width of the logic gates. These ultra short pulses are
inverted, delayed and passed to the gate of the MOS transistors. These MOS transistors
switches on and off for a short time defined by the gate pulse width. During the switching, a
capacitor (CP) gets the charges pumped in from the supply and pumped out to the ground of the
circuit resulting in generation of a GMP. This capacitor also filters out the DC portion of the
pulse and the output is seen across the resistance (RL≈50Ω). This technique is also a power
efficient technique because it consumes power only during the switching and keeps itself idle
for rest of the time. Only logic gates were used in this design which also implies smaller
P a g e | 39
dimension of the device. Figure 5.6 depicts the wave shapes at different stages of the circuit
shown in figure 5.5.
VDD
Cp
(b)
Data
(d)
Delay Block
RL
(a)
Delay Block
(c)
Figure 5.5: The proposed IR based UWB transmitter for medical imaging
(a)
(b)
(c)
(d)
Figure 5.6: The wave shapes at different stages of the proposed circuit shown in figure 5.5.
P a g e | 40
5.2.2 UWB TRANSMITTER CIRCUIT FOR DATA COMMUNICATION
To facilitate data communication using the same topology described above, both transition of
high to low and low to high must be defined by the transmitted data pulse. As BPSK
modulation technique yields better BER performance, the transition of data pulses from high to
low can be defined by positive GMP and transition from low to high by negative GMP. As
mentioned in chapter 03, this sort of modulation will ensure lower BER than OOK. Figure 5.7
depicts the modified circuit for this technique. This circuit uses the same technique with an
extra circuit to generate negative GMP for data transition from low to high. This topology only
generates pulses during data transition and it does not require any external clock. So it
completely eliminates any additional circuitry for data synchronization with external clocks.
The pulses generated by IR UWB transmitter occupy a bandwidth of several gigahertz.
VDD
Data
(a)
(b)
(c)
Delay Block
VDD
Delay Block
(g)
Cp
RL
(f)
(e)
(d)
Figure 5.7: The proposed IR based UWB transmitter for data communication
Positive GMP can be produced by using same logic operation as described in previous section.
To generate negative GMP for pulse transition from low to high another logic combination was
used. A 2-input NAND gate was used for this purpose. NAND gate produced a short pulse
through logic operation using the data bit and inverted delayed data bit. The delay block was
used to delay and feed the pulses to the output. The short pulses cause the discharging of Cp by
triggering NMOS first, then charging by triggering PMOS. This result in an output pulse
shaped like negative GMP.
This type of pulse modulation technique is only dependent on gate size of the logic gates which
determine the delay. This makes this technique extremely easy to be implemented in any
CMOS technology. The pulse generation is independent of slew rate of input data bits because
the circuit entirely operates on logic basis. This flexibility also facilitates wide data rate
P a g e | 41
variation for data communication. Simulated results shown in chapter 07 will demonstrate the
proposed circuits‘ functionality.
(a)
(b)
(c)
(e)
(f)
(g)
Figure 5.8: The wave shapes at different stages of the proposed circuit for data communication
shown in figure 5.7
5.3 POWER DISSIPATION
The total power consumption in a CMOS inverter chain has three components. The dynamic
power dissipation arising from the charging and discharging of the capacitance CL at a
switching frequency of f Hz is given by
Pdynamic = fCLVDD
(5.1)
P a g e | 42
The short circuit power is consumed when both the transistors are on simultaneously with a
peak current Ipeak and if tSC represents the time when both the devices are conducting then this
power is given by
Psc =fVDDIpeaktSC.
(5.2)
Finally, the static power dissipated by the leakage current of the transistor Ileak is calculated
from
Pstat = VDDIleak
(5.3)
So the overall power dissipation for a single inverter can be written as:
Ptotal = Pdynamic + Psc + Pstat.
(5.4)
The calculations shown in equations are derived for a single inverter. For a cascaded inverter
the delay and the power losses would be approximately doubled.
The proposed circuit design is composed of logic gates, delay blocks composed with inverters
and charge pump stages. The delay blocks are CMOS inverters and power dissipation through
this MOS transistors are dominated by operating frequency or switching frequency. Higher
operating frequency will yield higher power consumption.
The logic gates and NOT gates draw power from the supply during switching or logic
evaluation. Other than that, these MOS transistors remain in cut-off or linear mode. Leakage
current is quite small in these sorts of operations making static power loss negligible.
The charge pump stage draws much current than logic combinational circuits. The short
circuit power consumption in this stage is quite high because, both NMOS and PMOS must be
kept on to ensure smooth transition of pulses from high to low value or low to high value.
5.4 PULSE WIDTH CHANGING BY WIDTH OF ACTIVE DEVICES
To develop an expression for propagation delay achieved by the delay block, we consider the
pair of inverters in delay block of figure 5.1. The overall delay of a single stage is equal to the
combined propagation delay of the two inverters. To calculate the delay produced by a single
inverter the saturation current of the transistors is considered as the average inverter current,
Iaverage=Isat=
=
n
2
n
2
(VGS  VTH ) 2
(VDD  VTH ) 2
Where, VTH
(5.5)
(5.6)
is the threshold voltage,  n is the transconductance gain, VGS is the gate to
source voltage of the transistor and VDD is the power supply voltage.
P a g e | 43
The NOR gate is designed larger rise time, tr and smaller fall time, tf. This can be designed by
changing the width of the MOS transistor. The following equation shows the delay,
3 𝐶𝐿
tr =
(5.7)
𝛽𝑛 𝑉𝐷𝐷
3𝐶𝐿
tf =
(5.8)
𝛽𝑝 𝑉𝐷𝐷
Here, CL is the output capacitive load. If we can increase decrease 𝛽𝑝 then tr will increase and
𝛽𝑛 increases then tf will decrease.
The propagation delay provided by the inverter can be derived using this current value as
shown in following equations,
𝐶𝐿
tpd=
(
𝐿𝑝
2 𝑉𝐷𝐷 𝐶𝑜𝑥 𝑊𝑝 𝜇 𝑝
= 1 (t pdLH  t pdHL )
2
+
𝐿𝑛
𝑊𝑛 𝜇 𝑛
)
(5.9)
(5.10)
Where, t pdLH and t pdHL are propagation delays for low to high and high to low transitions of the
output signal respectively, CL is the intermediate or load capacitance,  p and  n are the gain
factors for PMOS and NMOS transistors, Cox is the oxide capacitance per unit area with µm, W
and L denoting the mobility of carriers, the width of the transistors and the length of the
transistors respectively. A similar expression can be obtained for the rise/fall time of each
inverter stage
tr ( f ) 
3CL
VDD  n  p
(5.11)
The equations from 5.5 to 5.11 demonstrates that the delay is dependent on  p and  n are the
gain factors for PMOS and NMOS transistors, CL is the intermediate or load capacitance which
are dependent on MOS transistors dimension. As all the transistors used here are minimum
length transistors, the delay is entirely dependent on MOS width.
P a g e | 44
CHAPTER 06
DESIGN OF UWB FRONT END RECEIVER
6.1 LNA TOPOLOGY AND DESIGN CHALLENGES
As silicon technology is more matured and available for designing microwave low noise
amplifiers used in wireless communication systems, there are many documented highfrequency wideband amplifiers using silicon transistors in CMOS technology. Traditional
topologies such as classic shunt feedback amplifiers fails to provide wide band input matching
at high frequency whereas distributed amplifiers consumes more power. Basic common gate
topology and balanced amplifiers provide better input matching at the cost of degradation of
overall noise performance [21].
Owing to extremely low effective isotropically radiated power (-41.3dBm/MHz), UWB signals
has a noise like behavior. This implies the need of developing an efficient Low Noise
Amplifier (LNA) to suppress the unwanted noise and amplify the desired signal at the frontend of the receiver. In the band of interest, the LNA must show good input impedance
matching with antenna and output matching at the output to minimize reflection from both
ends. The forward gain must be high enough to amplify the received signal with minimal
power consumption. The noise figure of LNA must be low enough to ensure better sensitivity
of the circuit. The circuit must show linearity up to a certain received power at input. In
practical design, it is difficult to achieve the best of all the aforementioned parameters because
of the complex dependency of one parameter to other. Moreover, all the radio frequency
circuits are extremely sensitive to parasitic in the circuit which can deteriorate the circuit
performance. Among all the LNA design techniques and topologies developed, powerconstrained simultaneous noise and input matching (PCSNIM) technique is one of the best to
ensure optimized power, noise and matching for narrow band applications [34]. PCSNIM
technique can be utilized for UWB LNA design by employing a chebyshev filter at input [21].
In this section, circuit analysis for different part of LNA design is discussed. The different
LNA topologies are summarized with their corresponding circuit analysis results to
demonstrate performance dependency on circuit parameters.
6.2 CIRCUIT ANALYSIS
6.2.1 INPUT MATCHING
A 3rd order T section bandpass chebyshev filter is used to ensure minimum reactive part of the
input impedance throughout the band to yield nearly optimum noise figure. Figure 6.1
illustrates a 3rd order T-section chebyshev filter.
Figure 6.1: 3rd Order T-section Chebyshev Filter
P a g e | 45
In order to cascade the chebyshev filter with inductively degenerated MOS amplifier, the
inductor connected to the MOS was divided into two, gate inductance, Lg and source
inductance, Ls. The MOS transistor was biased by a diode connected MOS, Mbias through L2
and Lg shown in fig 6.2. The value of overdrive voltage is selected to match
𝑔𝑚 𝐿𝑠
(𝐶𝑝 + 𝐶𝑔𝑠 )≈Rs=50Ω. Capacitor Cp, is used to match required capacitance.
VDD
X
Lg
Cgd
RL
+
Zin
gV
m gs
Cgs C p
Vgs
Rb
LL
Rb
-
(b)
Ls
C buf
M3
C out
M2
V out
M4
C1
L1
VGM,4
Lg
X
M1
Cp
Rs
ID1
VDD
L2
C2
Vs
R bias
Ls
(a)
Mbias
Cbias
Figure 6.2: Simplified circuit for wideband LNA(a) and small signal model for M1 transistor(b)
6.2.2 NOISE ANALYSIS
In a cascade amplifier, if the voltage gain provided by two active devices M 1 and M2 are VGM1
and VGM2 respectively, then the overall noise figure of the LNA can be determined from
Ferries law-
NF  NFM1 
NFM 2  1
VGM1
.....
(6.1)
From equation (2), it can be realized that overall NF is determined by NFM1 and noise
contribution from each stage gets decreased by the factor related to the gain of previous stages.
For the simplicity of the calculation, all the capacitors and resistors are assumed lossless and
do not contribute to noise.
P a g e | 46
To analyze the noise contribution of CS stage ( NFM1 ) of the cascade structure, we have to
consider the MOS transistor noise model. The noise voltage, en can be expressed as sum of
correlated, ec and uncorrelated component, eu to noise current, in.
en=ec+eu
(6.2)
The following fig 6.3 illustrates the MOS transistor‘s noise sources with two noise current
sources. ind is the drain noise current due to carrier thermal agitation and ing is the induced gate
noise due to fluctuating channel charges at gate terminal.
Lg
X
2
en
+
Vgs
gmVgs
2
Zs
Zin
ig
- Cgs
2
id
Cp
+
Vgs
gmVgs
2
in
- Cgs
Cp
(b)
(a)
Ls
Ls
Figure 6.3: Noise model for MOS transistor, M1. (a) M1 noise sources (b) Input-referred
equivalent noise generator.
This two noise generators can be replaced by an input referred noise generators shown in Fig
6.3,
jCt
inm  ig 
id
(6.3)
gm
enm  j Lsig 
(1  j Ls Ct )
id
gm
(6.4)
2
2
The induced gate and drain noise current noise power spectral densities are i g ( ) and i d ( )
respectively, [20]
2
i g ( )  4kT  g g
(6.5)
2
i d ( )  4kT  g d 0
(6.6)
2 2
where, g g   C gs / 5gd 0 . In [21], considering the loading effect of local feedback inductor
(Ls), the noise figure equation was reduced to
F ( )  1 
R
1
 s
Gu Rs Rn
 2 g d 0 1  2 | c |   / 5   2 / 5
Gu 
.

 2 (1 | c |2 ) / 5
1
 2 gd 0
1
Rn 
. 2 2

 C t (1  2 | c |   / 5   2 / 5 )
1
(6.7)
(6.8)
(6.9)
where, δ and γ are process dependent parameters,gdo1 is channel conductance at zero drain to
source, c is the correlation coefficient, α=gm1/gd01 and Rs is the real part of source impedance.
From the equations, it can be realized that NF can be reduced if g m is increased whereas gm is
dependent on overdrive voltage, width and drain current. This yields that best NF will result in
more power consumption. Within limited power constraint, an optimum value for MOS
P a g e | 47
transistor width and overdrive voltage must be selected to ensure best overall noise
performance. All the classical theories also demonstrate that noise will be minimized is real
part of source impedance, Rs=Ropt, where, Ropt=√(𝑅𝑛 /𝐺𝑢 ). All the detailed calculations are
presented in Appendix A.
The NF of the cascode structure can be expected to be worsening because of noise contribution
from the different elements in the circuit. Noise contribution from cascode MOS transistor,
load resistance, parasitic resistance, biasing circuit and buffer amplifiers (if used) can result in
increase of noise figure. Furthermore, the limited quality factor of the inductors used in the
input filter introduces some noise contribution at input.
6.2.3 LINEARIZATION
VDD
Id
VsgC
M2
VgsB
M1
VGN
IsB IsC
M3
IsA
VgsA
Figure 6.4: Cascode amplifier with PMOS IIP3 booster
The non-linear behavior of MOS transistor originates from relationship between its voltage
and current. The drain current and gate to source voltage relationship of a MOS transistor can
be modeled using the power series expression:
id  gm1vgs  gm 2v 2 gs  gm3v3 gs  ...
(6.10)
Where, g mi is the ith-order derivative of the dc characteristics (I d-Vgs) and gm3, the 3rd
nonlinearity obtained by the 3rd order derivative of the dc characteristics. The expression of
IIP3 in terms of g mi :
IIP3  1.334
g m1
g m3
(6.11)
In the cascade amplifier shown in figure 1, 6 and 10, the nonlinear current generated in CS
MOS transistor is fully transferred to next stage if the following stage acts as current buffer. By
utilizing derivative superposition (DS) technique, which nullifies the negative 3 rd order
derivate of a MOS transistor dc characteristics by paralleling an auxiliary MOS transistor
biased near a weak inversion region with multiple gated transistor technique. In [26] and [27],
additional PMOS and NMOS respectively, were used to selectively absorb IMD3 current
component so that only fundamental current components can be delivered to the output. The
following analysis adopts the guideline showed in [26].
P a g e | 48
Figure 6.4 shows the implementation of DS technique where a PMOS is connected in parallel
with CG NMOS. In this technique, the width of the PMOS and the gate bias voltage are
selected to nullify g m3. The currents through each MOS can be written with the aid of equation
(6.10)
isB  g mB1vgsB  g mB 2v 2 gsB  g mB 3v3 gsB
(6.12)
isC  g mC1vgsC  g mC 2v 2 gsC  g mC 3v 3 gsC
(6.13)
and the relationship between vgsA and vgsB can be written into power series expression:
vgsA  K1vgsB  K 2v 2 gsB  K3v3 gsB
(6.14)
where, the value of K1 is negative. Equations (12-14) were written up to third order. Applying
KCL at point a and b, it can be written that id=isA. From equation (6.12-6.14), the expression
for small signal current passes through M1, isA is reduced to:
isA  ( gmA1  K1 gmC1 )vgsA  ( gmA2  K12 gmC1 )v2 gsA  ( gmA3  K13 gmC 3 )v3 gsA
(6.15)
and without any auxiliary PMOS, isA can be written as:
isA  gmA1vgsA  gmA2v2 gsA  gmA3v3 gsA
(6.16)
Equation (10), (15) and (16) implies that, by modulating the value of gmc3 by varying gate bias
voltage and width of the PMOS, IIP3 can be improved. This technique lakes any initiative to
minimize the 2nd order nonlinearity, but uses least numbers of devices. As the PMOS is biased
near weak inversion, the power consumption and noise contribution from this transistor can be
neglected.
6.2.4 OUTPUT MATCHING
To minimize the output reflection coefficient (S22<-10dB) within the band of interest, the
output impedance seen from the output must match the load at output. In figure 6.2, the output
is seen through capacitor and the output impedance matches the load most at resonance
frequency determined by the load inductance (LL) and output capacitance (Cout). A resistance in
series with load inductor is used to contribute to the real part of output impedance and increase
headroom voltage for output. When a buffer stage is used boost the response, the output
impedance can be varied by varying the drain current through the buffer. Fig 6.2(a) illustrates a
wideband LNA design with a the buffer stage. The output impedance seen from the source of
M3 can be varied by changing the gate voltage of M4. This technique provides the flexibility to
match the output impedance as well as gain in time of need.
P a g e | 49
6.3 LNA TOPOLOGIES
The major challenges in designing a wideband LNA are:
(i) Forward gain degradation (decreases in S21) which necessitates some techniques to
compensate the gain roll-off.
(ii) Lower S11 and S22 ensuring better matching at input and output.
(iii) Increase in |S12| which will reduce the forward gain and increase the possibility of
oscillation and instability
(iv) NF degradation at high frequencies.
(v) Lower power consumption
To address these challenges in the design of a wideband LNA, several topologies and circuit
techniques have been proposed in the literature.
6.3.1 CLASSICAL NOISE MATCHING TECHNIQUE (CNM)
Figure 6.5: Schematic of an LNA topology in cascode structure(a) and simplified small signal
model (b) of the LNA for calculating input referred noise.
In this technique, the LNA is designed for minimum noise figure F min by presenting the
optimum noise impedance Zopt to the given amplifier, which is typically implemented by
P a g e | 50
adding a matching circuit between the source and the input of the amplifier. By using this
technique, the LNA can be designed to achieve NF equal to Fmin of transistor, the lowest NF
that can be obtained with given technology. However, due to the inherent mismatch between
Zopt and Zin* (where Zin* is the complex conjugate of the amplifier input impedance), the
amplifier can experience a significant gain mismatch at the input. Therefore, the CNM
technique typically requires compromise between the gain and noise performance.
Figure 6.5-(a) shows a CS-CG cascode LNA topology, which is one of the most popular
topology due to its wide bandwidth, high gain, and high reverse isolation. In the given
example, the selection of the cascode topology simplifies the analysis, and the gate-drain
capacitance can be neglected. Fig. 6.5-(b) shows the simplified small-signal equivalent circuit
of the cascode amplifier for the noise analysis including the intrinsic transistor noise model. In
Fig. 6.5-(b), the effects of the common-gate transistor M2 on the noise and frequency response
are neglected because the noise calculations is done only for input referred noise demonstrated
in section 6.2.2.
The noise calculation for the small signal model shown in fig 6.5(b) can be expressed as:
Rn0 
 1
 gm
(6.17)
0
 Cgs
Yopt
Fmin  1 


2
2
(1  c )  jCgs (1   c
)
5
5
2 
5 T
2
 (1  c )
(6.18)
(6.19)
Where, Rn represents the noise resistance, the optimum noise admittance, Y opt, the minimum
noise factor, Fmin. In Eq. (7) the cutoff frequency ωT is equal to gm/Cgs, and α ≡ gm/gdo is unity
for long-channel devices and decreases as channel length scales down. The noise is calculated
modeling two noise sources for thermal and flicker noise [34].
6.3.2 BALANCED AMPLIFIER
Balanced amplifier consists of two amplifiers in parallel and two 3dB Lange or hybrid
couplers. The basic operation is as follows:
P a g e | 51
Figure 6.6: Block diagram of balanced amplifier
The input signal is split into two quadrature components (equal but with a 90ophase shift) by
the input hybrid coupler. The two quadrature signals are then amplified using two identical
LNAs. The output coupler combines the output signals of the two amplifiers by introducing an
additional 90o phase shift, thus bringing them in phase again.
The advantage of this architecture is that it possesses a very good matching at the input and
output ports and continues to operate even if one of the amplifiers fails to function. However,
this architecture suffers from the increased power consumption of two amplifiers, increased
circuit size, and the bandwidth reduction caused by the couplers.
6.3.3 DISTRIBUTED AMPLIFIER
Distributed amplifiers employ a topology in which the gain stages are connected such that their
capacitances are separated, yet the output currents still combine in an additive fashion. Seriesinductive elements are used to separate capacitances at the inputs and outputs of adjacent gain
stages. The resulting topology, given by the inter lying series inductors and shunt capacitances,
forms what is essentially a lumped-parameter artificial transmission line. The additive nature of
the gain dictates a relatively low gain; however, the distributed nature of the capacitance allows
the amplifier to achieve very wide bandwidths[6]. The advantage of this architecture comes
from the fact that the input capacitances of these amplifiers are distributed in an LC network
which allows for the realization of amplifiers with large bandwidths. In fact, the series
inductive elements and capacitances of MOS devices form an artificial transmission line,
which allows the flow of the signal to the end of the gate line. The signal fed to the gate of the
MOS device is transferred to the drain line through the trans-conductance (gm) of the device. If
the phase velocity on the gate and drain lines are identical, then the signals at the output add in
the forward direction as they arrive at the output. Many wideband LNAs in CMOS have been
P a g e | 52
Figure 6.7: Basic Distributed Amplifier[6]
6.3.4 NEGATIVE FEEDBACK WIDEBAND LNA
The classical approach to satisfy the required impedance matching at the input of a wideband
LNA is to employ negative feedback. This technique will provide a flat gain and a very small
VSWR at the input and output ports, and also it reduces the sensitivity of the circuit to the
MOS device parameters. However, as discussed in the analysis of shunt feedback amplifier, the
feedback circuitry may increase the minimum NF and reduce the maximum achievable gain.
Different topologies have been proposed. One of the most popular variations of negative
amplifier is the shunt-series amplifier, symbolically shown in Fig. 6.8.1
Figure 6.8.1: Shunt-series amplifier [20]
Main problem associated with this circuit is that resistive negative feedback network continues
to contribute to the thermal noise and fails to maintain input impedance that is equal to
optimum noise impedance, Zopt. As a consequence, the overall noise performance of the circuit
degrades. Nonetheless, the broadband capability of this circuit is fairly enough to compensate
for higher noise figure.
P a g e | 53
As resistance is considered as a source for thermal noise which contributes to higher noise
figure, noiseless element like capacitor can be used instead of resistance.
6.3.5 SIMULTANEOUS NOISE AND INPUT MATCHING (SNIM) TECHNIQUE
As feedback techniques are often adopted in designing low-noise amplifiers in order to shift
the optimum noise impedance Zopt to the desired point, series feedback has been preferred to
obtain simultaneous noise and input matching without the degradation of NF. Especially, the
series feedback with inductive source degeneration, which is applied to the common-source or
cascode topology, is widely used for narrow band applications. Fig 6.8 illustrates the topology.
Figure 6.8.2: Schematic for CS-CG cascode LNA topology in SNIM technique (a) and small
signal model of the structure (b)[26]
Assuming that the inductors are lossless, the noise figure can be written as,
1
0
0
Z opt
 1/ Yopt



 
Cgs  

j
(1


c
)
2

5 
5

(1

c
)


 1
Rn  Rn0 
 gm
(6.20)
(6 .21)
P a g e | 54
Fmin  1 
2 
5 T
2
 (1  c )
0
Zopt  Zopt
 sLS
(6.22)
(6.23)
Whereas the input impedance can be written as,
Zin  sLS 
g L
1
 m S
sCgs
Cgs
(6.24)
In this topology, noise and power consumption can be minimized by propoer selection of
transconductance of the common source MOS transistor. The input impedance can also be
matched to Zopt and source impedance by changing value of transconductance and MOS width.
P a g e | 55
CHAPTER 07
SIMULATION RESULTS AND PERFORMANCE ANALYSIS
The proposed transmitter and front end receiver circuit are designed in IBM 130nm CMOS RF
technology and simulated in Cadence Spectre and HSPICE RF. In this chapter, the design for
proposed transmitter and receiver are discussed and simulated results are demonstrated to
prove circuits‘ functionality.
7.1 PROPOSED UWB TRANSMITTER
As discussed in chapter 05, the proposed IR based UWB transmitter is logic and delay based
design. The ultra short pulses are generated by delay and logic operations. Variable delay
elements have widespread applications in high speed very large scale integrated (VLSI)
circuits. Their use includes applications like voltage controlled oscillators (VCOs), digital
delay locked loops (DLLs), pulse width control loops (PWCLs), phase locked loops (PLLs),
time to digital converters (TDCs). But the difficulty arises when an engineer tries to design a
wideband delay element suitable for UWB schemes. TR-UWB systems use ultra-short pulses
(in the range of hundreds of picoseconds) to transmit information so that average power
density remains very low.
The propagation delay of a logic operation depends on the width of the logic gates and load
capacitance. The transistors used for delay operation have the aspect ratio (W/L) of 20 and the
delay provided by each NOT gate was calculated to be ~30ps. This information is utilized
during the circuit design.
In order to design an IR-based UWB transmitter for medical imaging, two important issues
were taken into consideration: the transmitter must be capable of transmitting pulse with in any
preferred time window and the pulse width of the transmitted pulses must be low enough to
cover a wide spectral range for better contrast in microwave imaging. In chapter 2, the property
of the UWB pulses were discussed for microwave imaging. The proposed design is capable of
generating UWB pulses of width of ~180ps. The PSD of this pulse illustrates its spectral
coverage from 5-10GHz within FCC unlicensed band.
Figure 5.6 and 5.7 illustrate the proposed transmitter designs. These designs consist of three
major components:
1. Delay Block
2. Logic Gates
2. Charge pump stage
7.1.1 DESIGN CONCERNS FOR THE DELAY B LOCK:
A major implementation concern of an IR-based UWB receiver is that it requires a wideband
delay element which would be able to handle ultra short UWB pulses. But delay elements for
P a g e | 56
UWB pulses suitable to be used in IR-based UWB transceivers are rarely reported because of
numerous design issues. If dual (positive and negative) power supply voltages are made
available for a basic inverter based delay element, which can result in displacement in the time
domain for input UWB pulses. There is also a complexity regarding dual power supply. The
proposed circuit uses a simple logic based delay element that does not need any dual power
supply and avoids all the complexity. The result will illustrate the amount of delay can be
controlled by varying four factors, MOS transistors width, length, number of stages used and
power supply.
The delay blocks are composed of NOT gates. As mention earlier, because of the large aspect
ratio, any pulse travelling through these gates are subjected to some propagation delay. The
propagation delay can be written by the following equation:
tpd=
𝐶𝐿
(
𝐿𝑝
2 𝑉𝐷𝐷 𝐶𝑜𝑥 𝑊𝑝 𝜇 𝑝
+
𝐿𝑛
𝑊𝑛 𝜇 𝑛
)
(7.1)
VDD
1.2
0
t1
t2
M2
M4
M1
M3
t1
t2
1.2
t 1+D t 2+D
0
0
M6
M5
VDD
1.2
0
1.2
1.2
M2
M4
M1
M3
t 1+D t 2+D
1.2
0
M6
0
M5
Figure 7.1: Delay Block used in transmitter design
where, the propagation delay is dependent on four factors mentioned above. If there is no fan
out effect, CL is the gate capacitance seen from the output of the not gate. All the transistors
used here are minimum length transistor, Cox is process and technology dependent. So, only
governing factor for delay is the width. Three NOT gates resulted in a delay of ~100ps, which
was enough to generate a short pulse through logic operation. Third not gate was used for both
delay and inversion of the pulse. Figure 7.1 shows the delay element used in the circuit: The
P a g e | 57
data bits used for input to the delay block and the output of the delay block are shown in fig
7.2. The figure also illustrates the function of delay blocks used before charge pump stages.
(a)
(b)
Figure 7.2 Data bits and delayed data bits(a), triangular and delayed triangular pulse (b) by
delay blocks used in the proposed circuit.
The delayed pulses and input pulses are fed to a NOR gate. NOR gate produces a ‘high‘ output
when it gets low as both of its input. The inverted delayed pulse and data pulse become zero
during the transition of data bit from ‗high‘ to ‗low‘. During this transition, the output of the
NOR gate becomes high and remains high as long as both inputs are low. The time duration of
the output pulse to be remained ‗high‘ is determined by the delay time provided by the delay
blocks. Fig 5.2 shows the input and output of the NOR gate. The NOR gate was designed to
provide equal rise and fall time, so that the output pulses become triangular in shape. Figure
7.3 depicts the output of NOR gate used in the circuit.
P a g e | 58
NOR gate Outpiut
Input Data
Delayed and Inverted Data
Figure 7.3 The input data bit, delayed data bits to NOR gate and output of the NOR gate. The
output pulse is generated during transition of input data bit from ‗1‘ to ‗0‘
7.1.2 DESIGN CONCERNS FOR THE CHARGE PUMP STAGE:
The output pulse of the NOR gates are now delayed and fed to the charge pump stage. The
main function of the delay block here is to introduce some delay between switching PMOS and
NMOS in the charge pump stage. The delay blocks were designed here with same aspect ratio
used in previous delay blocks. To generate a positive GMP, the PMOS is turned on first. In
order to get a sine like pulse, the inverted delayed pulse is used to switch on NMOS. Timing of
pulses is crucial here as the shape, amplitude and pulse width are dependent on pulse arrival.
VDD
Cp
Delay Block
Figure 7.4: Charge pump stage of the proposed circuit.
The charging PMOS and discharging NMOS are minimum length transistors. Their aspect ratio
is higher (~50) in order to increase conductance of the transistor channels. As mobility of
electrons is higher than holes, the PMOS has larger width than NMOS. Figure 8.4 shows the
charge pump stage of the circuit.
P a g e | 59
The purpose of the output capacitor (Cp) is to store the charge during charge pumping through
PMOS from supply and pump out the charge through NMOS to ground. The capacitor also
filters out the DC portion.
Cp =.1pF
Settling time
Cp =.5pF
Settling time
C p =2.5pF
Settling time
Figure 7.5: The effect of output capacitor on the output pulse settlement after switching ‗on‘
the circuit. Larger the value of the capacitor result in larger the pulse amplitude with cost of
larger settling time.
The value of Cp has effects on the output pulses transient response and pulse amplitude. Higher
value of Cp will result in higher pulse amplitude but it would take much time for settling the
baseline for pulse oscillation because of RC time constant where R=load resistance(50Ω). The
value of Cp was selected as ~0.5pF. Figure 7.5 illustrates the effect of Cp on output.
7.1.3 DESIGN CONCERNS FOR THE DATA COMMUNICATION :
In wireless body area network, it is common to design a system where sensors intend to
communicate between each other. Different communication technique has been
proposed[melody]. But in all cases, different band was used for low speed and high speed
communication. This would extend the complexities in design and different LNAs must be
used for different types of communication.
To reduce design complexity, the proposed transmitter can be used for data communication.
But the proposed design only generates pulses during transition of information bits. This
implies that the modulation technique would become OOK. OOK has low BER for short
distance communication but suffers from high BER for long distance communication. BPSK
P a g e | 60
modulation ensures better BER for long distance communication. Recent published works on
IR d transmitted reference based transmitter for radar communication show a new way for long
distance communication.
For short and long distance communication, the new circuit in figure 5.7 in chapter 5 has been
proposed. The function of this transmitter is to generate a negative GMP and positive GMP
during transition of information bits from ‗low to high‘ and ‗high to low‘ respectively. This
way, the modulation can be translated as transmitted reference based BPSK modulated
communication. This modulation technique will yield less BER than OOK. Apart from that,
both high and low speed communication can be done by this type of transmitter. In figure 5.7
demonstrates the proposed circuits‘ architecture. In CMOS 130nm RF technology, both low
threshold (LVT) MOS and nominal threshold (FET) MOS can be used for this design. LVT
MOS is faster than FET.
From the proposed circuit utilizes the same principle described for medical imaging transmitter
design. Two logic combinational circuits are used for generating two pulses during positive
and negative transition of the information bit respectively. The positive GMP is generated
using same technique described before. The negative GMP is generated applying the same
principle, where the pulse are delayed and fed to the charge pump stage one after another.
The delay block used here in before logic gates are different than the blocks used before charge
pump stage because of fan out effect. That is why the aspect ratio ~22 to reduce the
propagation delay of the delay blocks.
The MOS transistors used in charge pump stage have higher aspect ratio (~100) yielding
higher peak-peak amplitude of the pulses. The output capacitor Cp (~2.5pF) ensured a balance
between settling time and pulse amplitude. All the MOS are used in the design have minimum
length.
7.2 SIMULATED RESULTS OF THE PROPOSED TRANSMITTER
All simulation results were done in Cadence Spectre and HSPICE RF using IBM 130nm
CMOS RF technology library files.
7.2.1 PULSE WIDTH AND PULSE AMPLITUDE
The output of the proposed model are simulated and seen across a 50Ω load resistance. The
output is a single ended output. The proposed model for microwave imaging produces a
positive GMP when supplied with a data bit. Figure 7.6 shows the output of the circuit. The
peak to peak amplitude of the GMP is 320mV and pulse width ~125ps. For data
communication, as discussed before, another circuit model is proposed and the simulated
output is demonstrated in figure 7.7.1. The average pulse width of negative and positive GMP
is ~183ps. The average peak to peak voltage is 390mV. The pulse shape of positive and
negative GMP is not identical, because two different blocks were used for generating the forementioned pulses.
P a g e | 61
Figure 7.6: Shape of the output pulse at proper scale for the proposed circuit of medial
microwave imaging.
Figure 7.7.1: Shape of the output pulse at proper scale for the proposed circuit of medical
microwave imaging and data communication.
7.2.2 POWER SPECTRAL DENSITY
The power spectral density of the pulse shown in fig 7.6 is shown in figure 7.7.2. The PSD
illustrates that the -3dB bandwidth of the GMP‘s are within 5-10GHz with center frequency
near 7.5GHz for series of UWB pulses. This has motivated to design a UWB front end receiver
operating from 5-10GHz.
-3dB marker
(a)
(b)
1G
5G
10G
15G
Figure 7.7.2: PSD of the (a) series of UWB pulses and (b) single UWB pulse for microwave
imaging
P a g e | 62
The proposed circuit for data communication and imaging produces two GMP pulses shown in
figure 7.7.1, hence shifting and spreading the spectrum shown in fig: 7.8. The center frequency
remained un-shifted and the -3dB band falls from 5GHz to 10.6GHz.
-3dB Marker
-3dB Marker
(a)
5G
0
10G
(b)
15G
5G
10G
15G
Figure 7.8: PSD of (a) series of UWB pulse and (b) two UWB GMPs for high speed data
communication and microwave imaging
7.2.3 POWER CONSUMPTION
The power consumption of the proposed circuit is dominated by two factors: short circuit
current flowing through the charge pumping MOS at the output and the rate of the data bit at
input. That is why the power consumption is expressed in terms of bit rate. The proposed
circuit in fig 5.6 consumes 0.38pJ/pulse with 1.2V supply found from the simulation. The
charging and discharging MOS transistors draw around ~4.5mA drain current during switching
operation.
As the modulation system for data communication proposed in chapter 05, two pulses are
needed to imply smallest information bit duration, so the power consumption has increased per
bit as. The power consumption is 0.68pJ/bit.
The power that will be delivered from transmitter circuit to
0.824mA*61.66mV≈51µW @ 1Gbps data rate which is equivalent to -13dBm.
antenna
is
7.2.4 PULSE RATE
As the pulses are only generated during transition of information bits, there is a flexibility of
data transmission rate. For microwave imaging, it is important to transmit pulses after predefined data frame. The proposed circuit shown in fig 5.6 was simulated with different data
rate in the results are shown in fig 7.9 and fig 7.10
The proposed circuit for data communication produces two GMPs, one at the start of data bit
and another at the end of the data bit. So highest bit rate that the circuit can utilize to produce
GMPs is ~4GHz. This circuit was simulated with lower data bit at input shown in fig: 7.11.
P a g e | 63
Figure 7.9: Pulse generation @5Gpulse per second. The data type at the input of this circuit is
NRZ data.
Figure 7.10: Pulse generation @100Mpulse per second. The data type at the input of this
circuit is NRZ data.
Figure 7.11: Pulse generation @4Gpulse per second. The data type at the input of this circuit is
NRZ data.
Monte Carlo simulation was performed to inspect the changes in pulse shape of the transmitter.
For medical imaging, the proposed circuit showed ~40ps variation in pulse width and ~50mV
P a g e | 64
Figure 7.12: Pulse generation @200Mpulse per second. The data type at the input of this
circuit is NRZ data.
variation in peak amplitude for 30 random process variations(Figure 7.12.1(a)). On the other
hand, the proposed transmitter for data communication showed ~35ps variation in pulse width
and ~40mV variation in peak amplitude (Figure 7.12.1(b)). Simulated results are depicted in
figure 7.12.1.
(b)
(a)
Figure 7.12.1: Monte Carlo Simulation results for proposed circuits
In table 7.1, the power consumption for different process corners are summarized. SS process
corner has shown higher power consumption due to higher short circuit current current
consumption.
Table 7.1: Power Consumption at different Process Corner
Corner
TT
FF
SS
FS
SF
Power @ 5Gbps
0.32455pJ/pulse
0.2978pJ/pulse
0.3442pJ/pulse
0.3422pJ/pulse
0.3044pJ/pulse
For low-cost systems that use energy detectors, an important criteria is the energy by pulses
relative to the pulse magnitude, η=
energy efficient design.
𝐸𝑛𝑒𝑟𝑔𝑦 𝑝𝑒𝑟 𝑝𝑢𝑙𝑠𝑒 (𝑝𝐽 )
. Lower value of η implies more
𝑝𝑒𝑎𝑘 −𝑝𝑒𝑎𝑘 𝑎𝑚𝑝𝑙𝑖𝑡𝑢𝑑𝑒 (𝑉)
P a g e | 65
7.3 PROPOSED UWB FRONT END RECEIVER
The proposed front-end receiver is composed of two cascade structure, a bandpass filter
cascaded with an LNA. An LC ladder based filter is used to obtain flat band characteristics
throughout the band of interest. The proposed LNA has a common source- common gate (CSCG) cascode structure with a second buffer stage for output impedance matching and improved
reverse isolation. Figure 6.2 illustrates the proposed design. The LNA is designed keeping the
following concerns in mind:
1. Filter type and response
2. Noise minimization
3. Linearity improvement
4. Improved reverse isolation
5. Lower power consumption
7.3.1 DESIGN CONCERNS FOR FILTER DESIGN AND CASCADE
In order to design a bandpass filter to operate within certain frequency, there are two obvious
choices: chebyshev filter and butterworth filter. A third order butterworth filter uses less
number of inductors and capacitors but unable to ensure a flat gain over the operational band.
On the other hand, 3rd order T-section chebyshev filter uses more inductors and capacitors than
butterworth filters but ensures flat gain over the operational band. This has worked as the
motivation for selecting chebyshev filter for filter designing. Although the design will consume
more die area, but as long as the transceiver will be used outside the body for imaging, the
device size constrain can be relaxed. Figure 7.13 illustrates the performance of butterworth and
chebyshev filter over the operational band whereas table 7.2 shows the values of inductors and
their corresponding dimensions and peak quality factors used in the circuit. The values of
resistance, inductors, capacitor and MOS width are summerized in table 7.3.
Id
Chebyshev Filter
Response
Butterworth Filter
Response
Figure 7.13: Comparison between the response of a chebyshev and butterworth filter
P a g e | 66
Inductor
L1
L2
Lg
Ls
LL
Table 7.2 Inductors used in the proposed LNA
Outer
Metal
Metal
Turns
Peak Q @
Dimension
Spacing
Width
GHz
220u
5u
11u
2
12.08
230u
5u
10u
1.25
17.047
250u
5u
10u
2
10.43
170u
5u
10u
1.25
22
300u
5u
15u
2.25
8.82
Inductance
1.264n
0.708n
1.594n
0.466n
2.082n
The chebyshev filter is cascaded with the inductively degenerated LNA to achieve the required
input impedance matching shown in fig 6.2. Considering the small signal model of an
inductively degenerated LNA, the input impedance of the amplifier shown in figure 7.14:
Lg
+
Zin
Vgs
gmVgs
C gs
CP
Ls
Figure 7.14: Input matching network
Z in ( s) 
g m Ls
1
 s( Ls  Lg ) 
s(Cgs  C p )
(Cgs  C p )

1
 s( Ls  Lg )  T Ls
s(Cgs  C p )
(7.1)
where, Cgs is gate-source capacitance the real part of Zin(s), ωTLs must be equal to the real part
of source impedance Rs, and T  gm / (Cgs  C p ) . To match the chebyshev filter requirements,
the values of C p  C1  Cgs and L1  Lg  Ls . The source-body capacitance is shorted out by
shorting source and body. The grate-drain capacitance can introduce some undesirable effect
which introduces multiple resonances with respect to equation (7.1) and the expected series
resonance gets shifted and determined by the resonance between Lg and Ct / (1   2 LsCt )  Cgd
[21].
Devices
M1
M2
M3
M4
Mbias
Table 7.3: List of device sizes used in the design
Value
Model
(W/L~1250)
nfet
(W/L~500)
nfet
(W/L~400)
nfet
(W/L~140)
nfet
(W/L~0.666)
nfet
Metal Layers
N/A
N/A
N/A
N/A
N/A
P a g e | 67
MBias1
L1, L2, Lg,
Ls, LL
C1,C2,Cp,Cbias
Rbias, RL, Rbias1
sub
(W/L~0.666)
1.264n, 0.708n, 1.594n,
0.466n, 2.082n
0.254p. 0.134p,0.065p, 2p
21.5K, 60, 10K
W=35u l=20u
nfet
Indp, single layer
Mimcap, vncap
Oppres, Silres
N/A
N/A
Metallization
MAIIE1
N/A
N/A
N/A
7.3.2 DESIGN CONCERNS AGAINST NOISE
The noise performance is related to the input impedance, Zin, source impedance, Zs and
optimum impedance, Zopt . To obtain the wideband noise and input matching, the source
impedance seen from the gate of the input transistor should be the complex conjugate of the
input impedance, Zin to deliver the maximum power and at the same time be equal to Zopt to
achieve NFmin. Thus the following four conditions should hold over the entire frequency band
of interest:
Re Z opt   Re Z s 
Im Z opt   Im Z s 
Re Z in   Re Z s 
Im Z in   Im Z s 
Combining the above criteria, simultaneous noise and input matching are achieved when,
Zin=Z*opt.
The value of Zin is analyzed in eqn 7.1 and Zopt. If they become equal, then the noise can be
optimized. But due to inductive terms in aforementioned equations, the inductive reactance
fades the effect of capacitance at high frequencies, further improving the noise performance.
The matching of Re{Zopt} and Re{Zin} in a wide frequency range is very challenging. This is
due to the fact that Re{Zopt} is frequency-dependant while Re{Zin}is constant and biasdependant (proportional to the cut-off frequency, ωT). However, Re{Zin} is also a function of
Ls and by the proper choice of this inductance we can optimize the circuit for wideband
operation. But because of the parasitic capacitance, Cgd, the expression for Re{Zin} becomes
complicated and slightly dependent on frequency. Figure 7.15 illustrated the dependency of NF
on Re{Zopt} and Re{Zin}.
P a g e | 68
Figure 7.15: Dependency of NF on Re{Zopt} and Re{Zin} derived from analytical expressions
7.3.3 DESIGN CONCERNS FOR CASCODE STRUCTURE
There are few cascode structures reported in literatures and published works. Cascode structure
ensures isolation between input and output isolation. Frequently used cascode structures are
common source-common drain(CS-CG)[21], common gate-common source (CG-CS)[23],
common source-common source cascode structure(CS-CS)[24].
CG-CS structure receives its input through CG structure where the input signal is just passed
through with proper input impedance matching. This type of structure has lower noise figure.
CS-CS structures are composed of two CS amplifiers one on another. The output of the first
stage is coupled to a second stage through a coupling capacitor and output is bypassed by
another bypass capacitor used in second stage. The main idea of using this type of structure is
to boost transconductance. A CS-CG structure can be converted to CS-CS structure with the
help of few capacitors and inductors. CS-CS structure demonstrates better voltage gain than
other structures. But it uses more inductors and capacitors thus increasing its die area.
CS-CG structures have a CG stage stacked over CS stage. CS stage does the main
amplification and CG stage just passes the amplified signal to the output. This type of structure
provides better reverse isolation. This structure is most commonly used structure for LNA
design. But this structure draws more biasing current thus increasing power consumption.
Figure 7.16 shows different type of cascode structures.
P a g e | 69
VDD
VDD
RL
RL
Vout
Vout
Vin
(b)
Ls
Vbias
V
Vgate
M1
Common
Gate
M1
Common
Source
(a)
Common
Source
M1
Vbias
LL
Common
Source
Vout
RL
LL
Common
Gate
Common
Source
LL
Vin
VDD
Ls
(c)
Ls
Vbias
Figure 7.16: Demonstrating (a) CS-CS cascode (b) CS-CG cascode and (c) CG-CS cascode
structures
7.3.4 DESIGN CONCERNS FOR LINEARITY IMPROVEMENT
In the traditional cascode LNA design, no matching has been considered between the commonsource stage and the common-gate stage. This is not desirable for the maximum power transfer
loss of the power directly affects the noise performance of the LNA, since both the input
impedance of the common-gate stage. Until now, the most efficient linearization method for a
CMOS LNA has been the derivative superposition (DS) technique. This method nulls the
negative third-order derivative of the main field-effect transistor‘s(FET‘s) dc transfer
characteristic(gm3) by paralleling the auxiliary FET biased near the weak inversion region with
multiple gated transistor (MGTR)method to the modified DS method. In [23], this technique
was used to improve linearity by paralleling an NMOS with CG amplifier, thus treating
nonlinear current through it. In [26], a PMOS was used parallel to CG structure to treat the
nonlinear current through introducing a third order derivative of drain current which is g m3
superimposing on the CG stage. The figure below illustrates the technique.
Figure 7.17: (a) Cascode amplifier with the folded cascode PMOS IIP3 booster. (b) Third-order
power series coefficients of drain current IsA at DC [26].
P a g e | 70
The figure 7.17(b) shows that the transconductance of PMOS varies alternatively with respect
to transconductance of NMOS in CG stage. As both NMOS and PMOS are in parallel
connection, the resultant transconductance will be settled to a constant value.
But this technique has following limitations:
 This technique lowers the gain.
 Additional PMOS will introduce additional noise
 The parasitic of PMOS can introduce some parasitic resonance
 Impedance matching can be affected by introduction of additional PMOS
 The cascading of PMOS should be done at the first amplifying stage because IIP# of
the overall circuit can be written as
1
1
K
K1



 .....................
IIP3 IIP31 IIP32 IIP33
(7.2)
Where, IIP31 is the IIP3 introduced by first stage, IIP32 is the IIP3 introduced by second stage
and following. K, K1 are constant associated with the gain of the preceding amplifier stages.
7.3.5 DESIGN CONCERN FOR OUTPUT M ATCHING
The output of the LNA is generally characterized with the behavior of the output node
to a 50Ω resistive termination. Both high Gain and wide Band-Width is desired in UWB
communication. But, unfortunately Gain and BW are inversely related. For this reason,
sometimes, Gain is required to sacrifice to get a wider BW. In all types of LNA, the output is
seen across an inductive load. The fig below shows the output drawn through Cout. Generally,
Series resistance, that is, the quality factor of LL governs the gain and BW of the LNA. That is
why a series resistance RL is used to reduce the quality factor at output, hence increasing the
bandwidth. Note that, RL is in the bias path of the amplifier. The value of RL must be limited to
reduce its contribution to overall noise.
VDD
RL
LL
Z
C out
M2
C1
L1
Vout
Zload
Lg
M1
CP
Rs
L2
Vs
I D1
C2
Vbias
Ls
Figure 7.18: Proposed LNA design without buffer stage.
The Q factor of LL is responsible for governing the gain and bandwidth of the LNA. RL,
in fact, controls the Q-factor of LL, which in turn determines the BW of the LNA.
P a g e | 71
RL
RLoad=Q L2R L
Cgd2
Cgd2
LL
LL
(b)
(a)
Z
Load
Z
Load
Figure 7.18.1: (a) The load impedance of the core Amplifier (b) Approximation of (a)
For mathematical demonstration of the Gain-Bandwidth adjusting technique [44], the
load impedance, Zload of the core amplifier is drawn in Fig 7.18.1:(a) where Cgd2 is the gatedrain capacitance of M2. Quality factor of LL is QL=(ωLL)/RL.
In Fig 7.18.1 (a),
Zload ( s) 

sLL  RL
s Cgd 2 LL  sRLCgd 2  1
2
sLL
s Cgd 2 LL  sRLCgd 2  1
2
And In Fig 7.18.1 (b),
sLL
Zload ( s)  2
s Cgd 2 LL  sLL / Rload  1
(7.3)
(7.4)
Fig 7.18.1(b) is an approximation of Figure 7.18.1(a), because comparing (7.3) and (7.4),
RL C gd 2 
LL
Rload
 Rload 
LL
RL C gd 2
 Rload 
L2
1
. L2 .RL
LL C gd 2 RL
 Rload  (c2 .
L2L
) RL  QL2 RL
RL2
Where, the center frequency is,
1
c 
LLCgd 2
(7.5)
P a g e | 72
And, QL is the quality factor of LL at the center frequency.
If the output matching circuit is good enough, such that Im{Zout}=0, then the gain of the
amplifier depends solely on Re{Zout}=Rload. From (6.4), as RL increases, Rload decreases and so
does the gain of the core amplifier. So, -3 dB cut-off points move away from the center
frequency giving rise to an increase in the Bandwidth of the amplifier. This is how RL can
adjust the Gain- Bandwidth of the circuit conveniently.
Maximum Gain and minimum BW is achieved when RL=0. Increasing RL increases the BW,
reducing the gain, thus keeping the Gain-Bandwidth product constant. So, Gain-Bandwidth can
be adjusted by simply choosing a suitable value of RL. Of course, NF will degrade slightly if
RL is increased.
Another approach to match Re{Zout}=Rload is to use a buffer stage shown in fig. 7.19, where
the MOS transistor M3 is biased in such a configuration so that the transconductance of the
transistor matches ~20mS. The output is seen from the source of this transistor, so the output
impedance can easily matched to 1/gm=Rload by changing bias current through M3. This can be
done if M3 is biased by a current mirror.
In this scenario, the value of RL and the value of output capacitance play a vital role. The value
of RL can be selected to put a zero that is outside the desired band[21] as well as increase the
BW. On the other hand, the value of output capacitance can be select to produce a resonance
point that is inside or outside the operational band.
7.3.6 DESIGN CONCERNS FOR OVERALL POWER CONSUMPTION
Power consumption, noise figure, linearity input and output impedance matching are related to
each other by the tranconductance of the MOS transistor. For any analog circuit design, low
power consumption is highly desired. But there is a complex relationship between all of these
parameters.

The input impedance is related to transcoductance by eqn . So to match
𝑔𝑚 𝐿𝑠
(𝐶𝑝 + 𝐶𝑔𝑠 )=50Ω, gm is tuned by changing bias current and width of the MOS
transistor. Power consumption is proportional to gm.

The noise figure demonstrated by equation 6.7-6.9 show that noise figure is dependent
on transcoductance and gate to source parasitic capacitance. An increase
transcoductance ensures lower noise figure but higher power consumption. So, the
width of the MOS, overdrive voltage and bias current must be selected by the tradeoff
between power consumption and noise figure. The following figure illustrates the
dependency of NF on width and bias current of a MOS transistor.
P a g e | 73
Figure 7.18.2: Dependency of noise on width and bias current of the amplifier [21]

1 dB compression point is inversely related to the gain of the circuit. The gain of a
cascode amplifier is dependent on the transconductance which in turn decides the
power consumption. So, lower gain can ensure improvement in 1 dB compression
point.
The output impedance is also dependent on transconductance as mentioned earlier.
7.4 SIMULATED RESULTS FOR THE PROPOSED LNA STRUCTURE
The main objective of UWB LNA used for microwave imaging is to increase the signal power
to a suitable level for signal processing and to overcome noise in preceding stages within
limited power consumption. The proposed LNA was been designed in IBM 130nm CMOS RF
technology and simulated in Cadence. Figure illustrates the schematic diagram of the proposed
CS-CG cascode LNA to operate within FCC unlicensed band from 4.8-10.9GHz as the -3dB
bandwidth of the transmitted pulses fall within this limit shown in figure 7.24.
Figure 7.19: Proposed front end LNA for UWB application
P a g e | 74
A 3rd order T-section chebyshev filter is cascaded with the cascode amplifyer to filter out the
received signal outside predefined band and pass the received signal to amplifier with a flat
gain. The values of inductors and capacitors used in the chebyshev filter are calculated for
matching a characteristics impedance, Z0=50Ω and a ripple of 0.15dC. Because of the gatedrain parasitic capacitance of M1, the resonance frequency gets shifted with respect to (1) and
determined by the resonance between Lg and Ct
/ (1   2 Ls Ct )  Cgd [21]. That is why Lg,
Ls and Cp values where changed to reduce the effect of Cgd. Fig 06 shows the input reflection
co-efficient in dB to demonstrate the input matching of the LNA in which S11<-10dB from 511GHz and the input is perfectly matched at 7.8GHz. The chebyshev filter ensure multiple
matching point of the input impedance to 50Ω, hence the input reflection parameter remains
less than -10dB for a wide band range.
Figure 7.20: Input and output reflection parameters. Input impedance is matched perfectly near
7.8GHz (S11<-27dB). As transconductance of M3 is matched to 20mS, the output reflection
parameter S22<-10dB from 5-14GHz. S22 shows minimum value near 10GHz because of
resonance between load inductor, LL and parasitic capacitance of M3 and M4.
P a g e | 75
Figure 7.21: Input impedance and output impedance of the proposed LNA. The input is
matched to 50Ω near 5G, 7.7G and 10.2GHz yielding improved scattering parameters.
To limit the noise contribution, the outer dimension, metal spacing and turns ratio of the
inductors were selected to peak Q near operating frequency during simulation. As from the
classical theory of noise, source resistance, R s=Rin=Ropt yield optimum noise, Ropt was
simulated. The noise figure of the MOS transistor, M1 is dependent on its
transconductance(gm), higher drain current would yield lower noise figure but higher power
consumption. So, to match the input impedance as well as lower the noise figure and power
consumption, the width of M1 was chosen 150µm. To limit the noise contribution from M2, the
width of M2 was chosen 60µm. The drain current, ID1 drawn from the source is ≈4.2mA
Figure 7.22: Noise Figure of the proposed LNA. NF(min) and NF coincides near 7GHz and
10GHz.
P a g e | 76
Figure 7.23: Equivalent Noise Resistance
from a supply voltage of 1.2V. For biasing the M1, a diode connected MOS transistor,
Mbias,(W/L≈.75, L>.13µm) in series with Rbias (~27.27KΩ)was used. Cbias(~2pF) was used to
provide an ac ground. The gate of M2 was connected to supply voltage. Figure 7.22
demonstrates the simulated noise performance of the proposed LNA. The noise figure
illustrates that the overall noise contribution is less than 4dB within operational band and it is
minimum (2.65dB) near 6.5GHz.
The fig 7.23 illustrates equivalent noise resistance of the proposed circuit. The equivalent noise
resistance, Rn is lower near the lower edge of the operational band but higher near the high
frequency which implies the contribution from noise resistance is higher at the lower edge of
the band.
The table 7.1 shows the peak quality factor and dimension of the inductors chosen for the
circuit to ensure higher quality factor of the inductors and to reduce contribution to input
referred noise.
The load inductance, LL was selected to ensure higher gain. Cabuf (~500fF) filters out the dc
part from the output of the cascode structure and fed the signal as an input to the buffer stage.
The value of RL (~60Ω) was chosen to contribute to the real part of output impedance and to
provide some voltage headroom through simulation. RL is also chosen to put a zero frequency
as close as possible to the lower edge of the band to improve gain at lower frequency. Another
important use of RL is gain bandwidth adjustment as described in this chapter.
The buffer stage was chosen to boost the gain. Figure 08 shows the forward gain(S21) of the
proposed LNA. The flat 12dB forward gain, S21 is seen from 5.2-9.8GHz. -3dB band width is
P a g e | 77
Figure 7.24: Forward gain and reverse isolation of the proposed LNA
found from 4.7-10.9GHz. In the buffer stage, a source follower structure was stacked on
common gate structure shown in figure 7.19. The width of M3 was chosen ~30µm with a drain
current ~1.4mA so that the output impedance seen from the output through C out matches 50Ω.
The gate voltage was supplied to M3 through RbM(~4kΩ). M4(W/L≈30/.36) was in current
mirror configuration with Mbias to reduce any additional circuit. Table 1 depicts the elements
used in the circuit. Figure 7.24 illustrates the output reflection co-efficient, S22 <-10dB from
4.5GHz-14GHz. S22 shows better performance (-18dB) near 10GHz because of the resonance
between LL and combination of Cabuf and other parasitic capacitances. The reverse isolation, S12
is <-40dB which demonstrates the superiority of cascode structure for isolation.
Figure 7.25: Transducer gain, GT, Available gain, GA and Power gain, GP of the proposed LNA
P a g e | 78
Figure 7.26: The group delay of the proposed circuit
The transducer gain, GT, available gain, GA and power gain, GP are shown in the fig 7.25. All
the gain curves demonstrate quasi-flat response within operational band yielding the
performance of the proposed LNA. The values of the three plots coincide at the center
frequency which indicates good matching for the design.
The group delay of the proposed LNA showed flat response from 6-10GHz where the group
delay is around 0.140ns. The group delay is much higher near the lower edge of the band and
~200ps near 5GHz. The group delay lowers near higher edge of the operational band shown in
fig 7.26.
Figure 7.27: Voltage Standing Wave Ratio at input and output (matched for 50Ω)
VSWR1 and VSWR2 are 1.05 and 1.6 respectively at 7.5GHz and. If VSWR is under 2,
antenna match is very good. Input reflection parameter is at minimum at VSWR 1.05 at
7.5GHz and <1.6 throughout the operational band.
P a g e | 79
1dB compression
point at -16.5dBm
Output Power in dBm
IIP3 2dBm @7.1GHz
IIP3 -7dBm @6.9GHz
IIP3 calculation at 7.0GHz
Input Power in dBm
Figure 7.28: Linearity performance of the proposed LNA
Figure 7.29: Variation in linearity of the Proposed LNA
The figure 7.28 the 1dB compression point is -16.5dBm. The linearity of the circuit was
simulated via two tone test. The figure illustrates the simulated result of the two tone test at
7GHz with 50MHz spacing between two tones. This test generates two third order
intermodulated 3rd order frequency component at 7.1GHz and 6.9GHz. The simulation should
the circuit performs better to isolate lower intermodulation frequency than higher
intermodulation frequency. The intermodulation intercept point IIP3 at 6.9GHz is ~2dBm but
at 7.1GHz is ~-7dBm. The figure 7.29 demonstrates the circuits‘ average linearity performance
at different frequency over the operational BW. The figure also shows variation of 1dB
P a g e | 80
compression point with respect to frequency. The circuit demonstrates better performance for
IIP2 and OIP2
The proposed LNA suffers from gain compression because of lack of voltage headroom. But
for impulse radio communication, received signal by the antenna will be much weaker (around
µW) and at that lower input, the LNA will work perfectly. If the supply voltage is increased to
1.5V, this would result in better IIP3 but higher power consumption. Figure 7.29.1 shows the
linearity performance at 1.5V supply voltage.
IIP3 @ 1.06dBm
IIP3 @
4.16dBm
1dB compression
point @ -14dB
(b)
(a)
Figure 7.29.1: (a)Linearity performance of the Proposed LNA @ 1.5V supply and (b)variation
of output with respect to different supply voltage.
The unconditional stability (B1f>0) and conditional stability (Kf>1) are shown in fig 7.30
demonstrating the circuits stability in the band of interest. Kf>1 and B1f>0 ensure that the
circuit is both conditional and unconditionally stable for the operating band of frequency. So
the proposed LNA will not be prone to oscillate because of white noise which may be present
in the circuit during power up.
P a g e | 81
Figure 7.30: Stability of the proposed LNA
The circuit draws ~4.2mA current for the core amplifier, ~1.5mA current in buffer stage @
1.2V supply voltage. The gate voltage supplied to CS MOS amplifier draws negligible current
(~ hundreds of µA). The current mirror used to bias M3 is a long channel MOS drawing very
small current of .12mA. The total current drawn from supply is 6mA. The total power
consumption of the proposed circuit is ~7.77mW. The figure below shows the variation in gain
with respect to supply voltage variation. The gain tends to saturate near 1.3V supply.
Figure 7.31: Variation of minimum noise figure and forward gain of the circuit due to change
in supply voltage
P a g e | 82
The figure 7.31 illustrates that ±5% variation in supply voltage can cause <±5% variation in
gain. The minimum noise figure seems to be below 3dB±10% within band of operation. This is
obvious because of dependency of NF and gain on transconductance of the MOS transistors.
Figure 7.32-7.35 illustrates the scattering parameter and noise figure at different corner of the
IBM 130nm CMOS process. The forward gain for fast-fast (FF) and fast-slow(FS) corner were
seen higher than slow-slow(SS) and slow-fast(SF) corners.
Figure 7.33-7.35 demonstrates the input-output matching parameters and noise figure at
different process corners. As the proposed LNA design contains only NFET transistors, for
both FF and FS corner, all the simulation yield same result, because at these corners, NFET
transistors are fast. Same is true for SF and SS corners. FF and FS corners yield best results for
forward gain and noise figure due to higher transconductance of transistors at these corners
where as SS and SF corners demonstrates better output matching at lower frequencies. Table
7.4 summarizes the consumption at each corner of 130nm CMOS process.
Figure 7.32: Forward gain of LNA at different process corner
P a g e | 83
Figure 7.33: Input matching parameter, S11 at different process corner
Figure 7.34: Output matching parameter at different process corners
P a g e | 84
Figure 7.35: Minimum noise figure for different process corner
Process
Corners
FF
FS
TT
SF
SS
Table 7.4: LNA’s Performance at Different Process Corner
Power
Corner Simulation at extreme corner cases
Consumption(overall)
S21(dB) S11(dB) S22(dB) NFavg(dB)
0
8.35mW
TT, 27 C
12.4
<-10
<-10
3.2
7.77mW
FF, 550 C
12.1
<-10
<-10
3.21
7.74mW
TT, 1000 C
9.89
<-11
<-11
4.1
0
8mW
SS, -55 C
9.9
<-9.0
<-8.7
4.285
7.5mW
SS, 850 C
9.3
<-10
<-9.9
3.95
The proposed design intended to use an auxiliary circuit for implementing DS method for
improving linearity of the circuit. But the auxiliary circuit used in the design was unable to be
functioned properly to boost IIP3 of proposed design. That is why it is left from the final
circuit design.
The figure 7.36 depicts the time domain response of the proposed LNA to a series of UWB
pulses. From the output, it can easily be realized that for both positive and negative Gaussian
monocycle pulses, the LNA produces distinct response.
P a g e | 85
Figure: 7.36: Time domain input(red) and output(green) of the front-end receiver
To illustrate the change in response due to random process variability, a Monte Carlo
simulation was performed under 30 random process variations. From the simulated results, it
was seen that the forward gain(variation±1.1dB) and noise figure(variation±.4dB) are quite
insensitive to process variations. Figure 7.37 depicts the responses derived from simulation.
Figure 7.37: Variation in S21 and minimum noise figure in dB due to random process variations.
From the simulated results, it was seen that gain reduction due to any process variation results
in rise in noise figure. This quite obvious, because increase in gain i.e. power consumption is
inversely proportional to noise.
P a g e | 86
The following tables 7.5 and 7.6; document the results for proposed design and compare the
simulated output with other published works.
TABLE 7.5: Tabulated Transmitter Response
Proposed Circuit for µ-wave Imaging
Parameter
Value
Supply Voltage
1.2V
Pulse Rate Variability
10Mbps-8Gbps
Pulse Width
125ps
-3dB Bandwidth
5-10GHz
Power Consumption
0.32pJ/pulse
Proposed Circuit for µ-wave Imaging and Data Communication
Supply Voltage
1.2V
Data Rate Variability
10Mbps-4.1 Gbps (normal FET)
1Mbps-7.15 Gbps (LVT FET)
Pulse Width
183ps(normal FET)/ 152ps(LVT FET)
-3dB Bandwidth
4.8-10.2GHz
Power Consumption
0.68pJ/bit
TABLE 7.6: Tabulated LNA Response
Parameter
Value
S21
S12
S11
S22
Band Width
Noise Figure
Input Impedance (real)
Input Impedance (imaginary)
Output Impedance (real)
Output Impedance (imaginary)
IIP3
OIP3
Kf
Power Consumption
12.1 dB±.3dB from 5.2G-9.8GHz
<-40 dB
<-10 dB
<-10 dB
6.2 GHz (4.7-10.9 GHz)
2.6 dB at 6.5GHz, average 3.2dB
50Ω at 5, 7.5 and 10 GHz
0 at 7.5 GHz
~39Ω throughout the band
<5Ω at 7.5 GHz
-2.03 dBm
9.6 dBm
> 20
7.70 mW(including current mirror and buffer
stage), 5.22mW(Core amplifier)
Table 7.7 summarizes the result of the IR based transmitter and depicts the comparison. The
simulated results of LNA are summarized and compared with published work illustrated in
table 7.8. The figure-of-merit (FOM) is defined here to evaluate the circuit performance as,
S (dB).Freq(GHz )
FOM  21
Power (mW ).NF (dB)
P a g e | 87
TABLE 7.7: SUMMARY OF THE PROPOSED TRANSMITTER AND COMPARISON OTHER
PUBLISHED WORK
CMOS Technology
0.18 µm [37]
0.18 µm [36]
0.13 µm [45]
0.13 µm [38]
90nm [39]
90nm[46]
0.13 µm(proposed)
Energy Consumption
(pJ/pulse)
4.7
18
9
125
1.42
1.8
0.68 (pJ/Bit)
Amplitude(Peak
-to-Peak in mV)
500
180
1420
450
502
107
390
Pulse Width
(ns)
0.8
3.5
0.46
0.6
0.06
0.15-0.35
0.183
η
(pJ/p-pV)
9.4
100
6.33
277.78
2.82
16.82
0.872
TABLE 7.8: SUMMARY OF THE PROPOSED LNA AND COMPARISON OTHER
PUBLISHED WORK
Specification
Data
Year
CMOS
Process(µm)
Supply (V)
BW (GHz)
S21(dB)
S11(dB)
S22(dB)
NF min(dB)
NFave (dB)
ICP (dBm)
IIP3 (dBm)
Pdiss(mW)
Stage
Operating
Range(GHz)
FOM
This Work
Simulated
2014
[23]
Measured
2009
[25]
Simulated
2008
[29]
Simulated
2012
[30]
Measured
2007
[31]
Measured
2011
[40]
Simulated
2013
[41]
Measured
2008
0.13
0.13
0.13
0.18
0.18, SiGe
0.13, SiGe
0.18
0.18
1.2
6.2
12.4
-11~-30
<-10
2.65
3.2
-16.8
-2.3
5.22*
2
1.8
6.6
11.7
<-9
N/A
3.6
<4
N/A
11.7
2.62*
2
1.5
1.3
12
<-8.5
N/A
2.2
2.2
N/A
16
17.4
1
0.6
7
14.6
<-9.6
<-10
3.7
4.6
<-20
-13.19
3.1
1
1.8
10
8
<-12
<-10
2.9
2.9
-13.1
-3.5
21.6
1
2.2
10
20
<-8.5
N/A
N/A
7.2
N/A
N/A
180
2
0.9
2
16
<-5
> -8
2.1
3.2
<-15
-4
13
1
1.8
6
9.3
-13~-20
N/A
4.4
5.5
-14
-4
27
1
4.7-10.9
1.5--8.1
0.8-2.1
3.1-10.6
0.1-11
8-18
3-5
21-27
7.18
6.25
0.627
7.93
1.43
0.233
2.334
1.96
*core amplifier power consumption
As the operational band of the transmitter and LNA includes couple Wi-Fi and WiMax UNII
band, this may cause interference with the transmitted UWB signals and drive the LNA to
saturation. In order to avoid these bands, UWB antenna can be designed to attenuate these
bands precisely. In [47], a special microstrip UWB antenna was designed with notches near
5.4GHz and 8GHz.
P a g e | 88
Figure 7.38: S-parameter of UWB microstrip antenna with in-band notches[47]
The antenna was incorporated with a notch filter to filter out the undesired frequency
components. Figure 7.38 illustrates S-parameter of such antenna. Using the same design
techniques, any undesired narrowband interfere with fractional band width <0.45%.
P a g e | 89
CHAPTER 08
CONCLUSION
This dissertation documents design of a clock independent UWB transmitter and front-end
receiver using IBM 130nm CMOS RF technology platform for medical application. In order to
develop chip based systems for microwave imaging or medical implants, there are three major
limitations which are size, power dissipation of the chip and FCC regulation. Thus, the main
objective of this work is to design a power efficient UWB system that can be used for both
microwave imaging and short distance high and low speed data communication between
medical sensors operating within the unlicensed band. The proposed method for UWB
transmitter facilitates generation of UWB pulses without the help of any external clock circuit,
thus reducing the system complexity both at transmitter and receiver end. The proposed
technique is power and space efficient because it consists of fewer MOS transistors and no
inductor. This is highly desirable for medical implants and sensors. The generated pulses are
Gaussian monocycle pulse and their spectral region occupy 4.8-10GHz band. In order to use
the same transmitter for microwave imaging and data communication between sensors, a new
modulation technique is proposed and bit error rate in AWGN (Additive White Gaussian
Noise) environment is also calculated to show the effectiveness of the proposed modulation
technique. On the receiver end, a front end receiver is designed for UWB microwave imaging
and data communication. A power constrained, source degenerated, cascode CS-CG UWB
LNA design technique is adopted here. Under power dissipation constraint, the source
degeneration inductance plays a significant role in the fulfillment of the simultaneous noise and
input matching requirement. Finally, combining all the requirements of low-power
consumption, low NF, and simultaneous noise and input matching, a step-by-step design
technique is developed. The LNA is cascaded with a chebyshev filter to ensure flat band
response from 5-10GHz. The input referred noise is calculated to investigate the dependency of
design parameters on noise. The cascode structure ensured better reverse isolation. The gain
bandwidth adjustment technique was used at the output to ensure flat response over the band of
operation. A second buffer stage is used to match the output to a desired load. The buffer stage
and core amplifier are biased through a current mirror. To ensure optimum performance,
simulations on scattering parameters, noise, stability, transient, and harmonic balance are
carried out. The trade-offs among different requirements and how these have affected the
circuit parameters are also discussed in brief. All relevant time domain analysis for transmitter
and receiver were performed. The simulated results are compared with recently published
work. The proposed design has demonstrated better performance in terms of power dissipation,
scattering parameter, noise performance and stability.
9.1 FUTURE WORK
The proposed transceiver system can sponsor the wireless body area network for overall
medical sensing application. As any attempt of signal processing is not included in this work,
the suggestion for future work on this dissertation will focus on the signal processing part of
the system. There are different techniques reported for UWB signal processing on µ-wave
imaging. Effective signal processing circuitry can pave the way to successful implementation
of these imaging and data communication techniques.
The proposed system can also be utilized for short distance high speed wireless
communication. Recently inter-chip and intra-chip high speed communication has drawn much
attention. This work can be extended to explore the possibility to be used for such
P a g e | 90
communication. The proposed transmitter can be subjected to jitter if it is used for ultra high
speed (>4.5GHz) wireless communication. The jitter can be reduced by adjusting efficient jitter
reduction processes. The proposed LNA can be used as front en receiver for communication
within unlicensed band.
To consider the proposed designs for fabrication, the circuit designs are need to be optimized
up to post layout simulation level and the architecture is yet to be evaluated in terms of
parasitic extraction.
P a g e | 91
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Zhao, Liwu Yang, and Yumei Zho,‖ 1.8 pJ/Pulse Programmable Gaussian Pulse
Generator for Full-Band Noncarrier Impulse-UWB Transceivers in 90-nm CMOS,‖ in
IEEE Transactions on Industrial Electronics, vol. 57, no. 5,MAY 2010, pp.1555-1562
[47] Pankaj Sarkar, Manimala Pal, Rowdra Ghatak, and Dipak R. Poddar,‖ Miniaturized UWB
Bandpass Filter with Dual Notch Bands and Wide Upper Stopband‖ in Progress In
Electromagnetics Research Letters, Vol. 38, 161-170, 2013
P a g e | 95
APPENDIX A
CLASSIC MOS DEVICE NOISE ANALYSIS
In this appendix, at first the approximated 2-port MOS noise model shown in Fig 6.3 and the
equation shown in equation 6.3-6.6, the expression for noise figure will be derived. The noise
equivalent resistance can be written as:
2
inm

2
nm
i
4 KT
Rn
 i nm
(A.1)
2
jCt
 ig 
id
gm
2
jCt
jCt
id 
id
gm
gm
2
 ig  2. ig
2
(A.2)
ing and ind can be replaced as:
2
i g ( )  4kT 
 2Ct2
g
2
m
 2Cgs 2
5gd 0
2
i d ( )  4kT
 2Ct2
g
2
m
 gd 0
and the correlation coefficient, c can be written in form of:
ig id  c 4kT 
 2Cgs 2
5gd 0
.4kT  g d 0
(A.3)
The noise equivalent conductance, Gn=1/Rn demonstrated as:
4 KT (
 2Cgs 2
5gd 0
Gn 

 2Ct2
g m2
 g d 0  2Cgs
4 KT
 Ct
5 gm
 2Cgs 2 / Ct2
Cgs
 2
 Ct ( 
 1  2 2 c
 gd 0
5
Ct



 2 gd 0
2
2
2
 2Ct 2 ( 2  2 p 2  1  2 c  p)
)
(A.4)
 Ct 2
 gd 0 )
5 g m
(A.5)
P a g e | 96
where,  and

are process dependent parameters,
to source, c  ing ( )i
  gm / gd 0 ,
*
2
gd 0
is channel conductance at zero drain
2
nd ( ) / i ng ( ).i nd ( ) is the correlation coefficient,    / 5 ,
p=Cgs/Ct
The equivalent noise source at input, in can be written in terms of correlated and uncorrelated
current sources, ic and iu:
2
 (ic  iu ) 2  ic2  iu2  2.ic .iu
inm
 ic2  iu2
2
(A.6)
2
2
2
 inm
( c )  inm
(1  c )
Resistance induced from uncorrelated noise sources, Ru
Ru  1

Gu
 4 KT
iu2
2

 2 gd 0
 2  2 p 2 (1  c )
( 2  2 p 2  1  2 c  p )
(A.7)
The correlation impedance, Zc :
*
enminm
Zc 
*
inminm
*
nm nm
i i
 4 KT (
= 4 KT
*
enminm

where,
Zc 
(A.8)

 2Cgs 2
5gd 0
 gd 0
2

 2Ct2
g m2
 g d 0  2Cgs
 Ct
5 gm
)
(A.9)
 2Ct 2 ( 2  2 p 2  1  2 c  p)
id *
inm  j Ls inm
gm
2
id *
 jCt

inm  4 KT (C gs
g d 0 )
5
gm
gm

(A.10)

(1  c p ) 1   2 Ls Ct (1  2 c p  p 2 2  2 )
jCt (1  2 c p  p 2 2  2 )
(A.11)
P a g e | 97
 Zc   jX c
From classical noise optimization theory[20], the optimum resistance for minimum NF;
Ropt 
Ru
Ru
 Rc2 
Gn
Gn
(A.12)
As from eqn (A.11), Rc=0, and Xopt=-Xc
So, the NF of the LNA can be written from the guideline provided by [20],

F  1

Rs


2
Ru   X c  Rs  X s Gn

 1

Rs

2

R  Rs Gn
 1 u
;[ for min imum noise]
Rs


Ru
 1
 Rs Gn

Rs

2
Ru  Z c  Z s Gn
2
 2 2 2 2 2

 2  2 p 2 (1  c )
F  1
 Ct (  p  1  2 c  p)  2 2 2

g m Rs 
(  p  1  2 c  p) 

(A.13)
(A.14)
P a g e | 98
APPENDIX B
BIT ERROR CALCULATION FOR THE PROPOSED MODULATION SYSTEM
The bit error rate calculation for a particular modulation system in Additive White Gaussian
Noise channel (AWGN) follows the following procedures:
1. Modeling the channel;
2. Considering the no of line of sight;
3. Considering the contribution of noise in terms of signal to noise ratio (SNR);
4. The demodulation process to recover the transmitted bits
5. Find out the percentage of error bits for different SNR and number of line of sight
The BER was calculated for 1Kbit at 1.3Gbit/s speed for different SNR and number line of
sights. The AWGN channel was modeled by following equation:
hc (t )   ai (t  ti )
n i
(B.1)
Where, ai is related to the attenuation introduced due to travelling through ith path and
ai=1/(distance for travelling via ith path)^gamma. The value of gamma varies from 1~2. t i is the
delay associated with distance for travelling via ith path. Equation 3.14 and 3.15 describes the
modulated Gaussian monocycle pulses. With the aid of the equation 3.14, 3.15 and C.1, the
received signal can be written as, r (t )  x(t )  hc (t )  n(t ) , where n(t) is the noise contribution
from the channel defined by SNR.
The flow chart for calculating BER is shown below:
The calculated bit error rate is shown in the following figure:
P a g e | 99
(a) For one line of sight
(b) For two line of sight
Figure B.1: BER for different SNR and line of sight
P a g e | 100
APPENDIX C
THE COMPONENTS USED IN DESIGNING LOW NOISE AMPLIFIER CIRCUIT
C.1 MOSFETS
IBM 130 nm Standard CMOS 8RF technology offers following types of MOSFETs,
a)
b)
c)
d)
e)
f)
g)
Normal 1.5 V MOSFETs
3.3 V MOSFETs
Zero Vt MOSFETs
Low Power MOSFETs
Double Gate MOSFETs
Triple Well MOSFETs
RF MOSFETs
etc.
(a)
(b)
Figure C.1: (a)Illustrates the variation in drain current and transconductance with respect to
gate-source voltage under different body-source voltage of a typical NFET, (b) shows the DC
characteristics of NFET
P a g e | 101
Both NMOS and PMOSs are available in above categories, among which, we choose
normal 1.5 V MOSFETs for the front-end LNA and the transmitter. This is because RF
MOSFETs are compact in design; consume more area than normal NFET or PFET and their
parasitic are unpredictable. These normal FETs are biased with a 1.2V voltage source rather
than 1.5V to reduce power consumption. Figure C.1 and C.2 show the variation in DC
characteristics with respect to gate voltage under different conditions for BSIM4 and PSP
models. Figure C.3 demonstrates the noise performance of the FETs with minimum feature
size with design corners.
(a)
(b)
Figure C.2: (a)Illustrates the variation in drain current and transconductance with respect to
gate-source voltage under different body-source voltage of a typical PFET, (b) shows the DC
characteristics of PFET
P a g e | 102
Figure C.3: Input referred noise voltages of NFET and PFET. PFET shows lower contribution
to noise at higher frequencies.
(a)
(b)
Figure C.4: (a)Illustrates the variation in drain current and transconductance with respect to
gate-source voltage under different body-source voltage of a typical LVT_NFET, (b) shows the
DC characteristics of LVT_NFET
P a g e | 103
The proposed transmitter can also be designed using LVT_NFET. Fig C.4 depicts the
characteristics of LVT_NFET. Lower threshold voltage of this type of MOS transistor helps to
switch the transistor faster but consumes more power.[25]
C.2 INDUCTORS
Inductors offered by IBM 130 nm CMOS 8RF are:
a) Normal single level inductor
b) Series stacked inductor
c) Parallel stacked inductor
d) Symmetric stacked inductor
The metal turns of a spiral inductor experience two significant loss mechanisms that increase
with frequency; skin effect loss and magnetic field induced proximity effect loss.
 Skin Effect Loss
The ac current in a metal conductor will flow increasingly on the surface as frequency
increases. This causes the conductor‘s effective resistance and inductance to undergo change
over frequency. At high frequencies, the resistance of a conductor increases as a function of the
square root of frequency and the inductance decreases slightly and then levels off.
 Proximity Effect Loss
In a spiral inductor the enhanced magnetic field that exist in the central portions of the spiral
tends to cause non-uniform current flow in the turns. Inner turns tend to have current flow only
on the innermost edge of each turn, while outer turns tend to carry current on their outermost
edges. This effect is frequency dependent as the magnetic field increases with increasing
frequency. The non-uniform current flow is typically called "proximity effect" and tends to
cause effective spiral resistance to rise faster than can be attributed to skin effect. In addition,
the net inductance will decrease due to an effective reduction in the radius of the spiral caused
by the current crowding to the innermost edge of the inner turns. This "proximity effect" can be
the dominant loss mechanism at frequencies of interest for multi-turn spirals.
 Loss Model
An improved method of implementing frequency dependent behavior in an inductor model is
to use an R-L ladder network to replace the series loss elements in the model. An appropriate
selection of element values can achieve the desired frequency dependent behavior while
maintaining the ability to simulate at all frequencies under all simulation conditions (without
needing to specify a frequency at which to calculate the loss). This is the technique that has
been implemented in the inductor model.
All these inductors are octagonal spiral. The design normal single level octagonal spiral
inductor which exhibits least parasitic effect and consume least space as well.
P a g e | 104
As discussed in previous sections and chapters, the quality factor of the inductor has some
importance to input referred noise and gain parameters. The following figures illustrate the
correlation between inductance and quality factors.
Figure C.5 illustrates the change in conventional quality factor with respect to frequency for a
single layer spiral inductor model with Back End of Line (BEOL) option. Figure C.6 shows the
change in Q and inductance L with respect to change in frequency.
Figure C.5: Variation in quality factor, Q with respect to frequency of a single layer inductor.
(a)
(b)
Figure C.6: (a) Variation in inductance and (b) quality factor, Q with respect to frequency of a
single layer inductor.[43]
P a g e | 105
C.3 CAPACITORS
Following capacitors were available in this technology,
a)
b)
c)
d)
e)
MIM (Metal Insulator Metal) Caps
Dual MIM Caps
High-K MIM Caps
Vertical Natural Caps
MOSFET Caps
Hi-K RF MIM Capacitor models yield best result, because they were of minimum parasitic
effect. But this model consumes more die area. Figure 9.4.1 illustrates the performance of HiK
MIM capacitor with respect to variation in voltage across it under different temperature. As,
the design required a capacitor below 60 fF, vncap models were chosen. All these capacitors
are three terminal capacitors where the third terminals were connected to substrate.
Figure C.7: Variation in capacitance under different voltage and temperature.[43]
Figure C.8 illustrates the variation in capacitance under different operating frequency. The
capacitor of these models is suitable up to tens of GHz. After that, the capacitor behaves
P a g e | 106
unpredictably. The quality factor decrease as frequency increase.MIM cap models are suitable
for capacitors around pF. Figure 8.4.3 and 8.4.4 show the change in capacitance of vncap
model under different bias/gate voltage at the third terminal. For design simplicity, the third
terminal of this type of capacitor was connected to ground through substrate.
Figure C.8: Variation in capacitance and quality factor with respect to width of the capacitor
models.[43]
P a g e | 107
Figure C.9: Variation in capacitance of ncap models under different voltage[43]
Figure C.10: Varactor model @ Vg=0V, Vsd=0V[43]
C.4 RESISTORS
Options for Resistors in this technology were:
a)
b)
c)
d)
e)
f)
g)
h)
N+ Diffused Resistance (opndres),
NWell Resistance (nwres),
P+ Polysilicon Resistance (opppcres),
RP Polysilicon Resistance (oprppres),
RR Polysilicon Resistance (oprrpres),
TaN BEOL Resistance (kxres)
L1 BEOL Resistance (l1res),
Silicided Resistance (silres).
The design used high resistor P+ Polysilicon Resistance (opppcres) models and low resistor
Silicided Resistance (silres) in the LNA and in the biasing circuit. In order to model a desired
value resistance, resistors were connected in series and parallel.
P a g e | 108
Figure C.11: Variation in resistance of a silres model resistance under different frequency[43]
Figure C.12: Variation in resistance of a P+ Polysilicon Resistance model resistance under
different frequency [43]