RESONANT TRANSITION SWITCHING WELDING POWER SUPPLY
N. Frohleke, H. Mundinger, S. Beineke, P. Wallmeier, H. Grotstollen
Institute for Power Electronics and Electrical Drives
University of Paderborn, FB 14 - LEA
Pohlweg 47-49, 33095 Paderborn, Germany
Phone: ( 4 9 ) 5251-603157
Fax: (+49) 5251-603157
E-mail: nfroel @lea.uni-paderborn.de
Abstract - By means of parallel connected passive networks,
charged only modestly by energy for commutations, resonant
transition switching is ensured for this modified full-bridge
topology used in a welding power supply. A new driving scheme
adapts the resulting power circuitry for both the droplet and the
short-circuiting transfer welding modes occurring in the gas
metal arc welding process. Less voluminous and cheaper magnetic components together with lower conduction losses of transistors and diodes yield a cost effective, light weight power supply for 12kW l350A. The analysis of the circuit, the design and
the verification results are presented.
I. 1NTRODUCTION
Modern high-frequency welding power supplies (WPS)
feature low weight, robustness, high reliability and flexibility
at reasonable prices. Publications [ 1,2,3,4] demonstrate that
above mentioned features have already been achieved for low
power welding supplies by soft switching topologies.
For the higher output power required for the gas metal arc
welding (GMAW, MIGMAG) process, however, classical
resonant converters are not optimal. Some of their drawbacks
are high reactive power, high conduction losses, and a bulky
tank inductor as revealed in [5,61.
In order to extend the range of high frequency WPS to
GMAW applicatioic (e.g. 10kW) without the need to parallel
smaller units, a modified resonant transition switching fullbridge is proposed. The proposed full-bridge is denoted as
Passive Resonant Commutated Poles (€33-PRCP), following
the terminology used i n [6].
The F€-PRCP is based on the pseudo-resonant full-bridge
[7], but avoids the high peak current charge of the commutation inductors by employing the transformers leakage inductance as well as a modified driving scheme. Thus the size of
the inductors, the current charge of the switches, and the
reactive power circulating in the converter is reduced.
Pulsed gas metal arc welding power supplies
While the gas tungsten arc welding (GTAW) process requires a constant welding current, the GMAW process needs
for a self-adjusting arc a power supply with a constant voltage
characteristic [8]. For puls-welding WPS, however, the
welding current is pulsed to steer the droplet transfer. Therefore, the power stage is embedded into a currentcontrol loop
as shown in fig. I . The set point for the current control-loop
0-7803-3932-0
is governed by a slower voltage control-loop, which is not
shown, with pulse-generating features. As the current set
point will change rapidly, the current control-loop must be
fast.
For a currentcontrolled power stage a current sensor is
obligatory. Hence, the new control scheme can use the current
signal at almost no cost penalty.
inverter (ZVS)
[deadtime adiustm.
t t
t t
I
set point
current controller
I
I
Fig. 1 Current-controlled welding power supply
The publication divides into the following major chapters:
After explaining the operating principle, a circuit analysis is
summarized, which leads to designs equations for the control
signals for two modes of operation. Finally selected simulation results are given for a WPS, which was built as an example prototype featuring the above mentioned characteristics
and additionally a good efficiency at 12kW/350A supplied
from 400V three-phase mains. Calculated and simulated results were proved by measurements.
11. OPERATING PRINCIPLE
The FB-PRCP shown in fig. 2 is made up of two legs A and
B, a center tapped transformer with a small discrete inductor
Ls, the output rectifier, and the output filter inductor Lf,
which includes the wire inductance. At steady state the
GMAW welding arc can be represented by a resistor Ra and a
voltage source V, [I].
Leg A consists of an IGBT-half-bridge (Qap, Qan) with
two freewheeling diodes (Dap, Dan), two small resonant
capacitors (Cap, Can), an inductor La and two auxiliary capacitors ( C I , C2). Leg B is identical but uses different part
values. The auxiliary capacitors are relatively large (a few
pF) and provide approximately half the input voltage. In
order to obtain constant relative current ripple, the filter inductance Lfis nonlinear.
As shown in fig. 3, the IGBTs are driven in a phase-shifted
615
Leg A
I
’ 1 i N +?’
State: P P - N P ( p a r t 1 I
Leg B
I
N P-N N
P P - N P (part! I
I
I
Qap
Qan
I
Q ~ P
Vin
,
,
1
Qbn
. .
:
,
,
.
,
,
NN-PN (partl)
N N - P N (part!=
,
I
. .
. .
~
I
I
I
PN-PP
.
I
,
,
.
.
I
0
I
I
I
I
I
I
Fig. 2 Power circuitiy of FB-PRCP with load equivalent
manner, so that each leg voltage (vcu, vcb) has a duty cycle of
about 50% and the volt-second-product across the transformer primary winding is proportional to the phase-shift dr.
The time duration tcu and tCh for commutation in legs A and B
depend on the load current and are adjusted to maintain ZVS
for the IGBTs.
I
I *I le
dt
.
Q ~ P
.
.
.
,
1
I
.
I
I
,
I
On
off
‘O‘la‘2a
‘ 3 ‘4
I
I
8
,
,
,
,
\
,
I
,
I
,
‘ 5 ‘68 ‘7a
‘8 ‘9
t
‘IO
Fig. 4 Typical waveforms at heavy load
On
Off
mutation intervals are named with a pair of two letters, according to the initial and final network state.
On
Qbn
I
,
On
Off
to
t3
t5
f
t,,
Fig. 3 Control signa’s for FB-PRCP
At light load, the energy stored in La and Lb ensures the
commutation of the resonant phase legs from rail to rail similar to the FB-ARCP [9] but without the need for switching
devices with symmeirical blocking capability and associated
driver and logic circuitry.
At heavy Zoad, the energy of Ls is predominantly utilized to
commutate the resonant capacitors. Compared to the conventional phase-shifted resonant full-bridge Ls is much smaller
for a given minimum load current, and therefore the effective
duty ratio - thus also the transformers turns ratio - can be
increased.
Fig. 4 shows typical waveforms for heavy load conditions.
These waveforms d o not differ much from the conventional
phase-shifted resonant full-bridge, as the commutations are
primarily based on ih. The conversion cycle divides into 10
topological states indicated in fig. 4 and fig. 6. Network state
PN denotes that Sap and Sbn conduct resulting in respective
transistor voltages. Note, that equal symbols NN or PP denote
freewheeling and NI’ or PN denote energizing states. Com-
111. CIRCUIT ANALYSIS
For steady state analysis of the FB-PRCP the following assumptions are set:
a) semiconductors, capacitors, and inductors are ideal, lossless
components
b) the transformer is replaced by its magnetizing inductance Lm
and an ideal transformer with tums ratio n. The leakage inductance is included in the value of inductance Is, and the
auxiliary capacitors CI, C2, C3, and C4 are equal and large
enough to be replaced by voltage sources of V,, /2
c) the phase-shift dr is restricted to values suitable for converter
operation
d) the output current is perfectly smoothed; this assumption
holds, since the welding currents ripple must be small
e) the commutations are fast enough that i h ( t ) , i d t ) , and i L d t )
are constant during commutation intervals.
With these assumptions, fig. 5 results, representing the
equivalent power circuit, where Ca = Cap+Can and Cb=
Cbp+Cbn. For simplification reasons the ideal transformer is
eliminated and the output current is transferred to the primary
as Zup= /,Z n, with n being the turns ratio of primary to secondary (np/ns).
616
Finally, at
uCrrbecomes zero and the energizing interval NP (fig. 6d)
begins.
C. Energizing interval NP: t 2 I t I t j
As soon as the freewheeling of the output rectifier ends at t / b
or t2,, power is transferred to the output as depicted in fig. 6d.
The voltage across Lm is divided due to the the ratio of
Lml(Lm+Ls), whereby with reduced LmlLs a lower output
voltage results. Hence, the designer should not reduce I,, by
selecting a small Lm.
At t3, which is identical to dt+to , the modulator terminates
the energizing interval NP by opening switch Sbp; ucb swings
from V,, to zero voltage.
Fig. 5 Equivalent circuit
A. Abbreviations and Acronyms
o,:=JZz,
w2:=J7GZ3%-
D. Commutation interval NP-NN: t3 5 t 5 t4
During commutation interval NP-NN fig. 6e represents the
power circuit. Assuming ib is approximately constant with
iLb(t3)= ILb3we obtain
B. Commutation interval PP-NP: to 5 t I t z
The analysis starts at the end of the freewheeling interval PP,
hence the initial conditions are as follows:
switches Sap (transistor) and Sbp (diode) are conducting
the output rectifier is freewheeling, thus shorting the transformers magnetizing inductor.
immediately after opening switch Sap at to fig. 6a represents
the power circuit. With assumption e) iLFand uca result as:
(1)
i , ( t ) = I , cos(wo(t - t o ) ) - I ~ o
and
ucu(t)= U,,, - Z o l , s i n ( ~ ~ ( t - t ~ ) ) .
(2)
Mode A: Under heavy load, i.e. if I , , is large, Can is rapidly
discharged and the freewheeling diode of San starts to
conduct at
1
tlu = -arcsin
i L s ( t )= -zZ2 sin(w2T)-lxb cos(w2t")+ I,,,
(7)
and ucb(t) = U,, c0s(02t")-Z2lxb sin(o2t") .
(8)
I
where
t:=t-t3.
Applying sin(x) z x and cos(x)
and (8) to
i,(t) = -~
Ls+ Lm
2
f for x << f simplifies (7)
t" - ( I a p + ILmO)
and uCb(t)= Ui,- - tIxb
.
Cb
The commutation NP-NN is completed as soon as
comes zero at
uin
tq =-
(10)
UCb
be-
Cb+t3 .
Ixb
(3)
WO
with circuit topology depicted in fig. 6b being active until
when iLr5 I, - lap. The freewheeling of the output rectifier
is terminated.
Mode B: If lap
is small, the freewheeling of the output rectifier ends before uca becomes zero. Thus, starting at
iLFand iLocompete charging and discharging Ca, as shown in
fig. 6c.
E. Freewheeling interval NN: t-, I t I
t5
During freewheeling intervals no power is transferred to the
output and the output current is freewheeling. Theoretically,
ih and ib would sustain their level at the end of commutation
interval NP-NN. In reality, however, the voltage drop across
the rectifier diodes will occur on the transformers primary
winding and add to the voltage drop across the switches to
reduce the currents, and, more inconvenient, cause losses! (A
lot of work has been invested on reducing these freewheeling
losses. For cost-sensitive applications with wide input voltage
range none of them seemed suitable. E.g. it is not possible to
use saturable inductors for inductors La and Lb, because even
small variations in their part value or the input voltage would
cause large variations of their peak current.)
617
F. Symmetries: 1 5 I t I t l o
The FB-PRCP is symmetrical i n regard to upper and lower
half of the legs, but, due to the driving scheme, there is no
symmetry between leg A and leg B.
The symmetry within the legs can be exploited to derive
waveforms for the time interval r5 to r/,]. Because the auxiliary
capacitors (CI...C4) balance the currents ,i and iLh, some
steady state peak values can be approximated as
transition topologies and have [nus been investigated i n more
detail. Fig. 7 shows the principle waveforms of ucu and iLrfor
heavy and light load modes.
In order to generate proper gating signals it is important to
find the respective mode for a given I,. When the mode of
operation changes from A to B, t l , and t l h are equal, and so
are rlh and t2h. Using (5)and (6) yields
where Iuphmis the limit of Iq between modes A and B.
As (12) is based on the assumption of short commutation
intervals, the currentli have to be chosen somewhat higher to
compensate for losses due to finite commutation time.
G. DC voltage tnznsfer ratio
For modes A and EI the average voltage across the rectifier
output can be derived as
r +dt
V, ( 0 ) = k .
- t l U ] for mode A and
(13)
- [%-
-
2 K,,
Lm
Ts n L s + Lm
is constant for a fixed input voltage. As t l b , tru, tZhrand t4 all
depend on the load current, ( 1 3) and (14) are evaluated numerically.
IV. DESIGN ASPECTS
The existence of two load dependent operation modes A
and B makes the FB-PRCP different from other resonant
for mode B, where k =
d) t2 ... 9
In mode B the inductor current I k o has to ensure the commutation of leg A. The critical Iup is denoted Iaphm,because at
lower output current Ih,, and at higher current iLs will accelerate the discharging of Ca. Thus, Ik0 has to be larger than
( Iuptim-IDnO),which leads to
a simple but important equation. In the original pseudoresonant full-bridge the peak current through La must be
above the maximum primary transformer current to achieve
soft switching. In the FB-PRCP, however, it can be chosen to
meet requirements set by the designer. Selecting a large Cu,
thus represents an effective turn off snubber, gives a low Z,,
affecting a high Izo on one hand, hence Ih0 must be high. On
the other hand, Iolo can be lowered by selecting a larger Ls for
the price of a smaller volt-seconds-product at the transformers
primary winding. Though a larger Ls will have a negative
effect on the converters overall performance, it limits dildt for
the transformers primary winding and the rectifier diodes as
positive side effects.
t) ' 4
e) '3 ... t4
...
'5
Fig. 6 Topological network states for the FB-PRCP and notation (network states for ts...tlo are not shown):
f5.4
Commutation NN-PN; initiated when
Commutation PP-NP; initiated when modulator
modulator opens Sun(with Sun comprising
opens Sup (with Sup comprising Qup and Dup)
Qan and Dan) at t5 = %Ts; part 1
at to; part 1
Commutation NN-PN; part 2; heavy load
Commutation PP-NP; part 2; heavy load
t6a..t7u
Commutation NN-PN; part 2; light load
Commutation PP-NP; part 2; light load
r6k..r7h
Energizing PN
Energizing NP
t7 ...t#
Commutation PN-PP; initiated when
Commutation NP-NN; initiated when modulator t8...t9
modulator opens Shn at t8 = %Ts + dt
opens Sbp at t j = dt
Freewheeling PP
Freewheeling NN
tg ...t / o
618
p,y:.;
large and switches of leg B should be gated as soon as the
voltage across them is zero.
vm .....,........ ....... ..
,_,,._
V. SIMULATIONS
t
v.m- xI a 20
.....4 ....: LI...CL..
I
I
Extensive PSpice simulations were performed before
building a prototype WPS for 350AllOkW from 400VAC
three-phase mains, out of which one is depicted i n fig. 9.
Circuit data were:
La= I20pH. Lb= 400pH. Cup= Can= Cbp= Cbn= IOnF, n= 12,
Ls= 12pH. Lm= 0.7mH,fs= 4OkHz, V,,= 540VDC
1.- - - - - - - II
I
I
2 '0,
..
I
2.0,
Fig. 7 Typical waveforms during commutation PP-NP
Delay times tca and f c b
First, switches and diodes were modeled by conduction losses
and linearized switching characteristics. Losses in passive
components were modeled by appropriate e m . Second,
manufacturer provided semiconductor models were used
using IGBT-half-bridge modules SKM7SGBIOfBN and
MUR20020CT rectifier diodes.
Delay times tca and tCh have to be adjusted according to the
load current. Hence, to maintain ZVS of leg A, Qan may not
be gated before ucu is zero; using (3) and (6) gives this minimum time delay tCa. An upper limit for rCa is given, because
the Qan must be gated before the anti-parallel diode Dan
stops to conduct. Typical ,,may-be-gated" and ,,has-to-begated" curves for legs A and B are depicted in fig. 8 for the
prototype converter as specified in chapter V. The gap between ,,may-be" and ,,has-to-be" of leg A can be controlled
by the inductance of La. The lower La, the bigger the gap,
and the larger the window of tolerable part values. Selecting a
wide gap even allows to use a constant tCa.
-6OOV
1
-
360~s
Fig. 9
*..-....
/ "may be gated" leg B
-.._ *...._.
-e...
....._..*. ......._
..__.+*..
3701.15
380~s
390~s
400~s
Simulation results at steady state and at nominal
operating point
VI. EXPERIMENTAL RESULTS
A prototype was built with part values given in chapter V,
I
I
I
1
b
using the UC387.5 controller and analog circuitry for the load
1OOA
200A
300A load current[A] dependent adjustment of the dead-times. Fig. 10a to fig. 10d
Fig. 8 Range for gating signals
show waveforms of transistor Voltages vca and VCb. resulting
voltage
at the transformers primary winding "Tr* and primary
For leg B there is only a minimum delay time t C b , Because
Current ~ L Fat different load conditions. Both vca and V C b beZrh depends o n Zq, the duration of the commutation interval
have like predicted by the analysis, providing ZVS for the
NP-NN can be estimated from (1 1) as
IGBTs. Note, the slight increase of ih during energizing
U,, .Cb
U,, .Cb
St4 -t3 5
.
(17)
states, mostly caused by the magnetizing current i b . A drop
ILb3 I L m O -IIapMar
ILb3 + I,,
of ih can be observed during freewheeling intervals, as preThe bigger the peak resonant current ILb3 the smaller the dicted by simulations. Values for diLr/dt are largely limited
maximum commutation time. Note, that based o n Zho the due to Ls, and almost n o voltage overshoot of the leg voltages
circuit would work even without any Lb. The commutation
is visible. Fig. 1Oe shows the zoomed commutation PP-NP,
time, however, would be very long and limit the usable range with ih reversing its direction before uco decays to zero. Nevfor dt, so that it is not suitable for proper circuit operation. ertheless, Cu is discharged and ZVS is maintained. Finally, in
Therefore, to obtain a large range for dr, Lb should not be too fig. 10f transistor voltage ucu and the auxiliary inductor cur"may be gated" leg A
**'*...
619
VIII. ACKNOWLEDGEMENT
The authors gratefully acknowledge the support by
Cloos SchweiRtechnik, Haiger, Germany.
IX. REFERENCES
[I] L. Malesani a. o., “Electronic Welder with High
Frequency Resonant Inverter, ” IEEE Trans. Industry
Applicutions, vol. 31, no. 2 , March/April 1995,
pp. 273-279.
[2] T.-F. Wu, a. o., “Analysis and Design of Variable
Frequency and Phase-Shift Controlled Series Resonant Converter Applied for Electric Welding Machines,” in Proceedings of the 1995 IECON, pp. 656661.
[3] P.C. Theron, a. o., “Welding power supplies using
Partial Series Resonant Converter,” in Proceedings
ofthe 1995 IECON, pp. 1319-1321.
[4] H. Pollock, J.O. Flower, “Series-parallel loadresonant converter for controlled-current arc welding
power supply,“ in IEE Proceedings-Electric Power
Applications, vol. 143, iss. 3, May 96, pp. 2 1 1-21 8.
[ 5 ] R.L. Steigerwald, a. o., “A comparison of high power
DC-to DC soft-switched converter topologies,” in
Proceedings of ?he 1994 IAS, pp. 1090-1096.
[6] R. Fanington, a. o., “Analysis of reactive power in
resonant converters,” in Proceedings of rhe 1992
PESC, pp. 191-205.
[7] O.D. Patterson, a. o., “Pseudo-Resonant Full-Bridge
DClDC Converter, ” IEEE Trans. on Power Electronics, vol. 6, no. 4, Oct. 1991.
Fig. 10 Measurements: a) at load current 20A, b) I50A, c) 250A, d ) 375A
(resulting in ari output power of 13kW), e ) commutation PP-NP at
20A, f) inductor current iL, at 1SOA
rent ib is shown, which proves that
during commutation intervals of leg A.
i
k
is almost constant
VII. CONCLUSION
A full-bridge topology for the gas metal arc welding process
is introduced featuring high power density in conjunction
with low costs. This is achieved by using resonant transition
switching of the four transistors and relatively low reactive
power processing due to parallel connected passive networks,
which are only modestly charged by the energy for commutations. The low cost fe.iture arises momentarily by omitting the
symmetrical blocking devices incl. drives of the branch for
commutation in the auxiliary resonant pole converter (whose
costs need to be weighted against the passive means for
commutation.) The so called passive resonant commutated
poles full-bridge (FB-PRCP)
is facilitated by a new low cost
driving scheme adapting the resulting power circuit according
to the load conditions.
620
[8] J.F. Lancaster, The Physics of Welding, Pergamon
Press, I.ed., 1984, p. 150.
[9] R.W. De Donker, J.P. Lyons, “The Auxiliary Resonant Commutated Pole Converter,“ in Proceedings of
the IAS 1990, pp. I 128-1235.