High-Frequency Soft-switching LCC Resonant
Current-fed DC/DC converter with High Voltage Gain
for DC Microgrid Application
Devendra Patil, Akshay K. Rathore, Senior Member, IEEE, Dipti Srinivasan, Senior Member, IEEE, and Sanjib K.
Panda, Senior Member, IEEE
Electrical and Computer Engineering
National University of Singapore
117583 Singapore
eledrp@nus.edu.sg; eleakr@nus.edu.sg; dipti@nus.edu.sg; eleskp@nus.edu.sg
Abstract— This paper proposes a high frequency soft switched
high voltage gain dc/dc converter for DC microgrid application.
The proposed converter employs a half-bridge resonant boost
converter at input and a voltage quadruple circuit at output.
Resonant boost converter is operated at frequency of 150 kHz to
gain advantage of low output voltage ripple and reduced
magnetics. Zero voltage turn-on is achieved for all switches. Zero
current turn-on and turn-off is achieved for all diodes. High
frequency film capacitors increase life time of the converter.
Voltage stress across switches is less and is clamped naturally
without external snubber circuit. Experimental converter rated at
300 W has been designed, and tested to verify the analysis, design
and demonstrate the performance of the proposed converter.
Keywords – High-frequency, Soft-switching,
converter, DC/DC converter, High gain
I.
Current-fed
INTRODUCTION
Microgrid is an evolving area that reduces tension on utility
by locally supplying power to load by acting like a local grid [1].
Recently, DC microgrid is becoming quite popular as several
energy sources i.e. solar photovoltaic (PV), fuel cells generate
DC power. Therefore, it requires less power conversion stages
and makes the system more efficient and reliable [2].
Fig. 1 shows a typical architecture of the DC microgrid. It has
a DC bus of 380V and various energy sources and energy
storage connected to DC bus. Generally, PV array generates low
voltage of around 40-50V. To interface PV with DC bus of
380V a gain of almost 10 times is required. Boost converter is
normally used for stepping up the DC voltage. However it has to
operate at extreme duty cycle in order to obtain high gain. High
duty cycle leads to high voltage stress across semiconductor
devices and therefore, the efficiency of the boost converter at
higher duty cycle is normally low [3].
In literature, several high voltage gain converters have been
reported [4-11]. Most of the hard switching converters have
problems of low efficiency, bulky magnetics due to low
switching frequency and high electromagnetic interference
(EMI) [12]. Soft-switching high voltage gain converters are
promising due to low EMI, high efficiency, and smaller
magnetics [12-16].
k,(((
In literature different soft-switching converters have been
reported. Park et.al [12] proposed a soft-switching dc/dc
converter, which has zero voltage switching (ZVS) turn-on and
zero current switching (ZCS) turn-off for diode. However
voltage stress across devices is high and voltage gain of the
converter is increased by 2x as compared to conventional boost
converter. Choi et. al [14] proposed a modified PWM method
to reduce the turn off current stress of switches and to achieve
ZCS for the output diode. However, the turn-off current is quite
high and lead to significant switching losses and limit the high
frequency operation of the converter. An integrated boost
converter with stacked structure at the output to increase the
output voltage was proposed in [17], which has features of ZVS
turn-on of the switches. Nevertheless, switching frequency
cannot be increased due to turn off switching losses of switch
and reverse recovery losses of the diode. A quasi resonant high
step-up dc/dc converter was proposed in [18] with reduced
turn-off switch current. However, voltage stress across devices
is higher and is not clamped. This paper proposes a
soft-switching LCC resonant based high voltage gain dc/dc
converter. The proposed converter has the following merits: (i)
ZVS turn-on for both switches, (ii) ZCS turn-on and turn-off for
all diodes, (iii) low output voltage ripple and electrolytic
capacitor-less converter, (vi) LCC resonant circuit may have a
gain above unity, (vii) Switch voltage is clamped without
additional snubber circuit.
II. OPERATION AND ANALYSIS OF PROPOSED CONVERTER
Fig. 2 shows the proposed converter, which is composed of (a)
integrated boost LCC resonant switching cell to integrate
source while achieving soft-switching of semiconductor
devices and (b) voltage multiplier circuit to increase the overall
voltage gain of the converter. The half-bridge LCC resonant
switching cell has two switches M1 and M2 and both are
switched complementarily with appropriate dead time. The
resonant inductor, Lr, parallel capacitor, CP and half bridge
capacitor, C4 forms a LCC resonant circuit. The output circuit is
a voltage quadruple circuit, which increases the gain by 4x. The
key steady-state operating waveforms of the proposed
converter are shown in Fig. 3.
Fig. 1. Typical configuration of DC microgrid.
Fig. 2. Proposed soft switched dc/dc converter
The converter has four operating modes explained below. The
equivalent circuits of operation are shown in Fig. 4.
Interval 1 (Fig. 4(a); t0 < t <t1): In this interval switches M1 and
M2 are off. At t=t0 parasitic capacitance C1 start discharging and
C2 start charging. At output side, diode D4 is conducting to
charge capacitor C5 and diode D6 is conducting to charge
capacitor C8. Power is transferred to the load by energy stored
in the output capacitor C7 and C8. At t=t1 capacitor C1 is
discharged completely and capacitor C2 is charged to Vin/1-D.
(a)
(b)
(c)
(d)
Fig. 3. Theoretical waveform of the proposed converter
At output side, capacitor C5 and C8 are charging. Output
capacitor C7 and C8 continue to supply power to load.
Interval 3 (Fig. 4(c); t2 < t <t3): In this interval, M1 is turned on
with ZVS. Resonant inductor current iLr starts flowing through
switch M1, capacitor C3, resonant inductor Lr and capacitor CP.
Input inductor current(iL) start decreasing linearly. At end of
this mode diode D4 and D6 turns off with zero current. The
current through various components is given by
ܸ஼ ሺ‫ݐ‬ଶ ሻെܸ஼௣ ሺ‫ݐ‬ଶ ሻ
݅௅௥ ൌ య
ή •‹ ‫ݓ‬௥ ሺ‫ ݐ‬െ ‫ݐ‬ଶ ሻሺͳሻ
ܼ௥
(e)
where, ‫ݓ‬௥ ൌ ට
஼య ା஼ು
௅ೝ ஼ೝ ஼య
௅ೝ ሺ஼య ା஼ು ሻ
and ‫ݖ‬௥ ൌ ට
஼య ஼ು
ܸ௜௡ െ ܸ஼య ሺ‫ݐ‬ଶ ሻ െ ܸ஼ర ሺ‫ݐ‬ଶ ሻ
ሺ‫ ݐ‬െ ‫ݐ‬ଶ ሻሺʹሻ
݅௅ ሺ‫ݐ‬ሻ ൌ ݅௅ ሺ‫ݐ‬ଶ ሻ െ
‫ܮ‬
(f)
Interval 4 (Fig. 4(d); t3 < t <t4): During this interval the switch
M1 is turned on. At output side all diodes are reverse biased.
The power to load is supplied by energy stored in output
capacitor C7 and C8. At t=t4 switch M1 is turned off.
Interval 5 (Fig. 4(e); t4 < t <t5): During this interval, switch M1
and M2 are turned off. In this interval, parasitic capacitor C1 is
charged and parasitic capacitor C2 is discharged. At output side
no diode is conducting and no power is transferred from input
to output. Power to load is supplied by capacitor C7 and C8. At
end of this interval capacitor C2 is discharged completely and
capacitor C1 is charged to Vin/1-D.
(g)
Interval 6 (Fig. 4(f); t5 < t <t6): At t=t5 diode D2 is forward
biased by difference of resonant inductor current iLr and input
inductor current iL. At output side all diodes are reverse biased
conducting.
(h)
Interval 7 (Fig. 4(g); t6 < t <t7): In this interval, switch M2 is
turned on with ZVS. Input inductor L starts charging linearly
with a slope. Resonant inductor Lr, capacitor CP and C4
resonant together. The resonant current through Lr is given by
݅௅௥ ൌ െ
௏಴ర ሺ௧ల ሻା௏಴೛ ሺ௧ల ሻ
௓ೝ
•‹ ‫ݓ‬௥ ሺ‫ ݐ‬െ ‫ ଺ݐ‬ሻሺ͵ሻ
Current through input inductor is given by
ܸ௜௡
ሺ‫ ݐ‬െ ‫ ଺ݐ‬ሻ(4)
݅௅ ሺ‫ݐ‬ሻ ൌ ݅௅ ሺ‫ ଺ݐ‬ሻ ൅
‫ܮ‬
(i)
Interval 8 (Fig. 4(h); t7 < t <t8): In this interval, switch M2 is in
on state. At t=t7 diode D3 gets forward biased. Energy stored in
capacitors CP and C5 is transferred to C7. Simultaneously,
output capacitor C7 and C8 supply power to load.
(j)
Fig. 4. Different operating modes of the proposed converter for the interval
shown in Fig.3
Interval 2 (Fig. 4(b); t1 < t <t2): At t=t1 parasitic capacitance
C1 is completely discharged and parasitic capacitance C2 is
completely charged. The difference of input inductor current iL
and resonant inductor current iLr passes through anti-parallel
diode D1 of switch M1 causing zero voltage across switch M1.
Interval 9 (Fig. 4(i); t8 < t <t9): During this interval, input
inductor L is charging through switch M2. At output side diode
D5 gets forward biased at t=t8.At this instant, capacitor CP starts
charging capacitor C6. Simultaneously, capacitor CP and C5
charges capacitor C7. At end of this interval diode D5 and D3 are
turned off with zero current.
Interval 10 (Fig. 4(j); t9 < t <t10): In this interval all output
diodes are in off state and power to load is transferred by output
capacitor C7 and C8. Switch M2 is in on state and input inductor
L is storing energy in it. At t=t10 switch M2 is turned off.
III. VOLTAGE CONVERSION RATIO
The total voltage gain of the converter is summation of the gain
of boost converter, LCC resonant cell, and voltage quadruple.
In case of unity gain of the LCC resonant cell, the total output
voltage is given by:
4Vin
(5)
VO
1 D
The voltage gain with respect to duty cycle is plotted in Fig. 5.
It is clear that the proposed converter has 2x higher gain than
conventional circuit [12].
TABLE I. SIMULATION AND EXPERIMENTAL PARAMETERS OF
PROPOSED CONVERTER
Parameters
L
C3 and C4
CP
Output voltage
PO, System power rating
Input voltage, Vin
Lr
C5 and C6
C7, C8 and C9
Switching frequency, fs
C3
M1 and M2
D1 and D2
Values
156ȝH
1ȝF
0.1ȝF
380V
300W
42V
13ȝH
10ȝF
20ȝF
150kHz
10ȝF
IPB200N25N3G
STTH30R04W
The design equations are derived to determine the components’
ratings.
1) Average current through
input inductor is given by
ܲை
‫ܫ‬௜௡ ൌ
ൌ ͹ǤͳͶʹ‫ܣ‬ሺ͸ሻ
ߟܸ௜௡
2) Duty cycle is selected at input voltage, i.e., Vin =42V and
full load based on maximum switch voltage rating.
‫ܦ‬ൌ
At input voltage D=0.5578.
௏೚ ିସ௏೔೙
௏೚
ሺ͹ሻ
3) Value of input boost inductor is given by
ܸ௜௡ ‫ܦ‬
‫ܮ‬ൌ
ሺͺሻ
ȟ‫ܫ‬௜௡ ݂௦
where, ǻ Iin is the boost inductor ripple current. For,
ǻIin=1A, L=156μH.
4) In order to determine the value of resonant inductor Lr the
ZVS condition for switch M1 and M2 has to be satisfied. The
value of resonant Lr for achieving ZVS condition for M1 is
given by
ଶ
ܸ
ሺ‫ܥ‬ଵ ൅ ‫ܥ‬ଶ ሻ ቀ ௜௡ ቁ ͳ െ ‫ ܦ‬ሺͻሻ
‫ܮ‬௥ ൐
ଶ
െ ݅௥ ሺ‫ݐ‬ଵ ሻଶ ‫ܫ‬௅௔௩௚
ZVS condition for switch M2 gives the value of Lr as
ଶ
ܸ
ሺ‫ܥ‬ଵ ൅ ‫ܥ‬ଶ ሻ ቀ ௜௡ ቁ
ͳ െ ‫ ܦ‬ሺͳͲሻ
‫ܮ‬௥ ൐
ଶ
‫ܫ‬௅௔௩௚
െ ݅௥ ሺ‫ݐ‬ଷ ሻଶ
Gain
Fig. 5. Voltage gain versus duty ratio of proposed converter and converter in
[12].
IV.
DESIGN OF CONVERTER
In this Section, converter design procedure is illustrated by a
design example for the following specifications: input
voltage, Vin=42V, output voltage VO=380V, output power,
PO=300W, switching frequency, fs=150kHz. The following
assumption are made:
(a) Efficiency of the converter is assumed 100%
(b)Voltage ripple across capacitor C3, C4, C5, C6, C7 and C8 to
be negligible.
(c) All the components are ideal and lossless.
Fig. 6 Graph showing various values of Lr to ensure ZVS for varying load
condition for switch M1 and M2
Equations (9) and (10) helps get the value of resonant inductor
Lr required to obtain ZVS for both switches M1 and M2. A curve
between the value of Lr required to achieve ZVS turn-on of
switch M1 and M2 and load power is plotted in Fig. 6. It should
be observed that high value of Lr is required to achieve ZVS at
light load. Therefore, it is difficult to achieve ZVS for switch
M1 at lower power level. As compared to M1, the value of Lr
required to achieve ZVS for low power level is quite less, so it
is easy to operate switch M2 in ZVS condition over wide load
range. From the plot value of Lr= 13μH is chosen to get ZVS
for switch M1 for load greater than 30%.
5) The resonant frequency of the LCC is given by
‫ݓ‬௥ ൌ ඨ
‫ܥ‬ଷ ൅ ‫ܥ‬௉
ሺͳͳሻ
‫ܮ‬௥ ‫ܥ‬௉ ‫ܥ‬ଷ
For resonant frequency of 146.4 kHz the C3=C4=1μF and
CP=0.1μF.
V.
(c)
SIMULATION AND EXPERIMENTAL RESULTS
The designed converter rated at 300 W was first simulated
using PSIM 9.03 to verify the analysis and design of the
proposed converter. Later, an experimental prototype was
designed and developed in the laboratory to demonstrate the
performance of the converter. The components’ values are
tabulated in Table I. Simulation waveforms match closely with
the theoretical operating waveforms shown in Fig. 3. When
both switches, M1 and M2 are in off state the difference of
current, iL1 and iLr flows through anti-parallel diode, D1 before
the switch, M1 is turned-on, causing ZVS turn-on as shown in
Fig. 7(a). Fig. 7(b) shows that the anti-parallel diode of the
switch, M2 conducts prior to its conduction resulting in its ZVS
turn on. The voltage across diode D5 is almost trapezoidal in
shape due to LCC resonant circuit and current through diode
naturally commute to zero with ZCS and no turn-off recovery
losses as shown in Fig. 7(c). Fig. 7(d) show the varying voltage
across capacitor, CP and the 90° phase shift leading current
through resonant inductor, Lr. Input voltage of 42V is given to
the converter to get output voltage of 380V with duty cycle of
M1 and M2 is 0.6 as shown in Fig.7(e).
(d)
(e)
Fig. 7. Simulation waveforms at Vin =42V and full load: (a) iM1, VM1 and VGSM1
(b) iM2, VM2 and VGSM2, (c) diode current VD5 and iD5 , (d) voltage across parallel
capacitor, VCP and current through resonant inductor current, iLr ,(e) )Input
voltage, boost inductor current, iL1 and output voltage, VO.
(a)
Fig. 8. Experimental prototype of proposed dc/dc C converter.
(b)
Fig. 8 shows the experimental prototype developed in the
laboratory. The details of the experimental converter are as
follows. MOSFET’s, IPB200N25N3G (M1 and M2); Diode,
STTH30R04W (D3-D6); Boost inductor, L1: CM610125 MPP
core, 17 turns, measured inductance =130μH; Resonant
inductor, Lr: PC47RM14Z-12, RM core, 6 turns. IR21814
driver ICs are used as a gate driver for gating the MOSFET’s.
The components’ ratings along with the selected components
are provided in Table I.
The converter is tested for input voltage of 42V at full load. The
experimental results are shown in Fig. 9. Fig. 9(a) shows the
ZVS turn on of M1. The anti-parallel body diode of switch M1 is
conducting before the switch starts conducting. Similarly, Fig.
9(b) confirms the ZVS operation of switch, M2. Output diodes
of voltage quadruple circuit turns-on and turn-off in ZCS
condition, thereby recovery losses are negligible as is shown in
Fig. 9(c). Fig. 9(d) shows the gate signal, drain source voltage
and drain current of M2. The resonant inductor current leads
resonant capacitor current by 90° and it is confirmed in Fig.
9(e). Fig. 9(f) shows the input voltage of 42V is stepped-up to
380V with small voltage ripple in the output voltage. Boost
inductor ripple current is also small.
(a)
(b)
(c)
(d)
(e)
(f)
Fig.9. Experimental waveforms at Vin =42V and full load: (a) VM1 (50 V/div) and iM1 (5 A/div), (b) VM2(50 V/div) and iM2 (5 A/div), (c) diode current VD5 (100V/div)
and iD5 (5A/div), (d) Gate source voltage across M2, VGSM2 (20V/div), VM2 (50V/div) and iM2 (5 A/div) (e)voltage across parallel capacitor, VCP (50V/div) and current
through resonant inductor current, iLr (10A/div) and (f) Input voltage, Vin (20V/div), boost inductor current, iL1 (2A/div) and output voltage, VO (200V/div).
[4]
[5]
[6]
[7]
[8]
Fig. 10. Plots of efficiency versus output power obtained from experiments.
Fig. 10 shows measured efficiency for different load conditions
at Vin=42V. While maintaining soft-switching of the devices,
the converter is able to achieve 92.5% efficiency at rated load.
Peak efficiency of 94% is obtained. Above 89% efficiency is
obtained down to 10% load.
VI. CONCLUSION
[9]
[10]
[11]
A high efficiency high frequency and high voltage gain dc/dc
converter is presented for DC microgrid application. The
proposed converter achieves ZVS for switches and ZCS for
diodes over a wide load range. Device voltage is clamped
without any external snubber. The proposed converter has the
following features: (i) ZVS turn-on for both switches, (ii)
Reduced turn-off current for both switches, (iii) ZCS turn-on
and turn-off for all diodes, (iv) High frequency magnetics, (v)
low output voltage ripple and electrolytic capacitor-less
converter, (vi) LCC resonant circuit has a gain above unity,
(vii) Switch voltage is clamped. Detailed steady-state
operation, analysis and design of the converter have been
presented. Experimental results are demonstrated for 300W
prototype. The theoretical analysis and simulation results
match with experimental results closely. Soft-switching (ZVS
of switches and ZCS of diodes) have been demonstrated as
claimed. High efficiency is obtained over wide load range.
[12]
[13]
[14]
[15]
[16]
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