LT1229/LT1230
Dual and Quad 100MHz
Current Feedback Amplifiers
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FEATURES
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DESCRIPTIO
The LT®1229/LT1230 dual and quad 100MHz current
feedback amplifiers are designed for maximum performance in small packages. Using industry standard pinouts,
the dual is available in the 8-pin miniDIP and the 8-pin SO
package while the quad is in the 14-pin DIP and 14-pin SO.
The amplifiers are designed to operate on almost any
available supply voltage from 4V (±2V) to 30V (±15V).
100MHz Bandwidth
1000V/µs Slew Rate
Low Cost
30mA Output Drive Current
0.04% Differential Gain
0.1° Differential Phase
High Input Impedance: 25MΩ, 3pF
Wide Supply Range: ±2V to ±15V
Low Supply Current: 6mA Per Amplifier
Inputs Common Mode to Within 1.5V of Supplies
Outputs Swing Within 0.8V of Supplies
These current feedback amplifiers have very high input
impedance and make excellent buffer amplifiers. They
maintain their wide bandwidth for almost all closed-loop
voltage gains. The amplifiers drive over 30mA of output
current and are optimized to drive low impedance loads,
such as cables, with excellent linearity at high frequencies.
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APPLICATIO S
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Video Instrumentation Amplifiers
Cable Drivers
RGB Amplifiers
Test Equipment Amplifiers
The LT1229/LT1230 are manufactured on Linear
Technology’s proprietary complementary bipolar process.
For a single amplifier like these see the LT1227 and for
better DC accuracy see the LT1223.
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
Loop Through Amplifier Frequency
Response
Video Loop Through Amplifier
R G1
3.01k
R F1
750Ω
R G2
187Ω
10
R F2
750Ω
0
NORMAL SIGNAL
–
3.01k
VIN –
1/2
LT1229
+
–
3.01k VIN+
12.1k
1/2
LT1229
+
12.1k
BNC INPUTS
HIGH INPUT RESISTANCE DOES NOT LOAD CABLE EVEN
WHEN POWER IS OFF
VOUT
GAIN (dB)
–10
–20
–30
1% RESISTORS
WORST CASE CMRR = 22dB
TYPICALLY = 38dB
–40
VOUT = G (VIN+ – VIN – )
R F1 = RF2
–60
R G1 = (G – 1) RF2
R F2
RG2 =
G–1
COMMON MODE SIGNAL
–50
10
100
1k
10k
100k
1M
10M 100M
FREQUENCY (Hz)
LT1229 • TA02
TRIM CMRR WITH RG1
LT1229 • TA01
1
LT1229/LT1230
W W
W
AXI U
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ABSOLUTE
RATI GS (Note 1)
Supply Voltage ...................................................... ±18V
Input Current ...................................................... ±15mA
Output Short Circuit Duration (Note 2) ......... Continuous
Operating Temperature Range
LT1229C, LT1230C ............................... 0°C to 70°C
LT1229M, LT1230M (OBSOLETE).. –55°C to 125°C
Storage Temperature Range ..................–65°C to 150°C
Junction Temperature
Plastic Package .............................................. 150°C
Ceramic Package (OBSOLETE) ................ 175°C
Lead Temperature (Soldering, 10 sec.)................. 300°C
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PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
TOP VIEW
OUT A
1
–IN A
2
8
V+
7
OUT B
6
–IN B
5
+IN B
A
+IN A
3
V–
4
B
LT1229CN8
LT1229CS8
S8 PART MARKING
N8 PACKAGE
S8 PACKAGE
8-LEAD PLASTIC DIP 8-LEAD PLASTIC SOIC
TJ MAX = 150°C, θJA = 100°C/W (N8)
TJ MAX = 150°C, θJA = 150°C/W (S8)
1229
J8 PACKAGE
8-LEAD CERAMIC DIP
TJ MAX = 175°C, θJA = 100°C/W (J8)
ORDER PART
NUMBER
TOP VIEW
OUT A
1
–IN A
2
+IN A
3
V+
4
+IN B
5
–IN B
6
OUT B
7
14 OUT D
13 –IN D
A
D
12 +IN D
11 V –
ORDER PART
NUMBER
LT1230CN
LT1230CS
10 +IN C
B
C
9
–IN C
8
OUT C
N PACKAGE
S PACKAGE
14-LEAD PLASTIC DIP 14-LEAD PLASTIC SOIC
TJ MAX = 150°C, θJA = 70°C/W (N)
TJ MAX = 150°C, θJA = 110°C/W (S)
J PACKAGE
14-LEAD CERAMIC DIP
TJ MAX = 175°C, θJA = 80°C/W (J)
ORDER PART
NUMBER
LT1230MJ
LT1230CJ
LT1229MJ8
LT1229CJ8
OBSOLETE PACKAGE
OBSOLETE PACKAGE
Consider the N Package for Alternate Source
Consider the N Package for Alternate Source
Consult LTC Marketing for parts specified with wider operating temperature ranges.
2
LT1229/LT1230
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Each Amplifier, VCM = 0V, ±5V ≤ VS = ±15V, pulse tested unless
otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
TA = 25°C
MIN
TYP
MAX
UNITS
±3
±10
±15
mV
mV
●
Input Offset Voltage Drift
IIN
+
Noninverting Input Current
TA = 25°C
±0.3
±3
±10
µA
µA
±10
±50
±100
µA
µA
●
IIN–
Inverting Input Current
µV/°C
10
●
TA = 25°C
●
en
Input Noise Voltage Density
f = 1kHz, RF = 1k, RG = 10Ω, RS = 0Ω
3.2
nV/√Hz
+in
Noninverting Input Noise Current Density
f = 1kHz, RF = 1k, RG = 10Ω, RS = 10k
1.4
pA/√Hz
–in
Inverting Input Noise Current Density
f = 1kHz
RIN
Input Resistance
VIN = ±13V, VS = ±15V
VIN = ±3V, VS = ±5V
CIN
Input Capacitance
Input Voltage Range
●
●
VS = ±15V, TA = 25°C
●
VS = ±5V, TA = 25°C
●
CMRR
Common Mode Rejection Ratio
Inverting Input Current
Common Mode Rejection
PSRR
VS = ±15V, VCM = ±13V, TA = 25°C
VS = ±15V, VCM = ±12V
VS = ± 5V, VCM = ±3V, TA = 25°C
VS = ±5V, VCM = ± 2V
VS = ±15V, VCM = ±13V, TA = 25°C
VS = ±15V, VCM = ±12V
VS = ±5V, VCM = ±3V, TA = 25°C
VS = ±5V, VCM = ±2V
●
●
pA/√Hz
25
25
MΩ
MΩ
3
pF
±13
±12
±3
±2
±13.5
V
V
V
V
55
55
55
55
69
±3.5
2.5
2.5
●
●
Noninverting Input Current
Power Supply Rejection
VS = ±2V to ±15V, TA = 25°C
VS = ±3V to ±15V
●
Inverting Input Current
Power Supply Rejection
VS = ±2V to ±15V, TA = 25°C
VS = ±3V to ±15V
●
60
60
dB
dB
dB
dB
69
●
VS = ±2V to ±15V, TA = 25°C
VS = ±3V to ±15V
Power Supply Rejection Ratio
32
2
2
10
10
10
10
80
µA/V
µA/V
µA/V
µA/V
dB
dB
10
50
50
nA/V
nA/V
0.1
5
5
µA/V
µA/V
3
LT1229/LT1230
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Each Amplifier, VCM = 0V, ±5V ≤ VS = ±15V, pulse tested unless
otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
AV
Large-Signal Voltage Gain, (Note 3)
VS = ±15V, VOUT = ±10V, RL = 1k
VS = ±5V, VOUT = ±2V, RL = 150Ω
ROL
Transresistance, ∆VOUT/∆IIN–, (Note 3)
VS = ±15V, VOUT = ±10V, RL = 1k
VS = ±5V, VOUT = ±2V, RL = 150Ω
VOUT
Maximum Output Voltage Swing, (Note 3)
VS = ±15V, RL = 400Ω, TA = 25°C
MIN
TYP
●
●
55
55
65
65
dB
dB
●
●
100
100
200
200
kΩ
kΩ
±12
±10
±3
±2.5
±13.5
30
65
125
mA
6
9.5
11
mA
mA
●
VS = ±5V, RL = 150Ω, TA = 25°C
●
IOUT
Maximum Output Current
RL = 0Ω, TA = 25°C
IS
Supply Current, (Note 4)
VOUT = 0V, Each Amplifier, TA = 25°C
Slew Rate, (Notes 5 and 7)
TA = 25°C
SR
Slew Rate
tr
Rise Time, (Notes 6 and 7)
BW
Small-Signal Bandwidth
tr
ts
700
V/µs
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 400Ω
TA = 25°C
2500
V/µs
100
MHz
Small-Signal Rise Time
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω
3.5
ns
Propagation Delay
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω
3.5
ns
Small-Signal Overshoot
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω
0.1%, VOUT = 10V, RF =1k, RG= 1k, RL =1k
15
%
45
ns
Settling Time
10
20
ns
Differential Gain, (Note 8)
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 1k
0.01
%
Differential Phase, (Note 8)
0.01
Deg
Differential Gain, (Note 8)
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 1k
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 150Ω
0.04
%
Differential Phase, (Note 8)
VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 150Ω
0.1
Deg
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: A heat sink may be required depending on the power supply
voltage and how many amplifiers are shorted.
Note 3: The power tests done on ±15V supplies are done on only one
amplifier at a time to prevent excessive junction temperatures when testing
at maximum operating temperature.
Note 4: The supply current of the LT1229/LT1230 has a negative
temperature coefficient. For more information see the application
information section.
Note 5: Slew rate is measured at ±5V on a ±10V output signal while
operating on ±15V supplies with RF = 1k, RG = 110Ω and RL = 400Ω. The
4
300
UNITS
V
V
V
V
±3.7
●
SR
MAX
slew rate is much higher when the input is overdriven and when the
amplifier is operated inverting, see the applications section.
Note 6: Rise time is measured from 10% to 90% on a ±500mV output
signal while operating on ±15V supplies with RF = 1k, RG = 110Ω and RL =
100Ω. This condition is not the fastest possible, however, it does
guarantee the internal capacitances are correct and it makes automatic
testing practical.
Note 7: AC parameters are 100% tested on the ceramic and plastic DIP
packaged parts (J and N suffix) and are sample tested on every lot of the
SO packaged parts (S suffix).
Note 8: NTSC composite video with an output level of 2VP.
LT1229/LT1230
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TYPICAL PERFOR A CE CHARACTERISTICS
Voltage Gain and Phase vs
Frequency, Gain = 6dB
160
6
90
140
GAIN
135
4
180
3
225
2
1
PHASE SHIFT (DEG)
5
VS = ±15V
RL = 100Ω
RF = 750Ω
–2
0.1
160
RF = 500Ω
120
RF = 750Ω
100
80
RF = 1k
60
40
RF = 2k
20
0
1
10
4
6
8
10
14
12
LT1229 • TPC01
21
180
45
160
135
18
180
17
225
16
15
VS = ±15V
RL = 100Ω
RF = 750Ω
12
0.1
1
–3dB BANDWIDTH (MHz)
90
GAIN
PHASE SHIFT (DEG)
VOLTAGE GAIN (dB)
0
19
14
10
100
RF = 250Ω
80
RF = 500Ω
RF = 750Ω
60
RF = 1k
40
RF = 2k
RF = 750Ω
60
RF = 1k
40
RF = 2k
0
2
4
6
8
10
14
12
16
18
0
2
4
6
8
10
14
12
– 3dB Bandwidth vs Supply
Voltage, Gain = 100, RL = 1kΩ
14
–3dB BANDWIDTH (MHz)
16
14
RF = 500Ω
10
RF = 1k
8
6
18
LT1229 • TPC06
90
12
16
SUPPLY VOLTAGE (±V)
RF = 2k
4
RF = 500Ω
12
RF = 1k
10
8
RF = 2k
6
4
2
0
0
2
4
6
8
10
12
14
16
18
SUPPLY VOLTAGE (±V)
LT1229 • TPC07
RF = 500Ω
80
18
PHASE SHIFT (DEG)
VOLTAGE GAIN (dB)
RF = 250Ω
20
2
FREQUENCY (MHz)
18
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
100
16
225
16
120
18
37
100
14
12
140
45
180
VS = ±15V
RL = 100Ω
RF = 750Ω
10
– 3dB Bandwidth vs Supply
Voltage, Gain = 10, RL = 1k
0
38
35
8
LT1229 • TPC03
– 3dB Bandwidth vs Supply
Voltage, Gain = 100, RL = 100Ω
36
6
LT1229 • TPC05
135
10
4
SUPPLY VOLTAGE (±V)
GAIN
1
2
SUPPLY VOLTAGE (±V)
120
0
39
32
0.1
0
0
100
PHASE
RF = 2k
RF = 1k
160
20
42
34
18
140
Voltage Gain and Phase vs
Frequency, Gain = 40dB
33
40
180
LT1229 • TPC04
40
16
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
FREQUENCY (MHz)
41
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
60
– 3dB Bandwidth vs Supply
Voltage, Gain = 10, RL = 100Ω
PHASE
13
80
LT1229 • TPC02
Voltage Gain and Phase vs
Frequency, Gain = 20dB
20
100
SUPPLY VOLTAGE (±V)
FREQUENCY (MHz)
22
RF = 750Ω
120
0
2
0
100
RF = 500Ω
140
20
–3dB BANDWIDTH (MHz)
–1
180
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
–3dB BANDWIDTH (MHz)
0
– 3dB Bandwidth vs Supply
Voltage, Gain = 2, RL = 1k
–3dB BANDWIDTH (MHz)
180
45
PHASE
–3dB BANDWIDTH (MHz)
0
7
8
VOLTAGE GAIN (dB)
– 3dB Bandwidth vs Supply
Voltage, Gain = 2, RL = 100Ω
0
0
2
4
6
8
10
12
14
16
18
SUPPLY VOLTAGE (±V)
LT1229 • TPC08
LT1229 • TPC09
5
LT1229/LT1230
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TYPICAL PERFOR A CE CHARACTERISTICS
Maximum Capacitance Load vs
Feedback Resistor
Total Harmonic Distortion vs
Frequency
10000
10
RL = 1k
PEAKING ≤ 5dB
GAIN = 2
1
0
1
2
0.01
VO = 7VRMS
–70
100
1k
FEEDBACK RESISTOR (kΩ)
10k
Output Short-Circuit Current vs
Junction Temperature
–1.5
–2.0
2.0
V – = –2V TO –18V
1.0
0.5
V–
–50 –25
0
25
50
75
70
–0.5
–1.0
RL = ∞
±2V ≤ VS ≤ ±18V
1.0
0.5
V–
–50 –25
100 125
OUTPUT SHORT CIRCUIT CURRENT (mA)
OUTPUT SATURATION VOLTAGE (V)
COMMON MODE RANGE (V)
LT1229 • TPC12
V+
V + = 2V TO 18V
0
25
75
50
100
Spot Noise Voltage and Current vs
Frequency
40
30
–50 –25
125
0
POWER SUPPLY REJECTION (dB)
en
+in
1
1k
10k
FREQUENCY (Hz)
100k
75 100 125 150 175
Output Impedance vs
Frequency
100
VS = ±15V
RL = 100Ω
RF = RG = 750Ω
60
POSITIVE
40
NEGATIVE
20
0
10k
VS = ±15V
10
1.0
RF = RG = 2k
RF = RG = 750Ω
0.1
0.01
100k
1M
10M
100M
0.001
10k
100k
1M
10M
100M
FREQUENCY (Hz)
FREQUENCY (Hz)
LT1229 • TPC16
50
LT1229 • TPC15
80
10
25
TEMPERATURE (°C)
Power Supply Rejection vs
Frequency
100
100
50
LT1229 • TPC14
LT1229 • TPC13
10
60
TEMPERATURE (°C)
TEMPERATURE (°C)
–in
100
FREQUENCY (MHz)
Output Saturation Voltage vs
Temperature
V+
1.5
10
LT1229 • TPC11
Input Common Mode Limit vs
Temperature
SPOT NOISE (nV/√Hz OR pA/√Hz)
1
100k
FREQUENCY (Hz)
LT1229 • TPC10
–1.0
3RD
–50
–60
10
–0.5
2ND
–40
VO = 1VRMS
0.001
3
VS = ±15V
VO = 2VP-P
RL = 100Ω
RF = 750Ω
AV = 10dB
–30
OUTPUT IMPEDANCE (Ω)
CAPACITIVE LOAD (pF)
VS = ±15V
100
–20
VS = ±15V
RL = 400Ω
RF = RG = 750Ω
DISTORTION (dBc)
VS = ±5V
TOTAL HARMONIC DISTORTION (%)
0.10
1000
6
2nd and 3rd Harmonic
Distortion vs Frequency
LT1229 • TPC17
LT1229 • TPC18
LT1229/LT1230
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TYPICAL PERFOR A CE CHARACTERISTICS
Settling Time to 10mV vs
Output Step
Settling Time to 1mV vs
Output Step
INVERTING
4
4
OUTPUT STEP (V)
6
VS = ±15V
RF = RG = 1k
0
NONINVERTING
8
6
2
10
–2
–4
9
8
INVERTING
SUPPLY CURRENT (mA)
NONINVERTING
8
OUTPUT STEP (V)
Supply Current vs Supply Voltage
10
10
2
VS = ±15V
RF = RG = 1k
0
–2
–4
INVERTING
–6
–6
–8
NONINVERTING
0
20
40
60
80
100
0
4
8
12
16
20
SETTLING TIME (µs)
SETTLING TIME (ns)
125°C
4
175°C
3
1
–10
–10
25°C
6
5
2
NONINVERTING
–8
INVERTING
–55°C
7
0
0
2
4
6
8
10
14
16
18
SUPPLY VOLTAGE (±V)
LT1229 • TPC20
LT1229 • TPC19
12
LT1229 • TPC21
W
W
SI PLIFIED SCHE ATIC
One Amplifier
V+
+IN
–IN
VOUT
V–
LT1229 • TA03
7
LT1229/LT1230
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APPLICATI
S I FOR ATIO
The LT1229/LT1230 are very fast dual and quad current
feedback amplifiers. Because they are current feedback
amplifiers, they maintain their wide bandwidth over a wide
range of voltage gains. These amplifiers are designed to
drive low impedance loads such as cables with excellent
linearity at high frequencies.
Feedback Resistor Selection
The small-signal bandwidth of the LT1229/LT1230 is set
by the external feedback resistors and the internal junction
capacitors. As a result, the bandwidth is a function of the
supply voltage, the value of the feedback resistor, the
closed-loop gain and load resistor. The characteristic
curves of Bandwidth versus Supply Voltage are done with
a heavy load (100Ω) and a light load (1k) to show the effect
of loading. These graphs also show the family of curves
that result from various values of the feedback resistor.
These curves use a solid line when the response has less
than 0.5dB of peaking and a dashed line when the response has 0.5dB to 5dB of peaking. The curves stop
where the response has more than 5dB of peaking.
limited by the gain bandwidth product of about 1GHz. The
curves show that the bandwidth at a closed-loop gain of
100 is 10MHz, only one tenth what it is at a gain of two.
Capacitance on the Inverting Input
Current feedback amplifiers want resistive feedback from
the output to the inverting input for stable operation. Take
care to minimize the stray capacitance between the output
and the inverting input. Capacitance on the inverting input
to ground will cause peaking in the frequency response
(and overshoot in the transient response), but it does not
degrade the stability of the amplifier. The amount of
capacitance that is necessary to cause peaking is a function of the closed-loop gain taken. The higher the gain, the
more capacitance is required to cause peaking. We can
add capacitance from the inverting input to ground to
increase the bandwidth in high gain applications. For
example, in this gain of 100 application, the bandwidth can
be increased from 10MHz to 17MHz by adding a 2200pF
capacitor.
+
VIN
Small-Signal Rise Time with
RF = RG = 750Ω, VS = ±15V, and RL = 100Ω
1/2
LT1229
VOUT
–
RF
510Ω
RG
5.1Ω
CG
LT1229 • TA05
At a gain of two, on ±15V supplies with a 750Ω feedback
resistor, the bandwidth into a light load is over 160MHz
without peaking, but into a heavy load the bandwidth
reduces to 100MHz. The loading has so much effect
because there is a mild resonance in the output stage that
enhances the bandwidth at light loads but has its Q
reduced by the heavy load. This enhancement is only
useful at low gain settings; at a gain of ten it does not boost
the bandwidth. At unity gain, the enhancement is so
effective the value of the feedback resistor has very little
effect. At very high closed-loop gains, the bandwidth is
Boosting Bandwidth of High Gain Amplifier with
Capacitance on Inverting Input
49
46
C G = 4700pF
43
40
GAIN (dB)
LT1229 • TA04
C G = 2200pF
37
34
CG = 0
31
28
25
22
19
1
10
100
FREQUENCY (MHz)
LT1229 • TA06
8
LT1229/LT1230
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APPLICATI
S I FOR ATIO
Capacitive Loads
The LT1229/LT1230 can drive capacitive loads directly
when the proper value of feedback resistor is used. The
graph Maximum Capacitive Load vs Feedback Resistor
should be used to select the appropriate value. The value
shown is for 5dB peaking when driving a 1k load at a gain
of 2. This is a worst case condition; the amplifier is more
stable at higher gains and driving heavier loads. Alternatively, a small resistor (10Ω to 20Ω) can be put in series
with the output to isolate the capacitive load from the
amplifier output. This has the advantage that the amplifier
bandwidth is only reduced when the capacitive load is
present, and the disadvantage that the gain is a function of
the load resistance.
Power Supplies
The LT1229/LT1230 amplifiers will operate from single or
split supplies from ±2V (4V total) to ±15V (30V total). It is
not necessary to use equal value split supplies, however,
the offset voltage and inverting input bias current will
change. The offset voltage changes about 350µV per volt
of supply mismatch, the inverting bias current changes
about 2.5µA per volt of supply mismatch.
Power Dissipation
The LT1229/LT1230 amplifiers combine high speed and
large output current drive into very small packages. Because these amplifiers work over a very wide supply range,
it is possible to exceed the maximum junction temperature
under certain conditions. To ensure that the LT1229 and
LT1230 remain within their absolute maximum ratings,
we must calculate the worst case power dissipation,
define the maximum ambient temperature, select the
appropriate package and then calculate the maximum
junction temperature.
The worst case amplifier power dissipation is the total of
the quiescent current times the total power supply voltage
plus the power in the IC due to the load. The quiescent
supply current of the LT1229/LT1230 has a strong negative temperature coefficient. The supply current of each
amplifier at 150°C is less than 7mA and typically is only
4.5mA. The power in the IC due to the load is a function of
the output voltage, the supply voltage and load resistance.
The worst case occurs when the output voltage is at half
supply, if it can go that far, or its maximum value if it
cannot reach half supply.
For example, let’s calculate the worst case power dissipation in a video cable driver operating on ±12V supplies that
delivers a maximum of 2V into 150Ω.
Pd (MAX) = 2VS IS (MAX) +  VS – VO (MAX) 


(
VO (MAX)
RL
)
2V
150Ω
= 0.168 + 0.133 = 0.301W per Amp
Now if that is the dual LT1229, the total power in the
package is twice that, or 0.602W. We now must calculate how much the die temperature will rise above the
ambient. The total power dissipation times the thermal
resistance of the package gives the amount of temperature rise. For the above example, if we use the SO8
surface mount package, the thermal resistance is
150°C/W junction to ambient in still air.
Pd (MAX) = 2 • 12V • 7mA + 12V – 2V •
Temperature Rise = Pd (MAX) RθJA = 0.602W •
150°C/W = 90.3°C
The maximum junction temperature allowed in the plastic
package is 150°C. Therefore, the maximum ambient allowed is the maximum junction temperature less the
temperature rise.
Maximum Ambient = 150°C – 90.3°C = 59.7°C
Note that this is less than the maximum of 70°C that is
specified in the absolute maximum data listing. If we must
use this package at the maximum ambient we must lower
the supply voltage or reduce the output swing.
As a guideline to help in the selection of the LT1229/
LT1230 the following table describes the maximum supply voltage that can be used with each part in cable driving
applications.
9
LT1229/LT1230
W
U
U
UO
APPLICATI
S I FOR ATIO
Large-Signal Response, AV = 2, RF = RG = 750Ω
Assumptions:
1. The maximum ambient is 70°C for the commercial
parts (C suffix) and 125°C for the full temperature
parts (M suffix).
2. The load is a double-terminated video cable, 150Ω.
3. The maximum output voltage is 2V (peak or DC).
4. The thermal resistance of each package:
J8 is 100°C/W
J is 80°/W
N8 is 100°C/W
N is 70°/W
S8 is 150°C/W
S is 110°/W
LT1229 • TA07
Maximum Supply Voltage for 75Ω Cable Driving Applications at
Maximum Ambient Temperature
PART
PACKAGE
MAX POWER AT TA
MAX SUPPLY
LT1229MJ8
LT1229CJ8
LT1229CN8
LT1229CS8
Ceramic DIP
Ceramic DIP
Plastic DIP
Plastic SO8
0.500W at 125°C
1.050W at 70°C
0.800W at 70°C
0.533W at 70°C
VS < ±10.1
VS < ±18.0
VS < ±15.6
VS < ±10.6
LT1230MJ
LT1230CJ
LT1230CN
LT1230CS
Ceramic DIP
Ceramic DIP
Plastic DIP
Plastic SO14
0.625W at 125°C
1.313W at 70°C
1.143W at 70°C
0.727W at 70°C
VS < ±6.6
VS < ±13.0
VS < ±11.4
VS < ±7.6
Larger feedback resistors will reduce the slew rate as will
lower supply voltages, similar to the way the bandwidth is
reduced.
Large-Signal Response, AV = 10, RF = 1k, RG = 110Ω
Slew Rate
The slew rate of a current feedback amplifier is not
independent of the amplifier gain the way it is in a traditional op amp. This is because the input stage and the
output stage both have slew rate limitations. The input
stage of the LT1229/LT1230 amplifiers slew at about
100V/µs before they become nonlinear. Faster input signals will turn on the normally reverse-biased emitters on
the input transistors and enhance the slew rate significantly. This enhanced slew rate can be as much as
2500V/µs.
The output slew rate is set by the value of the feedback
resistors and the internal capacitance. At a gain of ten with
a 1k feedback resistor and ±15V supplies, the output slew
rate is typically 700V/µs and – 1000V/µs. There
is no input stage enhancement because of the high gain.
10
LT1229 • TA08
Settling Time
The characteristic curves show that the LT1229/LT1230
amplifiers settle to within 10mV of final value in 40ns to
55ns for any output step up to 10V. The curve of settling
to 1mV of final value shows that there is a slower thermal
contribution up to 20µs. The thermal settling component
comes from the output and the input stage. The output
contributes just under 1mV per volt of output change and
the input contributes 300µV per volt of input change.
Fortunately, the input thermal tends to cancel the output
thermal. For this reason the noninverting gain of two
configurations settles faster than the inverting gain of one.
LT1229/LT1230
U
W
U
UO
APPLICATI
S I FOR ATIO
Crosstalk and Cascaded Amplifiers
The amplifiers in the LT1229/LT1230 do not share any
common circuitry. The only thing the amplifiers share is
the supplies. As a result, the crosstalk between amplifiers
is very low. In a good breadboard or with a good PC board
layout the crosstalk from the output of one amplifier to the
input of another will be over 100dB down, up to 100kHz
and 65dB down at 10MHz. The following curve shows
the crosstalk from the output of one amplifier to the
input of another.
Amplifier Crosstalk vs Frequency
OUTPUT TO INPUT CROSSTALK (dB)
120
VS = ±15V
AV = 10
RS = 50Ω
RL = 100Ω
110
100
90
80
70
60
50
10
100
1k
10k
100k
1M
10M 100M
FREQUENCY (Hz)
LT1229 • TA12
The high frequency crosstalk between amplifiers is
caused by magnetic coupling between the internal wire
bonds that connect the IC chip to the package lead frame.
The amount of crosstalk is inversely proportional to the
load resistor the amplifier is driving, with no load (just
the feedback resistor) the crosstalk improves 18dB. The
curve shows the crosstalk of the LT1229 amplifier B
output (Pin 7) to the input of amplifier A. The crosstalk
from amplifier A’s output (Pin 1) to amplifier B is about
10dB better. The crosstalk between all of the LT1230
amplifiers is as shown. The LT1230 amplifiers that are
separated by the supplies are a few dB better.
When cascading amplifiers the crosstalk will limit the
amount of high frequency gain that is available because
the crosstalk signal is out of phase with the input signal.
This will often show up as unusual frequency response.
For example: cascading the two amplifiers in the LT1229,
each set up with 20dB of gain and a –3dB bandwidth of
65MHz into 100Ω will result in 40dB of gain, BUT the
response will start to drop at about 10MHz and then flatten
out from 20MHz to 30MHz at about 0.5dB down. This is
due to the crosstalk back to the input of the first amplifier.
For best results when cascading amplifiers use the LT1229
and drive amplifier B and follow it with amplifier A.
UO
S
Single 5V Supply Cable Driver for Composite Video
The transistor’s base is biased by R1 and R2 at 2V. The
emitter of the transistor clamps the noninverting input of
the amplifier to 1.4V at the most negative part of the input
5V
R1
3k
R4
1.5k
C3
47µF
2N3904
R2
2k
C2
1µF
C1
1µF
+
This circuit amplifies standard 1V peak composite video
input (1.4VP-P) by two and drives an AC coupled, doubly
terminated cable. In order for the output to swing
2.8VP-P on a single 5V supply, it must be biased accurately. The average DC level of the composite input is a
function of the luminance signal. This will cause problems
if we AC couple the input signal into the amplifier because
a rapid change in luminance will drive the output into the
rails. To prevent this we must establish the DC level at the
input and operate the amplifier with DC gain.
(the sync pulses). R4, R5 and R6 set the amplifier up with
a gain of two and bias the output so the bottom of the sync
pulses are at 1.1V. The maximum input then drives the
output to 3.9V.
+
VIN
1/2
LT1229
R3
150k
–
R5
750Ω
C4
1000µF
+
TYPICAL APPLICATI
R6
510Ω
R7
75Ω
VOUT
R8
10k
LT1229 • TA11
11
LT1229/LT1230
U
PACKAGE DESCRIPTIO
J8 Package
8-Lead CERDIP (Narrow .300 Inch, Hermetic)
(Reference LTC DWG # 05-08-1110)
CORNER LEADS OPTION
(4 PLCS)
0.005
(0.127)
MIN
0.023 – 0.045
(0.584 – 1.143)
HALF LEAD
OPTION
0.045 – 0.068
(1.143 – 1.727)
FULL LEAD
OPTION
0.405
(10.287)
MAX
8
7
6
5
0.025
(0.635)
RAD TYP
0.220 – 0.310
(5.588 – 7.874)
1
2
3
0.300 BSC
(0.762 BSC)
4
0.200
(5.080)
MAX
0.015 – 0.060
(0.381 – 1.524)
0.008 – 0.018
(0.203 – 0.457)
0° – 15°
0.045 – 0.065
(1.143 – 1.651)
NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE
OR TIN PLATE LEADS
0.014 – 0.026
(0.360 – 0.660)
0.125
3.175
MIN
0.100
(2.54)
BSC
J8 1298
J Package
14-Lead CERDIP (Narrow .300 Inch, Hermetic)
(Reference LTC DWG # 05-08-1110)
0.005
(0.127)
MIN
0.785
(19.939)
MAX
14
13
12
11
10
9
8
0.220 – 0.310
(5.588 – 7.874)
0.025
(0.635)
RAD TYP
1
2
3
4
5
6
7
0.300 BSC
(0.762 BSC)
0.200
(5.080)
MAX
0.015 – 0.060
(0.381 – 1.524)
0.008 – 0.018
(0.203 – 0.457)
0° – 15°
0.045 – 0.065
(1.143 – 1.651)
NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE
OR TIN PLATE LEADS
0.014 – 0.026
(0.360 – 0.660)
OBSOLETE PACKAGES
12
0.100
(2.54)
BSC
0.125
(3.175)
MIN
J14 1298
LT1229/LT1230
U
PACKAGE DESCRIPTIO
N8 Package
8-Lead PDIP (Narrow .300 Inch)
(Reference LTC DWG # 05-08-1510)
0.400*
(10.160)
MAX
8
7
6
5
1
2
3
4
0.255 ± 0.015*
(6.477 ± 0.381)
0.300 – 0.325
(7.620 – 8.255)
0.009 – 0.015
(0.229 – 0.381)
(
0.065
(1.651)
TYP
+0.035
0.325 –0.015
8.255
+0.889
–0.381
0.130 ± 0.005
(3.302 ± 0.127)
0.045 – 0.065
(1.143 – 1.651)
)
0.125
(3.175) 0.020
MIN (0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
0.100
(2.54)
BSC
N8 1098
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
S8 Package
8-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
8
7
6
5
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
SO8 1298
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0°– 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.014 – 0.019
(0.355 – 0.483)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
2
3
4
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
13
LT1229/LT1230
U
PACKAGE DESCRIPTIO
S Package
14-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
0.337 – 0.344*
(8.560 – 8.738)
14
13
12
11
10
9
8
0.228 – 0.244
(5.791 – 6.197)
0.150 – 0.157**
(3.810 – 3.988)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
3
4
5
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
6
7
0.004 – 0.010
(0.101 – 0.254)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
14
2
0.050
(1.270)
BSC
S14 1298
LT1229/LT1230
U
PACKAGE DESCRIPTIO
N Package
14-Lead PDIP (Narrow .300 Inch)
(Reference LTC DWG # 05-08-1510)
0.770*
(19.558)
MAX
14
13
12
11
10
9
8
1
2
3
4
5
6
7
0.255 ± 0.015*
(6.477 ± 0.381)
0.130 ± 0.005
(3.302 ± 0.127)
0.300 – 0.325
(7.620 – 8.255)
0.045 – 0.065
(1.143 – 1.651)
0.020
(0.508)
MIN
0.065
(1.651)
TYP
0.009 – 0.015
(0.229 – 0.381)
+0.035
0.325 –0.015
0.005
(0.125)
MIN 0.100
(2.54)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
BSC
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
(
+0.889
8.255
–0.381
)
0.125
(3.175)
MIN
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
0.018 ± 0.003
(0.457 ± 0.076)
N14 1098
15
LT1229/LT1230
UO
TYPICAL APPLICATI
S
Single Supply AC Coupled Amplifiers
Noninverting
5V
0.1µF
Inverting
+4.7µF
5V
10k
10kΩ
+
VIN
10k
1/2
LT1229
+
VOUT
0.1µF
–
4.7µF
+
51Ω
+4.7µF
1/2
LT1229
VOUT
–
510Ω
RS
VIN
LT1229 • TA09
4.7µF
+
AV = 11
BW = 600Hz TO 50MHz
10kΩ
51Ω
510Ω
510Ω
AV =
10
RS + 51Ω
BW = 600Hz TO 50MHz
LT1229 • TA10
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1227
Single 140MHz CFA
Single Version of the LT1229
LT1395/LT1396/LT1397 Single/Dual/Quad 400MHz CFA
16
Linear Technology Corporation
SOT-23, MSOP-8 and SSOP-16 Packaging
122930fb LT/CP 0801 1.5K REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com
 LINEAR TECHNOLOGY CORPORATION 1992