IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004
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A 3–10-GHz Low-Noise Amplifier With Wideband
LC-Ladder Matching Network
Aly Ismail and Asad A. Abidi, Fellow, IEEE
Abstract—Reactive matching is extended to wide bandwidths
using the impedance property of LC-ladder filters. In this paper,
we present a systematic method to design wideband low-noise
amplifiers. An SiGe amplifier with on-chip matching network
spanning 3–10 GHz delivers 21-dB peak gain, 2.5-dB noise figure,
and 1-dBm input IP3 at 5 GHz, with a 10-mA bias current.
Index Terms—Amplifier noise, low-noise amplifier (LNA), noise
figure (NF), SiGe amplifier, ultrawideband (UWB), wideband
matching.
Fig. 1. Spectrum allocated for UWB communication.
I. INTRODUCTION
W
IDEBAND systems have recently received a great deal
of interest due to their potential for high-speed wireless communication [1]–[3]. However, there are new challenges
to be faced for a feasible transceiver implementation. Ultrawideband (UWB) is a wireless technology that transmits an
extremely low-power signal over a wide swath of radio spectrum [1]–[3]. UWB systems operate across a wide range of
frequencies from 3.1 to 10.6 GHz, as shown in Fig. 1. Conventional RF circuits fail to meet the needs of the UWB receiver. The dynamic range of wideband amplifiers that work
well in other applications like high-speed optical transceivers
may not be good enough for UWB systems. In fact, the lownoise amplifier (LNA) design is one of the biggest challenges.
To interface with the antenna and preselect filter, the LNA input
impedance should be close to 50 across the band from 3 to
10 GHz. The inductor-degenerated LNA used in conventional
wireless receivers offers the right properties [4]–[11], but only
in a narrow band around a single frequency [12]–[14]. A resistor feedback wideband amplifier cannot provide sufficiently
low noise figure (NF) (2 3 dB) and high gain ( 20 dB) while
consuming low power. A new methodology to extend the reactively matched LNAs to wide bandwidths is presented. It is
shown that this approach satisfies the tough system requirement
of a UWB system with moderate power consumption.
To provide some background, Section II shows that resistive
feedback amplifiers cannot provide the required performance
while consuming low power. Section III briefly describes the
design tradeoffs in the narrowband inductively degenerated amplifier. The concept of wideband impedance matching is introduced in Section IV and used to extend the bandwidth of the nar-
Manuscript received April 19, 2004; revised July 10, 2004.
A. Ismail is with Skyworks Solutions, Inc., Irvine, CA 92612 USA (e-mail:
aismail@icsl.ucla.edu).
A. A. Abidi is with the Integrated Circuits and Systems Laboratory, Electrical
Engineering Department, University of California, Los Angeles, CA 90095
USA.
Digital Object Identifier 10.1109/JSSC.2004.836344
Fig. 2.
Schematic of a resistive feedback amplifier.
rowband LNA. The design methodology as well as some practical considerations are discussed in Sections V and VI. Experimental results for a wideband fully integrated SiGe amplifier
are presented in Section VII.
II. RESISTIVE FEEDBACK AMPLIFIERS
In the design of LNAs in wireless receivers, there are several common goals. These include low NF of the amplifier,
reasonable gain with sufficient linearity, a stable 50- input
impedance, and low power consumption, which is needed in
portable systems. Satisfying all of the design goals of the UWB
systems is particularly difficult because of the broad bandwidth
compared to conventional wireless receivers. To demonstrate
that, let us consider the resistive feedback amplifier circuit
shown in Fig. 2. The input resistance is given by
(1)
where
is the small-signal transconductance gain of the NPN
transistor.
The voltage gain is given by
0018-9200/04$20.00 © 2004 IEEE
(2)
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IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004
In the absence of any other losses, this produces the LNA
noise factor
(6)
Fig. 3. Equivalent circuit for the inductively degenerated transistor.
and, neglecting the base and emitter parasitic resistance, the
noise factor is given by
It is noted that the noise figure improves quadratically with
for a given resonance frethe transistor unity gain frequency
. Clearly, due to the noiseless inductive degeneration
quency
, (5) and (6) are decoupled. To obtain the best noise figure,
. Practically, the
the transistor should be biased to the peak
parasitic base and emitter resistances will degrade the obtained
noise factor. Taking into account these losses, the noise factor is
(7)
In this case, the IIP3 is given by
(3)
IIP
where
is the 50- source resistance. The required linearity,
typically measured in terms of the third-order intercept point
(IIP3), is specified by
IIP
(4)
Intuitively, the higher the , the larger the loop gain, which
means more current
improves linearity [19]. A larger
consumption. However, for high-frequency operation, more
current consumption is usually needed to drive the parasitic
capacitances and obtain enough gain. This results in a little
. The voltage gain given by (2)
flexibility in the choice of
and
for a given . As a result,
sets a relation between
the noise factor and input resistance are coupled because,
and
.
as shown in (1) and (3), they both depend on
Because of this tradeoff, it is generally difficult to achieve an
arbitrarily low noise factor for an input impedance of 50 with
a reasonable current consumption [2], [20]. To achieve more
design flexibility, ways to decouple the noise factor from the
input impedance are needed [21], [37]. Inductor degeneration
in common-emitter LNAs was introduced by Van der Ziel
and Strutt to generate the real part needed to match the input
impedance, resulting in an improvement in the output SNR
[21]. By decoupling the input impedance from the noise factor,
this topology allows the optimization of the dynamic range
with reasonable power consumption [22]. Consider the circuit
results in an
shown in Fig. 3. The degeneration inductance
equivalent input resistance given by
(5)
is the unity gain frequency of the transistor. The rewhere
sulting equivalent circuit of the degenerated transistor consists
and
, as shown in Fig. 3. The input impedance
of the ,
and
. In
is purely resistive at the resonance frequency of
practice, an off-chip inductor
is added in series to align the
series resonance frequency with the desired frequency of operation.
IIP
IIP
(8)
The dynamic range of the LNA is defined as
DR
IIP
(9)
This means that, at a constant impedance match, the dynamic
range is proportional to the bias current. Thus, a strong reactive
feedback merely slides the dynamic range by lowering the noise
can also
figure and intercept point together [22]. Note that
be used to trade some noise figure to IIP3 and vice versa.
In some cases, the minimum degeneration inductance is
limited because of packaging considerations. For a wirebond
package, the minimum degeneration inductance even in the case
of multiple parallel downbonds cannot be lower than around 0.5
nH because of mutual inductance. At first sight, (5) would limit
to obtain a 50- input resistance
the maximum allowable
and thus effecting the noise factor. However, passive and
elements transform down a real resistance in a narrow band of
frequencies. This process is better visualized using the Smith
Chart [2], [14]–[18]. As shown in Fig. 4, a transformation ratio
(a) is used in the matching network to lower the input resistance
to 50 . As a result, for a given degeneration inductance ,
the resulting noise factor is given by
(10)
. Note that, compared with (6), the noise
where
. This means that the use
factor improves only linearly with
of advanced packaging techniques like flip-chip, which allow
lower degeneration inductances, can potentially improve further
the noise figure. This also means that differential LNAs with
integrated degeneration inductors have an additional degree of
flexibility compared to their single-ended counterparts in wirebond packages. Recently, some narrowband differential LNAs
were reported with very low noise figures at 5–6 GHz [24], [25].
ISMAIL AND ABIDI: A 3–10-GHz LNA WITH WIDEBAND LC–LADDER MATCHING NETWORK
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Fig. 5. Doubly terminated two-port network.
Fig. 4. narrowband impedance transformation using passive elements.
Alternatively, the noise factor can be expressed as
Fig. 6.
Second-order low-pass ladder filter section.
(11)
where is the quality factor of the input matching network. The
because more voltage
noise factor improves with a higher
gain is seen across the input capacitance of the transistor.
The input impedance is resistive only in a narrow bandwidth
around the resonance frequency
. To obtain a
wideband impedance matching, the of the matching circuit
should be significantly lowered. This will largely degrade the
noise figure which defeats the purpose. As a result, this type of
amplifier cannot be used for wideband applications.
Fig. 7. Input impedance of the low-pass filter versus frequency.
III. WIDEBAND IMPEDANCE MATCHING
Wideband impedance matching was first introduced by Bode
[26] in 1945 and Fano [27] in 1950 to enhance the bandwidth
of antennas. Fano has derived a complete set of integrals that
predicts the gain–bandwidth restrictions for lossless matching
networks terminated in an arbitrary load impedance [27].
Fano’s broadband method is hence a natural solution to extend
the bandwidth of narrowband circuits. As a result, conceptually,
it should be possible to extend the bandwidth of the narrowband
LNA. However, the antenna is well defined prior to the design
of the matching circuit. In the course of the design of the
wideband LNA, both the amplifier and the matching circuit
are unknown. The matching circuit as shown previously can
affect the circuit performance. In other words, the bandwidth
enhancement using Fano’s broadband matching criteria of an
optimally designed narrowband amplifier will not necessarily
result in the optimum wideband amplifier. What is needed
is a systematic methodology that will allow us to design the
optimum wideband amplifier side by side with the matching
circuit. To gain some insight, we turn to the ladder filters.
Consider the doubly terminated two-port network with an
and a load resistance
shown in Fig. 5
input resistance
, the
[28]. We define the (input/output) transfer function
loss (in nepers), and the phase (in radians) by
(12)
It is noted that
(13)
is the maximum power available from the source
where
and
is the actual power dissipated in
. In the case
of impedance matching, as the two-port network elements are
in the
all lossless, the loss is equal to zero and
passband. Under this condition, the input impedance seen from
the source terminal is real and equal to . Let us consider the
simple doubly terminated second-order low-pass filter shown in
Fig. 6. By choosing the values of and so that
(14)
the input impedance is mainly resistive and equal to up to
, the low-pass filter cut-off frequency in Fig. 7. In the stopband, the impedance is mainly reactive and no signal propagates
through the circuit. Using the low-pass to bandpass transfor, the series inductor is
mation
transformed to a series LC and the shunt capacitor to a parallel
LC. The fourth-order doubly terminated bandpass filter shown
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Fig. 8.
IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004
Fourth-order bandpass ladder filter used for impedance matching.
Fig. 10. Different currents flowing in the bandpass filter.
which implies that the voltage across the input capacitance of
the transistor rolls off at 20 dB/decade versus frequency. The
noise factor of the new amplifier is
(20)
Fig. 9.
Input impedance of the bandpass filter versus frequency.
where
and
are the input-referred current and voltage
sources of the amplifier. The resulting noise factor expression is
in Fig. 8 is obtained. Here the input impedance across the arto
is uniform and equal to , as
bitrary passband from
shown in Fig. 9.
The fractional bandwidth ( ) is defined as
(15)
Since
, the bandpass filter can be seen as a low-pass
filter section interleaved with a high-pass filter section. In this
case, the matching circuit elements are chosen so that
and
which worsens quadratically with frequency. This is mainly because the gain from the input voltage to the drive voltage across
the transistor rolls off with frequency.
The IIP3 of the amplifier is equal to the IIP3 of the degenerated transistor multiplied by 2 because of the potential divider
at the input across . IIP3 can be estimated by studying the
feedback applied across the transistor due to the inductive degeneration [30].
, the amplifier
Neglecting the effect of the nonlinearity of
is hence given by
(16)
(17)
Interestingly, the right part of the bandpass filter looks similar to the equivalent circuit of the inductively degenerated transistor in Fig. 3. Therefore, the bandpass filter can embed the
inductively degenerated transistor and obtain the desired input
impedance, as shown in Fig. 8. This results in a broadband
impedance matching of the inductively degenerated transistor.
Now let us consider the simplified circuit in Fig. 10. In the
passband where impedance is matched [28], the current entering
and resistive. For maxthe left port of the ladder filter is
imum power transfer, all of this current must flow into the termination resistor at the right, which means that
(18)
From (17), this means that
(19)
(21)
where
is the loop gain due to degeneration and
is
the transistor thermal voltage. This means that, unlike the noise
factor, IIP3 gets better with frequency. Equation (17) also implies that
(22)
This determines the bias current. Now, to minimize the noise
factor, from (20),
should be as large as possible. As shown in
of the SiGe HBT is around 90 GHz. HowFig. 11, the peak
of 60 GHz for two reasons.
ever, the transistor is biased at an
First, it is not recommended to operate the transistor at the peak
current density which can markedly degrade the performance
due to process variations. Second, the base resistance which is a
main contributor to the noise factor is large for small transistors
3 GHz and match
and is traded off with . From (22), for
to 50 , the bias current will be 10 mA and is 7 .
ISMAIL AND ABIDI: A 3–10-GHz LNA WITH WIDEBAND LC–LADDER MATCHING NETWORK
Fig. 11.
density.
Plot of the unity gain frequency of the transistor versus its current
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Fig. 13. Plot showing the transistor IP3 dependence on the voltage headroom
across it.
Fig. 12. Alternative bandpass ladder filter for unequal load and source
resistances.
When using wirebond connections to the package, operation
needs some sort of impedance transformation. Unat large
fortunately, we cannot use the same impedance transformer network that is customary in narrowband amplifiers. What is required here is a doubly terminated bandpass filter with unequal
loads. In a similar fashion, the amplifier will replace the series RLC in the new filter. To achieve this wideband impedance
transformation between the source and load resistances, the cirand
[29], as
cuit should include two additional inductors
shown in Fig. 12. When realized on-chip, the loss resistance of
the inductors will degrade the noise figure. To obtain the best
noise figure with the simplest matching circuit, it was chosen to
use an on-chip degeneration inductance that leads directly to a
50- input resistance.
The weak in-band signal suffers from intermodulation
distortion from both in-band and out-of-band blockers. The
input matching network which is a bandpass filter attenuates
out-of-band blockers. The higher the filter order, the more these
blockers are attenuated. Since there is no way to detune parasitic
reactances, we implement the remaining matching components
on-chip. To limit the extra noise contributed by inductor loss,
no more than two inductors are used in the matching circuit.
into account, the resulting
Taking the coupling capacitor
network is a fifth-order bandpass filter. In this case, in-band
blockers will dominate the voltage swing across the input
capacitance of the transistor and specify the amplifier IIP3.
are
As shown in Figs. 13 and 14, the amplifier IIP3 and
across the transistor. For this reason,
sensitive to the bias
1 V to obtain the best IIP3 and .
Fig. 14. Plot showing the transistor f dependence on the voltage headroom
across it.
Fig. 15.
Wideband voltage gain equalization using an inductive load.
The overall amplifier gain should be flat across the passband.
The output current from the transistor rolls off inversely with
. An inductive load equalizes the voltage gain
frequency, like
to a constant value across the passband, as shown in Fig. 15. The
parasitic capacitance across the inductor should be minimized
should be suffito ensure self resonance beyond 10 GHz.
ciently low so that the inductive region of the impedance spans
the passband.
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Fig. 16.
IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004
Simulated noise figure versus frequency for different values of r .
IV. SOME PRACTICAL CONSIDERATIONS
In a typical LNA, the base parasitic resistance contributes
significantly to the output noise. Taking into account, the resulting noise figure is
Fig. 17.
Complete schematic of the prototype LNA.
Fig. 18.
Die photograph of the LNA chip.
(23)
which means that will add noise uniformly at all frequencies.
Clearly, efforts to lower the base resistance will pay off with a
lower possible noise figure.
Another point of considerations is related to the high-pass
which is specified by
and , as shown
cut-off frequency
is the parasitic capacitance of the pad and of
in (16) and (17).
is specified by the input resistance,
the on-chip inductor .
to a frequency higher
as shown by (16) and (17). To push
should be less than 300 fF. As mentioned prethan 10 GHz,
viously, the quality factor of
should be sufficiently high so
as not to degrade the noise figure of the circuit. There is a maximum quality factor and a self-resonance frequency associated
also
with the optimally designed spiral implementing .
causes another problem. The assumed frequency roll-off for the
is 20 dB/decade. However, at the approach of
,
transistor
the roll-off is more rapid because of the filter transition band. It
beyond 10 GHz. As a
is difficult in this technology to push
result, the noise figure increases significantly at high frequencies, as shown in Fig. 17. Now the base resistance also affects
the input resistance
(24)
This means that, for a higher base resistance, smaller
is
higher
needed for matching. Thus, can be used to push
since, from (17),
and thus lower the in-band
roll-off. As a result, higher noise figure at lower frequencies
can be traded for lower noise figure at higher frequencies, as
shown in Fig. 16. In this way, the signal-weighted wideband
noise figure can be optimized.
Another tradeoff is in the choice of the coupling capacitor
. If
is too large, its bottom plate capacitance will add
, which effects
. If it is too small, the bandpass filter
to
will not meet the specification on
.
was chosen to be
the smallest possible value in order not to increase the parasitic
. The load resistance of the amplifier
was
capacitance
increased to equalize the voltage gain transfer function to a flat
response.
V. FINAL CIRCUIT
The final wideband LNA circuit is shown in Fig. 17. The
matching circuit consists of the capacitor , which is the parasitic capacitance at the input node including the pad in parallel
in series with the
with the on-chip spiral inductor , and
on-chip spiral inductor . The cascode transistor improves reverse isolation and lowers Miller multiplied capacitance. The
in seload of the amplifier consists of the on-chip inductor
ries with the polysilicon resistor
. An emitter follower inserted for measurement purposes only buffers the output to an
external 50- resistance. This emitter follower is not present
when the LNA is a part of a fully integrated UWB receiver. All
of the bias voltages and currents are generated on-chip from a
bandgap reference. The amplifier draws 10 mA and the emitter
follower draws 15 mA from a 3-V supply.
The circuit was fabricated in the Jazz Semiconductor SiGe
0.18- m BICMOS process. The die photograph is shown in
Fig. 18. The total die area is 1.8 mm . All of the spiral inductors and MIM capacitors were very carefully modeled across
the band of interest and compared to measurement data from
test chips. Interconnects were modeled as transmission lines and
ISMAIL AND ABIDI: A 3–10-GHz LNA WITH WIDEBAND LC–LADDER MATCHING NETWORK
back annotated in the final simulations. The value of the spiral
inductors and capacitors were adjusted to equalize the response
after modeling the interconnects. Large MIM capacitors were
used to ac ground the supply line and the base node of the cascode transistor. Spiral inductors were separated to lower the mutual coupling and placed close to the pads. Short interconnects
connect the amplifier to the emitter follower with least parasitic
capacitance. Ground was connected on metal 1 under the RF
pad to prevent substrate noise from coupling to the amplifier
RF input.
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TABLE I
SUMMARY OF THE MEASUREMENT RESULTS
VI. LINEARITY CONSIDERATIONS
Linearity is an important measure to the ability of the circuit
to handle large signals. In narrowband amplifiers, the linearity is
typically described with the third-order intercept point IP3 and
the 1-dB compression point (CP). A wideband circuit may need
other ways of specification because one large signal can compress the gain at all frequencies [18], [19]. In a UWB receiver,
the LNA encompass several subbands as well as the WLAN signals in the middle of the band spanning from 5.1 to 5.8 GHz.
Equation (21) gives an estimate for the in-band IIP3. Linearity
improves at higher frequency because the matching circuit amplifies less. Therefore, the worst-case linearity is at the lower end
of the spectrum. Three different measurements indicate in-band
and out-of-band effects. The first two measurements are the
in-band IIP3 and the out-of-band CP, respectively, and they are
both swept across frequency. IP3 was measured using two tones
that are separated by a 100 MHz and of equal amplitude. In-band
CP was measured by sweeping the amplitude of a single tone to
detect the small-signal gain compression of the signal. The third
measurement shows the effect of cross-band compression due to
out-of-band blockers. Cross-band compression is defined as the
signal power in one band that causes 1-dB degradation in the
small-signal gain in another band. To measure this, the amplitude of a tone is increased to monitor the small signal gain compression of a 3.4-GHz weak tone which lies at the lower end
of the spectrum. An out-of-band IP3 measurement at 5.4 and
5.6 GHz which lie in the WLAN band was added for completion and to show that the same LNA can receive both UWB and
WLAN. The measurement results are shown in the next section.
Due to the second-order nonlinearity of the transitor, the base
current is modulated at the beat frequency between the blockers.
Flowing to ground through a large low-frequency impedance
from the base of the transitor, this effect worsens the nonlinearity of a bipolar LNA and degrades IIP3 [36]. Fortunately, the
offers a low impedance path at dc from the
shunt inductor
base of the transistor to ground, and hence suppresses this effect.
VII. MEASUREMENTS
The circuit was contacted with Pico G-S-G RF probes for
measurements. The noise figure was measured using an Agilent
N8975A noise figure analyzer. A summary of the measurement
results is shown in Table I. Fig. 19 shows the measured insertion
gain versus the simulated gain. The measured gain has a peak
of 21 dB from 3 to 8 GHz and rolls of by 4 dB at 10 GHz.
We believe that the small departure from simulations is due to
inaccuracy in the high-frequency modeling of the interconnects.
Fig. 19.
Measured and simulated amplifier gain versus frequency.
Fig. 20.
Measured and simulated noise figure versus frequency.
Fig. 20 shows both the measured and simulated noise figure.
The measured noise figure agrees well with simulations with a
minimum of 2.5 dB from 3 to 6 GHz and rising to 4.2 dB at
10 GHz. Input impedance matching is characterized in Fig. 21,
. The measured
which shows the measured and simulated
is also close to simulations and is below 10 dB from 2.2
to 8 GHz, worsening only slightly to 9 dB at 10 GHz. Fig. 22
shows the input-referred IP3 of the amplifier while swept versus
frequency. As expected, the linearity improves as the frequency
goes higher. Fig. 23 shows both the in-band and out-of-band
compression points while swept versus frequency. Cross-band
compression is measured by sweeping the blocking tone with
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IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004
TABLE II
COMPARISON WITH THE STATE-OF-THE-ART REPORTED
WIDEBAND AMPLIFIERS
Fig. 21.
Measured and simulated S
versus frequency.
with a conventional wideband amplifier based on resistor feedback. A systematic methodology was presented to optimize the
performance of the wideband LNA. The concept is validated
through the measurement results of an SiGe wideband amplifier that achieves a superior performance with moderate power
consumption.
REFERENCES
Fig. 22. Measured IIP3 versus frequency.
Fig. 23.
Measured in-band and out-of-band CP versus frequency.
frequency while keeping the weak signal at 3.4 GHz. As expected, both in-band and out-of-band CPs increase at higher frequencies. At 3.4 GHz, the cross-band CP, which is a two-tone
test, is around 3 dB less than the in-band compression, which is
a one-tone test as predicted by theory [19]. A summary of the
rest of the measurements is also included in Table I. Table II
compares the performance of the amplifier with the reported
state-of-the-art amplifiers.
VIII. CONCLUSION
A new method that exploits the unique properties of ladder
filters was used to transform the well-known single-frequency
reactive matching circuit used in a low-noise tuned amplifier
into a wideband matching circuit. A higher gain and lower
noise is obtained over a wide frequency range than is possible
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Aly Ismail received the B.Sc. (with honors) degree
and the M.S. degree in electrical engineering from
Cairo University, Giza, Egypt, in 1998 and 2000, respectively. He is currently working toward the Ph.D.
degree in electrical engineering at the University of
California, Los Angeles (UCLA).
He joined the Integrated Circuits and Systems Laboratory, UCLA, as a Research Assistant in 2000. In
2002, he joined Conexant systems, Newport Beach,
CA, where he worked on designing RF integrated circuits for cellular systems. He is currently with Skyworks Solutions, Irvine, CA. His research interests include RF integrated circuit
and analog/mixed signal circuit design.
Mr. Ismail received the Analog Devices Inc. (ADI) Outstanding Student Designer Award in 2003.
Asad A. Abidi (S’75–M’80–SM’95–F’96) received
the B.Sc. (with honors) degree from Imperial College, London, U.K., in 1976, and the M.S. and Ph.D.
degrees in electrical engineering from the University
of California, Berkeley, in 1978 and 1981, respectively.
He was at Bell Laboratories, Murray Hill, NJ,
from 1981 to 1984 as a Member of Technical Staff in
the Advanced LSI Development Laboratory. Since
1985, he has been with the Electrical Engineering
Department, University of California, Los Angeles
(UCLA), where he is a Professor. He was a Visiting Faculty Researcher at
Hewlett Packard Laboratories in 1989. His research interests are in CMOS RF
design, data high-speed analog integrated circuit design, conversion, and other
techniques of analog signal processing.
Dr. Abidi was the Program Secretary for the International Solid-State Circuits Conference from 1984 to 1990, and General Chairman of the Symposium on VLSI Circuits in 1992. He was Secretary of the IEEE Solid-State Circuits Council from 1990 to 1991. From 1992 to 1995, he was Editor of the
IEEE JOURNAL OF SOLID-STATE CIRCUITS. He has received an IEEE Millennium Medal and the 1988 TRW Award for Innovative Teaching and the 1997
IEEE Donald G. Fink Award. He was a corecipient of the Best Paper Award at
the 1995 European Solid-State Circuits Conference, the Jack Kilby Best Student
Paper Award at the 1996 International Solid-State Circuits Conference (ISSCC),
the Jack Raper Award for Outstanding Technology Directions Paper at the 1997
ISSCC, and the Design Contest Award at the 1998 Design Automation Conference, and received an Honorable Mention at the 2000 Design Automation
Conference.