JOURNAL OF COMPUTERS, VOL. 9, NO. 2, FEBRUARY 2014
337
A Novel Unity Power Factor Hysteresis Current
Control for SPMSM Using Switching Table
Mingdi Fan
School of Automation, Northwestern Polytechnical University, Xi’an, Shaanxi, 710072, P. R. China
Email: fanli1998@163.com
Hui Lin and Bingqiang Li
School of Automation, Northwestern Polytechnical University, Xi’an, Shaanxi, 710072, P. R. China
Email: {linhui, libingqiang}@nwpu.edu.cn
Abstract—In this paper, a novel unity power factor (UPF)
hysteresis current control method for surface-mounted
permanent magnet synchronous motor (SPMSM) drive
system is presented. The optimum voltage vectors, which
are selected from a switching table according to the magnitude and phase of the stator current vector, are applying to
the voltage source inverter (VSI). The UPF operation for
SPMSM is achieved by maintaining the angle between the
stator current vector and stator flux vector at 90°. Compared with traditional direct torque control (DTC), the
proposed method eliminates the disadvantages of the actual
stator flux and torque estimation without deterioration of
the dynamic properties of DTC. The current regulator is
realized by using the hysteresis controller and the switching
table while the vector control (VC) employs complicated
coordinate rotating transformation, PI controller, and pulse
width modulation (PWM). So, the proposed structure and
the algorithm are easier to implement comparing with VC.
It has been found that with the proposed scheme SPMSM
drive shows adequate dynamic torque performance and
considerable torque ripple reduction.
Index Terms—unity power factor, permanent magnet synchronous motor, direct torque control, vector control,
switching table, voltage source inverter1
I.INTRODUCTION
The permanent magnet synchronous motors (PMSMs)
are popular in industrial AC motor drives due to high
torque-to-current ratio, large power-to-weight ratio, high
efficiency, high power factor and robustness, etc.
Two most widespread AC motor control schemes are
vector control (VC) and direct torque control (DTC). The
VC was invented for induction motors by Blaschke in
1972 [1], which is also known as field oriented control
(FOC). Takahashi et al. [2] and Depenbrock [3] invented
the DTC scheme for the control of induction motors in
the mid of 1980’s. In view of the successful application
of the induction motor, VC [4-7] and DTC [8-10] have
been applied to the PMSMs. Nowadays PMSM drives
with VC and DTC are available on the market with several producers, different solutions and performance. Both
Manuscript received June 14, 2013
Corresponding author: fanli1998@163.com
© 2014 ACADEMY PUBLISHER
doi:10.4304/jcp.9.2.337-344
VC and DTC can provide excellent static and dynamic
performance for PMSM drive. However, their principles
are significantly different. Reference [11] proposed the
simulations of the motor dynamic response, parameter
sensitivity and system complexity using VC and DTC
respectively. The two control schemes were experimentally compared with a same hardware platform in Reference [12]. The simulations of the motor transient response and torque ripple were analyzed [13-15].
A unity power factor (UPF) control is of paramount
importance in applications where energy saving and cost
of electricity are critical, such as home appliances, fans,
pumps, hybrid electric and electric vehicle motor drives
[16]. Currently, the UPF operation for PMSM always
utilizes VC [17, 18, 19]. Reference [17] designed a vector
control system for PMSM based on stator flux orientation
to obtain UPF. In Reference [18], based on controlling the
d-axis stator current component, UPF operation method
for PMSM was implemented using a digital signal processor (DSP) TMS320F2808 board, and power factor
value was close to unity (greater than 0.94). A method of
sensor-less VC with UPF for surface mounted permanent
magnet synchronous motor (SPMSM) was presented in
Reference [19].
Compared with VC, DTC has advantages as it is simple owing to the absence of PWM current controller and
reference frame transformation, good robustness to parameter variation and external disturbances. Most importantly, it also has better dynamic response [20,21]. In
the traditional DTC, the reference magnitude of the stator
flux vector is a constant, which is approximately equal to
the rotor flux magnitude. As a result, the power factor is
low, approximately equal to 0.77 [22].
In this paper, based on the comparative analysis of the
operation principle of VC and DTC, a stator current control scheme with switching table is presented, which ensures UPF operation for the PMSM drives. In the proposed algorithm, the magnitude of the stator current vector, the angle between the stator current vector and the
stator flux vector, and the stator current vector position
are used to select the optimum voltage switching vectors
applying to the voltage source inverter. To achieve the
speed control, only a PI controller, two hysteresis con-
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JOURNAL OF COMPUTERS, VOL. 9, NO. 2, FEBRUARY 2014
trollers, a switching table, two current sensors, and a
speed sensor are required.
Some criteria to evaluate the performance of PMSM
drives are proposed in this paper. They are used to compare the three control schemes (VC, DTC and the proposed control scheme) in both steady-state and transient
operating conditions. For the sake of comparison, the
three control schemes are implemented in the same simulation environment.
II.CONTROL SCHEMES ANALYSIS
Without loss of generality and for the sake of clarity,
the following assumptions are made in the derivation [23].
1) Saturation is neglected. 2) The induced EMF is sinusoidal. 3) Eddy currents and hysteresis losses are negligible. 4) There are no field current dynamics. 5) There is no
cage on the rotor.
With these assumptions, the space vector diagram for
PMSM is shown in Fig. 1, where αβ is the two-phase
JJG
stationary frame, dq is the two-phase rotary frame, us is
JJG
JG
the stator voltage vector, is is the current vector,ψ s is the
JJJG
stator flux vector andψ f is the permanent magnet rotor
JJG
flux vector, Es is the electromotive force (EMF), Ls is
the stator inductance, Rs is the stator winding resistance,
p is the pair of poles, ωr is the mechanical rotor angular
speed, ωe is the electrical rotor angular speed, θ i is the
phase of stator current in αβ frame, θ r is the rotor position angle, θ s is the stator flux angle, δ is the load
angle, γ is the torque angle and θψ i is the flux current
angle, i.e. the angle between stator current vector and
stator flux vector.
JJG
us
JG
Rs is
JJG
Es
JG
is
q
β
JJG
ψs
JG
Ls is
i
i
δ
O
θ s JJJG
ψf
r
ωe = pωr
d
α ( A)
Figure 1. Space vector diagram for PMSM
The voltage and flux equations used to model PMSM
in stationary frame can be derived as follows:
JJG
JJG
JG JJG
JG dψ s
(1)
us = Rs is + Es = Rs is +
dt
JJG
JG JJJG
ψ s = Ls is + ψ f
(2)
JG
JJG JJJG JJJG
JJG
di
Es = Esi + Esm = Ls s + jωeψ s
dt
© 2014 ACADEMY PUBLISHER
(3)
JG
JJG
JJJG
dis
is the induced EMF ( Esi ) and jωr ψ s is
where: Ls
dt
JJJG
the motional EMF ( Esm ). Note that, the motional EMF is
the product of rotor angular speed and stator flux.
Input to the VC or DTC is the reference torque Te* ,
which is set either by the user or by a superimposed control (e.g. speed control). The basic electromagnetic torque
equations of SPMSM (for SPMSM: d-axis stator inductance is equal to q-axis stator inductance, i.e. Ld = Lq = Ls )
for VC and DTC can be obtained as:
Te =
3 JJJG
3 JJJG JG
p ψ f iq = p ψ f is sin γ
2
2
JJJG JJG
3 ψ f ψs
sin δ
Te = p
2
Ls
(4)
(5)
Note that, for VC, as long as the amplitude and phase
of the stator current vector are accurately controlled, the
electromagnetic torque can be effectively controlled. For
DTC, the control objects are the amplitude and phase of
the stator flux vector.
As shown in Fig. 1, an orthogonal control grid, formed
by the direct axis d and the quadrature axis q, is used in
VC. The stator current is decomposed into two parts using coordinate transformation matrix (three-phase stationary frame ABC to two-phase rotary frame dq ): d-axis
stator current id and q-axis stator current iq . id aligning
JJJG
with the rotor flux vector ψ f may be seen as the
flux-producing current, and iq which has perpendicular
JJJG
phase relationship with ψ f may be seen as the
torque-producing current. The regulators of id and iq are
usually enforced through the inverter with one of the
popular methods: hysteresis control, pulse width modulation (PWM), and space vector modulation (SVM).
Therefore, the VC is exploited to make the PMSM drive
system a high-performance drive system with independent control of its stator flux and electromagnetic torque.
Unlike VC, the DTC does not require any current regulators, coordinate transformation, PWM or SVM. The
principle of DTC is to directly select voltage vectors according to the difference between the reference and the
actual value of the torque and the flux linkage. Thus, the
torque and flux errors are compared in hysteresis comparators respectively. Depending on the comparators a
voltage vector is selected from the switching table. In
spite of its simplicity, DTC allows a good torque control
in steady-state and transient operating conditions to be
obtained. Compared to VC, DTC might be preferred for
high dynamic applications, but it shows higher current
and torque ripple. The performance of DTC strongly depends on the quality of the estimation of the actual stator
flux and torque.
The proposed scheme in this paper utilizes the magnitude of the stator current vector, the angle between the
stator current vector and the stator flux vector, and the
JOURNAL OF COMPUTERS, VOL. 9, NO. 2, FEBRUARY 2014
339
stator current position to select the optimum voltage
switching vectors from the switching table. Similar to VC,
the basic idea is to control the amplitude and phase of
stator current vector. Similar to DTC, it is based on the
two-phase stationary frame without PWM and SVM as
well as without decoupling of the stator current. The stator current vector is regulated through the selection of
optimum space voltage vectors from the switching table
with two hysteresis controllers, and then excellent dynamic torque response is achieved. The new control
scheme can eliminate the disadvantages of depending on
the actual stator flux and torque estimation without deterioration of the dynamic properties of DTC.
The basic electromagnetic torque equations of SPMSM
for the proposed scheme can be alternately derived as:
3 JJJG JG
3 JJG JG
Te = p ψ f is sin γ = p ψ s is sin θψ i
2
2
(6)
JG
The relative position between stator current vector is
JJG
and the stator flux vector ψ s are shown in Fig. 1. θψ i is
JG
JJG
the angle between the is and ψ s . A control strategy
can be synthesized to keep the flux current angle θψ i at
JJG
JJG
90°. The EMF vector Es leads ψ s by 90°. Controlling
JJG
JG
is to lead ψ s by the same angle results in the coinciJJG
JG
dence of Es and is . If the stator resistance Rs is neJJG
glected, the stator terminal voltage will be equal to Es ,
and the relative phase between them is zero. Therefore,
the angle between the stator current vector and stator
voltage vector is zero, resulting in the UPF operation of
the SPMSM drive.
III.UPF- HYSTERESIS CURRENT CONTROL
A. Stator Current Vector
In proposed stator current direct control, the parameters are
calculated in the αβ two-phase stationary frame. The
stator current iα and iβ can be obtained by iA and iB
measured by current sensors using coordinate transformation matrix as [24]:
⎡1
⎡iα ⎤
⎢ ⎥ 2⎢
⎢iβ ⎥ = 3 ⎢ 0
⎢1 2
⎢i ⎥
⎣0⎦
⎣
−1 2 ⎤ ⎡ iA ⎤
⎥
3 / 2 − 3 / 2 ⎥ ⎢⎢ iB ⎥⎥
12
1 2 ⎥⎦ ⎢⎣ −iA − iB ⎥⎦
As shown in Fig. 1, the flux current angle θψ i can be
derived as:
JG
Ls is
θψ i = γ − δ = θi − θ r − arcsin JJJG
ψf
To achieve the UPF, the flux current angle should be
controlled at 90º, so the reference torque angle γ * is related with the load angle δ as:
JG
Ls is π
π
*
γ = δ + = arcsin JJJG +
(11)
2
2
ψf
B. Voltage Vector Selection
According to (1) and (3), the derivative of the stator
current vector can be derived as:
JG JJG
JG
JJG
dis us − Rs is − jωeψ s
(12)
=
dt
Ls
which determines the direction and the rate of the stator
current vector change. Graphical illustration of formula
(12) is shown in Fig 2.
JJJG
Esm is the motional EMF which can be obtained by
multiplying the mechanical rotor angular speed by stator
flux. As shown in Fig. 2, the effect of stator voltage vector on stator current vector in proposed control scheme is
similar to the effect of stator voltage vector on stator flux
vector in DTC, i.e. the direction of the derivative of the
stator current is the same with the set stator voltage vector, under the condition that the motional EMF is zero
(motor speed is zero), or can be ignored. However, usually the motional EMF cannot be ignored; so the direction
of the derivative of the stator current will not be the same
with the set stator voltage vector. When the motor speed
is high, due to the presence of the motional EMF, the
stator voltage vector selection may be unable to make
JG
is and γ increase at the same time, and may also be unJG
able to make is increase and γ decrease. In order
u3
u2
−1 2
(7)
JJJG
Esm
where: iA and iB are stator phase current, iα and iβ
θi = arctan
© 2014 ACADEMY PUBLISHER
iβ
iα
(9)
JG
is
JG
Rs is
u4
are stator current in αβ frame. i0 is zero-sequence current. Then the amplitude and phase of the stator current
are derived as [24]:
JG
is = iα2 + iβ2
(8)
(10)
u5
u1
JG JJG
JG JJJG
d is us − Rs is − Esm
=
dt
Ls
u6
Figure 2. Typical moving direction of current vector locus
to specifically analyze the effect of stator voltage vector
on the amplitude and phase of the stator current vector in
the presence of motional EMF, the stator current vector
340
locating between the stator voltage vector u1 and u2 is
taken for an example, as shown in Fig. 3.
According to Fig. 3, the stator current vector is located
between the stator voltage vectors u1 and u2 , O is the
JJJG
starting point of Esm , u1 and u2 . E and D are the respective terminal points of u1 and u2 . A and C are the
respective midpoints of u2 and u1 . Taking A as the
center and OD as the diameter, the purple circle is drawn.
As the angle inscribed in a semicircle is always a right
JJJG
angle, if the terminal of Esm is located in this purple
circle, the magnitude of the stator current vector will be
increased by u2 and vice versa. Similarly, Taking A as
the center and OE as the diameter, the brown circle is
JJJG
drawn. If the terminal of Esm is located in this brown
circle, the magnitude of the stator current vector will be
JJJG
increased by u1 and vice versa. The Esm of a rated
JJG
speed PMSM motor is normally equal to (0.35 ~ 0.85) us
JJJG
[24]. It means that the terminal of Esm usually does not
exceed the area (O-A-D-B-E-C-O).
Based on the above analysis, this area can be divided
into the following three sub area. Area I (O-A-B-C-O): if
JJJG
the terminal of Esm is located in area I, u2 can make
the current vector amplitude and γ increase, while u1 can
make the current vector amplitude increase and γ deJJJG
creases. Area II (A-D-B-A): if the terminal of Esm is
located in area II, u2 can make the current vector amplitude and γ increase which is similar with in area I,
but u1 will make the current vector amplitude and γ deJJJG
crease. Area III (C-B-E-C): if the terminal of Esm is
located in area III, u1 can make the current vector amplitude increase and γ decrease which is similar with in
area I, but u2 will make the current vector amplitude
decrease and γ increase.
In this paper, a twelve sectors control method of stator
current vector is presented for the construction of the
switching table. The twelve sectors are shown in Fig. 4.
As discussed before, the voltage vector switching table
is adopted, as presented in Table I. In order to discuss the
reasonability of Table I, taking the sector θ1 as example,
u2 will make the current vector amplitude decrease and
JJJG
γ increase where the terminal of Esm is located in area
III as shown in Fig. 3(c). Although that is not suitable
with the outputs of the hysteresis controllers, considering
the discrete nature of the VSI, u2 is still the best choice
compared with the other voltage vectors. Furthermore,
from Fig. 3(c) we can see that, u2 makes the current
vector amplitude decrease slowly and it will make γ increase quickly to reach the upper limit of the hysteresis
controller, and then u1 should be chosen to increase the
current vector amplitude. Sectors θ 2 ~ θ12 are similar
© 2014 ACADEMY PUBLISHER
JOURNAL OF COMPUTERS, VOL. 9, NO. 2, FEBRUARY 2014
with sector θ1 . So setting the hysteresis controllers reasonable, the current will be well regulated by selecting
voltage vectors from Table I.
u2
Esm
u1
I (O-A-B-C-O)
D
u2
Ⅱ
A
B
Esm
Ⅰ
O
Ⅲ
u1
C
E
(b) II (A-D-B-A)
u2
Esm
u1
(c) III (C-B-E-C)
Figure 3. Effect of stator voltage vector on stator current vector
θ5
θ4
u3 (010)
θ3
u2 (110)
θ6
θ2
θ1
u1 (100)
u4 (011)
θ12
θ7
θ8
u5 (001)
θ9
u6 (101)
θ10
θ11
Figure 4. Sectors partition for current vector
JOURNAL OF COMPUTERS, VOL. 9, NO. 2, FEBRUARY 2014
341
TABLE II.
PARAMETERS OF SPMSM USED FOR SIMULATION
TABLE I.
SWITCHING TABLE OF VOLTAGE VECTORS
Hi Hγ θ1
θ2
θ3
θ4
θ5
θ6
θ7
θ8
θ9
θ10
θ11
θ12
1
0
1
0
u2
u1
u4
u6
u3
u2
u4
u6
u3
u2
u5
u1
u4
u3
u5
u1
u4
u3
u6
u2
u5
u4
u6
u2
u5
u4
u1
u3
u6
u5
u1
u3
u6
u5
u2
u4
u1
u6
u2
u4
u1
u6
u3
u5
1
0
u2
u1
u3
u5
C.
Control Diagram
The control diagram of the proposed control scheme is
shown in Fig. 5.
JG *
is
Hi
γ* =δ +
Hγ
π
2
JG
is
γ
θ4
θ5
θ3
V1 θ1
θ 7 V4
θ8
θ2
V2
V3
θ6
θ12
V6
V5
θ9
θ11
θ10
θi
θi = arctan ( isβ isα )
isα = iA
δ = arcsin( Ls is ψ f )
isβ =
γ = θi − θ r
1
( iA + 2iB )
3
JG
is = isα 2 + isβ 2
iA
iB
θr
Figure 5. Control diagram of the presented scheme
The proposed control scheme can be described as follows:
zThe amplitude and phase of the stator current vector
is obtained using current sensors and speed sensor
by (7) (8) (9) and (10).
zThe reference amplitude of stator current is set either
by the user or by a superimposed control.
zTo achieve the UPF, the reference torque angle γ *
is obtained from (11).
zThe equation (12) describes the derivative of the stator current vector.
zThe inputs of the hysteresis controllers are the difference between the reference and the actual value
of the stator current amplitude and the torque angle.
Depending on the outputs of hysteresis controllers a
voltage vector is selected from the switching table
(Table I).
zThe electromagnetic torque can be calculated from
(6).
IV.RESULTS AND DISCUSSION
In order to investigate the performance of SPMSM
drive under VC, DTC or the presented scheme, simulation models are constructed using Matlab/Simulink software package. Three control schemes are implemented in
the same SPMSM drive using the same implementation
conditions.
The step size of simulation time is 5μs.
The SPMSM is Y-connected with parameters as in Table II. The VSI used in simulation is IGBT inverter with
+150 to -150 dc link voltage. The maximum switching
frequency of the IGBT is set at 10 kHz.
© 2014 ACADEMY PUBLISHER
Rated DC voltage
Un
300 (V)
Rated speed
ωn
2000 (r/min)
Rated torque
Ten
8 (Nm)
Number of pole pairs
p
4
Stator winding resistance
Rs
0.9585 (Ω)
Stator inductance
Ls
5.25 (mH)
Permanent magnet flux
ψf
0.1827 (Vs)
Moment of inertia
J
0.0006329 (kgm2)
Friction constant
B
0.0 (Nms)
The speed loop proportional and integral parameters
(in parallel PI form) of DTC are 0.2 and 0.5 respectively.
The band width of the torque hysteresis controller is
0.05Nm, and the band width of the flux hysteresis controller is 0.001Vs.
The VC control scheme is based on the control strategy
id=0. The speed loop proportional and integral parameters
are 0.1 and 100 respectively. The d, q axis current loops
proportional and integral parameters are 100 and 50 respectively.
The proposed control scheme is based on the control
strategy UPF. The speed loop proportional and integral
parameters are 0.1 and 50 respectively. The band width of
the current hysteresis controller is 0.05A, and the band
width of the torque angle hysteresis controller is 2º.
In order to compare the performance of the proposed
with DTC and VC, the following criteria are selected.
Power factor (PF), power ripple factor (PRF) and copper losses ( PCu ) are the criteria that concern the power
performance, such that:
PF =
PRF =
P
2
P +Q
2
=
P
S
Pmax − Pmin
× 100%
Pm ean
JG 2
PCu = 1.5*Rs is
(13)
(14)
(15)
where: P is the active power, Q is the reactive power, S is
the apparent power. In this paper, the 3-phase instantaneous active power P and reactive power Q are calculated
using the following equations [25]:
P = Va * I a + Vb * I b + Vc * I c
Q=
3
(Vbc * I a + Vca * I b + Vab * I c )
3
(16)
(17)
where: Va , Vb and Vc are the instantaneous stator phase
voltages; Vab , Vbc and Vca are the instantaneous stator
line voltages; I a , I b and I c are the instantaneous stator
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JOURNAL OF COMPUTERS, VOL. 9, NO. 2, FEBRUARY 2014
phase currents. In a Y-connected SPMSM, the phase currents are equal to the line currents.
Torque ripple factor (TRF) and torque per ampere rate
(TPA) are the criteria that concern the torque performance of the SPMSM drives, such that:
TRF =
Te _ max − Te _ min
Te _ m ean
TPA =
× 100%
Te JG
is
(18)
TABLE III.
POWER PERFORMANCE OF SPMSM
Q
(Var)
463.1
S
(VA)
1812.1
PF
PRF
PCu (W)
VC
P
(W)
1751.9
0.967
6.9%
76.4
DTC
1758.3
985.7
2015.8
0.872
10.3%
82.8
proposed
1754.3
3.7
1754.3
1.00
3.1%
78.8
(19)
The simulation results of the three control schemes are
shown in Fig. 6 and Fig. 7. The power performance is
simulated with 2000r/min closed speed loop, 8Nm load
torque, while torque dynamic response is simulated with
open speed loop. The active power, reactive power and
copper losses of the three control schemes are shown in
Fig. 6. The power factors and power ripple factors can be
calculated from (13) and (14), and the power performance
is shown in table III. The striking feature is the low apparent power requirement in the proposed control, as the
stator flux linkages decrease with stator current increasing, limiting the stator voltage requirement.
Fig. 7(a), (b) and (c) show the torque performance of
the SPMSM drive with VC (id=0), DTC and the proposed
control scheme respectively. Note that, the rise time of
the motor torque from 2Nm to 8Nm with VC is 2.46ms,
while it is 1.11ms with DTC, and 1.40ms with the proposed control. The TRF and TPA can be calculated from
(18) and (19). The TRF has the amplitude of 8.1 percent
in the case of the VC, and 13.5 percent in the case of the
DTC, while no more than 10.4 percent in the case of the
proposed control. So the proposed control scheme is a
tradeoff between torque ripple and torque dynamic,
which shows lower torque ripple similar to VC and excellent torque dynamic similar to DTC.
T
8
TPA = JGe =
= 1.09 Nm A
is 7.32
(a) VC (id=0)
T
8
TPA = JGe =
= 1.03 Nm A
7.74
is
(b) DTC
0.83Nm
1.40ms
TRF = 0.83/8 = 10.4%
T
8
TPA = JGe =
= 1.07 Nm
A
7.47
is
Figure 6. Power performance with rated load torque
(c) The proposed control
Figure 7. Torque dynamic response with open speed loop
© 2014 ACADEMY PUBLISHER
JOURNAL OF COMPUTERS, VOL. 9, NO. 2, FEBRUARY 2014
As shown in Fig. 7, the TPA of VC, DTC and the proposed control are 1.09Nm/A, 1.03Nm/A and 1.07Nm/A
respectively. The TPA of VC is more than the others,
because the VC control scheme is based on the control
strategy id=0 which results in the maximum torque per
ampere operation in SPMSM.
As shown in Fig. 6, Fig. 7 and Table III, the proposed
control scheme improves the torque dynamic performance without deterioration of the torque smoothness
properties of VC, and as compared with DTC, it shows
considerable torque ripple reduction, and achieves UPF to
improve the power performance. Because of the difference in power factor, for equivalent power output, it can
be inferred that the proposed control will have apparent
power compared to the other two. Lower apparent power
means lower volt–amp rating of the inverter, lower volume filter, and higher efficiency in SPMSM drives as
compared with the other two.
Fig. 8 and Fig. 9 show the transient response and
steady states of the proposed control with 2000 r/min
343
closed speed loop, and 8Nm load torque. It is noticed that
with the proposed control, the motor current is fully controlled to track its reference curve both at transient response and steady states, and the flux current angle is
fixed at 90° which can make the SPMSM drive achieve
UPF.
V.CONCLUSION
To achieve the high-performance of PMSM drive system, the stator current control based on UPF is presented.
The basic idea of the control is to calculate the parameters in the two-phase stationary frame, regulate the magnitude and the phase of the stator current vectors in hysteresis controllers by choosing proper voltage vectors
from the switching table. So that satisfactory dynamic
response of the torque and UPF can be expected.
The proposed control scheme implements the current
regulator without decoupling the stator currents, throwing
aside the complicated coordinate rotating transformation.
The essential parameters are stator currents of three
phases and the rotor position. As a result, the structure
and the algorithm of the proposed control scheme are
simple and easy to implement.
The simulation results of the proposed control scheme
show that the SPMSM drive system has good performance during transient response and steady-state operations. It is a combination of the advantages of VC and
DTC, and it can be applied to the servo systems that high
precision, high dynamic and high power factor are needed.
REFERENCES
Figure 8. Steady state of SPMSM with the presented control
Figure 9. Locus circle of the stator flux and current with
the presented control
© 2014 ACADEMY PUBLISHER
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Mingdi Fan was born in HeZe City,
Shandong, China, in 1987. He obtained
bachelor degree in Northwestern Polytechnical University, China, in 2008.
From 2010 to 2011, he worked as a visiting scholar in Kassel University, Germany. He is now a doctoral student at
Northwestern Polytechnical University.
His major research fields are AC servos,
nonlinear motor control and wind power
technology.
E-mail: fanli1998@163.com
Hui Lin was born in 1957. He received
the Ph.D. degree in control theory and
applications from Northwestern Polytechnical University in 1993. From 1989
to 1990, he worked as a visiting scholar
in University of Braunschweig, Germany.
From 2003 to 2004, he worked as a visiting scholar in University of Michigan,
USA. Now, he works as a professor and
Ph.D. supervisor in Northwestern Polytechnical University. His research interests include iterative
learning control, control theory and control engineering, and
motion control technology.
E-mail: linhui @nwpu.edu.cn
Bingqiang Li was born in 1982. He
received the M.S. degree in power electronics and power drives and Ph.D. degree in control science and engineering
from Northwestern Polytechnical University in 2007 and 2010, respectively.
From 2010 to 2012, he worked as a
post-doctor in Postdoctoral Research
Station of Electrical Engineering in
Northwestern Polytechnical University. Now, he works as a
lecturer in Northwestern Polytechnical University. His research
interests include iterative learning control, power electronics
and power drives, and AC servos.
E-mail: libingqiang@nwpu.edu.cn