Comparison of Switching and Conducting Performance of
SiC-JFET and SiC-BJT with a State of the Art IGBT
W.-Toke Franke, Friedrich W. Fuchs
INSTITUTE OF POWER ELECTRONICS AND ELECTRICAL DRIVES
CHRISTIAN-ALBRECHTS-UNIVERSITY OF KIEL
Kaiserstr. 2
24143 Kiel, Germany
Phone: +49 431 8806106
Fax: +49 431 8806103
Email: tof@tf.uni-kiel.de, fwf@tf.uni-kiel.de
URL: http://www.tf.uni-kiel.de
Acknowledgments
This work has been supported by the company TranSiC providing the SiC bipolar transistors BitSiC1206
and the company SiCED provinding the SiC JFET.
Keywords
<<Power semiconductor device>>, <<SiC-device>>, <<Bipolar device>>, <<JFET>>,
<<IGBT>>, <<Silicon Carbide>>.
Abstract
Silicon Carbide (SiC) power semiconductors being actually in development are promising devices for
the future. To outline their characteristics the switching and conducting performance of a SiC-JFET and
a SiC-BJT are investigated and compared to a state of the art Si-IGBT. The power losses, the switching
times and the efforts for the driving circuits are investigated. The focus is put on the influence of the junction temperature on the power losses of the investigated devices. Therefore, 1200 V / 6 A devices have
been used. The BJT and JFET show some advantages concerning their total losses and their temperature
range.
Introduction
Until today, MOSFETs and IGBTs are the first choice for switching devices in power electronic circuits
for most applications. Typical applications are inverters for driving electrical machines or rectifiers
feeding the grid from renewable energy sources or power supplies. For these applications there is a
demand for more efficient inverters. There are different ways to achieve this: One way is to use power
semiconductor devices with good conducting behavior to reduce the losses during on-state. In this case
the devices designed in this way have a bad switching performance meaning high switching losses that
leads to low switching frequencies and hence to large filters creating losses as well. Another approach
is to use devices with good switching performance so that high switching frequencies and small filters
are possible. However in this case the conducting losses increase and the system efficiency is reduced
as well. In recent years new promising devices which seem to overcome the disadvantages mentioned
above and combine the advantages of silicon MOSFETs and IGBTs are under investigation and start to
be introduced into the market [1]. To achieve this, the substrate of these devices is no longer silicon
but silicon carbide is used. The wide band gap allows higher blocking voltages at thinner depletion
layer [2][3][4][5][6]. That leads to a reduction of the on-state losses as well as of the switching losses
compared to IGBTs. Different companies intensify their research effort on silicon carbide switches.
Some made SiC-schottky diodes commercially available. Others are just before to commercialize SiC
power bipolar transistors (BJT) and JFETs. The intention of this paper is to evaluate and compare the
characteristics of these two SiC power semiconductors with a Si IGBT. The investigated characteristics
are switching and conducting losses, switching and delay time and temperature dependency which are
most important for application engineering [7][8]. Therefore, the relevant parameters of a SiC-JFET,
SiC-BJT and a comparable Si-IGBT are measured and the mentioned characteristics are calculated and
compared.
The paper is structured as following: In the next section the method of measuring is explained. After that
measurements are presented which show the conducting and switching behavior as well as the switching
times versus the junction temperature and moreover the gate driver are discussed. In the following
section the results are compared and evaluated concerning the power losses and the complexity of their
gate drivers [9][10][11]. At the end there is the conclusion.
Method of Measuring
Setup of the Test rig
For all measurements, it is important that the device under test (DUT) is turned on for only a very short
time to eliminate self heating effects during the on-state [12]. The junction temperature is controlled by
a heating plate. For measuring the static losses as conducting and driving losses the device is connected
to an adjustable current source. For this experiment, low voltage probes to measure vGS / vBE and vDS
/ vCE are used to achieve the highest possible resolution. The device is turned on for 100 µs and the
measurements take place after 95 µs to ensure that the currents and voltages are in steady state.
For the dynamic parameters, the well-known double pulse test is used [13]. The test setup comprises a
variable high dc-voltage source, a buck converter and an inductive load. The switch of the buck converter
is the device under test. As a freewheeling diode, a SiC diode is used. This test circuits allows turning the
device under test at a defined current and voltage on and off. Therefore, two pulses drive the transistor.
The length of the first pulse defines the collector current for the switching, since the current increases
linear because of inductance. At the end of the first pulse, the turn-off behavior can be studied. After 10
µs the device is switched on again to study the turn on behavior under load conditions. While the device
is turned off the current through the inductance stays nearly constant by flowing through the freewheeling
diode. A list of the applied measurement equipment is shown in Table I
Table I: List of measurement equipment
Scope
Current probes
Voltage probe (conducting behavior)
Voltage probe (switching behavior)
Tektronix DPO 4054 500 MHz, 2,5 GS/s
Tektronix TCP 0030
Tektronix P6139A 500MHz
Tektronix P5205 100 MHz
Definition of parameters and calculation of power losses
Fig. 1 shows an ideal switching operation of an IGBT to define the switching times. The upper waveform
is the gate signal while the lower waveforms describe the voltage and the current of the IGBT. The turnon delay time td(on) is defined as the time the gate voltage reaches 10 % of its maximum voltage to the
collector current has risen to 10 % of its on value. The duration the current takes to reach 90 % of its
on value is defined as the rise time tr . The sum of td(on) and tr is the turn-on time ton . The turn-off delay
time td(o f f ) starts than the gate voltage falls to 90 % and ends than the collector current has dropped to
90 % of its on value. The fall time t f is the duration it takes for the current to drop from 90 % to 10 % of
its on value. As for the turn-on time the sum of td(o f f ) and t f is defined as the turn-off time to f f . For the
bipolar transistor and the JFET tr , ton , t f and to f f are defined in the same way as for the IGBT. Only the
start of td(on) and td(o f f ) is different according to the different gate signals. For the BJT the delay times
start when the base current has risen to 10 % and ends when it has fallen to 90 %, respectively. Since
the JFET is turned off between -15.6 V and -23 V depending on the device, the start of the delay time is
defined to 90 % for td(on) and to 10 % for td(o f f ) of the turn-off gate voltage.
The conducting power losses and the switching energy are calculated based on the measured waveforms
by the following equations:
Pcon = u (t) i (t)
(1)
Z
Eon , Eo f f =
u (t) i (t) dt
tsw,on ,tsw,o f f
(2)
Figure 1: Definition of switching parameters
There tsw,on and tsw,o f f are the times from the beginning of the rise of current or voltage to the end of
the fall of voltage or current, respectively. The integration is realized by summing up all products of the
discrete measuring points and multiplying them by the sample rate.
Measurements
All measurements have been done at comparable conditions and measurements methods of voltages,
currents and temperatures as well as the test circuit. Only the drivers are different since all devices need
specific gate / base signals. However all drivers are tuned to have the same rise and fall time of their
output signals at 25 ◦ C and no load current and voltage. The adjustment is done by gate resistors and
capacitors. A typical dc voltage for a 1200 V power semiconductor in an inverter is 600 V. The characteristic waveforms of a turn-on and a turn-off event at 600 V, 6 A and the maximum specific operating
temperature of the device are shown in figure 2 to 7.
For the IGBT and the JFET load current and voltage and the gate voltage are illustrated. For the BJT the
base current is illustrated in addition. The load current waveform during turn-on has the characteristic
reverse recovery peak of the diode for all devices. Because of the SiC diode the current peak is low and
short. The gate signals have some ringing. The ringing increases with the switched voltage. Especially
for the JFET but also a little bit for the IGBT the ringing influences the switching behavior so that the
falling value has a staircase-shaped waveform. This leads to an increase of the fall time. The reason
for that is electromagnetic coupling between the gate signal and the load. By a further optimization of
the gate driver this effect might be reduced. For the BJT such a coupling is not recorded since the base
current is less EMC sensitive.
Another effect has to be regarded before the detailed analysis of the measurements is started. Fig. 3, 5
and 7 show an unexpected behavior of the current while turning the device off. The current is dropping,
while the voltage is increasing. The reason for that is the capacitance between the winding of the inductor
of the buck converter. This capacitance is discharged due to the voltage change across the inductor and
causes a negative current through the DUT that can be seen on the scope. Measurements of the parasitic
capacitance of the inductance by the resonant frequency method [14] [15] lead to a resonant frequency
of 51,9 kHz and with L = 22 mH the capacitance is calculated:
Cpar =
1
= 427 pF
2π fres L
(3)
By means of the current drop calculated parasitic capacitance is
Cpar,meas =
idrop ∆t
= 590 pF
∆u
(4)
It can be concluded that the parasitic capacitance of the inductance causes a large percentage of the
current drop. The remaining parasitic capacitances are located in the test circuit and in the cables to the
inductor.
Figure 2: Turn-on waveform of the IGBT at 600 V, 6 A Figure 3: Turn-off waveform of the IGBT at 600 V, 6 A
and 150 ◦ C. Channel 1: vCE , Ch 2: iC , Ch 3:vGE , time and 150 ◦ C. Channel 1: vCE , Ch 2: iC , Ch 3:vGE , time
base: 40ns/div
base: 200ns/div
Figure 4: Turn-on waveform of the JFET at 600 V, 6 A Figure 5: Turn-off waveform of the JFET at 600 V, 6 A
and 180 ◦ C. Channel 1: vDS , Ch 2: iD , Ch 3:vGS , time and 180 ◦ C. Channel 1: vDS , Ch 2: iD , Ch 3:vGS , time
base: 40ns/div
base: 40ns/div
Figure 6: Turn-on waveform of the BJT at 600 V, 6 A Figure 7: Turn-off waveform of the BJT at 600 V, 6 A
and 180 ◦ C. Channel 1: vCE , Ch 2: iC , Ch 3:vBE , Ch 4:iB , and 180 ◦ C. Channel 1: vCE , Ch 2: iC , Ch 3:vBE , Ch 4:iB ,
time base: 40ns/div
time base: 40ns/div
Figure 8: IGBT: Conducting losses vs. Temperature at
iC = 4 A (blue solid) and iC = 6 A (red dotted)
Figure 9: IGBT: Switching losses vs. Temperature at
iC = 4 A (blue solid) and iC = 6 A (red dotted)
Analysis of the Si-IGBT Performance
For reference measurement the IGBT IKW08T120 (1200 V, 8A)[16] with low current rating is chosen.
This IGBT uses the fieldstop technology that leads to thin n− basis and low conducting losses. The
waveforms of the current and voltages while turning on and off are shown in 2 and 3, respectively.
The conducting losses as shown in figure 8 top for 4 A (blue solid) and 6 A (red dotted) are almost
constant with the temperature. However the power losses increase with the current. The conducting
losses are almost doubled while the collector current is only increased by 50%. The reason for that is
that an IGBT has not only a threshold voltage VCE,sat but also a collector emitter resistance rCE defined
as
vCE = VCE,sat + rCE,on iC
(5)
As the temperature depending waveforms of rCE,on and VCE,sat in figure 8 middle and bottom show, these
values are temperature depending. While the resistance is increasing the threshold voltage is decreasing
with the temperature. Knowing rCE and VCE,sat the conducting losses are calculated as:
PIGBT,con = VCE,sat īC + rCE i˜C
2
(6)
Figure 9 illustrates the switching energy of the IGBT for 4 A and 6 A. It can be concluded that the turnoff losses are by a factor of 8 higher than the turn-on losses. The reason for the low turn-on losses is the
use of a SiC-diode as freewheeling diode that has a low reverse recovery peak current. The high turn-off
losses originate in the long fall time of the current as discussed later and the losses of the tail current.
The current dependence of the turn-off energy is linear for all temperatures while the turn-on energy
increases faster than linear. The turn-off energy increases constantly with the temperature from 612 µJ at
25 ◦ C for 6 A to 1130 µJ at 150 ◦ C. The turn-on energy is almost independent of the temperature. The
total switching energy has a similar characteristic as the turn-off energy because Eo f f is dominating the
switching losses.
Figure 10 illustrates the characteristic turn-on and turn-off times versus the temperature for iC = 4 A and
iC = 6 A. All times show an increase with the temperature. Characteristic for an IGBT is the long turnoff delay time compared to the turn-on delay time. The fall time has a strong temperature dependence
since it increases at 6 A from 70 ns at room temperature to 211 ns at 150 ◦ C. The fall time is for all
temperatures below 15 ns. This results in long turn-off and short turn-on times. In applications the long
turn-off time determines the required dead time.
Analysis of the SiC-JFET Performance
For the SiC-JFET a normally on device from SiCED [17] is investigated. This specific JFET has a pinchoff voltage of -18.5 V. The waveforms of the current and voltages while turning on and off are shown in
4 and 5, respectively.
The conducting losses are illustrated in the upper part of figure 12 for 4 A (blue solid) and 6 A (red
dotted). Up to 100 ◦ C the conducting losses increase moderately but increase faster beyond. The conducting losses increase quadratic with the drain current since the JFET has no threshold voltage. The
drain source resistance in conducting state rDS,on increases linear from 0.4 Ω to 1.1 Ω at 200 ◦ C.
The switching energies of the JFET are shown in figure 13. Both, the turn-off and the turn-on energies slightly decrease with the temperature. The turn-off and turn-on energies have similar magnitude
Figure 10: Characteristic times of the IGBT
Figure 11: Characteristic times of the JFET
Figure 12: JFET: Conducting losses vs. Temperature
at iC = 4 A (blue solid) and iC = 6 A (red dotted)(top),
rDS,on (bottom)
Figure 13: JFET: Switching losses vs. Temperature at
iC = 4 A (blue solid) and iC = 6 A (red dotted)
between 380 µJ and 330 µJ for Eo f f (6 A) and 300 µJ and 230 µJ for Eon (6 A). The influence of the
temperature on the switching times referring to figure 11 is low. The turn-off and turn-on delay time is
with 15 to 25 ns low and does not depend on the drain current. The fall time of iD is current depending, a
higher current also leads to higher fall time. This also explains the faster increase of the Eo f f , as observed
before, by referring to (2). The rise time is almost independent of the drain current. The current rise is
twice as fast as the current fall. Regarding figure 13 and figure 11 a relation between rise and fall time
and switching energy is observed. A longer rise or fall time leads to a higher turn-on and turn-off time,
respectively.
Analysis of the SiC-BJT
For the SiC bipolar transistor the BitSiC1206 from TranSiC [18] is investigated. The waveforms of the
currents and voltages while turning on and off are shown in 6 and 7, respectively.
Figure 14 shows the calculated conducting losses of the BJT at 4 A (blue solid) and 6 A (red dotted) for
a base current of 1.2 A as a function of the temperature. The green dashed waveform is also measured at
6 A but the base current is adjusted between 310 mA and 800 mA, so that the total losses are minimized.
The first plot shows the conducting losses of the load current circuit, the second the losses caused by
the driving current. The third plot shows the total losses, calculated as the sum of the first and second
plot. It is clearly observable that the conducting losses of the load current increase linear until 170 ◦ C.
After that they start to increase faster. The conducting losses of the loss optimized waveform are slightly
higher because the base current is smaller and hence the collector-emitter resistor is larger. It can be also
observed that both waveforms at 6 A converge with the increase of the temperature because of the rise of
iB that reaches similar values at high temperatures.
The driving losses that occur in the device are almost constant for iB = 1.2 A with the temperature. If
the loss-optimized base current is applied, the losses are at low temperatures only 20% of the losses at
a constant base current of 1.2 A and increase up to 60% at 250 ◦ C. The total conducting losses are the
sum of the losses mentioned before. Here it can be observed that the loss optimized base current leads to
slightly lower losses at all temperatures. The differences are 3.5 W at room temperature and decrease to
1.8 W at 250 ◦ C that corresponds to 29% and 4.3% of the total conducting losses. Regarding the current
Figure 14: BJT: Conducting losses vs. Temperature
of BJT with iB = 1.2A at iC = 4 A (blue solid) and
iC = 6 A (red dotted) and iC = 6A and iB adjusted for
maximum current gain (green dashed)
Figure 15: BJT: Switching losses vs. Temperature
of BJT at iC = 4 A (blue solid) and iC = 6 A (red dotted)
Figure 16: BJT: Current gain vs. temperature for 4
A(blue) and 6 A with iB = 1A (red dotted) and iB optimized (green dashed)
Figure 17: Characteristic times of the BJT
gain β illustrated in fig. 16 (upper) and defined as
β=
iC
iB
(7)
it is observed that β is below 5 for iB = 1.2 A. For the loss optimized base current a maximum gain
of 20.3 at room temperature is achieved, that decreases to 7.7 at 250 ◦ C. The maximum current gain of
40 is achieved at room temperature and a base current of 60 mA. However for this operating point the
conducting losses are very high and hence this is not applicable.
From these results it can be concluded that there is a relation between the maximum operating temperature and the current gain. If the typical junction temperature during operation is known the required base
current for a loss optimization can be read from fig. 16 (lower).
In figure 15 the switching energy is illustrated. Again the blue waveform describes the switching energy
at Ic = 4 A and the red dotted one at 6 A. While the turn-off energy decreases with the temperature,
the turn-on energy is nearly constant up to 185 ◦ C and starts to decrease afterwards. Another interesting
point is that both the turn-off and the turn-on energies increase slower than proportional with the collector
current. The reason for that is that higher current results in a faster rise and fall times of the voltage
because the charge carrier are cleared out faster. The total switching energy is defined as the sum of the
turn-off and turn-on energies.
Figure 17 shows the turn-off and turn-on times of the BJT. The temperature dependence is diverse for the
different values. While the turn-off delay time and turn-off time show an increase with the temperature
the fall time of the current is almost constant. Regarding the turn-on event, all times decrease with the
temperature especially after 180 ◦ C and the collector current dependence is negligible small. For the
turn-off a higher collector current leads to a reduction of td(o f f ) and to f f and to a slightly enhancement
of t f .
Figure 18: IGBT driver: RG =
10 Ω
Figure 19: BJT driver: RB = 10 −
34 Ω, CB = 3, 3 nF
Figure 20: JFET driver: R2 = 12 kΩ,
RG = 1Ω, C3 = 15 nF
Driver
For all tested devices very fast gate drivers have been designed. For the galvanic isolation magnetic
coupler (isoloop) IL712 for the signal and galvanic isolated dc-dc converter (SIM2-1212D SIL7) for
the power are utilized. The over current detection is realized by a current sensing single channel driver
(IR 2127). The specific gate or base signals are processed by an ultra fast low side MOSFET driver
(IXDD414) as proposed in [19]. By gate and base resistors and capacitors the same rise and fall time of
the gate voltage and the base current is adjusted.
The specific gate and base circuits are illustrated in figure 18 to figure 20. For the IGBT the gate voltage
is between -5 and 15 V and the driver is connected to the IGBT via a gate resistor RG . The negative
voltage is achieved by a zener diode and a resistor. The voltage across the IXDD414 is 20 V. The zener
diode allows a voltage drop of 15 V and the remaining voltages drops across the resistor. Regarding the
virtual source potential as ground the gate signal is switched between +15 V and -5 V. The capacitors C1
and C2 are for stabilization of the voltage. For the BJT there is an adjustable base resistor RB to limited
the base current and in parallel a small capacitor CB for a fast rise and fall time of the base signal.
High efforts are necessary for the gate voltage driver of the JFET since the JFET is blocking at gate
voltage between -15 V and -22 V depending on the device and the voltage should not be lower than a
few volts from its pinch-off voltage. The JFET starts conducting above this voltage and has its lowest
on-resistance at +1 V. To realize these specifications a virtual source potential as done for IGBT driver
is created by a resistance, a zener diode and the two capacitors as in figure 20. The value of the zener
diode is 1.2 V so that the gate signal is switched between +1.2 V and -22.8 V. However -22.8 V at the
gate of devices with high pinch-off voltage might destroy the gate, so that this voltage has to be limited.
In [9] an effective circuit to limit the gate voltage individual for the specific JFET is proposed. The diode
D1 ensures a fast turn-on of the device. For the turn-off the diode is blocking and to discharge the gate
capacitance fast the current is flowing through the capacitance C3 . While the JFET is turned-off the large
resistor R2 limits the gate voltage, since a small leakage current through the gate capacitor starts to flow
than the pinch-off voltage is exceeded. To adjust the rise and fall time of the gate voltage a gate resistor
RG is used.
Comparison of the Devices and the Driving Efforts
All three investigated devices have a maximum blocking voltage of 1200 V. The chip sizes of the JFET
and the BJT are similar, the chip size of the IGBT is slightly larger. The rated continuous current at
125 ◦ C is 6 A for the IGBT and the BJT. For the JFET only the pulsed current of 10 A is known. But both
SiC devices have the same chip size and are mounted in a TO-220 housing, while the IGBT is mounted
in a TO-247. Even though not all parameters of the devices are exactly the same due to availability and
their different operating modes they are similar enough for comparison. Therefore the total losses of the
IGBT, BJT and JFET defined as the sum of switching, conducting and for the BJT also of driving losses
are calculated. Since the switching losses are linear to the switching frequency figure 21 to 23 show
the total losses versus the temperature and the frequency at 4 A (lower blue surface) and 6 A (upper
red surface). For the BJT the 6 A surface is with the loss optimized base current. For comparison it is
assumed that the devices are operating at a case temperature of 80 ◦ C and have a thermal junction case
resistance of Rth,JC = 1.7 K/W. That leads to following maximum power dissipations at 25 ◦ C ambient
temperature:
• Ptot (T j = 150 ◦C) = 41, 1 W
• Ptot (T j = 200 ◦C) = 70, 6 W
• Ptot (T j = 250 ◦C) = 100 W
Figure 21: IGBT: Total power
losses vs. temperature and frequency at 6 A (red upper surface)
and 4 A (blue lower surface
Figure 22: BJT: Total power losses
vs. temperature and frequency at 6
A (red upper surface) and 4 A (blue
lower surface
Figure 23: JFET: Total power
losses vs. temperature and frequency at 6 A (red upper surface)
and 4 A (blue lower surface
Figure 24: Comparison of the total losses at 150 ◦ C, 40 kHz and 6 A and 600 V
For the IGBT it is concluded that the total power losses at fs = 100 kHz are almost doubled from 25 ◦ C to
150 ◦ C. The absolute loss increase from room temperature to 150 ◦ C is ∆Ptot (100 kHz) = 53 W. Applying
the maximum power dissipation of 41,1 W as a basis, a theoretical maximum switching frequency of 25
kHz is feasible.
The temperature dependence of power losses of the BJT is with ∆Ptot (100 kHz) = 12 W from 25 ◦ C to
150 ◦ C much smaller. Assuming also a maximum power dissipation of 41,1 W a maximum switching
frequency of 48 kHz is possible if the junction temperature is limited to 150 ◦ C. If junction temperatures
of 200 or even 250 ◦ C are allowed the maximum switching frequency increases to 100 kHz and above.
The power losses of the JFET increase from 25 ◦ C to 150 ◦ C as for the BJT with ∆Ptot (100 kHz) = 12
W. Since the JFET has high fraction of conducting power losses, the theoretically maximum switching
frequency at 150 ◦ C is with 14.2 kHz low. Allowing a junction temperature of 200 ◦ C the maximum
switching frequency increases to 45 kHz.
Figure 24 shows the distribution of the power losses for 150 ◦ C, 40 kHz, 600 V and 6 A, where the BJT
has the lowest total losses and the Si-IGBT has the highest. However the power distribution of the losses
is very different. While the IGBT is very optimized concerning its conducting losses, the JFET has the
lowest switching losses. For the BJT the switching and conducting losses are uniformly distributed if the
driving losses are added to the conducting losses. That leads to the conclusion that the IGBT is optimized
for low switching frequencies, while the JFET is preferred for very high frequencies. However because
its high RDS,on the conducting losses are high, so that the current has to be derated. The BJT is able to
carry its rated current over a wide switching frequency range.
The efforts for the driver of the JFET are very high because of the device specific pinch-off voltage,
the required negative voltage and the protection circuit. That means that as long as there is a drainsource-voltage across the JFET the driver must provide a negative gate voltage, even when the main
power supply has a failure. It is also not possible to use the dc-link voltage for the driver supply, because
usually first the drivers have to operate and after that the dc link can be charged. The current drawback
of the device depending pinch-off voltage will be overcome by future development of the JFET in the
opinion of the authors. In case that the SiC-JFETs get widely used in power electronic application, it
is to expect that there will be special JFET driving IC on the marked as it is for IGBTs and MOSFETs.
The efforts for driving the BJT is, compared to an IGBT, higher because a base current is required. This
current is proportional to the collector current. That means that the driver has to be able to deliver 5 to
10 % of the transferred power. For comparing the losses the driving losses cannot be neglected. For the
IGBT the driver appears to be simple since standard components that has already been developed for this
purpose can be used.
Conclusion
Measurement results of a SiC-BJT (BitSiC1206, TranSiC), a JFET (SiCED) and a state of the art IGBT
(IKW08T120, Infineon) are presented, analyzed and compared. The experimental setup is described;
measurements of the switching and conducting behavior are taken and analyzed. Based on these results
a comparison of the selected three devices is done focusing on the switching, conducting and driving
losses as well as the total losses. Other aspects as the different performances concerning switching and
delay time and driving efforts are evaluated. It can be concluded that the BJT has the lowest total losses
around the chosen operating point (600 V, 6 A, 40 kHz), although a certain base current in steady state is
necessary to drive the device. The JFET has the lowest switching losses but high conducting losses that
strongly reduces the total efficiency. The total losses of SiC-devices show a low temperature dependence,
the switching losses decrease with the temperature.
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