Progress in holographic video displays based on guided-wave
acousto-optic devices
Quinn Y. J. Smithwick, Daniel E. Smalley, V. Michael Bove, Jr., and James Barabas
Object-Based Media Group, MIT Media Laboratory, Room E15-368, 20 Ames St., Cambridge, MA
USA 02142-1308
ABSTRACT
The novel frequency-multiplexed modulator architecture of the MIT Mark III holo-video display poses a significant
challenge in generation of appropriate video signals. Unlike in our previous work, here it is necessary to generate a
group of adjacent single-sideband RF signals; as this display is intended to be manufacturable at consumer-electronics
prices we face the added requirement of compact and inexpensive electronics that are compatible with standard PC
graphics processors. In this paper we review the goals and architecture of Mark III and then describe our experiments
and results in the use of a hardware/software implementation of Weaver’s single-sideband modulation method to upconvert six 200MHz baseband analog video signals to a set of RF signals covering a nearly contiguous 1GHz range. We
show that our method allows efficient generation of non-overlapping signals without aggressive filtering.
Keywords: synthetic holography, 3-D display, holographic video, acousto-optic devices, RF signal processing
1. INTRODUCTION
In an effort to develop an inexpensive holographic video display of high image quality, we have developed a modified
Scophony architecture which centers on a LiNbO3 2-D guided-wave device and eliminates the high-speed horizontal
scanning mirror that normally is required to provide a stationary image in a Scophony display but that adds complexity
and limits the scalability of such an architecture. [1] Our method – which employs a fixed helical mirror or an equivalent holographic optical element (HOE) – is related to a discretely-stepped-mirror architecture that was proposed by
Son, et al. in conjunction with a pulsed laser source. [2]
Fig. 1. MIT Media Laboratory Mark III holo-video display architecture.
The Mark III display (Figure 1) is a proof-of-concept monochrome horizontal-parallax-only (HPO) display based on
guided-wave devices we have developed and fabricated. It is intended to point the way toward holo-video displays for
consumers, which means that (unlike our earlier systems) it must be compact, manufacturable for at most a few hundred
dollars, and capable of being driven by standard graphics hardware of a PC or game console (rather than specialized
hardware). Such a display would be able to take advantage of the large amount of 3-D visual data now underlying gaming applications and on-line virtual worlds as well as other consumer applications such as home-design CAD software.
While Mark III will be able to replay any prerecorded bandlimited horizontal-parallax-only hologram of suitable resolution, for real-time computation, generation, and display, Mark-III uses diffraction-specific holographic encoding. [3]
In order to achieve the space-bandwidth product needed for a standard-definition-television (SDTV) resolution image,
with a reasonable view angle (25° for Mark III), size (80mm), and refresh rate (30Hz), the light modulator must have a
horizontal bandwidth of at least 1GHz. Current higher-performance graphics processors typically use three (RGB)
channel, 8-bit, 400Mpixel/second RAMDACs for analog output, corresponding to an analog bandwidth of 200MHz per
channel. A dual-head graphics card therefore has a 1.2GHz bandwidth, more than enough to generate the required signals, though subdivided into six 200MHz baseband channels. With the exception of a DC offset and programmable
Hblank and Vblank regions, the signals can be arbitrary. The 0.35V DC offset is due to the analog output’s standard
VGA 0-0.7V voltage range. The Hblank and Vblank regions are relatively brief and remove only portions of views. As
we have seen in previous implementations, there is negligible effect on the viewing experience. [4]
Although a guided-wave device can have a very high overall bandwidth it is difficult to make an individual transducer
have a bandwidth of more than a few hundred MHz. Thus our guided-wave devices have five horizontal channels
(transducers) corresponding in bandwidth and frequency spacing to the bandwidths of the output channels of a dualhead graphics chip (e.g. NVIDIA Quadro FX 4500). A sixth vertical channel is used in conjunction with the special
HOE to eliminate the need for horizontal scanning mirrors.
Scophony displays require the upconversion of baseband video to the operating frequency range of the light modulator,
typically not as a standard amplitude-modulated (AM) signal but instead as a single-sideband (SSB) signal. There are a
number of well-known methods for doing such upconversion (as we discuss below) but in our system we have to generate not just a single such signal but rather five adjoining and non-overlapping signals corresponding to the five horizontal channels of the guided-wave device, and doing so inexpensively poses a significant signal-processing challenge. In
this paper we present a solution to this problem which uses inexpensive circuitry and makes efficient use of the graphics
processor.
For the horizontal inputs to the guided-wave device, five 200MHz baseband signals need to be shifted in frequency (upconverted) to adjacent frequency bands covering 200MHz to 1.2GHz. Frequency bands must abut without overlap. Unwanted upconversion artifacts cause frequency bands to overlap creating spurious images and views in the holodisplay.
The vertical input to the device requires a constant amplitude chirp signal between 460-490MHz. This signal doesn't
carry holographic information, but is needed to drive the vertical scan’s sweep. A single channel from the graphics card
is sufficient to cover this bandwidth, but must be upconverted to the appropriate frequency range (see Figure 2).
Fig. 2. R, G, and B channels from a dual-head graphics card are treated as independent 200MHz signals and SSB upconverted appropriately for the channels of the guided-wave device. The challenge is to make the sidebands abut but not overlap.
2. UPCONVERSION
In this section we discuss the conversion of the baseband signals to the appropriate frequency ranges. This topic is well
known to radio engineers, but likely less so to holographers; understanding the effects of the upconversion on the appearance of holo-video images entails a bit of mental exercise no matter what one’s background.
2.1. Double-sideband AM
The standard means of upconversion is by mixing – multiplying the baseband signal by a high frequency sinusoidal
carrier. [5] The baseband signal will be shifted in frequency by the carrier. A mirror image of the baseband signal about
the carrier frequency will also be produced. This is a double-sideband AM signal (DSB-AM) and results from the positive and negative frequencies of the baseband signal becoming centered on the carrier frequency (see Figure 3).
In practical terms, each modulated chirp signal represents a section of a converging lenslet producing a portion of a real
emitter with variable views. An emitter is recreated using five frequency adjacent chirps with each chirp representing
only one-fifth (1/5) of the total views. The unwanted sideband frequencies extend each chirp and its field of view with a
mirror-flipped set of views. The extended chirp will overlap with adjacent chirps producing ghost images in different
views. There will thus be five ghost images for each emitter. These ghost images are undesirable, and thus we need an
upconversion method that results in only a single sideband. There are three common methods of creating singlesideband signals, and we will discuss them in the next three sections.
Fig. 3. Frequency upconversion by mixing the baseband video signal with a sinusoidal carrier results in dual sidebands, and thus
ghost views.
2.2. Single-sideband modulation by filtering
In the filtering technique (Figure 4), a highpass or bandpass filter is used to remove the lower sideband (the mirrored
image) after mixing. Ideally, this would be a “brick wall” filter. Practically, we must be able to suppress the lower sideband by 50dB within 7MHz. A bright view on the edge of one frequency band would be totally dropped to one gray
level within the first view of the adjacent frequency band. Note that one view of the ghost image may still partially exist
overlapping the adjacent view.
As the frequency range increases, the filtering must become more aggressive to achieve the same amount of suppression
within 7MHz. This becomes not only expensive, but impractical at higher frequencies. For example, for the frequency
band ending at 1GHz, we would need to drop 50dB within 0.7% of the cutoff frequency to reject the lower sideband.
Fig. 4. Single-sideband generation through filtering requires filters to remove the unwanted sideband created during mixing. Filters
must be increasingly (and impractically) aggressive at higher modulation frequencies.
2.3. Single-sideband modulation by phase techniques
Phase techniques use two real signals, in-phase (I) and quadrature-phase (Q) signals, to represent mixing of complex
signals. Real signals consist of sinusoids with both positive and negative frequencies responsible for the double sideband after mixing. Complex signals can have purely positive or negative frequencies, and can therefore have single
sidebands after upconversion. Unfortunately, complex signals can’t be implemented directly, but need two real signals
(I/Q) to represent their real and imaginary parts. I/Q signals have opposite-signed negative frequencies. The I/Q signals
are upconverted by mixing with quadrature carriers (cosine and sine signals at the same frequency) and then added. The
lower sidebands (which are the negative frequencies upconverted) are cancelled leaving a single (upper) sideband signal
(Figure 5).
Fig. 5. Phase-based SSB generation uses a Hilbert filter to create I and Q signals which are multiplied by two quadrature carriers.
Their sum cancels out the unwanted sideband.
Saying that I and Q signals have opposite-signed negative frequencies is equivalent to saying that the quadrature-phase
signals have all frequency components shifted by 90° from their in-phase counterparts. A cosine wave in the in-phase
signal becomes a sine wave in the quadrature-phase signal. A quadrature signal can be created from an in-phase signal
by using a Hilbert filter. A delay is usually applied to the in-phase signal to account for the delay of the Hilbert filter.
Unfortunately, Hilbert filters can only be approximated and aren’t effective at low frequencies.
Alternatively, the quadrature pair can be computed directly in the graphics chip; that is, a cosine chirp and a sine chirp
can be generated on two separate channels. Unfortunately, this is not an efficient encoding as the I/Q pair would require
two 200MHz baseband signals to produce a single 200MHz SSB signal. The graphics chip (or chips) would need to
provide 2GHz of output bandwidth to create our needed 1GHz of upconverted signals. This method requires two carriers and two mixers as well as an adder to upconvert each I/Q pair.
2.4. Weaver’s Third Method
In Weaver’s Third Method, [6] (Figure 6) the I/Q pair is created by downconverting the baseband signal using two
quadrature carriers so the resulting desired sideband is centered at DC, while the undesired sideband is at higher frequencies. The downconverted pair is filtered to remove the undesired sideband. Note that one view of the ghost image
may still partially exist, extending into and overlapping the adjacent view. The filtered downconverted I/Q pair is upconverted to the appropriate frequency range using quadrature carriers to create a single sideband signal. This process
requires four carriers, four mixers, two filters, and an adder.
Fig. 6. Weaver’s Third Method of SSB generation first downconverts and lowpass filters the baseband signal to generate an I/Q pair.
Single-sided chirps become double-sided chirps. The I/Q pair is then upconverted with quadrature carriers and summed to cancel the
unwanted sideband.
In our case, the downconversion and filtering process produces two 100MHz upper sideband signals with the original
higher frequencies folded. The single sided chirp functions (DC–fmax) become double sided chirp functions (fmax/2 to DC
to fmax/2) after downconversion and ideal filtering of the lower sideband. Because the filtering acts on downconverted
baseband signals, the filter is the same regardless of the final upconverted frequency range. The lower sidebands can be
filtered using a gentler filter, as 7MHz is 7% of the 100MHz bandwidth. However, expecting 50dB down in 7% of the
frequency range from an analog filter is still difficult and expensive. The upconversion process results in a higher frequency single sided chirp.
2.5. Precomputed Weaver method
To ease the analog hardware requirements, we propose to compute the effects of downconversion and filtering in software. The digital mixer is a simple multiplication operation, and achieving sharp cutoffs with digital filters is relatively
simple with the slight disadvantage of computational cost and filter delay. Since we are downconverting, the bandwidths of the signals remain in the bandwidths of the graphics card channels. We can convert the result of the digital
mixing and filtering to an analog signal output from the graphics card.
However, we don’t actually have to do the digital mixing and filtering, as the signals we are downconverting are known
in advance – chirp functions. We can perform the mixing and filtering analytically by generating the chirps as they
would appear after the downconversion and filtering. The single-sided chirp functions become double-sided chirp functions after downconversion and ideal filtering of the lower sideband. The effective “filtering” of the undesired sideband
will be perfect, since only the desired sideband is computed. Therefore, we can directly modulate a pair of 100MHz
double-sided quadrature chirps in software, output them as an analog signal, then do a single I/Q mixing upconversion
(Figure 7). This would remove the need for two carriers, two filters, and two mixers. The encoding is more efficient
than in the computed phase-method discussed above, as now two 100MHz signals are needed to generate one 200MHz
signal. Unfortunately, each 100MHz signal comes from a separate 200MHz bandwidth channel of the graphics card,
thus wasting half the channel’s bandwidth.
Fig. 7. Precomputed Weaver’s Third Method analytically computes the I/Q pair that would result from downconversion and filtering,
saving a modulation and filtering stage.
To avoid this wasted bandwidth, two 200MHz double-sided quadrature chirps (which each fill a 200MHz bandwidth
channel) are generated to create one 400MHz SSB signal. This signal is fed into two of the guided-wave device transducers in series. Each transducer responds only to frequencies in its passband, allowing the remaining signal to pass to
the next channel. We can take advantage of the fact that the guided-wave device is designed for wide bandwidth rather
than brickwall bandpass behavior. In the overlapping bandpass regions, each transducer will take a portion of the energy.
Since there are an odd number of horizontal channels, the unpaired 200MHz channel will be upconverted using standard
mixing and the double-sided signal filtered. To use the upper sideband and to avoid overlapping adjacent channels, this
200MHz channel must be the lowest frequency band. The remnants of the lower sideband will be a DC–200MHz signal,
the majority of which is outside the lowest transducer’s passband. The portion within the passband will produce a virtual emitter whose location is outside the field of view of the real emitters 200MHz–1.2GHz and thus is not interfering.
The remaining 200MHz graphic card channel is used for the 60MHz bandwidth chirp used by the vertical scanner. This
is upconverted to a double-sideband signal (370–490MHz). The input signal is not generated as 60MHz baseband signal, but rather a 40–200MHz signal to provide a 280MHz empty band between sidebands. Either a separate filter or the
inherent bandpass filter nature of the guided-wave device should filter out the lower sideband.
Using the precomputed I/Q Weaver method, the undesired sideband is fully suppressed without the need for filtering
and the graphic card bandwidth is fully utilized. The precomputation of the I/Q pair removes the need for two mixers
and two lowpass filters for each Weaver modulator, while the use of 200MHz I/Q signals fills the graphics card’s channels and removes the need for two entire Weaver modulators.
3. IMPLEMENTATION
3.1. Hardware
To implement the precomputed I/Q Weaver Modulator, we use an Analog Devices ADL5385 50MHz to 2200MHz
Quadrature Modulator. The ADL5385 takes two differential baseband inputs (I/Q pair) and mixes them with two quadrature carriers, then sums the results together. The quadrature carriers are derived on the ADL5385 from an external
local oscillator (LO) input with twice the desired carrier frequency. This ADL5385’s generation of the quadrature carriers is advantageous not only in reducing the additional hardware in creating quadrature carriers, but also in assuring the
accuracy of those carriers. If the two carriers are not exactly in quadrature, then the cancellation of the sidebands will
not be complete.
Fig. 8. One of three small PC boards used in our display for upconversion via the precomputed Weaver method (actual size about
7.5cm square).
As the ADL5385 requires differential baseband inputs, an AD8131 High Speed Differential Driver (400MHz full power
bandwidth) converts the single-ended graphics card output to a differential signal. The graphic card’s output first passes
through a capacitor to remove any DC component (as a standard VGA signal is DC–0.7V). This highpass filter can be
made to block signals well below 7MHz, so that the majority of that view is passed.
The local oscillator (LO) is provided by an Analog Devices ADF4360-XX family Integrated Synthesizer and voltage
controlled oscillator (VCO) with Fox Crystals FOX924B 16MHz TCXO crystals. The ADF4360’s use an integer N
divider to divide down the integrated VCO output and an integer R divider to divide down the local oscillator. The two
divided signals are locked together with an onboard phase-locked loop (PLL) thus providing a highly stable high frequency output. Selection of the N and R divider values and hence the output frequency are controlled by internal registers programmed via I2C.
An Atmel Atmega-32 microcontroller is used to program the ADF4360 at startup. The microcontroller uses common
clock and data pins for all the ADF4360’s, and controls an enable line for each ADF4360.
The outputs of the ADL5385 are sent to a Minicircuits HELA-10 RF amplifier then to the SAW transducers.
The use of standard wide-band integrated components eases the RF circuit design and reduces costs. As the ADL5385
can handle 50MHz to 2200MHz, the same chip can be used for all three frequency ranges. Likewise, the single-ended to
differential conversion takes place on baseband signals and has a 400MHz bandwidth, so all three modulators can use
the same ADL5385 Differential Driver. The ADF4360-XX family have similar pinouts, except for the need for an external filter loop and inductor to set the frequency. The programming registers of the ADF4360-XX family are also the
same, so programming and layout are very similar. The majority of the PCB layout can be reused for different frequency ranges with only slight modifications for the PLL filters and frequency selection inductors for the ADL5385.
This is important as the effectiveness of RF PCB design is dependent on the proper layout. See Figure 8 for the
800MHz to 1.2GHz precomputed I/Q Weaver Modulator PCB.
3.2. Implementation
For verification of the hardware, we programmed the NVIDIA Quadro FX 4500 graphics card using OpenGL to output
a ramped-amplitude 200MHz double-sided cosine chirp on one channel and a ramped-amplitude 200MHz double-sided
sine chirp on another. This signal is sent to the quadrature mixer to be upconverted to the 200–400MHz band. After
quadrature upconversion and summation, this produced a ramped 400MHz single-sided chirp, as desired.
Viewed on a spectrum analyzer, after upconversion a ramped 400MHz bandwidth chirp is observed from 200–400MHz.
However, we observed a –3dB droop at 400MHz (consistent with the DVI/VGA specification), as well as a general sinc
envelope due to the sample and hold of the DAC. Both of these can be accommodated in software by predistorting the
inputs to the mixer, although some grayscale level is lost. The reconstruction image is also apparent.
3.3. Reconstruction filter
The reconstruction image just mentioned is not related to the DSB/SSB modulation process, but rather to the video digital-to-analog conversion. Because we are operating at the Nyquist limit of the DAC, the reconstruction image is abutted
to the upconverted signal. The reconstruction filter on the DAC has a very gradual attenuation, definitely not –50dB in
7MHz. Although this is a similar problem to the SSB creation using filtering, the filtering occurs only at the 200MHz
range before mixing.
Coincidentally, 200MHz is the same frequency as US television channel 11. We can buy a channel deletion brick wall
lowpass filter (Microwave Filter Company) to remove all channels above channel 11 (i.e. all frequencies above
200MHz). These filters would go between the graphics card outputs and the single ended to differential conversion. The
brick wall filter can achieve greater than –50dB within 7MHz at 200MHz cutoff frequency. Unfortunately, the cost of
these filters is prohibitive, at over $900 per filter, and one filter is required for each of the six graphic card output channels.
We can reduce the requirements of the filter, say to –50dB within 14MHz at 200MHz cutoff frequency, then commercially-available filters come down in price to $195 per filter for low quantities, although this is still too expensive for
our project. So although it is possible to replace the graphic card’s DAC reconstruction filter with a more aggressive
one, the added expense does not fit into the goals of our project.
3.4. A revised design
In order to achieve our design goals, we must sacrifice using the graphics card to generate the vertical scanner’s signal.
Fortunately, the vertical scanner’s chirp is not modulated, but rather a constant known signal. It can be created using
direct digital synthesis (DDS) based on low cost Analog Devices AD9954 400MSPS Direct Digital Synthesizers. Al-
though it is possible to directly create a chirp in the 430–490MHz range by using a DDS, VCO and a PLL (Analog Devices AD9858 1GSPS DDS), we have opted to do DDS in the 30–90MHz range and upconvert to the 430–490MHz
range using phase upconversion techniques previously described.
The guided-wave device can be modified to have six (instead of five, as discussed above) transducers covering the same
1GHz bandwidth. The 200MHz channel previously used to generate the vertical scanning signal can be reallocated to
the added horizontal channel, and the 1GHz signal divided into 6 channels. This would allow 166MHz per channel for
signal, with a 68MHz guard band between each signal and its first image. An inexpensive ($6) lowpass filter (Minicircuits) can easily remove the first image (Figure 9). The filtered signal pairs can then be upconverted using quadrature
mixing to achieve single sideband signals without any images.
Fig. 9. Precomputed Weaver modulation plus image filtering adds a guard band so that inexpensive filters can reject the reconstruction image.
4. CONCLUSIONS
We have shown that multiband wide-bandwidth holo-video signals can be produced using a single standard dual head
graphics card and precomputed I/Q Weaver modulation. The precomputed I/Q Weaver modulator has comparable sideband rejection ability and hardware requirements of a phase-based modulator but uses the graphics card’s bandwidth
maximally efficiently. Our implementation uses standard RF integrated circuits, easing circuit design and enabling
compact, inexpensive, and power-efficient drive circuitry. Experiments showed the expected rejection of unwanted
sidebands. Realizable filters exist that can remove the abutted reconstruction image due to the operation of the DAC at
the Nyquist limit, but they are too expensive for use in this project; thus we developed an improved technique. In the
latter case we use direct digital synthesis to generate the vertical scanner signal, reallocating the freed graphics-card
bandwidth for a guard band in six horizontal channels. This allows inexpensive filters to remove the reconstruction image.
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