AN11618
SSL523XT buck-boost controller
Rev. 2 — 31 March 2016
Application note
Document information
Info
Content
Keywords
SSL5231T, SSL5235TE, SSL5236TE, SSL5237TE, buck-boost converter,
non-isolated LED controller/driver, retrofit SSL, LED, Boundary
Conduction Mode (BCM), Discontinuous Conduction Mode (DCM), low
Total Harmonic Distortion (THD), external switch, constant current
Abstract
This application note describes how to design a dimmable single mains
buck-boost converter with low-THD performance for general lighting
applications. The SSL5231T IC requires an external MOSFET. The
SSL5235TE, SSL5236TE, and SSL5237TE ICs have integrated
MOSFETs.
The mains dimmable SSL523XT LED driver supports wall dimmers.
The SSL523XT incorporates an integrated overvoltage protection and an
internal temperature protection.
The SSL523XT is a cost-effective solution for mains dimmable
applications.
AN11618
Silergy Corp.
SSL523XT buck-boost controller
Revision history
Rev
Date
Description
v.2
20150430
second issue
Modifications:
v.1
20150305
AN11618
Application note
•
Text and graphics have been updated throughout the application note.
first issue
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AN11618
Silergy Corp.
SSL523XT buck-boost controller
1. Introduction
This application note describes how to design a mains phase-cut dimmable single mains
buck-boost converter in a low THD configuration for general lighting applications using the
SSL523XT.
This driver operates in BCM using on-time control and valley detection for efficient
switching. This application note describes the design considerations for the SSL5231T
(external MOSFET) and the SSL5235TE, SSL5236TE, and SSL5237TE (integrated
MOSFET).
The SSL523XT platform is intended for SSL retrofit applications and fixtures. It is
optimized for use in cost-effective, high-efficiency driver solutions for high-voltage LED
strings or LED modules.
Further information and design tools can be found on the NXP web site. They are also
available through your local sales office.
2. Basic operation theory
Before going into detail on the SSL523XT applications, it is important to have a basic
knowledge of buck-boost converters.
The basic components of the buck-boost converter consist of an inductor, a switch that
controls the inductor current, and a rectifying diode. Operation alternates between
connecting the inductor to the source voltage to store energy in the inductor and
discharging the energy into the load. See the application note AN11060, "TEA172X 5 W to
11 W Power Supply/USB charger" (Ref. 5) for more information about buck-boost design
considerations.
See the application note AN11136, "SSL2109/SSL2129AT controller for SSL applications"
(Ref. 6) for more information about external MOSFET design considerations.
AN11618
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AN11618
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SSL523XT buck-boost controller
3. Functional description
The SSL5231T application requires an external MOSFET. The SSL5235TE, SSL5236TE,
and SSL5237TE driver ICs incorporate an internal high-voltage power MOSFET.
The SSL5231T, SSL5236TE, and SSL5237TE ICs are intended for low-mains as well as
high-mains applications. The SSL5235TE IC is only intended for low-mains applications.
The SSL523XT IC has the following features:
• Switch-mode buck-boost controller with power-efficient BCM operation including:
– Minimal reverse recovery losses in rectifying diode
– Zero Current Switching (ZCS) for switch turn-on
– Zero voltage or valley switching for switch turn-on
– Minimum inductance value and size for the inductor
•
•
•
•
•
High Power Factor (PF; > 0.9) and low-THD mode
High efficiency (up to 90 % depending on the output power and the output current)
Ultra-low IC current during operation (< 150 A)
Mains phase-cut dimmable (see AN11136; Ref. 6)
Fast output transient response through cycle-by-cycle current control, preventing
overshoots and undershoots in the LED current
• Internal protections:
– UnderVoltage LockOut (UVLO)
– Leading-Edge Blanking (LEB)
– Cycle-by-cycle OverCurrent Protection (OCP)
– Short Winding Protection (SWP; see AN11136; Ref. 6)
– Internal OverTemperature Protection (OTP)
– Output Short Protection (OSP)
– Output OverVoltage Protection (OVP)
See the SSL523XT data sheets for more information (Ref. 1 to Ref. 4).
AN11618
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AN11618
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SSL523XT buck-boost controller
4. Step-by-step design procedure
This section provides a step-by-step guide for designing a buck-boost converter
application with the SSL523XT.
Remark: The derivation of the equations applied is beyond the scope of this application
note. Where values used in formulas are application-specific, reasonable estimates have
been made.
4.1 Basic configuration
The converter using the SSL523XT is a source switched, BCM, on-time controlled
buck-boost system. Figure 1 shows the basic application diagram of a typical buck-boost
application for the SSL5235TE, SSL5236TE, and SSL5237TE with an internal
high-voltage power MOSFET, driving a single LED string.
The mains voltage is rectified, buffered, and filtered in the input section. It is connected via
the driver IC to inductor L4.
The internal high-voltage switching transistor is connected as a switched source. The
GATE pin (pin 1) is connected to the VCC pin (pin 3) of the IC.
At switch-on, the source of the internal high-voltage power FET is connected to the
(floating) IC-ground via the internal low-voltage power MOSFET and the external sense
resistor R13.
When the internal low-voltage power MOSFET is switched off, the internal high-voltage
power MOSFET cannot conduct current via the source terminal anymore. So the voltage
at the source of the high-voltage MOSFET starts to increase. The gate is kept at a
constant DC voltage. As a consequence, the gate-source voltage starts to decrease.
When the gate-source voltage drops to below the gate-source threshold voltage value (of
the high-voltage MOSFET), the internal power MOSFET is switched off.
This converter operates at the boundary between Continuous Conduction Mode (CCM)
and Discontinuous Conduction Mode (DCM). When the internal source switch is switched
on at t0 (see Figure 11), the inductor current IL builds up from zero proportionally with Vbus
during source switch on-time. Energy is stored in the inductor. When the internal source
switch switches off, IL flows through the rectifying diode D3 and the output capacitor. The
current through the inductor drops proportionally in time with the value of the fixed LED
voltage. When IL reaches zero, and after a short delay caused by valley detection, a new
switching cycle is started. Valley detection reduces the switching losses significantly (see
Section 4.5).
AN11618
Application note
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AN11618
Silergy Corp.
SSL523XT buck-boost controller
L3
BD1
MD5S
R9
750 kΩ
R7
180 kΩ
R10
750 kΩ
R8
180 kΩ
2
U1
R2
C2
47 nF
400 V
5
6
4
R11
22 kΩ
7
DRAIN
8
VLED
90 V
36 mA
IC
9
COMP
330 Ω
TP
R12
910 kΩ
3
2
1
GATE
RV1
VDRS07H275BSE
C4
47 nF
400 V
DIM
2
1
GND
4
3
MST1A250V
DEMOVP
X1
MKDS
1.5/2
1
+
ISNS
F1
Vbus
3.3 mH
VCC
230 V (AC)
10 W
PF > 0.9
D3
ES1J
C6
2.2 μF
6.3 V
R16
150 kΩ
R15
5.6 kΩ
C7
47 nF
50 V
C5
2.2 μF
50 V
R17
160 kΩ
R13
8.2 Ω
L4
4.7 mH
GNDL
LEDN
K2
LED
connections
C9
220 μF
100 V
LEDP
aaa-017683
(1) The dots of inductors L3 and L4 indicate the start of the winding.
(2) n.m. = not mounted.
Fig 1.
SSL5235TE, SSL5236TE, and SSL5237TE basic application diagram
Remark: In Figure 1, the LED string is connected to the side of buffer capacitor C4 that
has negative potential. It prevents that the LEDs have a high-frequency voltage swing
equal to the drain-source voltage swing. The LED assembly is relatively large with
extended wires and a heat sink. The assembly has substantial capacitive coupling
regarding its surroundings. In combination with a high-frequency voltage swing, the
capacitive coupling has a negative effect on efficiency and EMC.
By measuring the inductor current IL4 via sense resistor R13, the on-time is regulated so
that the averaged ISNS voltage Vintregd(AV)ISNS) during the off-time of the main switch is
regulated to 300 mV. The average output current Iout(AV) is calculated with Equation 1
V
I out  AV  = 0.3
------------R13
(1)
4.2 Input section
The SSL5231T, SSL5236TE, and SSL5237TE platforms support AC mains voltages of
100 V, 120 V, or 230 V. The SSL5235TE is a dedicated driver for low-mains applications
only. For all applications, the input section incorporates:
•
•
•
•
AN11618
Application note
The rectifying stage
Protection against overvoltage
Protection against overcurrent and inrush peak current
Buffer circuit with EMI filter
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AN11618
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SSL523XT buck-boost controller
4.2.1 Mains OverVoltage Protection (OVP)
The AC mains input voltage is rectified with diode bridge BD1. The Metal Oxide Varistor
(MOV) device (RV1) is added for overvoltage protection. All components of the input
stage must withstand the voltage at which RV1 operates. The protection voltage level is
calculated with Equation 2:
V RV 1 =
2  V mains  nom   
(2)
Where:
•  = 1.45 for TRIAC dimmable applications and then rounded up to the next 10 V
(NEMA SSL7A table 3 (Ref. 12)).
The internal high-voltage MOSFET must withstand a higher voltage: The output voltage
must be added to the protection voltage level.
V DRAIN =
2  V mains  nom    + V out
(3)
4.2.2 Mains OverCurrent Protection (OCP)
The primary protection against overcurrent is a fuse or fused resistor that breaks down
when the current is too high. When a fuse is selected, choose a value that handles the
inrush current while still providing protection. In practice, a slow fuse with a rating of 1 A to
1.5 A is sufficient. If a fused resistor is selected, the minimum value for the series resistor
for inrush current protection (see Figure 4) can be calculated with Equation 4. Typically,
for most diode bridge rectifiers, the IFSM peak forward surge current is about 20 A.
R fuse =
V mains  max 
2  ---------------------------I FSM
(4)
Example:
At Vmains = 230 V (AC), +20 %, Vmains(max) = 276 V (AC), the calculated value
R1 + R2 = 19.4  The practical value for Rfuse = 20 . So 10  for each resistor.
To meet the inrush current and dimmer requirements, series resistors R1 and R2 are
placed to dampen the (repetitive) mains input current peaks.
In addition to the resistive value, the continuous power dissipation is important. The power
dissipation depends on the power consumption of the complete circuit. For sinusoidal
input current, the power dissipation for resistors R1 and R2 can be calculated with
Equation 5.
2
P tot
P R1 = R1  -----------------2
V mains
(5)
With phase-cut dimmers, especially leading-edge dimmers, the dissipation is much
higher. The dissipation depends on the required output power, the buffer capacitance, and
the dimming angle.
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Application note
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AN11618
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SSL523XT buck-boost controller
Example:
• Vmains = 230 V (AC)
• Ptot = 4 W
• R1 = R2 = 330 
The result: The total dissipated power in resistors R1 and R2 is 2  100 mW.
4.2.3 Buffer circuit with EMI filter
The buffer with EMI filter circuit consists of capacitors C2 and C4 and inductor L3.
The circuit has a dual functionality:
• Filtering ripple current due to converter operation ensuring compliance with legal
standards and regulations for mains conducted emissions.
• Storing energy to supply the converter during the mains voltage zero crossings.
4.2.3.1 EMI filter
The combination of inductor L3 and capacitors C2 and C4 creates a pi-filter that helps to
filter out high-frequency currents caused by converter operation. Although a single filter
stage is often not sufficient to reach the limits defined by the legal regulations, it helps to
achieve the requirements. The cut-off frequency of this filter must be at least a factor of
two below the converter switching frequency.
1
f cutoff = -----------------------------------2   L3  C2
(6)
Capacitor C4 merely acts as a high-frequency filter capacitor to suppress the
high-frequency currents to the converter. Inductor L3 and capacitor C2 act as a low-pass
filter towards the mains.
Remark: To dissipate the high-frequency energy and block unwanted oscillations, use a
low-frequency, absorbent soft ferrite material, such as 3S1 (Ferroxcube) or 3W1200
(Würth Elektronik) for this inductor. For more information, see application note AN11532,
“SSL50x1 4 W to 25 W high-efficiency LED driver” (Ref. 7).
4.2.4 Inrush current and surge protection
4.2.4.1 Surge current protection
To prevent that voltage spikes on the mains damage the SSL523XT controller, some
surge protection is required. To reduce voltage spikes, the input series resistors R1 and
R2 (see Figure 1) or an active damper for higher LED power (PLED  10 W) are required.
Figure 2 shows an example of an active damper circuit. For more information, see the
AN11533 application note (Ref. 8).
AN11618
Application note
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Rev. 2 — 31 March 2016
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AN11618
Silergy Corp.
SSL523XT buck-boost controller
Vbus
820 kΩ
680 Ω
active
damper
820 kΩ
latch
rectified
mains
100 kΩ
10 nF
TSM1N60
150 nF
10 Ω
200 Ω
aaa-016840
Fig 2.
Active damper circuit for dimmable LED-drivers exceeding 10 W
When the TRIAC of the dimmer is activated and just before the zero crossings, the current
(if present) flows through R2, causing a relatively high dissipation during a short period.
When the rectified mains voltage is high enough, capacitor C5 is charged and MOSFET
M1 starts conducting. The current flows through M1 and resistor R7. Only a small
remaining part is still flowing though R2. The result is that the voltage across the damper
is relatively low when the damper is active. So, during almost the complete conduction
time of the TRIAC of the dimmer, the dissipation is relatively low. The active damper limits
the dissipated power in the total damper circuit by switching from a high series resistor to
a low series resistor, when the current is high.
4.2.4.2 Inrush current limiting
Resistors R1 and R2 limit the inrush current. Use special pulse withstanding resistors like
the EMC2 types Welwyn Components). These resistors also have a specified reliable
fusing capability. In case of an overload current, the EMC22 resistors fuse safely without
burning or emitting incandescent particles. After fusing, the resistance is at least 100
times greater than the original nominal value.
Figure 3 shows the maximum voltage across the resistor for an IEC61000-4-5 1.2/50 s
pulse (See application note AN11532, “SSL50x1 4 W to 25 W high-efficiency LED driver”
(Ref. 7)).
AN11618
Application note
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AN11618
Silergy Corp.
SSL523XT buck-boost controller
aaa-015695
700
Vpeak
(V)
600
500
400
300
200
100
0
0
10
20
30
40
50
60
70
Resistance (Ω)
IEC61000-4-5 1.2/50 s pulse performance Welwyn Components EMC2 resistors
Fig 3.
4.2.5 Low-THD application
Figure 4 shows the basic low-THD application diagram. The buffer capacitors can only
absorb a very limited part of the pulse energy. To prevent overvoltage on the drain of
switching MOSFET Q1 or on the DRAIN pin (see Figure 1), a Metal Oxide Varistor (MOV;
RV1) is required.
L3
VBUS
3.3 mH
230 V (AC)
10 W
PF > 0.9
BD1
MD5S
F1
X1
MKDS
1.5/2
1
+
4
3
R9
750 kΩ
R7
180 kΩ
R10
750 kΩ
R8
180 kΩ
MST1A250V
U1
TP
5
6
DRAIN
8
7
SSL5231T
9
R12
910 kΩ
C2
47 nF
400 V
COMP
4
R11
22 kΩ
C6
2.2 μF
6.3 V
R16
150 kΩ
R15
5.6 kΩ
C7
47 nF
50 V
3
2
C5
2.2 μF
50 V
1
GATE
C4
47 nF
400 V
ISNS
R1
330 Ω
VCC
R2
330 Ω
DIM
2
DEMOVP
RV1
VDRS07H275BSE
GND
2
1
VLED
90 V
36 mA
D3
ES1J
K2
LED
CONN.
R13
8.2 Ω
1
R17
160 kΩ
L4
4.7 mH
GNDL
R19
150 kΩ
C9
220 μF
100 V
2
aaa-016415
(1) The dots of inductors L3 and L4 indicate the start of the winding.
(2) n.m. = not mounted.
Fig 4.
SSL5236TE basic low-THD application diagram
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AN11618
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SSL523XT buck-boost controller
The MOV is a variable resistor, comparable with a bidirectional Zener diode with a series
resistor. As a result, the clamping voltage increases when the clamping current increases.
For a 275 V (RMS) MOV, the specification gives the following values:
• 473 V at 1 mA
• 710 V at 10 A (8/20 s pulse)
Remark: To limit excessive increase of the clamping voltage, it is important to limit the
value of the surge current through the MOV.
The most straight forward method is using pulse withstanding series resistors (see R1 and
R2 in Figure 4).
Example:
• 1.2/50 s pulse = 325 V + 500 V = 825 V total mains voltage including surge
• 550 V actual MOV clamping voltage at the peak of the surge
• 275 V resulting voltage across resistors R1 and R2
From Figure 3 we can conclude that a 10  2 W EMC2 is operated within specification
with a 10 % margin.
Although a MOV is used to clip surge voltage, the current through the mains filter inductor
L3 shows a damped oscillation caused by the mains input voltage step. The result is an
oscillating voltage on capacitor C4, with a peak voltage that is slightly higher than the
voltage on C2 in Figure 5.
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Application note
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SSL523XT buck-boost controller
(1) The filter inductor 'sees' a voltage step from 325 V to 530 V on the input. As a result the inductor
boosts up Vbus to 630 V. The first peak of Vrect equals the 275 V (AC) MOV clamping voltage.
(2) This part is zoomed in and displayed in the lower image.
(3) Saturating mains filter inductors.
C1 (yellow): Voltage on capacitor C2 = 100 V/div
C2 (red): Voltage on capacitor C4 = 100 V/div
C3 (blue): L3 EMI inductor current = 1 A/div
C4 (green): mains input surge voltage = 200 V/div
Fig 5.
AN11618
Application note
EMI inductor voltage during surge
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SSL523XT buck-boost controller
To reduce the oscillating voltage across inductor L3 and the corresponding increased
voltage on capacitors C4 and C2, a PESD diode can be connected in parallel to inductor
L3. Figure 6 shows the measuring results.
Fig 6.
EMI inductor voltage during surge with PESD in parallel with L3
The large voltage step from 325 V to 530 V is still present across the filter inductor, but the
boosted energy is now limited by the PESD diode. The peak voltage across capacitor C4
is now almost equal to the 275 V (AC) MOV clamping voltage (520 V).
If a larger inductor value is used for inductor L3, increasing the voltage of the PESD diode
to 24 V or 30 V may be required. Otherwise, the diode can start conducting during normal
operation (see application note AN11532, “SSL50x1 4 W to 25 W high-efficiency LED
driver” (Ref. 7)).
4.2.6 Dimmer compliance requirements of the input stage
The input stage must support most existing wall dimmers. To fulfill the TRIAC dimmer
latching requirements, the input stage includes a latching circuit consisting of resistors R5
and R6 and capacitor C3 (see Figure 7).
See for more information about the basics of the latching circuit and a detailed design
procedure the AN11533 application note (Ref. 8).
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SSL523XT buck-boost controller
L3
VBUS
2.2 mH
BD1
MD5S
L1
R1
X1
MKDS
1.5/2
10 Ω
fusistor
1
3
3.3 mH
R3
n.m.
CX1
RV1
10 nF VDRS07H275BSE
400 V
4.7 kΩ
2
R5
270 Ω
2W
1
+
4
2
C2
10 nF
400 V
C3
150 nF
400 V
C4
22 nF
400 V
L2
R2
10 Ω
fusistor
R6
270 Ω
2W
3.3 mH
R4
4.7 kΩ
GNDL
Fig 7.
aaa-016372
Input stage equipped with latching circuit
In practice, some additional filtering with inductors L1 and L2 (and capacitor CX1) is
required (see Figure 7). To dampen parasitic oscillations, resistors R3 and R4 are added.
Resistors R1 and R2 (fuse resistors) act as a fuse in case of overcurrent. So a separate
fuse is omitted.
4.3 Buck-boost converter inductor dimensioning
Ideally, a direct relation exists between the total stroke time and the converter frequency.
The inductor value can be derived easily when the converter frequency is chosen:
•
1
f sw = ------t sw
•
t sw = t on + t off
V
L4
• I peak = t on  -------in• V bus  max  = 2  V mains
Section 4.3.1 presents a comparison between a calculation and the corresponding
measuring results.
4.3.1 Low-THD configuration - inductor calculations
To ensure a low-THD figure, the mains input current of the converter must be almost
sinusoidal and free of higher harmonics. To ensure a high power factor, the mains input
current must also be in phase with the mains voltage. As a result, to limit recharging pulse
currents, the capacitance value of buffer capacitors C2 and C4 must be relatively small.
The energy absorption of the converter is regarded as a sinusoidal modulated power sink
(fripple = 2 fmains).
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SSL523XT buck-boost controller
The LEDs mounted in series determine the output voltage. The output voltage is
considered constant over a mains cycle. As a result, the converter current to the output
capacitor is modulated with a frequency of twice the mains frequency. The high-frequency
filtered converter current varies between 0 A and twice the nominal LED current
(see Figure 8).
I converter  max  = 2  I LED  AV 
(7)
HF filtered output current
to output cap. + LEDs
rectified
filtered
mains
current
lconv_max
lLED_av
CONVERTER
time
aaa-016373
Fig 8.
Converter filtered output current as a function of mains phase
To set the average LED output current to the specified value, the on-time of the switch is
regulated very slowly (time constant > 20 ms). So the on-time is kept almost constant over
a mains cycle.
The triangular wave shape and the switching times, determine the maximum peak
converter current (see Figure 9):
• ton: The on-time of the MOSFET
• toff: The time the secondary diode is conducting
• tring: The ringing time, required for optimal valley switching
lpeak(max)converter
lconverter(max)
ton
toff
tring
ton
aaa-016375
Fig 9.
Converter wave form; Diode D3 current
t on + t off + t ring
I peak  max converter = 2  2   -------------------------------------  I LED  AV 

t
(8)
 t on + t off + t ring   f sw = 1
(9)
off
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SSL523XT buck-boost controller
Combining Equation 8 and Equation 9 gives:
I LED  AV 
I peak  max converter = 4  -------------------------- f sw  t off 
(10)
When a relatively large buffer capacitor is used, the bus voltage Vbus is constant during
one half-frequency cycle. When the buffer capacitor has a relatively low value, the buffer
voltage shows a low-frequency voltage modulation. The resulting on-time is adapted to
deliver the correct average output current.
The calculated converter current must be considered at low mains and high mains. The
correction factor represents the lower bus voltage at the moment of the peak inductor
current:
V bus  min 
2  -------------------------------------------------------- V bus  max  + V bus  min  
(11)
Where:
• Vbus(max) is the maximum bus voltage at the peak of the high mains voltage
• Vbus(min) is the minimum bus voltage at the peak of the low mains voltage
Combining Equation 10 and Equation 11 gives:
V bus  min 
I LED  AV 
I peak  max converter = 8  ---------------------------  -------------------------------------------------------- f sw  t off   V bus  max  + V bus  min  
(12)
The calculated Ipeak(max)converter value represents the maximum peak current value,
required for the inductor design.
At the peak of the mains, the relationship between ton and toff can be calculated with
Equation 13.
2  V mains  nom   t on = V LED  t off
(13)
Where:
V
+V

bus  max 
bus  min 
• V mains  nom  = ---------------------------------------------------------
2  2
1  V
--bus  max  + V bus  min    t on = V LED  t off
2
(14)
Equation 9 can be rewritten:
1
1
t on + t off + t ring  = t sw = ---------  t on = --------- – t off – t rin
f sw
f sw
(15)
The result of Equation 15 can be substituted into Equation 14. The result (Equation 16):
V bus  max  + V bus  min 
t off =  --------------------------------------------------------------------------------------   t sw – t ring 
V
+
V
+
2

V

bus  max 
bus  min 
LED
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Substituting this outcome in Equation 12 results in:
8  I LED  AV   t sw  V bus  min    V bus  max  + V bus  min  + 2  V LED 
I peak  max converter = ----------------------------------------------------------------------------------------------------------------------------------------------------------------------- (17)
2
  V bus  max  + V bus  min     t sw – t ring  
By default a 1 s tring can be chosen for converters with the SSL523XT IC.
For low-power buck-boost converters (5 W to 10 W) however, the following factor based
on the outcome of some ring time measurements gives a better estimate.
6.8
t ring  s  = ---------------------------------------------- I LED  AV    mA 
(18)
Substituting Equation 18 in Equation 17 results in an improved estimate:
8  I LED  AV   t sw  V bus  min    V bus  max  + V bus  min  + 2  V LED 
I peak  max converter = ----------------------------------------------------------------------------------------------------------------------------------------------------------------------- (19)
2
6.8
V

+ V bus  min     t sw – ------------------------------bus

max



 I LED  AV   
di
Based on the general formula U = L  ----- , the inductance of inductor L can be calculated
dt
with Equation 20.
V LED  t off
L = -------------------------------------------I peak  max converter
(20)
Combining this formula with Equation 16 and Equation 19 results in an expression for
calculating the buck-boost inductance value.
V bus  max  + V bus  min 
 t sw – t ring 
L = V LED   --------------------------------------------------------------------------------------  ------------------------------------------------V
 I peak  max converter 
bus  max  + V bus  min  + 2  V LED 
6.8
V
+ V bus  min     t sw – ------------------------------- 
 bus  max 

 I LED  AV   
L = V LED  ---------------------------------------------------------------------------------------------------------------------------------------------------  V bus  max  + V bus  min  + 2  V LED    I peak  max converter  
3
6.8
  t – ------------------------------ 2  f  V
sw
bus  max  + V bus  min   
  sw

 I LED  AV  
L = V LED  ----------------------------------------------------------------------------------------------------------------------------------------------------------------2
  V bus  max  + V bus  min  + 2  V LED   8  I LED  AV   V bus  min  
(21)
Equation 12 gives the maximum current the buck-boost inductor must handle. Table 1
gives the results.
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Table 1.
Measured and calculated inductor current values
At Ipeak(L4)
At Vmains(peak)
Measured
Calculated
Measured
Calculated
L (mH) at 10.6 W
1.5
1.44
1.5
1.45
I (mA) at 10.6 W
620
626
620
624
L (mH) at 4 W
4.7
4.36
4.7
4.56
I (mA) at 4 W
239
215
216
210
4.3.1.1 High-frequency ripple voltage on the buffer capacitor Vbus(min) and Vbus(max)
calculation
By default the buffer capacitance for the low-THD application must be relatively small.
As a result a high-frequency ripple voltage is noticeable. It influences the set point of the
converter.
Buffer capacitor C4 is charged with a current through inductor L3.
During the on-time, energy is drawn from buffer capacitor C4 (partly via L3). It is
transferred to the LED output via the buck-boost inductor L4.
The peak mains voltage at 230 V (RMS) is about 325 V.
Due to the RMS losses of the input stage, the voltage slightly decreases.
The total impedance of the input stage (series resistors and parasitic resistance of filter
inductors; see Figure 13) is approximately 33 .:
• The average mains current at the mains peak voltage for the measured 10 W
Low-THD system is about 62 mA. The result is a total resistive voltage drop Vresistive
of approximately 2 V.
• The voltage drop (Vbridge) across the diode bridge of about 2 V must also be
considered.
• Additionally, the other filtering capacitors of the input stage Vfilter cause the peak
mains voltage to decrease about 2 V. The reason for the decrease is the voltage drop
across the parasitic series resistance of the buffer capacitor.
At the peak of the mains, the average voltage of capacitor C4 (VC4(AV)) is:
V bus  AV  =
2  V mains – V res – V bridge – V filter = 319 V
(22)
For low-watt systems (about 4 W), a resistive high-loss input smoothing stage can be
used. It can result in a voltage decrease up to 50 V).
The minimum and maximum voltages can be calculated with Equation 23:
2 1
2
1
2  P LED  AV  = f sw   ---  C4  V bus  max  – ---  C4  V bus  min  
2

2
(23)
At the peak of the mains, the power delivered by the converter for a low-THD system is
twice the average LED power.
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The average buffer voltage VC4(AV) is known. The minimum and maximum voltages can
be calculated by substituting the known values in Equation 24.
2 1
2
1
P LED  AV  = f sw   ---  C4  V bus  max  – ---  C4  V bus  min  

2
2
(24)
2 1
2
1
P LED  AV  = f sw   ---  C4  V bus  AV  – ---  C4  V bus  min  
2

2
 V bus  min  = 297 V and V bus  max  = 340
Due to voltage overshoot (switching frequency-related ringing of the EMI filter), the
Vbus(max) voltage can be higher than the peak rectified mains voltage.
Table 2 shows the results.
Table 2.
Measured and calculated buffer voltages at 230 V mains voltage
At Vmains(peak)
Measured
Calculated
Vbus(min) at 10.6 W
299
297
Vbus(max) at 10.6 W
339
340
Vbus(min) at 4 W
304
295
Vbus(max) at 4 W
304
302
4.3.2 Summary of formulas for inductor and inductor current
The inductor calculation and the corresponding peak inductor current at the top of the
mains voltage can be calculated with Equation 25 and Equation 26.
2
3
6.8
V LED   t per – -------------------------------  f sw   V bus  max  + V bus  min  


 I LED  AV  
L4 = -----------------------------------------------------------------------------------------------------------------------------------------------------------2
 V bus  max  + V bus  min  + 2  V LED   8  I LED  AV   V bus  min 
peak  max converter
(25)
8  I LED  AV   t per  V bus  min    V bus  max  + V bus  min  + 2  V LED
= ------------------------------------------------------------------------------------------------------------------------------------------------------------------------(26)
2 
6.8

 V bus  max  + V bus  min    t per – ------------------------------
 I LED  AV  
Vbus(AV) can be calculated with Equation 27:
V bus  AV  =
2  V mains – V res – V bridge – V filter
(27)
Where:
P
V mains
mains
• V resistive = 2   R1 + R2 + R L1, L2, L3   ---------------
• Vbridge = 2 V (typical)
• Vfilter = 2 V (typical; additional losses)
Based on Equation 24 the Vbus(max) and Vbus(min) can be calculated with Equation 28 and
Equation 29:
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V bus  max  =
P LED  AV  
2 
V
+ 2  ---------------------- bus  AV  
f  C4  
(28)
V bus  min  =
P LED  AV  
2 
V
– 2  ---------------------- bus  AV  
f  C4  
(29)
sw
sw
4.4 Dimming control
When the phase-cut mains voltage is measured via DIM pin, DIM voltage modulates the
internal reference voltage. For the average output current control, the VI(ISNS) voltage is
measured during the secondary stroke, when diode D3 is conducting. The dimmed output
current supplied to the LED (IO(dim)) can be calculated with Equation 30:
V dim  itg  AV  ISNS 
I O  dim  = -----------------------------------------R13//R14
(30)
Remark: In general, Vdim is not derived from Vrect but from Vbus. Deriving Vdim from Vbus
results in a reduced dimming ratio due to the remaining bus capacitor charge at the mains
zero crossings. This effect must be considered during the design phase.
Vdim(itg)AV(ISNS)
(mV)
310
30
0.23
0.65
2.1 DIM (V)
aaa-014939
Fig 10. Dimming control transfer function
For the minimum specified mains voltage during undimmed conditions, the voltage at the
DIM pin (pin 8 of the IC) must have a voltage of at least 2 V + 10 % margin, 2.2 V
(see Figure 10). A value of 27 k is chosen for R11. The resulting value for R9 + R10 can
now be calculated with Equation 31:
   2---------2-  U


mains  min  – V DIM  1.1
  
 R9 + R10  = R11  ----------------------------------------------------------------------------------------------- V DIM  1.1 
(31)
To ensure stability and limit flutter during dimmed operation, resistor R12 limits the
dimming range. Resistor R12 is connected to the VCC and injects a small preset current of
about 11 A into resistor R11.
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Without compensation, the value for R9 + R10 is approximately 2.2 M for a minimum
mains voltage of 207 V. Taking the 11 A preset current into account, choose two
resistors in series of 1.3 M each.
For capacitor C6, choose a value of about 2.2 F. This value results in a time constant of
at least 50 ms.
4.5 Valley detection
When IL has decreased to zero at t3, the inductor voltage starts to oscillate around the 0 V
level with amplitude Vout and frequency (fring) (see Figure 11). Valley detection is
integrated in the SSL523XT. It senses when the inductor voltage reaches its lowest level
(valley) via the DEMOVP pin connection. For a 60 V LED application, the DEMOVP pin is
typically connected to resistors R15 (5.6 k), R16 (150 k), and R17 (160 k). See
Section 4.9.5 for calculation of these resistor values. If the valley is detected, the internal
source switch is switched on again. As a result, the switch-on switching losses are
minimal.
VSW
VOUT
0
VLEDP
valley
VIN
magnetization
demagnetization
IL
0
1
t0
2
t1
3
t2
4
t3
t00
T
aaa-014940
Fig 11. Buck-boost waveforms and valley detection
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4.6 Start-up current
The source switching topology drives a high-voltage power MOSFET.
The gate is connected to the VCC. The charge and discharge towards the gate of the
MOSFET balance each other. So there is no impact on the average required VCC current
during operation. To reduce the rising edge switching slope steepness of the source of the
MOSFET of the SSL5231T application, resistor R18 and diode D1 are inserted. The rising
edge switching slope influences the EMI spectrum. Capacitor C5 carries the positive and
negative pulse gate currents of the external MOSFET (see Figure 4).
Resistors R7 and R8 provide the supply current to the IC. The IC draws additional start-up
current (ICC(startup)) just before VCC reaches the start-up voltage level (Vstartup). The
consumed supply current in operation is therefore lower than during start-up conditions.
The hysteresis in supply current prevents that VCC becomes too low resulting in lamp
flicker, when operating the lamp close to the minimum operating voltage. The generated
VCC supply current varies when the mains voltage is increased or decreased. The lower
supply current prevents lamp flicker when the mains voltage is increased or decreased
slowly. Figure 12 shows the ICC input current waveform at start-up. When the converter is
operating in normal operation, the additional current sink Icc(startup) is turned off.
Vstartup
VVCC
VCC
lCC(startup)
lCC(oper)
ICC
VGATE
aaa-014941
Fig 12. Start-up current waveform
Resistors R7 and R8 must source the total current before the Vstartup threshold of the
converter is exceeded.
The total resistance of resistors R7 and R8 can be calculated with Equation 32:
2  V mains – V startup
R7 + R8  -------------------------------------------------------I CC  oper  + I CC  startup 
(32)
Where:
• Vmains  70 % or 80 % of Vmains(typ)
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A suitable value for R7 + R8 is 400 k for high mains applications. Taking the voltage
rating into account, two 1206 SMD resistors of 180 k in series, capable to withstand
200 V each, can be used. In combination with a 2.2 F VCC capacitor (C5), the start-up
time is < 0.2 s. Especially during dimming, the chosen values of VCC capacitor C5
(2.2 F) and resistors R7 and R8 are crucial for stable operation.
4.7 Leading-Edge Blanking (LEB)
To prevent false detection of overcurrent, a leading-edge blanking time following
switch-on is implemented. When the internal source-switch turns on, a short current spike
can occur because of the capacitive discharge of voltage over the drain and the source. It
is disregarded during the LEB time (tleb).
4.8 Reduction of switching losses at low mains voltages
Close to the mains-voltage zero crossings, the bus voltage is low. The system cannot
deliver much energy to the LEDs. The switching losses become dominant. To limit the
losses and improve the efficiency, the switching frequency is limited to 25 kHz (typical) in
the IC when the current sense level VI(min)ISNS is not met. The control is set to maximum
off-time (toff(max)) switching.
4.9 Protections
The following protections are implemented in the IC:
•
•
•
•
•
•
UnderVoltage LockOut (UVLO)
Cycle-by-cycle OverCurrent Protection (OCP)
Internal OverTemperature Protection (OTP)
Cycle-by-cycle maximum on-time protection
Output OverVoltage Protection (OVP)
Output Short Protection (OSP)
All protections are non-latched. They lead to a safe restart of the converter.
4.9.1 UnderVoltage LockOut (UVLO)
When the voltage on the VCC pin drops to below Vth(UVLO) ( 10 V), the IC stops
switching. An attempt is made to restart IC when the VCC > Vstartup ( 15 V).
4.9.2 Cycle-by-cycle OverCurrent Protection (OCP)
The SSL5231T contains a built-in peak current detector. It triggers when the voltage at the
ISNS pin reaches the peak level VI(max)ISNS. A resistor connected to the ISNS pin senses
the current through the inductor IL. The maximum current in inductor IL(max) can be
calculated with Equation 33:
V I  max  ISNS
I L  max  = ---------------------------R13//R14
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The sense circuit is activated after the LEB time (tleb). It automatically provides protection
for the maximum LED current during operation because the LED current is half the peak
current by design. The propagation delay ( 100 ns) exists between overcurrent detection
and the actual source-switch switch-off. Due to this delay, the actual peak current is
slightly higher than the OCP level set by the resistor in series with the ISNS pin.
4.9.3 OverTemperature Protection (OTP)
When the internal OTP function is triggered at Tpl(IC) = 150 °C (typical), the converter
stops switching. The IC resumes switching when the IC temperature drops to below
Tpl(IC)rst (= 118 °C (typical)).
4.9.4 Cycle-by-cycle maximum on-time protection
Measuring the inductor current IL using the voltage across sense resistor Rsense regulates
the on-time. The maximum on-time is limited to a fixed value ton(max) (= 15.5 s (typical)).
It protects the system and the IC when the ISNS pin is shorted or the system works at very
low mains (brownout protection).
4.9.5 Output OverVoltage Protection (OVP)
Measuring the voltage at the DEMOVP pin during the secondary stroke gives an accurate
output OVP. The DEMOVP triggering level is 1.81 V (typical). The resistive divider
connected to the LEDP node, the DEMOVP pin, and the GND pin. The DEMOVP pin sets
the maximum LED voltage (VLED(max)), which can be calculated with Equation 34. A value
of 5.6 k is chosen for resistor R15.
V LED  max 
V th  ovp 
------------------------------------------ = -----------------R15 + R16 + R17
R15
(34)
An internal counter prevents false OVP detection because noise can be present on the
DEMOVP pin. After three continuous cycles with a DEMOVP pin voltage exceeding the
OVP level, OVP is triggered.
When OVP is triggered, a restart sequence begins. A discharge current (ICC(dch)  3.7 mA)
is enabled and discharges VCC to below Vrst(latch) (= 6.2 V (typical). When Vrst(latch) is
reached, the IC is reset and starts up again.
4.9.5.1 Attention point for the output OVP protection
For very low inductance values of the buck boost inductor, the secondary stroke time can
become shorter than the transformer ringing suppression time tsup(xfmr_ring) during open
output condition. As a result, the output OVP detection does not operate. The output
overvoltage voltage is not protected adequately anymore (at peak of mains). To prevent
malfunction of the output OVP, the switching frequency must be reduced to < 60 kHz by
increasing the buck-boost inductor value.
If reducing the switching frequency is not possible or desired, a small Zener diode can be
connected across the LED output voltage terminals (V LED  max   V Zener  V o  cap ).
The output capacitor value can also impact the correct operation of the output OVP
protection. When the output capacitance is low, the output capacitor voltage increases
during the secondary stroke. The result is that the secondary stroke time is reduced. If the
secondary stroke time reduces below the tsup(xfmr_ring) time, the output OVP protection fails
to operate.
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4.9.6 Output Short Protection (OSP)
The converter operates in Discontinuous Conduction Mode (DCM) and Boundary
Conduction Mode (BCM). A new cycle is only started after the inductor current has
become zero. Demagnetization is detected through measuring the voltage on the
DEMOVP pin. When the DEMOVP pin voltage drops to below the demagnetization level
(Vdet(demag)) of 18 mV (typical)) and a valley is detected, a new cycle starts. During the
output short, the converter regulates the adjusted output current and the on-time is
reduced to a safe value by this feedback. The reduced on-time in combination with a very
long demagnetization time prevents any damage or excessive dissipation of the
converter.
To prevent false demagnetization detection, a blanking time (tsup(xfmr_ring); 1.5 s (typical))
is implemented at the start of the secondary stroke.
4.10 Supply management
The IC starts up when the voltage at the VCC pin exceeds Vstartup. The IC locks out (stops
switching) when the voltage at the VCC pin drops to below Vth(UVLO). The hysteresis
between the start and stop levels allows the VCC capacitor to supply the IC during
zero-crossings of the mains voltage.
The SSL523XT incorporates an internal band gap referenced clamping circuit on the VCC
pin. The clamp limits the voltage on the VCC supply pin to the maximum value
Vclamp(VCC) = 15.8 V. For a proper autorestart, the maximum current of the external
resistor must be lower than the limiting value ICC(dch) (= 3.7 mA). For a high-mains
application without external VCC Zener diode, the resistance of the external resistor
R7 + R8 must be  75 k. During switch-off of the power MOSFET, a significant current is
supplied via the MOSFET through the SW-pin to the VCC capacitor. The voltage across
the VCC capacitor may not increase to a value exceeding the absolute maximum ratings
of the IC.
See Section 10 for more information on the supply current generation tips and tricks.
4.11 The external power MOSFET for the SSL5231T application
In a SSL5231T buck-boost application, the external power MOSFET Q1 must handle a
voltage of at least the sum of the mains voltage and the output voltage. In practice, the
power MOSFET is a 600VDS type. The peak current handling capability is about 1 A for
10 W applications (Vmains = 230 V; VLED = 80 V). So a SPU2N60C3 type of MOSFET or
equivalent can be chosen to limit the on-state losses. A smaller MOSFET dissipates more
because of the increased on-state losses.
During the switch-off process of the external high-voltage power MOSFET, the parasitic
drain-source capacitance Cds is charged. The charge current flows via the drain
connection through Cds to the SW-pin of SSL523XT. During the off-time, a diode internally
connects this pin to the VCC pin. So the Cds_par charge current charges the VCC capacitor
C5.
Depending on the parasitic capacitances of the external power MOSFET, oscillations may
occur. An additional capacitor of about 1 nF across the gate-source terminals of the power
MOSFET prevents these oscillations. A higher capacitor value is possible, but the losses
increase because of the capacitor charge and discharge currents.
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The gate is kept at a constant DC voltage using series resistor R18. Resistor R18 is
connected to the VCC pin (pin 6) of the controller IC.
For an increased switch-off performance, a speed-up diode D1 can be applied
(see Figure 13). To limit the amount of switching EMI, avoid that the application switches
off too quickly. Preventing that the application switches off too quickly can be achieved by
connecting a resistor in series with diode D1.
Especially during deep dimming operation, the additional charging of the VCC capacitor
(C5) by the parasitic capacitor Cds_par is essential for a stable operation. The generated
VCC supply current largely depends on the total (parasitic) capacitance between the
DRAIN and SOURCE pins of power MOSFET Q1.
For a smaller MOSFET or a MOSFET with lower drain-to-source capacitance, an
additional 30 pF capacitor in parallel to the DRAIN-SOURCE connection of power
MOSFET Q1 is required to ensure stable VCC supply.
If ICC is too low, the UVLO causes the IC to switch off. Immediately after the VCC
capacitor is recharged to the VCC restart level, the IC starts operating again. The switch
on/off phenomenon causes an instable light output of the connected LEDs.
See Section 10 for more options about supply current generation. Figure 13 shows the
schematic.
L3
Vbus
2.2 mH
n.m.
220 Ω
shorted
R4
C4
22 nF
400 V
RV1
VDRS07H275BSE
4.7 kΩ
R10
1.3 MΩ
R8
180 kΩ
U1
1
2
SW
R2
VLED
60 V
140 mA
n.c
2
3
C3
150 nF
400 V
8
R12
910 kΩ
C2
10 nF
400 V
7
6
R22
51 k
5
R11
27 kΩ
R6
270 Ω
2W
C6
2.2 μF
6.3 V
R16
150 kΩ
R15
5.6 kΩ
C7
47 nF
50 V
R17
160 kΩ
C5
2.2 μF R13
50 V 3.9 Ω
L4
1.5 mH
R23
2.7 kΩ
2W
1
Q3B
FMBM5551
R14
5.6 Ω
C9
220 μF
100 V
n.m.
R19
150 kΩ
n.m.
C10
220 μF
100 V
R26
5.6 Ω
R25
R27
5.6 Ω
LEDM
secondary side bleeder
GNDL
2
R24
2.7 kΩ
2W
Q3A
FMBM5551
D3
ES1J
n.m.
C8
22 pF
50 V
K2
LED
connector
LEDP
4
SSL5231T
ISNS
n.m.
CX1
10 nF
400 V
Q1
SPU02N60C3
R18
R3
4.7 kΩ
L2
3.3 mH
10 Ω
fusistor
R7
180 kΩ
GND
2
R9
1.3 MΩ
BAS316
DEMOVP
10 Ω
fusistor
4
3
R5
270 Ω
2W
DIM
1
1
+
VCC
BD1
MD5S
R1
X1
MKDS
1.5/2
D1
n.m.
L1
3.3 mH
COMP
230 V (AC)
10 W
PF > 0.9
n.m.
D4
1N4148W
aaa-016370
The dots of inductors L3 and L4 indicate the start of the winding.
n.m. = not mounted.
Fig 13. Complete application diagram
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4.12 Output bleeder
Using the output bleeder principle can improve the dimming ratio of the IC. The output
bleeder circuit must be designed in alignment with the nominal LED voltage and current. A
correctly designed output bleeder circuit delivers the nominal output current with very
limited losses to the LED. At deep dimmed condition, the output bleeder bleeds almost all
the current, resulting in a reduced LED current. The dimming ratio is increased. The
additional bleeding current ensures dimming compatibility. Figure 14 shows the
schematic.
VLED = 60 V at 140 mA
LEDP
R22
R23
2.7 kΩ
2W
R24
2.7 kΩ
2W
K2
LED
CONN.
2
1
Q3A
FMBM5551
Q3B
FMBM5551
R26
5.6 Ω
R25
R27
5.6 Ω
n.m.
D4
1N4148W
LEDM
aaa-016416
n.m. = not mounted
Fig 14. Output bleeder schematic
Transistor Q3B acts as a reference voltage for the base of transistor Q3A Vref(Q3A). The
LED current (ILED) is directed through resistor R27, resulting in a voltage drop proportional
to the actual LED current. At low LED currents, the voltage drop across resistor R27 is
relatively low. So the BE junction of transistor Q3A is forward biased and starts
conducting.
• If the bias current through R25 is neglected, the current IR26 represents the bleeder
current Ibleeder.
• The current through resistor R27 represents the sum of the LED current and the
bleeder current.
The equilibrium condition can be calculated with Equation 35:
V ref  Q3  = V BE  Q3 A  + I bleeder  R26 +  I LED + I bleeder   R27
(35)
The bleeder current can be calculated with Equation 36:
 – V ref  Q3  – V BE  Q3 A  – I LED  R27 
I bleeder = --------------------------------------------------------------------------------------------- R26 + R27 
(36)
For a 60 VLED/10 W system, the secondary bleeder preferably starts bleeding for LED
currents below 80 mA (see Figure 15 to Figure 17).
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aaa-016417
100
ILED
(mA)
80
60
40
20
0
0
0.2
0.4
0.6
0.8
time (s)
1
Fig 15. Output side bleeder: 10 W; LED current as function of linear increasing converter
current
aaa-016418
10
Ibleeder
(mA)
8
6
4
2
0
0
0.2
0.4
0.6
0.8
time (s)
1
Fig 16. Output side bleeder: 10 W; bleeder current as function of linear increasing
converter current
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aaa-016419
600
Pbleeder
(mW)
500
400
300
200
100
0
0
0.2
0.4
0.6
0.8
1
time (s)
1.2
Fig 17. Output side bleeder: 10 W; bleeder dissipation as function of linear increasing
converter current
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5. Design examples with the buck-boost SSL523XT driver IC
In this section, the buck-boost IC applications are shown. The SSL523XT can also
operate in non-isolated flyback applications. However, these circuits are outside the scope
of this document.
The SSL5231T IC is a source-driven controller IC with an external MOSFET. So the
source current of approximately 800 mA (peak) limits the output power. It is targeted for
designs  10 W.
The SSL5235TE, SSL5236TE, SSL5237TE ICs are source-driven driver ICs with an
integrated MOSFET. So the source current of approximately 800 mA (peak) limits the
output power. The SSL5235TE, SSL5236TE, SSL5237TE IC versions are targeted for
designs up to 12 W, 7 W, and 10 W respectively.
Table 3.
Buck-boost applications
SSL5231DB1246
SSL5236DB1249
SSL5237DB1247
230 V; 10 W; A19; eco-LTHD
230 V; 4 W; dual; candle;
eco-THD
120 V; 7 W; PF > 0.9; LTHD
form factor:
A19
candle
candle
user manual:
UM10861 (Ref. 11)
UM10847 (Ref. 10)
UM10846 (Ref. 9
demo board
specifications:
demo board
photographs
top view:
bottom view
IC type:
SSL5231T
SSL5236TE
SSL5237TE
mains voltage:
230 V
230 V
120 V
input power:
10 W
4W
8W
power factor:
0.9
0.9
0.95
THD:
-
-
18 %
LED voltage:
60 V
36 V
54 V
Efficiency:
84 %
76 %
83 %
output ripple:
27 %
25 %
-
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6. PCB layout constraints DEMOVP pin
The DEMOVP pin is a sensitive pin. To ensure that OVP is activated at the correct trigger
voltage, a proper layout of the part of the printed-circuit board connected to the DEMOVP
is essential.
• Limit the value of the resistor between the DEMOVP pin and IC ground to a value
 5.6 k.
• Shield the DEMOVP track and make the corresponding DEMOVP loop area of as
small as possible.
The track connected to the DEMOVP pin must be kept short and must have a small loop
area (see Figure 18). The track between resistors R16 and R17 (if two resistors are
connected in series) must also be kept short and small.
L3
BD1 1
MD5S +
R7
180 kΩ
R10
750 kΩ
R8
180 kΩ
U1
2
R2
R1
330 Ω
330 Ω
TP
R12
910 kΩ
5
6
COMP
R11
22 kΩ
DRAIN
8
VLED
90 V
36 mA
IC
9
4
C2
47 nF
400 V
7
3
2
1
GATE
RV1
VDRS07H275BSE
C4
47 nF
400 V
DIM
2
1
GND
4
3
MST1A250V
DEMOVP
X1
MKDS
1.5/2
R9
750 kΩ
ISNS
F1
Vbus
3.3 mH
VCC
230 V (AC)
10 W
PF > 0.9
D3
ES1J
C6
2.2 μF
6.3 V
R16
150 kΩ
R15
5.6 kΩ
C7
47 nF
50 V
C5
2.2 μF
50 V
R13
8.2 Ω
R19
150 kΩ
K2
LED
connections
C9
220 μF
100 V
1
2
L4
4.7 mH
R17
160 kΩ
GNDL
aaa-016421
The dots of inductors L3 and L4 indicate the start of the winding.
n.m. = not mounted.
Fig 18. DEMOVP keep-out area
On the PCB, the DEMOVP track must be kept away from all parts and tracks that are
outside the area of the circle (see Figure 18).
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HSO8
TSOP6
GND
110 kΩ
110 kΩ
DEMOVP
5.6 kΩ
aaa-015704
Fig 19. IC_GND guard track shielding the DEMOVP pin
If it is not possible to achieve the above, then use a guard track around the DEMOVP pin
(see Figure 19) for the SO8 package. Do not close the guard track because it picks up
magnetic fields. The small stub traces under the resistor create additional parasitic filter
capacitance below the DEMOVP resistors. The parasitic filter capacitance helps to filter
disturbance at the DEMOVP pin.
To prevent that noise is picked up, capacitor C7, connected to the COMP pin of the IC,
must be routed as close as possible to the COMP and GND pins.
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7. ElectroMagnetic Interference
Shielded I-core inductors perform well but are not cost-effective. E-core inductors also
perform well but are not always 'off the shelf' and must be ordered.
Figure 20 shows the conducted EMI measurement for the 10 W SSL5231T demo board
mounted with a standard non-shielded I-core drum.
Fig 20. EMI measurements on the 10 W SSL5231T demo board: Non-shielded I-core drum
inductor
Unshielded I-core inductors can perform well, but this performance depends on the
inductor used. It is possible to end up with 6 dB to 20 dB more disturbance in the
conducted EMI measurement because of poor inductor performance. Some inductor
manufactures that produce coils that perform well with SSL523XT are Renco, Coilcraft,
TDK, and Würth Elektronik.
If I-core inductors are used, test the layout with a known inductor, which has proven to
fulfill the EMI requirements in previous applications.
To reduce coupling as much as possible, keep the distance between the inductor of the
EMI filter and the buck-boost inductor on the PCB long.
Additionally, the inductors can be mounted perpendicular to each other. The cores always
face each other at a 90 degree angle, X-axis, Y-axis, and Z-axis (see Figure 21; Z-axis is
not shown).
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EMI filter inductor
buck-boost inductor
aaa-015709
Fig 21. Suggested inductor mounting of two inductors close to each other
For optimal EMI performance the exact location and orientation is essential:
• The start of winding of buck-boost inductor L4 (see Figure 13; the wire closest to the
core) must be connected to the switching side of the converter, in this case IC ground.
The outer part of the winding acts as shield for the inner part of the winding, which is
operated at a high frequency high voltage. The inductor shields itself using the outer
part of the winding connected to the DC bus voltage. If an e-core is used for inductor
L4, connecting the end of the winding (so not the start of the winding) to the switching
side of the converter can be advantageous. The E-core has a better shielding
performance than the I-core. However, the core volume is much larger. The inner
winding also has a significant capacitive coupling to the inner leg of the relatively large
E-core. For the lowest overall EMI performance, the preferred connection of inductor
L4 must be checked using EMI measurements.
• The start of the winding of filter inductor L3 (see Figure 7; the wire closest to the core)
must be connected to the bridge rectifier. The outer part of the winding shields the
inner part of the winding, which is connected to the bridge rectifier.
• The start of the windings of filter inductors L1 and L2 (see Figure 13; the wire closest
to the core) must be connected to the input resistors R1 and R2. The outer part of the
windings of inductors L1 and L2 shield the inner part of the corresponding windings,
which are connected to the mains input via R1 and R2.
• If vertical mounting is chosen for all inductors, the winding direction of the inductors
must be equal to limit mutual interference of the inductors as much as possible.
Some manufactures use a dot marking on the inductor. Other manufacturers use a long
and a short wire to indicate the start of winding.
Würth Elektronik indicates the start of the winding in the following way:
• A dot marks the starting pin of the winding
or
• The shortest wire marks the starting pin of the winding.
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8. Appendix 1: Tips and tricks: Floating controller signal measurement
To prevent a short circuit, an isolated electronic power source or a variac must power the
Device Under Test (DUT).
For the SSL523x, SSL502x, and the SSL503x series ICs in buck and buck-boost
configurations, the IC ground is floating. The IC-ground connection is high-frequency
switching regarding the bridge-rectifier-ground.
The power supply of the application (electronic power source, or variac) has a (relatively)
low impedance to mains ground, especially for high-frequency (> 20 kHz) signals.
When measuring with an oscilloscope on a high-frequency switching circuit, the probe
ground connection must be connected to the rectified mains connection or another
potential with a DC or a mains frequency-related AC offset voltage (see Figure 22).
The oscilloscope is correctly connected: The oscilloscope grounding is connected to
nodes without high-frequency voltages.
ISOLATED
POWER
SUPPLY
IC
IC GND
probe
OSCILLOSCOPE
aaa-016424
The red wires indicate connections with high-frequency/high-voltage signals.
Fig 22. Measuring method using the negative rectified mains voltage as ground
If an oscilloscope ground is connected to the IC-ground, the IC sees a huge capacitive
load in its switching node. This capacitive load can severely damage the application.
Even a differential probe or a hand-help multimeter can have too much capacitance to
ground. Measurement values may not be reliable.
Remark: Switching nodes (red): Do not create a (large) capacitive load to mains ground.
8.1 Common-mode filter connected to mains input of floating ground
application
To be able to measure regarding IC-ground, a large common-mode filter with sufficient
inductance value can be placed in the mains input (L = 33 mH or even more). After this
modification, the oscilloscope ground can be connected to IC-ground and IC pin voltages
can be measured.
Remark: If instabilities occur after connecting the oscilloscope, the inductance of the
applied common-mode mains filter must be increased for reliable scope measurements.
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Remark: As long as the additional common-mode filter is connected, no reliable
measurements with the system connected to a dimmer can be done. The common-mode
filter has a significant impact on the dimmed behavior.
ISOLATED
POWER
SUPPLY
IC
IC GND
probe
OSCILLOSCOPE
aaa-016425
The red wires indicate connections with high-frequency/high-voltage signals.
Fig 23. Measuring method using the floating IC ground as ground
Remark: Switching nodes (red): Do not create a (large) capacitive load to mains ground.
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9. Appendix 2: Tips and tricks: VCC supply for floating controllers
9.1 Measurement of the clamping current on the VCC pin
Especially for ICs with an internal VCC clamping circuit, it is relevant to know how much
power the IC internal VCC clamp dissipates.
The maximum IC clamping current rating is 8.8 mA. For higher VCC currents, the VCC
voltage increases to a value that is too high. The result can be damage to the IC ESD
protection. In general, if the ESD protection is damaged, the IC VCC voltage does not
reach the start-up level anymore.
Especially for low IC supply currents at small conduction angles, UVLO may be activated.
The converter starts cycling, resulting in a blinking LED.
A direct measurement with a common-mode choke does not work when a dimmer is
added (see Section 8; the common-mode filter affects dimming).
Therefore, a different measurement setup is required. An optocoupler in series with a
10 V Zener diode is switched to the VCC after IC start-up. When the switch is closed, VCC
is clamped to about 11 V.
The VCC clamping current now flows through the input diode of the optocoupler, instead
of through the IC. The optocoupler transfer characteristic is measured initially. Based on
the measured optocoupler output current over the full dim range, the VCC clamping
current can be calculated.
Figure 24 shows that to generate a DC optocoupler input current, an additional RC filter is
placed between VCC and the optocoupler.
Remark: The optocoupler Current Transfer Ratio (CTR) is not constant over input current.
Current spikes may influence the measurements.
DIMMER
POWER
SUPPLY
VCC
A
10 V
9V
IC
IC GND
aaa-016426
Fig 24. Measuring method of the VCC clamping current
Additional buffering of the VCC supply voltage can influence the dimming behavior. When
measuring the optocoupler device current with an oscilloscope, a filter capacitor in parallel
with the series connected Zener diode and optocoupler can be connected.
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The clamping current can be calculated with the voltage across R1 and the optocoupler
transfer curve. Make sure that the peak clamping current into the IC does not exceed the
maximum ICC rating of 8.8 mA.
Fig 25. Setup for the VCC clamping current measurement
In the graph below the optocoupler input current is plotted as function of the output current
(optocoupler used is KT101Y, but any optocoupler works. Make sure to measure the
transfer curve).
To simplify the calculation of the VCC clamping current (Iclamp), a trend line can be added
in Excel. For example, for an optocoupler output current of 5000 A, the VCC clamp
current is 930 A ( 26009  5000
0.6903
= 930 ).
Remark: The optocoupler CTR is temperature-dependent. Make sure that the transfer
curve is measured at the same temperature as the Iclamp testing temperature. If the
optocoupler is replaced for any reason, the transfer ratio must be remeasured as the
optocoupler CTR typically spreads significantly for the device within one batch. So the
spread over different batches is even worse.
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aaa-016438
104
Iin
(μA)
700
CTR
(%)
600
(1)
103
500
(2)
(3)
400
102
300
1 kΩ
200
10
lout
9V
100
1
1
10
102
103
0
105
104
Iout (μA)
A
A
lin
Var
aaa-016434
(1) Iout (A)
(2) CTR (%)
(3) Iin (A)
a. Curves
b. Circuit
Fig 26. Optocoupler transfer characteristic of the measured optocoupler device
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10. Appendix 3: Tips and tricks: VCC take-over supply generation
For controller ICs with a bleeder resistor supplying the ICC supply current, a take-over
supply may be required to prevent the converter from cycling. Cycling causes a blinking
LED at low conduction angles. The VCC pin is supplied by at least two sources:
• The bus voltage supplies the (start-up) current, which can be calculated with:
 V bus – V CC 
I CC  startup  = --------------------------------.
 R7 + R8 
• The charge current of the Cds capacitor of the high-voltage switching transistor:
The Cds capacitor is charged in series with the VCC capacitor during the high-voltage
MOSFET switch-off process. When the Cds capacitor is discharging, it is directly
connected to the IC ground. The ICC supply current generated by the switching, is
calculated with:I CC  switching  =  V bus + V LED – V CC   C ds  f sw. Make sure that the
total II(VCC) remains below the limit of 8.8 mA.
Three alternatives are discussed below.
10.1 VLED
R1
VCC
IC
C1
IC GND
D1
aaa-016435
Fig 27. VCC take-over supply generation using VLED
Capacitor C1 is charged to the LED voltage during the secondary stroke.
 V LED – 15 
The additional current to VCC can be calculated: I = ----------------------------- . For example, if
R1
VLED = 30 V, resistor R1 can be 60 k for I = 250 A.
Remark: Diode D1 must be a high-voltage diode (low-current is OK without capacitor C1
the additional current has to be multiplied by the toff/tcycle). A small reduction in LED
current and the line/load regulation occurs in this solution.
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10.2 Charge pump for deep-dimming applications
10.2.1 Basic implementation of charge pump
C1
VCC
D2
D1
IC
IC GND
aaa-016436
Fig 28. VCC take-over supply generation using a charge pump
During deep-dimming operation, the VCC voltage can reach the UVLO level. As a result
the IC enters the restart mode, which can cause some visible flicker. To improve
deep-dimming stability, the charge pump can be applied in deep-dimming applications.
The low-voltage diodes D1 and D2 can be a dual diode package. Capacitor C1 is a
high-voltage capacitor.
Choose a capacitance for capacitor C1 that is not too high. The maximum capacitance
value for C1 can be calculated in the following way: the clamp current of the IC
(II(VCC) = 2.6 mA), divided by the maximum bus voltage (Vbus(max) = Vmains(peak)) and
divided by the corresponding operating frequency (fsw):
I I  VCC 
C1 = --------------------------------------------------------------------------------.
 V bus  max  + V LED – V CC   f sw
A capacitance that is too high can overload the VCC clamping circuit, especially during
non-dim conditions (II(VCC) = 8.8 mA).
Component values for a 120 V mains circuit:
• C1 = 47 pF to 68 pF
To limit charge and discharge current peaks, a resistor (R2) can be mounted in series
with C1.
• R2 = 4.7 k (1206)
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10.2.2 Charge pump using internal diode connected to the SSL5231T SW-pin
C1
R2
R1
SW
VCC
IC
IC GND
aaa-016860
Fig 29. Charge pump using SSL5231T internal switch and diode connected to SW-pin
The low-voltage diodes D1 and D2 can be omitted by connecting the charge pump
capacitor to the SW pin of the IC, using the internal diode and MOSFET switch
(see Figure 29). Capacitor C1 is a high-voltage capacitor. To limit the charge and
discharge current peaks, the series resistor R2 is added.
For increased dimmer compatibility, the start-up bleeder resistor R1 can be placed in
parallel to the charge pump capacitor C1. The result is that the VCC capacitor cannot be
discharged through the bleeder resistor during the secondary stroke anymore.
This principle can be used in combination with an IC with an external high-voltage
MOSFET (charge pump connected to the SW pin).
Choose a capacitance for capacitor C1 that is not too high. The maximum capacitance
I I  VCC 
value for C1 can be calculated with:C1 = --------------------------------------------------------------------------------.
 V bus  max  + V LED – V CC   f sw
A capacitance that is too high can overload the VCC clamping circuit, especially during
non-dim conditions (II(VCC) = 8.8 mA).
Component values for a 120 V mains circuit:
• C1 = 47 pF to 68 pF (charge pump capacitor)
• R1 = 180 k (1206) (VCC start-up resistor)
• R2 = 4.7 k (1206) (charge current limiting resistor)
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10.3 Coupled inductor
During normal operation, a coupled separated winding on the main inductor, like the
auxiliary winding of flyback applications (see Figure 30), can generate the VCC supply.
VCC
IC
D1
IC GND
aaa-016437
Fig 30. VCC take-over supply generation using a coupled inductor
V LED
Choose a winding ratio that results in: -----------------------------------= V CC . VCC must be between 11 V
winding ratio
and 15 V (UVLO < VCC < Vclamp).
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11. Abbreviations
Table 4.
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Application note
Abbreviations
Acronym
Description
BCM
Boundary Conduction Mode
CCM
Continuous Conduction Mode
CTR
Current Transfer Ratio
DCM
Discontinuous Conduction Mode
DUT
Device Under Test
EMC
ElectroMagnetic Compatibility
EMI
ElectroMagnetic Interference
ESD
ElectroStatic Discharge
LEB
Leading-Edge Blanking
LED
Light-Emitting Diode
MOSFET
Metal-Oxide Semiconductor Field-Effect Transistor
OCP
OverCurrent Protection
OSP
Output Short Protection
OTP
OverTemperature Protection
OVP
OverVoltage Protection
PF
Power Factor
SSL
Solid-State Lighting
SWP
Short Winding Protection
THD
Total Harmonic Distortion
TVS
Transient Voltage Suppression
UVLO
UnderVoltage LockOut
ZCS
Zero Current Switching
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12. References
[1]
SSL5231T data sheet — Mains dimmable buck-boost LED driver IC
[2]
SSL5235TE data sheet — Mains dimmable buck-boost LED driver IC 5 Ohm/550 V
[3]
SSL5236TE data sheet — Mains dimmable buck-boost LED driver IC
20 Ohm/700 V
[4]
SSL5237TE data sheet — Mains dimmable buck-boost LED driver IC
10 Ohm/700 V
[5]
AN11060 application note — TEA172X 5 W to 11 W Power Supply/USB charger
[6]
AN11136 application note — SL2109T/AT/SSL2129AT controller for SSL
applications
[7]
AN11532 application note — SSL50x1 4 W to 25 W high-efficiency LED driver
[8]
AN11533 application note — SSL5301 mains phase cut dimmable LED driver
[9]
UM10846 user manual — SSL5237DB1247 120 V dimmable A19 driver board
[10] UM10847 user manual — SSL5236DB1249 230 V/4 W non-isolated buck-boost
candle-form dimmable LED driver dual demo board
[11] UM10861 user manual — SSL5231DB1246 230 V/10 W buck-boost dimmable A19
LED driver demo board
[12] NEMA SSL7A — Phase Cut Dimming for Solid-State Lighting: Basic Compatibility
AN11618
Application note
Silergy Corp. Confidential- Prepared for Customer Use Only
Rev. 2 — 31 March 2016
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